AD810

a FEATURES High Speed 80 MHz Bandwidth (3 dB, G = +1) 75 MHz Bandwidth (3 dB, G = +2) 1000 V/ s Slew Rate 50 ns Settling Time to 0.1% (VO = 10 V Step) Ideal for Video Applications 30 MHz Bandwidth (0.1 dB, G = +2) 0.02% Differential Gain 0.04 Differential Phase Low Noise 2.9 nV/√Hz Input Voltage Noise 13 pA/√Hz Inverting Input Current Noise Low Power 8.0 mA Supply Current max 2.1 mA Supply Current (Power-Down Mode) High Performance Disable Function Turn-Off Time 100 ns Break Before Make Guaranteed Input to Output Isolation of 64 dB (OFF State) Flexible Operation Specified for 5 V and 15 V Operation 2.9 V Output Swing Into a 150 Load (VS = 5 V) APPLICATIONS Professional Video Cameras Multimedia Systems NTSC, PAL & SECAM Compatible Systems Video Line Driver ADC/DAC Buffer DC Restoration Circuits Low Power Video Op Amp with Disable AD810 CONNECTION DIAGRAM 8-Pin Plastic Mini-DIP (N), SOIC (R) and Cerdip (Q) Packages OFFSET NULL –IN +IN –VS 1 2 3 4 TOP VIEW AD810 8 7 6 5 DISABLE +V S OUTPUT OFFSET NULL PRODUCT DESCRIPTION The AD810 is a composite and HDTV compatible, current feedback, video operational amplifier, ideal for use in systems such as multimedia, digital tape recorders and video cameras. The 0.1 dB flatness specification at bandwidth of 30 MHz (G = +2) and the differential gain and phase of 0.02% and 0.04° (NTSC) make the AD810 ideal for any broadcast quality video system. All these specifications are under load conditions of 150 Ω (one 75 Ω back terminated cable). The AD810 is ideal for power sensitive applications such as video cameras, offering a low power supply current of 8.0 mA max. The disable feature reduces the power supply current to only 2.1 mA, while the amplifier is not in use, to conserve power. Furthermore the AD810 is specified over a power supply range of ± 5 V to ± 15 V. The AD810 works well as an ADC or DAC buffer in video systems due to its unity gain bandwidth of 80 MHz. Because the AD810 is a transimpedance amplifier, this bandwidth can be maintained over a wide range of gains while featuring a low noise of 2.9 nV/√Hz for wide dynamic range applications. 0.10 0.20 GAIN = +2 RF = 715Ω RL = 150Ω fC = 3.58MHz 100 IRE MODULATED RAMP 0.18 0.16 0.14 0.12 0.10 GAIN PHASE 0.08 0.06 0.04 0.02 0 15 0 0.09 PHASE SHIFT – Degrees PHASE –90 1 –135 VS = ±15V –180 GAIN –1 ±2.5V –2 VS = ±15V –3 –4 –5 1 10 100 FREQUENCY – MHz 1000 ±5V –270 ±5V –225 DIFFERENTIAL GAIN – % –45 0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01 CLOSED-LOOP GAIN – dB 0 ±2.5V 0 5 6 7 8 9 10 11 12 13 14 SUPPLY VOLTAGE – ± Volts Closed-Loop Gain and Phase vs. Frequency, G = +2, RL = 150, RF = 715 Ω Differential Gain and Phase vs. Supply Voltage REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 DIFFERENTIAL PHASE – Degrees GAIN = +2 RL = 150Ω AD810–SPECIFICATIONS (@ T = +25 C and V = A S 15 V dc, RL = 150 Min 40 55 40 50 13 15 unless otherwise noted) Min 40 55 40 50 13 15 AD810S1 Typ Max 50 75 80 65 22 30 16 350 1000 50 125 0.02 0.04 0.04 0.045 –61 6 7.5 1.5 4 15 0.8 2 1.0 0.2 80 72 56 50 0.4 60 0.3 3.5 1.0 100 88 64 60 0.1 72 0.05 2.9 13 1.5 ± 2.5 ± 12 ± 2.5 ± 12.5 ± 12 30 ±3 ± 13 ± 2.9 ± 12.9 150 60 15 2.5 10 40 2 6 15 Units MHz MHz MHz MHz MHz MHz MHz V/µs V/µs ns ns % % Degrees Degrees dBc mV mV µV/°C µA µA MΩ MΩ dB dB dB dB µA/V dB µA/V nV/√Hz pA/√Hz pA/√Hz V V V V V mA mA Ω MΩ Ω pF dB Parameter DYNAMIC PERFORMANCE 3 dB Bandwidth Conditions (G = +2) RFB = 715 (G = +2) RFB = 715 (G = +1) RFB = 1000 (G = +10) RFB = 270 (G = +2) RFB = 715 (G = +2) RFB = 715 VO = 20 V p-p, RL = 400 Ω RL = 150 Ω RL = 400 Ω 10 V Step, G = –1 10 V Step, G = –1 f = 3.58 MHz f - 3.58 MHz f = 3.58 MHz f = 3.58 MHz f = 10 MHz, VO = 2 V p-p RL = 400 Ω, G = +2 TMIN–TMAX VS ±5 V ± 15 V ± 15 V ± 15 V ±5 V ± 15 V ± 15 V ±5 V ± 15 V ± 15 V ± 15 V ± 15 V ±5 V ± 15 V ±5 V ± 15 V ± 5 V, ± 15 V ± 5 V, ± 15 V AD810A Typ Max 50 75 80 65 22 30 16 350 1000 50 125 0.02 0.04 0.04 0.045 –61 1.5 2 7 0.7 2 0.1 dB Bandwidth Full Power Bandwidth Slew Rate2 Settling Time to 0.1% Settling Time to 0.01% Differential Gain Differential Phase Total Harmonic Distortion INPUT OFFSET VOLTAGE Offset Voltage Drift INPUT BIAS CURRENT –Input +Input OPEN-LOOP TRANSRESISTANCE OPEN-LOOP DC VOLTAGE GAIN COMMON-MODE REJECTION VOS ± Input Current POWER SUPPLY REJECTION VOS ± Input Current INPUT VOLTAGE NOISE INPUT CURRENT NOISE INPUT COMMON-MODE VOLTAGE RANGE OUTPUT CHARACTERISTICS Output Voltage Swing3 0.05 0.07 0.07 0.08 0.05 0.07 0.07 0.08 TMIN–TMAX TMIN–TMAX TMIN–TMAX VO = ± 10 V, RL = 400 Ω VO = ± 2.5 V, RL = 100 Ω TMIN–TMAX VO = ± 10 V, RL = 400 Ω VO = ± 2.5 V, RL = 100 Ω TMIN–TMAX VCM = ± 12 V VCM = ± 2.5 V TMIN–TMAX TMIN–TMAX TMIN–TMAX f = 1 kHz –IIN, f = 1 kHz +IIN, f = 1 kHz ± 5 V, ± 15 V ± 5 V, ± 15 V ± 15 V ±5 V ± 15 V ±5 V ± 15 V ±5 V ± 5 V, ± 15 V ± 4.5 V to ± 18 V 65 ± 5 V, ± 15 V ± 5 V, ± 15 V ± 5 V, ± 15 V ±5 V ± 15 V ± 2.5 ± 12 ± 2.5 ± 12.5 ± 12 40 1.0 0.3 86 76 56 52 5 7.5 5 10 3.5 1.2 100 88 64 60 0.1 72 0.05 2.9 13 1.5 ± 3.0 ± 13 ± 2.9 ± 12.9 150 60 15 0.4 0.3 RL = 150 Ω, TMIN–TMAX RL = 400 Ω RL = 400 Ω, TMIN–TMAX TMIN–TMAX Open Loop (5 MHz) +Input –Input +Input f = 5 MHz, See Figure 43 See Figure 43 Short-Circuit Current Output Current OUTPUT RESISTANCE INPUT CHARACTERISTICS Input Resistance Input Capacitance DISABLE CHARACTERISTICS4 OFF Isolation OFF Output Impedance ±5 V ± 15 V ± 15 V ± 15 V ± 5 V, ± 15 V ± 15 V ± 15 V ± 15 V 2.5 10 40 2 64 (RF + RG) 13 pF 64 (RF+ RG) 13 pF –2– REV. A AD810 Parameter Turn On Time Turn Off Time Disable Pin Current Min Disable Pin Current to Disable POWER SUPPLY Operating Range Quiescent Current TMIN–TMAX Power-Down Current 5 Conditions ZOUT = Low, See Figure 54 ZOUT = High Disable Pin = 0 V VS Min AD810A Typ Max 170 100 50 290 30 Min AD810S1 Typ Max 170 100 50 290 30 Units ns ns µA µA µA ±5 V ± 15 V ± 5 V, ± 15 V ± 2.5 ± 3.0 75 400 75 400 TMIN–TMAX +25°C to TMAX TMIN ±5 V ± 15 V ± 5 V, ± 15 V ±5 V ± 15 V 6.7 6.8 8.3 1.8 2.1 ± 18 ± 18 7.5 8.0 10.0 2.3 2.8 ± 2.5 ± 3.5 6.7 6.8 9 1.8 2.1 ± 18 ± 18 7.5 8.0 11.0 2.3 2.8 V V mA mA mA mA mA NOTES 1 See Analog Devices Military Data Sheet for 883B Specifications. 2 Slew rate measurement is based on 10% to 90% rise time with the amplifier configured for a gain of –10. 3 Voltage Swing is defined as useful operating range, not the saturation range. 4 Disable guaranteed break before make. 5 Turn On Time is defined with ± 5 V supplies using complementary output CMOS to drive the disable pin. Specifications subject to change without notice. TOTAL POWER DISSIPATION – Watts Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V Internal Power Dissipation2 . . . . . . . Observe Derating Curves Output Short Circuit Duration . . . . Observe Derating Curves Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . ± VS Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V Storage Temperature Range Plastic DIP . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +125°C Cerdip . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C Small Outline IC . . . . . . . . . . . . . . . . . . . –65°C to +125°C Operating Temperature Range AD810A . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C AD810S . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C Lead Temperature Range (Soldering 60 sec) . . . . . . . +300°C NOTES 1 Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum raring conditions for extended periods may affect device reliability. 2 8-Pin Plastic Package: θJA = 90°C/Watt; 8-Pin Cerdip Package: θJA = 110°C/Watt; 8-Pin SOIC Package: θJA = 150°C/Watt. ABSOLUTE MAXIMUM RATINGS 1 MAXIMUM POWER DISSIPATION The maximum power that can be safely dissipated by the AD810 is limited by the associated rise in junction temperature. For the plastic packages, the maximum safe junction temperature is 145°C. For the cerdip package, the maximum junction temperature is 175°C. If these maximums are exceeded momentarily, proper circuit operation will be restored as soon as the die temperature is reduced. Leaving the device in the “overheated” condition for an extended period can result in device burnout. To ensure proper operation, it is important to observe the derating curves. 2.4 2.2 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 –60 8-PIN SOIC –40 –20 0 20 40 60 80 8-PIN MINI-DIP 8-PIN CERDIP 8-PIN MINI-DIP 100 120 140 ESD SUSCEPTIBILITY AMBIENT TEMPERATURE – °C ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 volts, which readily accumulate on the human body and on test equipment, can discharge without detection. Although the AD810 features ESD protection circuitry, permanent damage may still occur on these devices if they are subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid any performance degradation or loss of functionality. ORDERING GUIDE Model AD810AN AD810AR AD810AR-REEL 5962-9313201MPA Temperature Range –40°C to +85°C –40°C to +85°C –40°C to +85°C –55°C to +125°C Package Description 8-Pin Plastic DIP 8-Pin Plastic SOIC 8-Pin Plastic SOIC 8-Pin Cerdip Package Option N-8 R-8 R-8 Q-8 Maximum Power Dissipation vs. Temperature While the AD810 is internally short circuit protected, this may not be sufficient to guarantee that the maximum junction temperature is not exceeded under all conditions. 0.1µF +VS 7 2 3 SEE TEXT 10kΩ 1 5 6 0.1µF AD810 4 –VS Offset Null Configuration REV. A –3– AD810 –Typical Characteristics MAGNITUDE OF THE OUTPUT VOLTAGE – ±Volts 20 MAGNITUDE OF THE OUTPUT VOLTAGE – ±Volts 20 15 NO LOAD 10 15 NO LOAD 10 RL = 150Ω 5 RL = 150Ω 5 0 0 5 10 15 SUPPLY VOLTAGE – ±Volts 20 0 0 5 10 15 SUPPLY VOLTAGE – ±Volts 20 Figure 1. Input Common-Mode Voltage Range vs. Supply Voltage 35 Figure 2. Output Voltage Swing vs. Supply 10 OUTPUT VOLTAGE – Volts p-p 30 ±15V SUPPLY 9 SUPPLY CURRENT – mA 25 20 15 10 ±5V SUPPLY 5 VS = ±15V 8 VS = ±5V 7 6 5 0 10 100 1k LOAD RESISTANCE – Ohms 10k 4 –60 –40 –20 0 20 40 60 80 100 120 140 JUNCTION TEMPERATURE – °C Figure 3. Output Voltage Swing vs. Load Resistance Figure 4. Supply Current vs. Junction Temperature 10 8 10 8 INPUT BIAS CURRENT – µA 6 NONINVERTING INPUT 4 2 0 –2 –4 –6 –8 –10 –60 –40 –20 20 40 60 80 100 120 140 INVERTING INPUT VS = ±5V, ±15V VS = ±5V, ±15V INPUT OFFSET VOLTAGE – mV 6 4 VS = ±5V 2 0 –2 –4 –6 –8 –10 –60 –40 –20 0 20 40 60 80 100 120 140 VS = ±15V 0 JUNCTION TEMPERATURE – °C JUNCTION TEMPERATURE – °C Figure 5. Input Bias Current vs. Temperature Figure 6. Input Offset Voltage vs. Junction Temperature –4– REV. A Typical Characteristics– AD810 250 120 SHORT CIRCUIT CURRENT – mA 100 VS = ±15V 150 OUTPUT CURRENT – mA 200 VS = ± 15V 80 60 VS = ± 5V 40 100 VS = ±5V 50 –60 –40 –20 0 +20 +40 +60 +80 +100 +120 +140 20 –60 –40 –20 0 +20 +40 +60 +80 +100 +120 +140 JUNCTION TEMPERATURE – °C JUNCTION TEMPERATURE – °C Figure 7. Short Circuit Current vs. Temperature Figure 8. Linear Output Current vs. Temperature 10.0 1M CLOSED-LOOP OUTPUT RESISTANCE – Ω OUTPUT RESISTANCE – Ω GAIN = 2 1.0 RF = 715Ω VS = ±5V 100k 10k VS = ±15V 0.1 1k 0.01 10k 100k 1M FREQUENCY – Hz 10M 100M 100 100k 1M FREQUENCY – Hz 10M 100M Figure 9. Closed-Loop Output Resistance vs. Frequency Figure 10. Output Resistance vs. Frequency, Disabled State 100 100 VS = ±5V TO ±15V 30 VS = ±15V OUTPUT VOLTAGE – Volts p-p 25 20 OUTPUT LEVEL FOR 3% THD RL = 400Ω INVERTING INPUT CURRENT NOISE 10 10 15 10 VS = ±5V 5 VOLTAGE NOISE NONINVERTING INPUT CURRENT NOISE 1 10 1 100k 0 100k 1M 10M FREQUENCY – Hz 100M 100 1k FREQUENCY – Hz 10k Figure 11. Large Signal Frequency Response Figure 12. Input Voltage and Current Noise vs. Frequency REV. A –5– CURRENT NOISE – pA/ Hz ± VOLTAGE NOISE – nV/ Hz AD810 –Typical Characteristics 100 90 80 70 60 50 40 30 20 10k 80 70 RF = 715Ω AV = +2 COMMON-MODE REJECTION – dB POWER SUPPLY REJECTION – dB 60 50 40 30 20 10 CURVES ARE FOR WORST CASE CONDITION WHERE ONE SUPPLY IS VARIED WHILE THE OTHER IS HELD CONSTANT 100k 1M FREQUENCY – Hz 10M 100M VS = ±15V VS = ±5V 100k 1M FREQUENCY – Hz 10M 100M 10k Figure 13. Common-Mode Rejection vs. Frequency Figure 14. Power Supply Rejection vs. Frequency –40 –40 HARMONIC DISTORTION – dBc VO = 2V p-p HARMONIC DISTORTION – dBc –60 RL = 100Ω GAIN = +2 2nd HARMONIC –60 ±15V SUPPLIES GAIN = +2 RL = 400Ω VS = ±5V –80 VOUT = 20V p-p –100 2nd HARMONIC 3rd HARMONIC –120 2nd 3rd VOUT = 2V p-p –80 3rd HARMONIC VS = ±15V 2nd –120 100 3rd –100 1k 10k 100k FREQUENCY – Hz 1M 10M –140 100 1k 10k 100k 1M 10M FREQUENCY – Hz Figure 15. Harmonic Distortion vs. Frequency (RL = 100 Ω) Figure 16. Harmonic Distortion vs. Frequency (RL = 400 Ω) 10 8 1200 RL = 400Ω OUTPUT SWING FROM ±V TO 0V 6 4 2 0 –2 –4 –6 –8 0.1% 0.01% 0.1% 0.01% 1000 SLEW RATE – V/µs 800 GAIN = –10 GAIN = +10 RF = RG = 1kΩ RL = 400Ω 600 GAIN = +2 400 –10 0 20 40 60 80 100 120 140 SETTLING TIME – ns 160 180 200 200 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 Figure 17. Output Swing and Error vs. Settling Time Figure 18. Slew Rate vs. Supply Voltage –6– REV. A Typical Characteristics, Noninverting Connection–AD810 RF 100 1V 20nS +VS 0.1µF VIN 90 RG 2 7 VO TO TEKTRONIX P6201 FET PROBE 6 VO RL 10 0% AD810 VIN HP8130 PULSE GENERATOR 50Ω –VS 3 4 0.1µF VO 1V Figure 19. Noninverting Amplifier Connection Figure 20. Small Signal Pulse Response, Gain = +1, RF = 1 kΩ, RL = 150 Ω, VS = ± 15 V 0 PHASE SHIFT – Degrees –45 –90 PHASE PHASE 1 GAIN = +1 RL = 1kΩ –45 –90 CLOSED-LOOP GAIN – dB 1 VS = ±15V 0 ±5V –1 –2 –3 –4 –5 1 10 100 FREQUENCY – MHz VS = ±15V ±5V ±2.5V GAIN –135 –180 –225 CLOSED-LOOP GAIN – dB VS = ±15V ±5V ±2.5V GAIN –135 –180 –225 –270 0 –1 –2 –3 –4 –5 VS = ±15V ±5V ±2.5V 1 10 100 FREQUENCY – MHz ±2.5V –270 1000 1000 Figure 21. Closed-Loop Gain and Phase vs. Frequency, G= +1. RF = 1 kΩ for ± 15 V, 910 Ω for ± 5 V and ± 2.5 V Figure 22. Closed-Loop Gain and Phase vs. Frequency, G= +1, RF = 1 kΩ for ± 15 V, 910 Ω for ± 5 V and ± 2.5 V 110 100 90 G = +1 RL = 150Ω VO = 250mV p-p RF = 750Ω 200 180 PEAKING ≤ 1dB 160 G = +1 RL = 1kΩ VO = 250mV p-p –3dB BANDWIDTH – MHz 80 70 60 50 40 30 20 –3dB BANDWIDTH – MHz PEAKING ≤ 1dB 140 120 100 80 60 40 20 RF = 1.5kΩ RF = 1kΩ RF = 750Ω PEAKING ≤ 0.1dB PEAKING ≤ 0.1 dB RF = 1kΩ RF = 1.5kΩ 2 4 6 8 10 12 14 16 18 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE – ±Volts SUPPLY VOLTAGE – ±Volts Figure 23. Bandwidth vs. Supply Voltage, Gain = +1, RL = 150 Ω Figure 24. –3 dB Bandwidth vs. Supply Voltage G = +1, RL = 1 kΩ REV. A –7– PHASE SHIFT – Degrees GAIN = +1 RL = 150Ω 0 AD810–Typical Characteristics, Noninverting Connection 100mV 100 90 20nS 100 1V 50nS VIN VIN 90 VO 10 0% VO 10 0% 1V 10V Figure 25. Small Signal Pulse Response, Gain = +10, RF = 442 Ω, RL = 150 Ω, VS = ± 15 V Figure 26. Large Signal Pulse Response, Gain = +10, RF = 442 Ω, RL = 400 Ω, VS = ± 15 V PHASE SHIFT – Degrees PHASE RL = 150Ω –45 –90 PHASE RF = 270Ω RL = 1kΩ –45 –90 –135 CLOSED-LOOP GAIN – dB CLOSED-LOOP GAIN – dB 21 20 19 GAIN 18 17 16 15 1 VS = ±15V ±5V ±2.5V VS = ±15V ±5V ±2.5V –135 –180 –225 –270 21 20 19 GAIN 18 17 16 15 1 VS = ±15V ±5V ±2.5V 10 100 FREQUENCY – MHz ±5V ±2.5V VS = ±15V –180 –225 –270 10 100 FREQUENCY – MHz 1000 1000 Figure 27. Closed-Loop Gain and Phase vs. Frequency, G = +10, RL = 150 Ω Figure 28. Closed-Loop Gain and Phase vs. Frequency, G = +10, RL = 1 kΩ 100 100 –3dB BANDWIDTH – MHz G = +10 RL = 150Ω VO = 250mV p-p G = +10 RL = 1kΩ VO = 250m V p-p –3dB BANDWIDTH – MHz 90 80 70 60 50 40 30 20 90 80 70 60 50 40 30 PEAKING ≤ 0.5dB RF = 232Ω PEAKING ≤ 0.5dB RF = 232Ω RF = 442Ω RF = 1kΩ PEAKING ≤ 0.1dB RF = 442Ω PEAKING ≤ 0.1dB 20 16 18 RF = 1kΩ 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts Figure 29. –3 dB Bandwidth vs. Supply Voltage, Gain = +10, RL = 150 Ω Figure 30. –3 dB Bandwidth vs. Supply Voltage, Gain = +10, RL = 1 kΩ –8– REV. A PHASE SHIFT– Degrees GAIN = +10 RF = 270Ω 0 GAIN = +10 0 Typical Characteristics, Inverting Connection– AD810 1V RF 100 20nS +VS VIN 90 0.1µF VO TO TEKTRONIX P6201 FET PROBE 6 0.1µF RL VO RG VIN HP8130 PULSE GENERATOR 2 7 AD810 3 4 VO 10 0% –VS 1V Figure 31. Inverting Amplifier Connection Figure 32. Small Signal Pulse Response, Gain = –1, RF = 681 Ω, RL = 150 Ω, VS = ± 5 V PHASE RL = 150Ω PHASE 90 1 45 VS = ±15V ±5V GAIN –2 –3 –4 –5 1 VS = ±15V ±5V ±2.5V ±2.5V –90 0 –45 RL = 1kΩ 90 45 1 CLOSED-LOOP GAIN – dB CLOSED-LOOP GAIN – dB 0 –1 0 –1 GAIN –2 –3 –4 –5 VS = ±15V ±5V ±2.5V VS = ±15V ±5V ±2.5V 0 –45 –90 10 100 FREQUENCY – MHz 1000 1 10 100 FREQUENCY – MHz 1000 Figure 33. Closed-Loop Gain and Phase vs. Frequency G = –1, RL = 150 Ω, RF = 681 Ω for ± 15 V, 620 Ω for ± 5 V and ± 2.5 V Figure 34. Closed-Loop Gain and Phase vs. Frequency, G = –1, RL = 1 kΩ, RF = 681 Ω for VS = ± 15 V, 620 Ω for ± 5 V and ± 2.5 V 100 90 G = –1 RL = 150 VO = 250mV p-p 180 G = –1 160 RL = 1kΩ VO = 250mV p-p –3dB BANDWIDTH – MHz –3dB BANDWIDTH – MHz 80 70 60 50 40 30 20 2 4 RF = 1kΩ RF = 681Ω RF = 500Ω PEAKING ≤ 1.0dB 140 120 100 80 60 40 20 PEAKING ≤ 1.0dB RF = 500Ω PEAKING ≤ 0.1dB RF = 649Ω RF = 1kΩ 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 PEAKING ≤ 0.1dB 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 Figure 35. –3 dB Bandwidth vs. Supply Voltage, Gain = –1, RL = 150 Ω Figure 36. –3 dB Bandwidth vs. Supply Voltage, Gain = –1, RL = 1 kΩ REV. A –9– PHASE SHIFT – Degrees GAIN = –1 PHASE SHIFT – Degrees 180 135 180 GAIN = –1 135 AD810 –Typical Characteristics, Inverting Connection 100mV 100 20nS 100 1V 50nS VIN 90 VIN 90 VO 10 0% VO 10 0% 1V 10V Figure 37. Small Signal Pulse Response, Gain = –10, RF = 442 Ω, RL = 150 Ω, VS = ± 15 V Figure 38. Large Signal Pulse Response, Gain = –10, RF = 442 Ω, RL = 400 Ω, VS = ± 15 V 180 GAIN = –10 PHASE RF = 249Ω RL = 150Ω 21 180 PHASE SHIFT – Degrees 135 90 45 0 PHASE RF = 249Ω RL = 1kΩ 135 90 45 0 CLOSED-LOOP GAIN – dB CLOSED-LOOP GAIN – dB 21 20 19 GAIN 18 17 16 15 1 VS = ±15V ±5V ±2.5V 10 100 FREQUENCY – MHz VS = ±15V ±5V ±2.5V 20 19 GAIN 18 17 16 15 1 10 100 FREQUENCY – MHz VS = ±15V ±5V ±2.5V VS = ±15V ±5V ±2.5V –45 –90 –45 –90 1000 1000 Figure 39. Closed-Loop Gain and Phase vs. Frequency, G = –10, RL = 150 Ω Figure 40. Closed-Loop Gain and Phase vs. Frequency, G = –10, RL = 1 kΩ 100 G = –10 RL = 150Ω VO = 250mV p- p 100 NO PEAKING G = –10 NO PEAKING –3dB BANDWIDTH – MHz 80 70 60 50 40 30 20 –3dB BANDWIDTH – MHz 90 90 80 70 60 50 40 30 RL = 1kΩ VO = 250mV p- p RF = 249Ω RF = 249Ω RF = 442Ω RF = 442Ω RF = 750Ω RF = 750Ω 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 20 2 4 6 8 10 12 14 SUPPLY VOLTAGE – ±Volts 16 18 Figure 41. –3 dB Bandwidth vs. Supply Voltage, G = –10, RL = 150 Ω Figure 42. –3 dB Bandwidth vs. Supply Voltage, G = –10, R L = 1 kΩ –10– REV. A PHASE SHIFT – Degrees GAIN = –10 Applications– AD810 GENERAL DESIGN CONSIDERATIONS PRINTED CIRCUIT BOARD LAYOUT The AD810 is a current feedback amplifier optimized for use in high performance video and data acquisition systems. Since it uses a current feedback architecture, its closed-loop bandwidth depends on the value of the feedback resistor. Table I below contains recommended resistor values for some useful closedloop gains and supply voltages. As you can see in the table, the closed-loop bandwidth is not a strong function of gain, as it would be for a voltage feedback amp. The recommended resistor values will result in maximum bandwidths with less than 0.1 dB of peaking in the gain vs. frequency response. The –3 dB bandwidth is also somewhat dependent on the power supply voltage. Lowering the supplies increases the values of internal capacitances, reducing the bandwidth. To compensate for this, smaller values of feedback resistor are sometimes used at lower supply voltages. The characteristic curves illustrate that bandwidths of over 100 MHz on 30 V total and over 50 MHz on 5 V total supplies can be achieved. Table I. –3 dB Bandwidth vs. Closed-Loop Gain and Resistance Values (RL = 150 ) As with all wideband amplifiers, PC board parasitics can affect the overall closed-loop performance. Most important are stray capacitances at the output and inverting input nodes. (An added capacitance of 2 pF between the inverting input and ground will add about 0.2 dB of peaking in the gain of 2 response, and increase the bandwidth to 105 MHz.) A space (3/16" is plenty) should be left around the signal lines to minimize coupling. Also, signal lines connecting the feedback and gain resistors should be short enough so that their associated inductance does not cause high frequency gain errors. Line lengths less than 1/4" are recommended. QUALITY OF COAX CABLE Optimum flatness when driving a coax cable is possible only when the driven cable is terminated at each end with a resistor matching its characteristic impedance. If coax were ideal, then the resulting flatness would not be affected by the length of the cable. While outstanding results can be achieved using inexpensive cables, some variation in flatness due to varying cable lengths is to be expected. POWER SUPPLY BYPASSING VS = 15 V Closed-Loop Gain +1 +2 +10 –1 –10 VS = 5 V Closed-Loop Gain +1 +2 +10 –1 –10 RFB 1 kΩ 715 Ω 270 Ω 681 Ω 249 Ω RG 715 Ω 30 Ω 681 Ω 24.9 Ω –3 dB BW (MHz) 80 75 65 70 65 –3 dB BW (MHz) 50 50 50 55 50 RFB 910 Ω 715 Ω 270 Ω 620 Ω 249 Ω RG 715 Ω 30 Ω 620 Ω 24.9 Ω Adequate power supply bypassing can be critical when optimizing the performance of a high frequency circuit. Inductance in the power supply leads can contribute to resonant circuits that produce peaking in the amplifier's response. In addition, if large current transients must be delivered to the load, then bypass capacitors (typically greater than 1 µF) will be required to provide the best settling time and lowest distortion. Although the recommended 0.1 µF power supply bypass capacitors will be sufficient in most applications, more elaborate bypassing (such as using two paralleled capacitors) may be required in some cases. The AD810 will operate with supplies from ± 18 V down to about ± 2.5 V. On ± 2.5 V the low distortion output voltage swing will be better than 1 V peak to peak. Single supply operation can be realized with excellent results by arranging for the input common-mode voltage to be biased at the supply midpoint. A 10 kΩ pot connected between Pins 1 and 5, with its wiper connected to V+, can be used to trim out the inverting input current (with about ± 20 µA of range). For closed-loop gains above about 5, this may not be sufficient to trim the output offset voltage to zero. Tie the pot's wiper to ground through a large value resistor (50 kΩ for ± 5 V supplies, 150 kΩ for ± 15 V supplies) to trim the output to zero at high closed-loop gains. OFFSET NULLING POWER SUPPLY OPERATING RANGE ACHIEVING VERY FLAT GAIN RESPONSE AT HIGH FREQUENCY Achieving and maintaining gain flatness of better than 0.1 dB above 10 MHz is not difficult if the recommended resistor values are used. The following issues should be considered to ensure consistently excellent results. CHOICE OF FEEDBACK AND GAIN RESISTOR Because the 3 dB bandwidth depends on the feedback resistor, the fine scale flatness will, to some extent, vary with feedback resistor tolerance. It is recommended that resistors with a 1% tolerance be used if it is desired to maintain exceptional flatness over a wide range of production lots. REV. A –11– AD810 CAPACITIVE LOADS LOAD CAPACITANCE – pF When used with the appropriate feedback resistor, the AD810 can drive capacitive loads exceeding 1000 pF directly without oscillation. By using the curves in Figure 45 to chose the resistor value, less than 1 dB of peaking can easily be achieved without sacrificing much bandwidth. Note that the curves were generated for the case of a 10 kΩ load resistor, for smaller load resistances, the peaking will be less than indicated by Figure 45. Another method of compensating for large load capacitances is to insert a resistor in series with the loop output as shown in Figure 43. In most cases, less than 50 Ω is all that is needed to achieve an extremely flat gain response. Figures 44 to 46 illustrate the outstanding performance that can be achieved when driving a 1000 pF capacitor. RF +VS 0.1µF 1.0µF RG 2 7 RS (OPTIONAL) 1000 VS = ±5V 100 VS = ±15V 10 GAIN = +2 RL = 1kΩ 1 0 1k 2k 3k 4k FEEDBACK RESISTOR – Ω Figure 45. Max Load Capacitance for Less than 1 dB of Peaking vs. Feedback Resistor 5V 100nS AD810 VIN RT –VS 3 4 6 1.0µF 0.1µF CL RL VO VIN 100 90 VOUT Figure 43. Circuit Options for Driving a Large Capacitive Load G = +2 VS = ±15V RL= 10kΩ CL = 1000pF 0% 5V CLOSED-LOOP GAIN – dB 9 6 3 0 –3 –6 –9 RF = 4.5kΩ RS = 0 Figure 46. AD810 Driving a 1000 pF Load, Gain = +2, RF = 750 Ω, RS = 11 Ω, RL = 10 kΩ DISABLE MODE RF = 750Ω RS = 11Ω 1 10 FREQUENCY – MHz 100 By pulling the voltage on Pin 8 to common (0 V), the AD810 can be put into a disabled state. In this condition, the supply current drops to less than 2.8 mA, the output becomes a high impedance, and there is a high level of isolation from input to output. In the case of a line driver for example, the output impedance will be about the same as for a 1.5 kΩ resistor (the feedback plus gain resistors) in parallel with a 13 pF capacitor (due to the output) and the input to output isolation will be better than 65 dB at 1 MHz. Leaving the disable pin disconnected (floating) will leave the AD810 operational in the enabled state. In cases where the amplifier is driving a high impedance load, the input to output isolation will decrease significantly if the input signal is greater than about 1.2 V peak to peak. The isolation can be restored back to the 65 dB level by adding a dummy load (say 150 Ω) at the amplifier output. This will attenuate the feedthrough signal. (This is not an issue for multiplexer applications where the outputs of multiple AD810s are tied together as long as at least one channel is in the ON state.) The input impedance of the disable pin is about 35 kΩ in parallel with a few pF. When grounded, about 50 µA flows out Figure 44. Performance Comparison of Two Methods for Driving a Large Capacitive Load –12– REV. A AD810 DIFFERENTIAL GAIN – % 0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01 0 5 6 7 8 9 10 11 12 13 14 SUPPLY VOLTAGE – ± Volts GAIN PHASE 0.16 0.14 0.12 0.10 0.08 0.06 0.04 0.02 0 15 When operated on ± 15 V supplies, the AD810 disable pin may be driven by open drain logic such as the 74C906. In this case, adding a 10 kΩ pull-up resistor from the disable pin to the plus supply will decrease the enable time to about 150 ns. If there is a nonzero voltage present on the amplifier's output at the time it is switched to the disabled state, some additional decay time will be required for the output voltage to relax to zero. The total time for the output to go to zero will generally be about 250 ns and is somewhat dependent on the load impedance. OPERATION AS A VIDEO LINE DRIVER Figure 49. Differential Gain and Phase vs. Supply Voltage +0.1 RL = 150Ω 0 –0.1 ±5V ±2.5 ±15V 715Ω 715Ω +VS NORMALIZED GAIN – dB The AD810 is designed to offer outstanding performance at closed-loop gains of one or greater. At a gain of 2, the AD810 makes an excellent video line driver. The low differential gain and phase errors and wide –0.1 dB bandwidth are nearly independent of supply voltage and load (as seen in Figures 49 and 50). +0.1 RL= 1k 0 ±15V –0.1 ±5V 0.1µF 7 2 75Ω CABLE VIN 3 75Ω 75Ω CABLE VOUT 75Ω AD810 4 0.1µF 6 ±2.5 1M 10M FREQUENCY – Hz 100M 100k 75Ω –VS Figure 50. Fine-Scale Gain (Normalized) vs. Frequency for Various Supply Voltages, Gain = +2, RF = 715 Ω 110 Figure 47. A Video Line Driver Operating at a Gain of +2 0 100 G = +2 RL = 150Ω VO = 250mV p-p RF = 500 PEAKING ≤ 1.0dB PHASE SHIFT – Degrees –3dB BANDWIDTH – MHz PHASE GAIN = +2 RL = 150Ω –45 –90 90 80 70 60 CLOSED-LOOP GAIN – dB 1 0 GAIN –1 ±2.5V –2 –3 –4 –5 1 10 100 FREQUENCY – MHz VS = ±15V ±5V ±2.5V VS = ±15V –135 –180 ±5V –225 –270 RF = 750 50 40 RF = 1k 30 20 2 4 6 8 10 12 PEAKING ≤ 0.1dB 1000 14 16 18 SUPPLY VOLTAGE - ±Volts Figure 48. Closed-Loop Gain and Phase vs. Frequency, G = +2, RL = 150, RF = 715 Ω Figure 51. –3 dB Bandwidth vs. Supply Voltage, Gain = +2, RL = 150 Ω REV. A –13– DIFFERENTIAL PHASE – Degrees of the disable the disable pin for ± 5 V supplies. If driven by complementary output CMOS logic (such as the 74HC04), the disable time (until the output goes high impedance) is about 100 ns and the enable time (to low impedance output) is about 170 ns on ± 5 V supplies. The enable time can be extended to about 750 ns by using open drain logic such as the 74HC05. 0.10 0.09 GAIN = +2 RF = 715Ω RL = 150Ω fC = 3.58MHz 100 IRE MODULATED RAMP 0.20 0.18 AD810 2:1 VIDEO MULTIPLEXER 750Ω +5V 0.1µF 2 VINA 3 75Ω 750Ω +5V 7 750Ω The outputs of two AD810s can be wired together to form a 2:1 mux without degrading the flatness of the gain response. Figure 54 shows a recommended configuration which results in –0.1 dB bandwidth of 20 MHz and OFF channel isolation of 77 dB at 10 MHz on ± 5 V supplies. The time to switch between channels is about 0.75 µs when the disable pins are driven by open drain output logic. Adding pull-up resistors to the logic outputs or using complementary output logic (such as the 74HC04) reduces the switching time to about 180 ns. The switching time is only slightly affected by the signal level. 500mV 100 90 AD810 4 8 –5V 6 0.1µF 75Ω 75Ω CABLE VOUT 75Ω 750Ω 0.1µF 500nS VINB 2 7 AD810 3 4 8 –5V 6 0.1µF 75Ω VSW 10 0% 74HC04 5V Figure 54. A Fast Switching 2:1 Video Mux Figure 52. Channel Switching Time for the 2:1 Mux –45 –50 0.5 –90 –135 –180 GAIN –225 –270 VS = ±5V CLOSED-LOOP GAIN – dB FEEDTHROUGH – dB 0 –0.5 –1.0 –1.5 –2.0 –2.5 –3.0 1 10 FREQUENCY – MHz –60 –70 –80 100 –90 1 10 FREQUENCY – MHz 100 Figure 53. 2:1 Mux OFF Channel Feedthrough vs. Frequency Figure 55. 2:1 Mux ON Channel Gain and Phase vs. Frequency –14– REV. A PHASE SHIFT – Degrees –40 PHASE 0 AD810 N:1 MULTIPLEXER A multiplexer of arbitrary size can be formed by combining the desired number of AD810s together with the appropriate selection logic. The schematic in Figure 58 shows a recommendation for a 4:1 mux which may be useful for driving a high impedance such as the input to a video A/D converter (such as the AD773). The output series resistors effectively compensate for the combined output capacitance of the OFF channels plus the input capacitance of the A/D while maintaining wide bandwidth. In the case illustrated, the –0.1 dB bandwidth is about 20 MHz with no peaking. Switching time and OFF channel isolation (for the 4:1 mux) are about 250 ns and 60 dB at 10 MHz, respectively. PHASE SHIFT – Degrees 0 PHASE –45 0.5 –90 –135 –180 GAIN –1.0 –1.5 –2.0 –2.5 VS = ±15V RL = 10kΩ CL = 10pF –225 1kΩ +VS 0.1µF 2 VIN, A 75Ω 7 AD810 3 0.1µF –VS +VS 1kΩ 8 4 33Ω 6 SELECT A 0.1µF 2 VIN, B 3 75Ω 0.1µF –VS +VS 0.1µF 2 VIN, C 7 33Ω RL CL 1kΩ 7 33Ω AD810 8 4 6 SELECT B VOUT CLOSED-LOOP GAIN – dB 0 –0.5 AD810 3 8 4 6 –3.0 1 10 FREQUENCY – MHz 100 75Ω 0.1µF –VS +VS 0.1µF 1kΩ SELECT C Figure 56. 4:1 Mux ON Channel Gain and Phase vs. Frequency –30 2 7 FEEDTHROUGH – dB –40 VIN, D 3 75Ω 0.1µF AD810 8 4 6 33Ω SELECT D –VS –50 –60 Figure 58. A 4:1 Multiplexer Driving a High Impedance –70 1 10 FREQUENCY – MHz 100 Figure 57. 4:1 Mux OFF Channel Feedthrough vs. Frequency REV. A –15– AD810 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). Plastic Mini-DIP (N) Package 8 PIN 1 1 4 0.30 (7.62) REF 0.035 ±0.01 (0.89 ±0.25) 5 0.31 (7.87) 0.39 (9.91) MAX 0.165 ±0.01 (4.19 ±0.25) 0.125 (3.18) MIN 0.018 ±0.003 (0.46 ±0.08) 0.10 (2.54) BSC 0.033 (0.84) NOM 0.18 ±0.03 (4.57 ±0.76) 0.011 ±0.003 (0.28 ±0.08) 15° 0° SEATING PLANE Cerdip (Q) Package 0.005 (0.13) MIN 0.055 (1.40) MAX 8 PIN 1 1 5 0.310 (7.87) 0.220 (5.59) 4 0.320 (8.13) 0.290 (7.37) 0.060 (1.52) 0.015 (0.38) 0.405 (10.29) MAX 0.200 (5.08) MAX 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.100 0.070 (1.78) 0.014 (0.36) (2.54) 0.030 (0.76) BSC 0.150 (3.81) MIN 0.015 (0.38) 0.008 (0.20) 15° 0° SEATING PLANE 8-Pin SOIC (R) Package 0.150 (3.81) 8 0.244 (6.20) 0.228 (5.79) PIN 1 1 5 0.157 (3.99) 0.150 (3.81) 4 0.020 (0.051) x 45° CHAMF 0.190 (4.82) 0.170 (4.32) 8° 0° 10° 0° 0.098 (0.2482) 0.075 (0.1905) 0.030 (0.76) 0.018 (0.46) 0.197 (5.01) 0.189 (4.80) 0.010 (0.25) 0.004 (0.10) 0.050 (1.27) BSC 0.102 (2.59) 0.094 (2.39) 0.019 (0.48) 0.014 (0.36) 0.090 (2.29) All brand or product names mentioned are trademarks or registered trademarks of their respective holders. –16– REV. A PRINTED IN U.S.A. C1737–24–10/92 0.25 (6.35)