AD8138ARZ

Low Distortion, Differential ADC Driver AD8138 Data Sheet PIN CONFIGURATION –IN 1 8 +IN VOCM 2 7 NC V+ 3 +OUT 4 AD8138 6 V– 5 –OUT NC = NO CONNECT Figure 1. TYPICAL APPLICATION CIRCUIT 5V 5V 499Ω VIN 499Ω + VOCM 499Ω AIN AIN – AVDD DVDD DIGITAL OUTPUTS ADC AD8138 AVSS VREF 01073-002 Easy to use, single-ended-to-differential conversion Adjustable output common-mode voltage Externally adjustable gain Low harmonic distortion −94 dBc SFDR at 5 MHz −85 dBc SFDR at 20 MHz −3 dB bandwidth of 320 MHz, G = +1 Fast settling to 0.01% of 16 ns Slew rate 1150 V/µs Fast overdrive recovery of 4 ns Low input voltage noise of 5 nV/√Hz 1 mV typical offset voltage Wide supply range +3 V to ±5 V Low power 90 mW on 5 V 0.1 dB gain flatness to 40 MHz Available in 8-Lead SOIC and MSOP packages 01073-001 FEATURES 499Ω Figure 2. APPLICATIONS ADC drivers Single-ended-to-differential converters IF and baseband gain blocks Differential buffers Line drivers GENERAL DESCRIPTION The AD8138 is a major advancement over op amps for differential signal processing. The AD8138 can be used as a single-ended-to-differential amplifier or as a differential-todifferential amplifier. The AD8138 is as easy to use as an op amp and greatly simplifies differential signal amplification and driving. Manufactured on the proprietary ADI XFCB bipolar process, the AD8138 has a −3 dB bandwidth of 320 MHz and delivers a differential signal with the lowest harmonic distortion available in a differential amplifier. The AD8138 has a unique internal feedback feature that provides balanced output gain and phase matching, suppressing even order harmonics. The internal feedback circuit also minimizes any gain error that would be associated with the mismatches in the external gain setting resistors. The differential output of the AD8138 helps balance the input to differential ADCs, maximizing the performance of the ADC. Rev. G The AD8138 eliminates the need for a transformer with high performance ADCs, preserving the low frequency and dc information. The common-mode level of the differential output is adjustable by a voltage on the VOCM pin, easily level-shifting the input signals for driving single-supply ADCs. Fast overload recovery preserves sampling accuracy. The AD8138 distortion performance makes it an ideal ADC driver for communication systems, with distortion performance good enough to drive state-of-the-art 10-bit to 16-bit converters at high frequencies. The high bandwidth and IP3 of the AD8138 also make it appropriate for use as a gain block in IF and baseband signal chains. The AD8138 offset and dynamic performance makes it well suited for a wide variety of signal processing and data acquisition applications. The AD8138 is available in both SOIC and MSOP packages for operation over −40°C to +85°C temperatures. Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©1999–2016 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD8138 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1  Definition of Terms .................................................................... 16  Applications ....................................................................................... 1  Theory of Operation ...................................................................... 17  Pin Configuration ............................................................................. 1  Analyzing an Application Circuit ............................................ 17  Typical Application Circuit ............................................................. 1  Setting the Closed-Loop Gain .................................................. 17  General Description ......................................................................... 1  Estimating the Output Noise Voltage ...................................... 17  Revision History ............................................................................... 2  The Impact of Mismatches in the Feedback Networks ......... 18  Specifications..................................................................................... 3  Calculating the Input Impedance of an Application ............. 18  ±DIN to ±OUT Specifications...................................................... 3  Input Common-Mode Voltage Range in Single-Supply Applications ................................................................................ 18  VOCM to ±OUT Specifications ..................................................... 4  ±DIN to ±OUT Specifications...................................................... 5  VOCM to ±OUT Specifications ..................................................... 6  Absolute Maximum Ratings............................................................ 7  Thermal Resistance ...................................................................... 7  ESD Caution .................................................................................. 7  Pin Configuration and Function Descriptions ............................. 8  Typical Performance Characteristics ............................................. 9  Test Circuits ..................................................................................... 15  Setting the Output Common-Mode Voltage .......................... 18  Driving a Capacitive Load......................................................... 18  Layout, Grounding, and Bypassing .............................................. 19  Balanced Transformer Driver ....................................................... 20  High Performance ADC Driving ................................................. 21  3 V Operation ................................................................................. 22  Outline Dimensions ....................................................................... 23  Ordering Guide .......................................................................... 23  Operational Description ................................................................ 16  REVISION HISTORY 3/16—Rev. F to Rev. G Changes to Setting the Closed-Loop Gain Section .................... 17 Changes to Figure 46 ...................................................................... 21 Changes to Figure 47 ...................................................................... 22 1/06—Rev. E to Rev. F Changes to Features.......................................................................... 1 Added Thermal Resistance Section and Maximum Power Dissipation Section ........................................................................... 7 Changes to Balanced Transformer Driver Section..................... 20 Changes to Ordering Guide .......................................................... 23 3/03—Rev. D to Rev. E Changes to Specifications ................................................................ 2 Changes to Ordering Guide ............................................................ 4 Changes to TPC 16 ........................................................................... 6 Changes to Table I ............................................................................ 9 Added New Paragraph after Table I ............................................. 10 Updated Outline Dimensions ....................................................... 14 7/02—Rev. C to Rev. D Addition of TPC 35 and TPC 36 .....................................................8 6/01—Rev. B to Rev. C Edits to Specifications ......................................................................2 Edits to Ordering Guide ...................................................................4 12/00—Rev. A to Rev. B 9/99—Rev. 0 to Rev. A 3/99—Rev. 0: Initial Version Rev. G | Page 2 of 24 Data Sheet AD8138 SPECIFICATIONS ±DIN TO ±OUT SPECIFICATIONS At 25°C, VS = ±5 V, VOCM = 0, G = +1, RL, dm = 500 Ω, unless otherwise noted. Refer to Figure 39 for test setup and label descriptions. All specifications refer to single-ended input and differential outputs, unless otherwise noted. Table 1. Parameter DYNAMIC PERFORMANCE −3 dB Small Signal Bandwidth Bandwidth for 0.1 dB Flatness Large Signal Bandwidth Slew Rate Settling Time Overdrive Recovery Time NOISE/HARMONIC PERFORMANCE 1 Second Harmonic Third Harmonic IMD IP3 Voltage Noise (RTI) Input Current Noise INPUT CHARACTERISTICS Offset Voltage Conditions Min Typ VOUT = 0.5 V p-p, CF = 0 pF VOUT = 0.5 V p-p, CF = 1 pF VOUT = 0.5 V p-p, CF = 0 pF VOUT = 2 V p-p, CF = 0 pF VOUT = 2 V p-p, CF = 0 pF 0.01%, VOUT = 2 V p-p, CF = 1 pF VIN = 5 V to 0 V step, G = +2 290 320 225 30 265 1150 16 4 MHz MHz MHz MHz V/µs ns ns −94 −87 −62 −114 −85 −57 −77 37 5 2 dBc dBc dBc dBc dBc dBc dBc dBm nV/√Hz pA/√Hz VOUT = 2 V p-p, 5 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 20 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 70 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 5 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 20 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 70 MHz, RL, dm = 800 Ω 20 MHz 20 MHz f = 100 kHz to 40 MHz f = 100 kHz to 40 MHz VOS, dm = VOUT, dm/2; VDIN+ = VDIN− = VOCM = 0 V TMIN to TMAX variation Input Bias Current Input Resistance Input Capacitance Input Common-Mode Voltage CMRR OUTPUT CHARACTERISTICS Output Voltage Swing Output Current Output Balance Error 1 TMIN to TMAX variation Differential Common mode ∆VOUT, dm/∆VIN, cm; ∆VIN, cm = ±1 V Maximum ∆VOUT; single-ended output ∆VOUT, cm/∆VOUT, dm; ∆VOUT, dm = 1 V −2.5 ±1 ±4 3.5 −0.01 6 3 1 −4.7 to +3.4 −77 7.75 95 −66 Harmonic distortion performance is equal or slightly worse with higher values of RL, dm. See Figure 17 and Figure 18 for more information. Rev. G | Page 3 of 24 Max +2.5 7 −70 Unit mV µV/°C µA µA/°C MΩ MΩ pF V dB V p-p mA dB AD8138 Data Sheet VOCM TO ±OUT SPECIFICATIONS At 25°C, VS = ±5 V, VOCM = 0, G = +1, RL, dm = 500 Ω, unless otherwise noted. Refer to Figure 39 for test setup and label descriptions. All specifications refer to single-ended input and differential outputs, unless otherwise noted. Table 2. Parameter DYNAMIC PERFORMANCE −3 dB Bandwidth Slew Rate INPUT VOLTAGE NOISE (RTI) DC PERFORMANCE Input Voltage Range Input Resistance Input Offset Voltage Input Bias Current VOCM CMRR Gain POWER SUPPLY Operating Range Quiescent Current Power Supply Rejection Ratio OPERATING TEMPERATURE RANGE Conditions Min f = 0.1 MHz to 100 MHz VOS, cm = VOUT, cm; VDIN+ = VDIN– = VOCM = 0 V –3.5 ∆VOUT, dm/∆VOCM; ∆VOCM = ±1 V ∆VOUT, cm/∆VOCM; ∆VOCM = ±1 V 0.9955 ±1.4 18 TMIN to TMAX variation ∆VOUT, dm/∆VS; ∆VS = ±1 V −40 Rev. G | Page 4 of 24 Typ Max Unit 250 330 17 MHz V/µs nV/√Hz ±3.8 200 ±1 0.5 −75 1 V kΩ mV µA dB V/V 20 40 −90 +3.5 1.0045 ±5.5 23 −70 +85 V mA µA/°C dB °C Data Sheet AD8138 ±DIN TO ±OUT SPECIFICATIONS At 25°C, VS = 5 V, VOCM = 2.5 V, G = +1, RL, dm = 500 Ω, unless otherwise noted. Refer to Figure 39 for test setup and label descriptions. All specifications refer to single-ended input and differential output, unless otherwise noted. Table 3. Parameter DYNAMIC PERFORMANCE −3 dB Small Signal Bandwidth Bandwidth for 0.1 dB Flatness Large Signal Bandwidth Slew Rate Settling Time Overdrive Recovery Time NOISE/HARMONIC PERFORMANCE 1 Second Harmonic Third Harmonic IMD IP3 Voltage Noise (RTI) Input Current Noise INPUT CHARACTERISTICS Offset Voltage Conditions Min Typ VOUT = 0.5 V p-p, CF = 0 pF VOUT = 0.5 V p-p, CF = 1 pF VOUT = 0.5 V p-p, CF = 0 pF VOUT = 2 V p-p, CF = 0 pF VOUT = 2 V p-p, CF = 0 pF 0.01%, VOUT = 2 V p-p, CF = 1 pF VIN = 2.5 V to 0 V step, G = +2 280 310 225 29 265 950 16 4 MHz MHz MHz MHz V/µs ns ns −90 −79 −60 −100 −82 −53 −74 35 5 2 dBc dBc dBc dBc dBc dBc dBc dBm nV/√Hz pA/√Hz VOUT = 2 V p-p, 5 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 20 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 70 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 5 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 20 MHz, RL, dm = 800 Ω VOUT = 2 V p-p, 70 MHz, RL, dm = 800 Ω 20 MHz 20 MHz f = 100 kHz to 40 MHz f = 100 kHz to 40 MHz VOS, dm = VOUT, dm/2; VDIN+ = VDIN– = VOCM = 0 V TMIN to TMAX variation Input Bias Current Input Resistance Input Capacitance Input Common-Mode Voltage CMRR OUTPUT CHARACTERISTICS Output Voltage Swing Output Current Output Balance Error 1 TMIN to TMAX variation Differential Common mode ∆VOUT, dm/∆VIN, cm; ∆VIN, cm = 1 V Maximum ∆VOUT; single-ended output ∆VOUT, cm/∆VOUT, dm; ∆VOUT, dm = 1 V −2.5 ±1 ±4 3.5 −0.01 6 3 1 −0.3 to +3.2 −77 2.9 95 −65 Harmonic distortion performance is equal or slightly worse with higher values of RL, dm. See Figure 17 and Figure 18 for more information. Rev. G | Page 5 of 24 Max +2.5 7 −70 Unit mV µV/°C µA µA/°C MΩ MΩ pF V dB V p-p mA dB AD8138 Data Sheet VOCM TO ±OUT SPECIFICATIONS At 25°C, VS = 5 V, VOCM = 2.5 V, G = +1, RL, dm = 500 Ω, unless otherwise noted. Refer to Figure 39 for test setup and label descriptions. All specifications refer to single-ended input and differential output, unless otherwise noted. Table 4. Parameter DYNAMIC PERFORMANCE −3 dB Bandwidth Slew Rate INPUT VOLTAGE NOISE (RTI) DC PERFORMANCE Input Voltage Range Input Resistance Input Offset Voltage Input Bias Current VOCM CMRR Gain POWER SUPPLY Operating Range Quiescent Current Power Supply Rejection Ratio OPERATING TEMPERATURE RANGE Conditions Min f = 0.1 MHz to 100 MHz VOS, cm = VOUT, cm; VDIN+ = VDIN– = VOCM = 0 V −5 ∆VOUT, dm/∆VOCM; ∆VOCM = 2.5 V ±1 V ∆VOUT, cm/∆VOCM; ∆VOCM = 2.5 V ±1 V 0.9968 2.7 15 TMIN to TMAX variation ∆VOUT, dm/∆VS; ∆VS = ± 1 V −40 Rev. G | Page 6 of 24 Typ Max Unit 220 250 17 MHz V/µs nV/√Hz 1.0 to 3.8 100 ±1 0.5 −70 1 V kΩ mV µA dB V/V 20 40 −90 +5 1.0032 11 21 −70 +85 V mA µA/°C dB °C Data Sheet AD8138 ABSOLUTE MAXIMUM RATINGS Table 5. Ratings ±5.5 V ±VS 550 mW −40°C to +85°C −65°C to +150°C 300°C 150°C Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability. THERMAL RESISTANCE θJA is specified for the worst-case conditions, that is, θJA is specified for the device soldered in a circuit board in still air. Airflow reduces θJA. In addition, more metal directly in contact with the package leads from metal traces through holes, ground, and power planes reduces the θJA. Figure 3 shows the maximum safe power dissipation in the package vs. the ambient temperature for the 8-lead SOIC (121°C/W) and 8-lead MSOP (θJA = 145°C/W) packages on a JEDEC standard 4-layer board. θJA values are approximations. 1.75 Table 6. Package Type 8-Lead SOIC/4-Layer 8-Lead MSOP/4-Layer θJA 121 145 Unit °C/W °C/W Maximum Power Dissipation The maximum safe power dissipation in the AD8138 packages is limited by the associated rise in junction temperature (TJ) on the die. At approximately 150°C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8138. Exceeding a junction temperature of 150°C for an extended period can result in changes in the silicon devices, potentially causing failure. 1.50 1.25 1.00 SOIC 0.75 MSOP 0.50 0.25 0 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90 100 110 120 AMBIENT TEMPERATURE (°C) Figure 3. Maximum Power Dissipation vs. Temperature ESD CAUTION Rev. G | Page 7 of 24 01073-049 MAXIMUM POWER DISSIPATION (W) Parameter Supply Voltage VOCM Internal Power Dissipation Operating Temperature Range Storage Temperature Range Lead Temperature (Soldering 10 sec) Junction Temperature The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). The load current consists of the differential and common-mode currents flowing to the load, as well as currents flowing through the external feedback networks and internal common-mode feedback loop. The internal resistor tap used in the common-mode feedback loop places a negligible differential load on the output. RMS voltages and currents should be considered when dealing with ac signals. AD8138 Data Sheet –IN 1 8 +IN VOCM 2 7 NC V+ 3 6 V– 5 –OUT +OUT 4 AD8138 NC = NO CONNECT 01073-004 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS Figure 4. Pin Configuration Table 7. Pin Function Descriptions Pin No. 1 2 Mnemonic −IN VOCM 3 4 5 6 7 8 V+ +OUT −OUT V− NC +IN Description Negative Input Summing Node. Voltage applied to this pin sets the common-mode output voltage with a ratio of 1:1. For example, 1 V dc on VOCM sets the dc bias level on +OUT and −OUT to 1 V. Positive Supply Voltage. Positive Output. Note that the voltage at −DIN is inverted at +OUT (see Figure 42). Negative Output. Note that the voltage at +DIN is inverted at −OUT (see Figure 42). Negative Supply Voltage. No Connect. Positive Input Summing Node. Rev. G | Page 8 of 24 Data Sheet AD8138 TYPICAL PERFORMANCE CHARACTERISTICS Unless otherwise noted, Gain = 1, RG = RF = RL, dm = 499 V, TA = 25°C; refer to Figure 39 for test setup. 6 6 VIN = 2V p-p CF = 0pF VIN = 0.2V p-p CF = 0pF 3 3 VS = +5V 0 GAIN (dB) VS = ±5V –3 –6 0 VS = ±5V –3 1 10 100 FREQUENCY (MHz) 1000 –9 01073-005 –9 1 Figure 5. Small Signal Frequency Response 10 100 FREQUENCY (MHz) Figure 8. Large Signal Frequency Response 6 6 VS = ±5V VIN = 0.2V p-p VIN = 2V p-p VS = ±5V 3 3 CF = 0pF CF = 0pF 0 GAIN (dB) GAIN (dB) 1000 01073-008 –6 CF = 1pF –3 0 CF = 1pF –3 –6 –6 100 10 FREQUENCY (MHz) 1 1000 –9 01073-006 –9 1 Figure 6. Small Signal Frequency Response 10 100 FREQUENCY (MHz) 1000 01073-009 GAIN (dB) VS = +5V Figure 9. Large Signal Frequency Response 0.5 30 VS = ±5V VIN = 0.2V p-p 0.3 CF = 0pF G = 10, RF = 4.99kΩ 20 VS = ±5V CF = 0pF VOUT, dm = 0.2V p-p RG = 499Ω GAIN (dB) –0.1 10 G = 2, RF = 1kΩ CF = 1pF G = 1, RF = 499Ω 0 –0.5 1 10 FREQUENCY (MHz) 100 Figure 7. 0.1 dB Flatness vs. Frequency –10 1 100 10 FREQUENCY (MHz) 1000 Figure 10. Small Signal Frequency Response for Various Gains Rev. G | Page 9 of 24 01073-010 –0.3 01073-007 GAIN (dB) G = 5, RF = 2.49kΩ 0.1 AD8138 Data Sheet –50 –60 VOUT, dm = 2V p-p RL = 800Ω VS = ±5V RL = 800Ω –60 HD3 (F = 20MHz) –70 HD2 (F = 20MHz) DISTORTION (dBc) DISTORTION (dBc) –70 HD2 (VS = +5V) –80 HD2 (VS = ±5V) –90 –80 –90 HD2 (F = 5MHz) –100 –100 HD3 (VS = +5V) HD3 (F = 5MHz) –110 –110 10 30 40 50 20 FUNDAMENTAL FREQUENCY (MHz) 60 70 –120 01073-011 0 0 Figure 11. Harmonic Distortion vs. Frequency 1 2 3 4 5 DIFFERENTIAL OUTPUT VOLTAGE (V p-p) Figure 14. Harmonic Distortion vs. Differential Output Voltage –60 –40 VOUT, dm = 4V p-p RL = 800Ω VS = 5V RL = 800Ω –50 –70 HD2 (F = 20MHz) DISTORTION (dBc) HD3 (VS = +5V) –60 DISTORTION (dBc) 6 01073-014 HD3 (VS = ±5V) –120 –70 HD2 (VS = +5V) –80 HD2 (VS = ±5V) –80 HD3 (F = 20MHz) –90 HD2 (F = 5MHz) –100 –90 HD3 (F = 5MHz) –110 HD3 (VS = ±5V) 0 10 20 30 40 50 FUNDAMENTAL FREQUENCY (MHz) 60 70 –120 01073-012 –110 0 1 2 3 DIFFERENTIAL OUTPUT VOLTAGE (V p-p) Figure 15. Harmonic Distortion vs. Differential Output Voltage Figure 12. Harmonic Distortion vs. Frequency –60 –30 VS = 3V RL = 800Ω VOUT, dm = 2V p-p RL = 800Ω FO = 20MHz –70 DISTORTION (dBc) –50 HD2 (VS = +5V) –60 HD3 (VS = +5V) –70 –80 HD3 (VS = ±5V) –90 HD2 (VS = ±5V) –80 –90 HD2 (F = 5MHz) –4 –3 –2 0 1 VOCM DC OUTPUT (V) –1 2 3 4 01073-013 –100 –100 HD3 (F = 20MHz) HD2 (F = 20MHz) –110 0.25 HD3 (F = 5MHz) 0.50 1.00 1.25 0.75 1.50 DIFFERENTIAL OUTPUT VOLTAGE (V p-p) 1.75 Figure 16. Harmonic Distortion vs. Differential Output Voltage Figure 13. Harmonic Distortion vs. VOCM Rev. G | Page 10 of 24 01073-016 –40 DISTORTION (dBc) 4 01073-015 –100 Data Sheet AD8138 –60 45 RL = 800Ω VS = 5V VOUT, dm = 2V p-p –70 INTERCEPT (dBm) DISTORTION (dBc) 40 HD2 (F = 20MHz) –80 HD3 (F = 20MHz) –90 HD2 (F = 5MHz) VS = ±5V 35 VS = +5V 30 –100 600 1000 RLOAD (Ω) 1400 25 01073-017 –110 200 1800 0 Figure 17. Harmonic Distortion vs. RLOAD 20 40 FREQUENCY (MHz) 60 Figure 20. Third-Order Intercept vs. Frequency –60 VS = ±5V VS = ±5V VOUT, dm = 2V p-p –70 VOUT, dm HD3 (F = 20MHz) VOUT– –90 VOUT+ HD2 (F = 5MHz) –100 V+DIN HD3 (F = 5MHz) –110 600 1000 RLOAD (Ω) 1400 5ns 01073-018 1V –120 200 01073-021 DISTORTION (dBc) HD2 (F = 20MHz) –80 80 1800 Figure 18. Harmonic Distortion vs. RLOAD Figure 21. Large Signal Transient Response 10 FC = 50MHz VS = ±5V CF = 0pF VOUT, dm = 0.2V p-p VS = ±5V –10 CF = 1pF –50 –90 40mV –110 49.5 49.7 49.9 50.1 FREQUENCY (MHz) 50.3 50.5 5ns Figure 19. Intermodulation Distortion Figure 22. Small Signal Transient Response Rev. G | Page 11 of 24 01073-022 –70 01073-019 POUT (dBm) –30 01073-020 HD3 (F = 5MHz) AD8138 Data Sheet VOUT, dm = 2V p-p CF = 0pF VS = ±5V VOUT, dm VS = +5V VS = ±5V F = 20MHz V+DIN = 8V p-p G = 3 (RF = 1500) 5ns 4V Figure 26. Output Overdrive Figure 23. Large Signal Transient Response VOUT, dm = 2V p-p VS = ±5V CF = 0pF 30ns 01073-026 400mV 01073-023 V+DIN VS = ±5V CF = 0pF CL = 10pF CL = 5pF CF = 1pF 5ns Figure 24. Large Signal Transient Response 400mV 2.5ns 01073-028 400mV 01073-024 CL = 20pF Figure 27. Large Signal Transient Response for Various Cap Loads (See Figure 40) –20 VS = ±5V ∆VOUT, dm/∆VIN, cm VS = ±5V CF = 1pF 200µV –30 VOUT, dm CMRR (dB) –40 –50 –60 4ns 01073-025 1V –80 1 10 100 FREQUENCY (MHz) Figure 28. CMRR vs. Frequency Figure 25. Settling Time Rev. G | Page 12 of 24 1k 01073-029 –70 V+DIN Data Sheet AD8138 –20 5.0 –40 VS = ±5V –50 –60 VS = +5V –70 100 10 FREQUENCY (MHz) 1 1k Figure 29. Output Balance Error vs. Frequency (See Figure 41) 2.5 VS = ±5V VS = +5V 0 VS = +3V –2.5 –5.0 –40 01073-031 BALANCE ERROR (dB) –30 –20 0 20 40 TEMPERATURE (°C) 60 80 100 01073-034 DIFFERENTIAL OUTPUT OFFSET (mV) VIN = 2V p-p Figure 32. Output Referred Differential Offset Voltage vs. Temperature –10 5 ΔVOUT, dm/ΔVS –20 4 –PSRR (VS = ±5V) BIAS CURRENT (µA) PSRR (dB) –30 –40 –50 –60 +PSRR (VS = +5V, 0V AND ±5V) –70 VS = ±5V, +5V 3 VS = +3V 2 1 10 100 FREQUENCY (MHz) 1k 1 –40 01073-032 –90 Figure 30. PSRR vs. Frequency –20 0 20 40 60 TEMPERATURE (°C) 80 100 01073-035 –80 Figure 33. Input Bias Current vs. Temperature 100 30 SINGLE-ENDED OUTPUT SUPPLY CURRENT (mA) 10 VS = +5V 1 10 FREQUENCY (MHz) VS = +5V 15 VS = +3V 100 5 –40 –20 0 60 20 40 TEMPERATURE (°C) 80 Figure 34. Supply Current vs. Temperature Figure 31. Output Impedance vs. Frequency Rev. G | Page 13 of 24 100 01073-036 0.1 1 VS = ±5V 20 10 VS = ±5V 01073-033 IMPEDANCE (Ω) 25 AD8138 Data Sheet 100 6 VS = +5V –3 –9 1 10 100 FREQUENCY (MHz) 1k 10 1.1pA/ Hz 1 01073-037 –6 10 Figure 35. VOCM Frequency Response 100 1k 10k FREQUENCY (Hz) 100k 1M Figure 37. Current Noise (RTI) 1000 VOUT, cm 400mV 5ns 100 10 5.7nV/ Hz 1 10 Figure 36. VOCM Transient Response 100 1k 10k FREQUENCY (Hz) Figure 38. Voltage Noise (RTI) Rev. G | Page 14 of 24 100k 1M 01073-040 INPUT VOLTAGE NOISE (nV/ Hz) VS = ±5V VOCM = –1V TO +1V 01073-038 GAIN (dB) 0 01073-039 INPUT CURRENT NOISE (pA/ Hz) VS = ±5V 3 Data Sheet AD8138 TEST CIRCUITS 499Ω RG = 499Ω 49.9Ω RG = 499Ω 499Ω 49.9Ω 499Ω RL, dm = 499Ω AD8138 24.9Ω 01073-003 24.9Ω RF = 499Ω 499Ω 24.9Ω 24.9Ω AD8138 24.9Ω CL 453Ω 499Ω 01073-027 499Ω AD8138 249Ω 499Ω Figure 41. Test Circuit for Output Balance Figure 39. Basic Test Circuit 49.9Ω 499Ω 249Ω Figure 40. Test Circuit for Cap Load Drive Rev. G | Page 15 of 24 01073-030 RF = 499Ω AD8138 Data Sheet OPERATIONAL DESCRIPTION DEFINITION OF TERMS Common-mode voltage refers to the average of two node voltages. The output common-mode voltage is defined as CF VOUT, cm = (V+OUT + V−OUT)/2 RF RG –OUT +IN VOCM AD8138 –DIN RG RL, dm VOUT, dm +OUT –IN RF CF 01073-041 +DIN Figure 42. Circuit Definitions Differential voltage refers to the difference between two node voltages. For example, the output differential voltage (or equivalently output differential-mode voltage) is defined as Balance is a measure of how well differential signals are matched in amplitude and exactly 180° apart in phase. Balance is most easily determined by placing a well-matched resistor divider between the differential voltage nodes and comparing the magnitude of the signal at the midpoint of the divider with the magnitude of the differential signal (see Figure 41). By this definition, output balance is the magnitude of the output common-mode voltage divided by the magnitude of the output differential mode voltage: VOUT, dm = (V+OUT − V−OUT) where V+OUT and V−OUT refer to the voltages at the +OUT and −OUT terminals with respect to a common reference. Rev. G | Page 16 of 24 Output Balance Error = VOUT , cm VOUT , dm Data Sheet AD8138 THEORY OF OPERATION The AD8138 differs from conventional op amps in that it has two outputs whose voltages move in opposite directions. Like an op amp, it relies on high open-loop gain and negative feedback to force these outputs to the desired voltages. The AD8138 behaves much like a standard voltage feedback op amp and makes it easy to perform single-ended-to-differential conversion, common-mode level-shifting, and amplification of differential signals. Also like an op amp, the AD8138 has high input impedance and low output impedance. Previous differential drivers, both discrete and integrated designs, have been based on using two independent amplifiers and two independent feedback loops, one to control each of the outputs. When these circuits are driven from a single-ended source, the resulting outputs are typically not well balanced. Achieving a balanced output has typically required exceptional matching of the amplifiers and feedback networks. DC common-mode level-shifting has also been difficult with previous differential drivers. Level-shifting has required the use of a third amplifier and feedback loop to control the output common-mode level. Sometimes the third amplifier has also been used to attempt to correct an inherently unbalanced circuit. Excellent performance over a wide frequency range has proven difficult with this approach. The AD8138 uses two feedback loops to separately control the differential and common-mode output voltages. The differential feedback, set with external resistors, controls only the differential output voltage. The common-mode feedback controls only the common-mode output voltage. This architecture makes it easy to arbitrarily set the output common-mode level. It is forced, by internal common-mode feedback, to be equal to the voltage applied to the VOCM input, without affecting the differential output voltage. ANALYZING AN APPLICATION CIRCUIT The AD8138 uses high open-loop gain and negative feedback to force its differential and common-mode output voltages in such a way as to minimize the differential and common-mode error voltages. The differential error voltage is defined as the voltage between the differential inputs labeled +IN and −IN in Figure 42. For most purposes, this voltage can be assumed to be zero. Similarly, the difference between the actual output commonmode voltage and the voltage applied to VOCM can also be assumed to be zero. Starting from these two assumptions, any application circuit can be analyzed. SETTING THE CLOSED-LOOP GAIN Neglecting the capacitors CF, the differential-mode gain of the circuit in Figure 42 can be determined to be described by VOUT , dm VIN , dm = S RF S RG This assumes the input resistors, RGS, and feedback resistors, RFS, on each side are equal. ESTIMATING THE OUTPUT NOISE VOLTAGE Similar to the case of a conventional op amp, the differential output errors (noise and offset voltages) can be estimated by multiplying the input referred terms, at +IN and −IN, by the circuit noise gain. The noise gain is defined as R GN = 1 +  F  RG     To compute the total output referred noise for the circuit of Figure 42, consideration must also be given to the contribution of the Resistors RF and RG. Refer to Table 8 for the estimated output noise voltage densities at various closed-loop gains. The AD8138 architecture results in outputs that are very highly balanced over a wide frequency range without requiring tightly matched external components. The common-mode feedback loop forces the signal component of the output common-mode voltage to be zeroed. The result is nearly perfectly balanced differential outputs of identical amplitude and exactly 180°apart in phase. Table 8. Gain 1 2 5 10 RG (Ω) 499 499 499 499 RF (Ω) 499 1.0 k 2.49 k 4.99 k Bandwidth −3 dB 320 MHz 180 MHz 70 MHz 30 MHz Output Noise AD8138 Only 10 nV/√Hz 15 nV/√Hz 30 nV/√Hz 55 nV/√Hz Rev. G | Page 17 of 24 Output Noise AD8138 + RG, RF 11.6 nV/√Hz 18.2 nV/√Hz 37.9 nV/√Hz 70.8 nV/√Hz AD8138 Data Sheet When using the AD8138 in gain configurations where RF/RG of one feedback network is unequal to RF/RG of the other network, there is a differential output noise due to input-referred voltage in the VOCM circuitry. The output noise is defined in terms of the following feedback terms (refer to Figure 42): β1 = RG RF + RG CALCULATING THE INPUT IMPEDANCE OF AN APPLICATION The effective input impedance of a circuit such as the one in Figure 42, at +DIN and –DIN, depends on whether the amplifier is being driven by a single-ended or differential signal source. For balanced differential input signals, the input impedance (RIN, dm) between the inputs (+DIN and −DIN) is simply RIN, dm =2 × RG for −OUT to +IN loop, and β2 = In the case of a single-ended input signal (for example if −DIN is grounded and the input signal is applied to +DIN), the input impedance becomes RG RF + RG for +OUT to −IN loop. With these defined, VnOUT , dm  β − β2  = 2VnIN ,VOCM  1   β1 + β2  RIN , dm where VnOUT, dm is the output differential noise, and VnIN ,VCOM is the input-referred voltage noise in VOCM. THE IMPACT OF MISMATCHES IN THE FEEDBACK NETWORKS As previously mentioned, even if the external feedback networks (RF/RG) are mismatched, the internal common-mode feedback loop still forces the outputs to remain balanced. The amplitudes of the signals at each output remains equal and 180° out of phase. The input-to-output differential-mode gain varies proportionately to the feedback mismatch, but the output balance is unaffected. Ratio matching errors in the external resistors result in a degradation of the ability of the circuit to reject input commonmode signals, much the same as for a four-resistor difference amplifier made from a conventional op amp. In addition, if the dc levels of the input and output commonmode voltages are different, matching errors result in a small differential-mode output offset voltage. For the G = 1 case, with a ground referenced input signal and the output common-mode level set for 2.5 V, an output offset of as much as 25 mV (1% of the difference in common-mode levels) can result if 1% tolerance resistors are used. Resistors of 1% tolerance result in a worstcase input CMRR of about 40 dB, worst-case differential mode output offset of 25 mV due to 2.5 V level-shift, and no significant degradation in output balance error.     RG   = RF 1−   2 × (RG + RF )   The input impedance of the circuit is effectively higher than it would be for a conventional op amp connected as an inverter because a fraction of the differential output voltage appears at the inputs as a common-mode signal, partially bootstrapping the voltage across the input resistor RG. INPUT COMMON-MODE VOLTAGE RANGE IN SINGLE-SUPPLY APPLICATIONS The AD8138 is optimized for level-shifting, ground-referenced input signals. For a single-ended input, this would imply, for example, that the voltage at −DIN in Figure 42 would be 0 V when the negative power supply voltage of the amplifier (at V−) is also set to 0 V. SETTING THE OUTPUT COMMON-MODE VOLTAGE The VOCM pin of the AD8138 is internally biased at a voltage approximately equal to the midsupply point (average value of the voltages on V+ and V−). Relying on this internal bias results in an output common-mode voltage that is within about 100 mV of the expected value. In cases where more accurate control of the output commonmode level is required, it is recommended that an external source, or resistor divider (made up of 10 kΩ resistors), be used. The output common-mode offset listed in the Specifications section assumes the VOCM input is driven by a low impedance voltage source. DRIVING A CAPACITIVE LOAD A purely capacitive load can react with the pin and bondwire inductance of the AD8138, resulting in high frequency ringing in the pulse response. One way to minimize this effect is to place a small capacitor across each of the feedback resistors. The added capacitance should be small to avoid destabilizing the amplifier. An alternative technique is to place a small resistor in series with the outputs of the amplifier, as shown in Figure 40. Rev. G | Page 18 of 24 Data Sheet AD8138 LAYOUT, GROUNDING, AND BYPASSING As a high speed device, the AD8138 is sensitive to the PCB environment in which it has to operate. Realizing its superior specifications requires attention to various details of good high speed PCB design. The first requirement is for a good solid ground plane that covers as much of the board area around the AD8138 as possible. The only exception to this is that the two input pins (Pin 1 and Pin 8) should be kept a few millimeters from the ground plane, and ground should be removed from inner layers and the opposite side of the board under the input pins. This minimizes the stray capacitance on these nodes and helps preserve the gain flatness vs. frequency. The power supply pins should be bypassed as close as possible to the device to the nearby ground plane. Good high frequency ceramic chip capacitors should be used. This bypassing should be done with a capacitance value of 0.01 µF to 0.1 µF for each supply. Further away, low frequency bypassing should be provided with 10 µF tantalum capacitors from each supply to ground. The signal routing should be short and direct to avoid parasitic effects. Wherever there are complementary signals, a symmetrical layout should be provided to the extent possible to maximize the balance performance. When running differential signals over a long distance, the traces on the PCB should be close together or any differential wiring should be twisted together to minimize the area of the loop that is formed. This reduces the radiated energy and makes the circuit less susceptible to interference. Rev. G | Page 19 of 24 AD8138 Data Sheet BALANCED TRANSFORMER DRIVER CSTRAY VUNBAL 52.3Ω PRIMARY CSTRAY NO SIGNAL IS COUPLED ON THIS SIDE Figure 43. Transformer Single-Ended-to-Differential Converter Is Inherently Imbalanced 499Ω CSTRAY 49.9Ω 499Ω The balance of a differential circuit can be measured by connecting an equal-valued resistive voltage divider across the differential outputs and then measuring the center point of the circuit with respect to ground. Since the two differential outputs are supposed to be of equal amplitude, but 180° opposite phase, there should be no signal present for perfectly balanced outputs. Figure 45 compares the transformer being driven singleendedly by a signal generator and being driven differentially using an AD8138. The top signal trace of Figure 45 shows the balance of the single-ended configuration, while the bottom shows the differentially driven balance response. The 100 MHz balance is 35 dB better when using the AD8138. +IN OUT– VUNBAL AD8138 OUT+ –IN 49.9Ω 500Ω 0.005% VDIFF 500Ω 0.005% 499Ω CSTRAY 499Ω Figure 44. AD8138 Forms a Balanced Transformer Driver 0 OUTPUT BALANCE ERROR (dB) The circuit in Figure 43 shows a Mini-Circuits® T1-6T transformer connected with its primary driven single-endedly and the secondary connected with a precision voltage divider across its terminals. The voltage divider is made up of two 500 Ω, 0.005% precision resistors. The voltage VUNBAL, which is also equal to the ac common-mode voltage, is a measure of how closely the outputs are balanced. 500Ω 0.005% SECONDARY V DIFF 500Ω 0.005% 01073-042 If the interwinding capacitance (CSTRAY) is assumed to be uniformly distributed, a signal from the driving source couples to the secondary output terminal that is closest to the driven side of the primary. On the other hand, no signal is coupled to the opposite terminal of the secondary because its nearest primary terminal is not driven (see Figure 43). The exact amount of this imbalance depends on the particular parasitics of the trans-former, but is mostly a problem at higher frequencies. SIGNAL IS COUPLED ON THIS SIDE VIA CSTRAY 01073-043 However, when driving the transformer in a single-ended manner, there is an imbalance at the output due to the parasitics inherent in the transformer. The primary (or driven) side of the transformer has one side at dc potential (usually ground), while the other side is driven. This can cause problems in systems that require good balance of the differential output signals of the transformer. The well-balanced outputs of the AD8138 provide a drive signal to each of the primary inputs of the transformer that are of equal amplitude and 180° out of phase. Therefore, depending on how the polarity of the secondary is connected, the signals that conduct across the interwinding capacitance either both assist the secondary signal of the transformer equally, or both buck the secondary signals. In either case, the parasitic effect is symmetrical and provides a well-balanced transformer output (see Figure 45). –20 –40 VUNBAL , FOR TRANSFORMER WITH SINGLE-ENDED DRIVE –60 –80 VUNBAL , DIFFERENTIAL DRIVE –100 0.3 1 10 FREQUENCY (MHz) 100 500 01073-044 Transformers are among the oldest devices used to perform a single-ended-to-differential conversion (and vice versa). Transformers can also perform the additional functions of galvanic isolation, step-up or step-down of voltages, and impedance transformation. For these reasons, transformers always find uses in certain applications. Figure 45. Output Balance Error for Circuits of Figure 43 and Figure 44 Rev. G | Page 20 of 24 Data Sheet AD8138 HIGH PERFORMANCE ADC DRIVING The signal generator has a ground-referenced, bipolar output, that is, it drives symmetrically above and below ground. Connecting VOCM to the CML pin of the AD9224 sets the output common-mode of the AD8138 at 2.5 V, which is the midsupply level for the AD9224. This voltage is bypassed by a 0.1 µF capacitor. The circuit in Figure 46 shows a simplified front-end connection for an AD8138 driving an AD9224, a 12-bit, 40 MSPS ADC. The ADC works best when driven differentially, which minimizes its distortion. The AD8138 eliminates the need for a transformer to drive the ADC and performs singleended-to-differential conversion, common-mode level-shifting, and buffering of the driving signal. The full-scale analog input range of the AD9224 is set to 4 V p-p, by shorting the SENSE terminal to AVSS. This has been determined to be the scaling to provide minimum harmonic distortion. The positive and negative outputs of the AD8138 are connected to the respective differential inputs of the AD9224 via a pair of 49.9 Ω resistors to minimize the effects of the switched-capacitor front end of the AD9224. For best distortion performance, it runs from supplies of ±5 V. For the AD8138 to swing at 4 V p-p, each output swings 2 V p-p while providing signals that are 180° out of phase. With a common-mode voltage at the output of 2.5 V, each AD8138 output swings between 1.5 V and 3.5 V. The AD8138 is configured with unity gain for a single-ended, input-to-differential output. The additional 23 Ω, 523 Ω total, at the input to −IN is to balance the parallel impedance of the 50 Ω source and its 50 Ω termination that drives the noninverting input. A ground-referenced 4 V p-p, 5 MHz signal at DIN+ was used to test the circuit in Figure 46. When the combined-device circuit was run with a sampling rate of 20 MSPS, the spurious-free dynamic range (SFDR) was measured at −85 dBc. +5V +5V 499Ω 3 499Ω 49.9Ω 2 523Ω + 5 49.9Ω 24 VINB 0.1µF 15 26 28 AVDD DRVDD VOCM AD8138 4 1 49.9Ω 23 VINA 6 0.1µF DIGITAL OUTPUTS AD9224 AVSS 16 25 SENSE CML 17 22 DRVSS 27 499Ω 01073-045 50Ω SOURCE 8 0.1µF –5V Figure 46. AD8138 Driving an AD9224, a 12-Bit, 40 MSPS ADC Rev. G | Page 21 of 24 AD8138 Data Sheet 3 V OPERATION The circuit in Figure 47 shows a simplified front-end connection for an AD8138 driving an AD9203, a 10-bit, 40 MSPS ADC that is specified to work on a single 3 V supply. The ADC works best when driven differentially to make the best use of the signal swing available within the 3 V supply. The appropriate outputs of the AD8138 are connected to the appropriate differential inputs of the AD9203 via a low-pass filter. The circuit was tested with a −0.5 dBFS signal at various frequencies. Figure 48 shows a plot of the total harmonic distortion (THD) vs. frequency at signal amplitudes of 1 V and 2 V differential drive levels. –40 –45 –50 The AD8138 is configured for unity gain for a single-ended input to differential output. The additional 23 Ω at the input to −IN is to balance the impedance of the 50 Ω source and its 50 Ω termination that drives the noninverting input. THD (dBc) –55 AD8138–2V –60 –65 The signal generator has ground-referenced, bipolar output, that is, it can drive symmetrically above and below ground. Even though the AD8138 has ground as its negative supply, it can still function as a level-shifter with such an input signal. AD8138–1V –70 –80 0 The output common mode is raised up to midsupply by the voltage divider that biases VOCM. In this way, the AD8138 provides dc coupling and level-shifting of a bipolar signal, without inverting the input signal. AD8138 523Ω 4 1 0.1µF 6 49.9Ω 20pF 49.9Ω 20pF 28 25 AVDD AINP AINN 26 61 59 2 DRVDD AD9203 AVSS DRVSS 27 1 DIGITAL OUTPUTS 57 AD8138–1V 55 AD8138–2V 53 51 499Ω 01073-046 10kΩ Figure 47. AD8138 Driving an AD9203, a 10-Bit, 40 MSPS Analog-to-Digital Converter 49 47 45 0 5 10 15 FREQUENCY (MHz) 20 Figure 49. AD9203 SINAD at −0.5 dBFS AD8138 Rev. G | Page 22 of 24 25 01073-048 2 5 0.1µF SINAD (dBc) 3 + 63 0.1µF 499Ω 8 25 Figure 49 shows the signal-to-noise-and-distortion (SINAD) under the same conditions as above. For the smaller signal swing, the AD8138 performance is quite good, but its performance degrades when trying to swing too close to the supply rails. 3V 10kΩ 499Ω 20 65 3V 49.9Ω 10 15 FREQUENCY (MHz) Figure 48. AD9203 THD at −0.5 dBFS AD8138 The low-pass filter between the AD8138 and the AD9203 provides filtering that helps to improve the signal-to-noise ratio (SNR). Lower noise can be realized by lowering the pole frequency, but the bandwidth of the circuit is lowered. 0.1µF 5 01073-047 –75 Data Sheet AD8138 OUTLINE DIMENSIONS 3.20 3.00 2.80 5.00 (0.1968) 4.80 (0.1890) 8 5 4.00 (0.1574) 3.80 (0.1497) 1 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) 6.20 (0.2440) 5.80 (0.2284) 8 3.20 3.00 2.80 1.75 (0.0688) 1.35 (0.0532) 1 0.50 (0.0196) × 45° 0.25 (0.0099) 5 5.15 4.90 4.65 4 PIN 1 0.65 BSC 0.51 (0.0201) COPLANARITY SEATING 0.31 (0.0122) 0.10 PLANE 0.95 0.85 0.75 8° 0.25 (0.0098) 0° 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067) 1.10 MAX 0.15 0.00 COMPLIANT TO JEDEC STANDARDS MS-012-AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 0.38 0.22 COPLANARITY 0.10 0.23 0.08 8° 0° 0.80 0.60 0.40 SEATING PLANE COMPLIANT TO JEDEC STANDARDS MO-187-AA Figure 51. 8-Lead Mini Small Outline Package [MSOP] (RM-8) Dimensions shown in millimeters Figure 50. 8-Lead Standard Small Outline Package [SOIC] (R-8) Dimensions shown in millimeters and (inches) ORDERING GUIDE Model 1 AD8138AR AD8138AR-REEL AD8138AR-REEL7 AD8138ARZ AD8138ARZ-RL AD8138ARZ-R7 AD8138ARM AD8138ARM-REEL AD8138ARM-REEL7 AD8138ARMZ AD8138ARMZ-REEL AD8138ARMZ-REEL7 1 Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 8-Lead SOIC 8-Lead SOIC, 13" Tape and Reel 8-Lead SOIC, 7" Tape and Reel 8-Lead SOIC 8-Lead SOIC, 13" Tape and Reel 8-Lead SOIC, 7" Tape and Reel 8-Lead MSOP 8-Lead MSOP, 13" Tape and Reel 8-Lead MSOP, 7" Tape and Reel 8-Lead MSOP 8-Lead MSOP, 13" Tape and Reel 8-Lead MSOP, 7" Tape and Reel Z = RoHS Compliant Part. # denotes RoHS compliant part may be top or bottom marked. Rev. G | Page 23 of 24 Package Option R-8 R-8 R-8 R-8 R-8 R-8 RM-8 RM-8 RM-8 RM-8 RM-8 RM-8 Branding HBA HBA HBA HBA# HBA# HBA# AD8138 Data Sheet NOTES ©1999–2016 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D01073-0-3/16(G) Rev. G | Page 24 of 24