AD8143ACPZ-REEL7

High Speed, Triple Differential Receiver with Comparators AD8143 Data Sheet IN–_B IN+_B FB_B REF_B DIS/PD VS– GND PIN CONFIGURATION High speed 160 MHz large signal bandwidth 1000 V/µs slew rate at G = 1, VO = 2 V p-p High CMRR: 65 dB at 10 MHz High differential input impedance: 5 MΩ Input common-mode range: ±10.5 V (±12 V supplies) User-adjustable gain Wide power supply range: +5 V to ±12 V Fast settling: 8 ns to 1% Disable feature Low offset: ±3.4 mV on 5 V supply 2 on-chip comparators Small packaging: 32-lead, 5 mm × 5 mm LFCSP GND FEATURES 32 31 30 29 28 27 26 25 GND 1 24 GND REF_G 2 23 OUT_B FB_G 3 22 OUT_G IN+_G 4 21 OUT_R 20 VS+ 19 COMPB_IN+ 18 COMPB_IN– 17 GND IN–_G 5 REF_R 6 AD8143 12 13 14 15 16 GND 11 COMPB_OUT 10 COMPA_OUT 9 COMPA_IN– 8 COMPA_IN+ GND IN–_R RGB video receivers Keyboard-video-mouse (KVM) Unshielded twisted pair (UTP) receivers A IN+_R 7 GND FB_R APPLICATIONS 05538-001 B Figure 1. GENERAL DESCRIPTION The AD8143 is a triple, low cost, differential-to-singleended receiver specifically designed for receiving redgreen-blue (RGB) signals over twisted pair cable. It can also be used for receiving any type of analog signal or high speed data transmission. Two auxiliary comparators are provided to receive digital or sync signals. The AD8143 can be used in conjunction with the AD8133 and AD8134 triple, differential drivers to provide a complete low cost solution for RGB over Category-5 UTP cable applications, including KVM. The excellent common-mode rejection (65 dB at 10 MHz) of the AD8143 allows for the use of low cost unshielded twisted pair cables in noisy environments. Rev. A The AD8143 has a wide power supply range from single +5 V supply to ±12 V, which allows for a wide common-mode range. The wide common-mode input range of the AD8143 maintains signal integrity in systems where the ground potential is a few volts different between the drive and receive ends without the use of isolation transformers. The AD8143 is stable at a gain of 1. Closed-loop gain is easily set using external resistors. The AD8143 is available in a 5 mm × 5 mm, 32-lead LFCSP and is rated to work over the extended industrial temperature range of −40°C to +85°C. Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©2005–2016 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD8143 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1  Overview ..................................................................................... 18  Applications ....................................................................................... 1  Basic Closed-Loop Gain Configurations ................................ 18  Pin Configuration ............................................................................. 1  Terminating the Input................................................................ 19  General Description ......................................................................... 1  Input Clamping........................................................................... 19  Revision History ............................................................................... 2  Printed Circuit Board Layout Considerations ....................... 20  Specifications..................................................................................... 3  Driving a Capacitive Load......................................................... 22  Absolute Maximum Ratings............................................................ 9  Power-Down ............................................................................... 22  Thermal Resistance ...................................................................... 9  Comparators ............................................................................... 22  ESD Caution .................................................................................. 9  Sync Pulse Extraction Using Comparators ............................. 22  Pin Configuration and Function Descriptions ........................... 10  Outline Dimensions ....................................................................... 24  Typical Performance Characteristics ........................................... 11  Ordering Guide .......................................................................... 24  Theory of Operation ...................................................................... 17  Applications Information .............................................................. 18  REVISION HISTORY 4/16—Rev. 0 to Rev. A Changes to Figure 3 and Table 6 ................................................... 10 Updated Outline Dimensions ....................................................... 24 Changes to Ordering Guide .......................................................... 24 10/05—Revision 0: Initial Version Rev. A | Page 2 of 24 Data Sheet AD8143 SPECIFICATIONS VS = ±12 V, TA = 25°C, REF = 0 V, RL = 150 Ω, CL = 2 pF, G = 1, TMIN to TMAX = −40°C to +85°C, unless otherwise noted. Table 1. Parameter DYNAMIC PERFORMANCE −3 dB Bandwidth Bandwidth for 0.1dB Flatness Slew Rate Settling Time Output Overdrive Recovery NOISE/DISTORTION Second Harmonic Third Harmonic Crosstalk Input Voltage Noise (RTI) Differential Gain Error Differential Phase Error INPUT CHARACTERISTICS Common-Mode Rejection Common-Mode Voltage Range Differential Operating Range Resistance Capacitance DC PERFORMANCE Open-Loop Gain Closed-Loop Gain Error Input Offset Voltage Test Conditions/Comments Min Max Unit VOUT = 0.2 V p-p VOUT = 2 V p-p VOUT = 0.2 V p-p VOUT = 2 V p-p, RL = 1 kΩ VOUT = 2 V p-p, 1% VOUT = 2 V p-p, 0.1% 260 160 45 1000 8 31 50 MHz MHz MHz V/µs ns ns ns VOUT = 2 V p-p, 1 MHz VOUT = 2 V p-p, 1 MHz VOUT = 1 V p-p, 10 MHz f ≥ 10 kHz NTSC, 200 IRE, RL ≥ 150 Ω NTSC, 200 IRE, RL ≥ 150 Ω −70 −80 −70 14 0.03 0.06 dBc dBc dB nV/√Hz % Degrees Differential Common-mode Differential Common-mode 90 65 28 ±10.5 ±2.5 5 3 2 3 dB dB dB V V MΩ MΩ pF pF VOUT = ±1 V DC 70 0.25 dB % mV µV/°C µA µA nA/°C µA nA/°C DC, VCM = −3.5 V to +3.5 V VCM = 1 V p-p, f = 10 MHz VCM = 1 V p-p, f = 100 MHz V+IN − V−IN = 0 V 86 −4.3 TMIN to TMAX Input Bias Current (+IN, −IN) Input Bias Current (REF, FB) Input Bias Current Drift Input Offset Current Input Offset Current Drift OUTPUT PERFORMANCE Voltage Swing Output Current Short Circuit Current COMPARATOR PERFORMANCE VOH VOL Hysteresis Width Input Bias Current Propagation Delay, tPLH Propagation Delay, tPHL Output Rise Time Output Fall Time Typ +4.3 15 −3.0 −4.6 TMIN to TMAX (+IN, −IN) (+IN, −IN, REF, FB) TMIN to TMAX −2.55 RLOAD = 1 kΩ −10.80 16 +10.82 40 107/147 3.135 Rev. A | Page 3 of 24 +1.45 ±3 Short to GND, source/sink Input driven low RL = 10 kΩ RL = 10 kΩ 25% to 75%, RL = 10 kΩ 25% to 75%, RL = 10 kΩ +3.0 +3.7 3.3 0.2 41 3.5 20 15 15 11 0.255 V mA mA V V mV µA ns ns ns ns AD8143 Parameter POWER-DOWN PERFORMANCE Power-Down VIH Power-Down VIL Power-Down IIH Power-Down IIL Power-Down Assert Time POWER SUPPLY Operating Range Quiescent Current, Positive Supply Quiescent Current, Negative Supply PSRR, Positive Supply PSRR, Negative Supply Data Sheet Test Conditions/Comments Min Typ VS+ − 1.5 VS+ − 2.5 1.0 800 0.5 PD = VCC PD = GND 4.5 44.0 37.0 −75 −82 DC DC Rev. A | Page 4 of 24 Max Unit V V µA µA µs 24 57.5 51.0 −71 −81 V mA mA dB dB Data Sheet AD8143 VS = ±5 V, TA = 25°C, REF = 0 V, RL = 150 Ω, CL = 2 pF, G = 1, TMIN to TMAX = −40°C to +85°C, unless otherwise noted. Table 2. Parameter DYNAMIC PERFORMANCE −3 dB Bandwidth Bandwidth for 0.1dB Flatness Slew Rate Settling Time Output Overdrive Recovery NOISE/DISTORTION Second Harmonic Third Harmonic Crosstalk Input Voltage Noise (RTI) Differential Gain Error Differential Phase Error INPUT CHARACTERISTICS Common-Mode Rejection Common-Mode Voltage Range Differential Operating Range Resistance Capacitance DC PERFORMANCE Open-Loop Gain Closed-Loop Gain Error Input Offset Voltage Test Conditions/Comments Min Max Unit VOUT = 0.2 V p-p VOUT = 2 V p-p VOUT = 0.2 V p-p VOUT = 2 V p-p, RL = 1 kΩ VOUT = 2 V p-p, 1% VOUT = 2 V p-p, 0.1% 230 130 45 1000 10 23 50 MHz MHz MHz V/µs ns ns ns VOUT = 1 V p-p, 1 MHz VOUT = 1 V p-p, 1 MHz VOUT = 1 V p-p, 10 MHz f ≥ 10 kHz NTSC, 200 IRE, RL ≥ 150 Ω NTSC, 200 IRE, RL ≥ 150 Ω −68 −82 −70 14 0.3 0.6 dBc dBc dB nV/√Hz % Degrees Differential Common-mode Differential Common-mode 90 65 28 ±3.8 ±2.5 5 3 2 3 dB dB dB V V MΩ MΩ pF pF VOUT = ±1 V DC 70 0.25 dB % mV µV/°C µA µA nA/°C µA nA/°C DC, VCM = −3.5 V to +3.5 V VCM = 1 V p-p, f = 10 MHz VCM = 1 V p-p, f = 100 MHz V+IN − V−IN = 0 V 84 −3.7 TMIN to TMAX Input Bias Current (+IN, −IN) Input Bias Current (REF, FB) Input Bias Current Drift Input Offset Current Input Offset Current Drift OUTPUT PERFORMANCE Voltage Swing Output Current Short Circuit Current COMPARATOR PERFORMANCE VOH VOL Hysteresis Width Input Bias Current Propagation Delay, tPLH Propagation Delay, tPHL Output Rise Time Output Fall Time Typ +3.7 15 −3.0 −4.3 TMIN to TMAX (+IN, −IN, REF, FB) (+IN, −IN, REF, FB) TMIN to TMAX −2.9 RLOAD = 150 Ω −3.53 16 Input driven low 10% to 90% 10% to 90% Rev. A | Page 5 of 24 1.9 ±3 +3.53 40 107/147 Short to GND, source/sink RL = 10 kΩ RL = 10 kΩ +2.7 +3.0 3.02 3.14 0.19 32 3.5 20 15 15 11 0.25 V mA mA V V mV µA ns ns ns ns AD8143 Parameter POWER-DOWN PERFORMANCE Power-Down VIH Power-Down VIL Power-Down IIH Power-Down IIL Power-Down Assert Time POWER SUPPLY Operating Range Quiescent Current, Positive Supply Quiescent Current, Negative Supply PSRR, Positive Supply PSRR, Negative Supply Data Sheet Test Conditions/Comments Min Typ VS+ − 1.5 VS+ − 2.5 1 230 0.5 PD = VCC PD = GND 4.5 39.0 34.5 −80 −80 DC DC Rev. A | Page 6 of 24 Max Unit V V µA µA µs 24 49.5 43.5 −74 −75 V mA mA dB dB Data Sheet AD8143 VS = 5 V, TA = 25°C, REF = +2.5 V, RL = 150 Ω, CL = 2 pF, G = 1, TMIN to TMAX = −40°C to +85°C, unless otherwise noted. Table 3. Parameter DYNAMIC PERFORMANCE −3 dB Bandwidth Bandwidth for 0.1dB Flatness Slew Rate Settling Time Output Overdrive Recovery NOISE Crosstalk Input Voltage Noise (RTI) INPUT CHARACTERISTICS Common-Mode Rejection Common-Mode Voltage Range Differential Operating Range Resistance Capacitance DC PERFORMANCE Open-Loop Gain Closed-Loop Gain Error Input Offset Voltage Test Conditions/Comments Min Max Unit VOUT = 0.2 V p-p VOUT = 2 V p-p VOUT = 0.2 V p-p VOUT = 2 V p-p, RL = 1 kΩ VOUT = 2 V p-p, 1% VOUT = 2 V p-p, 0.1% 220 125 45 1000 10 23 50 MHz MHz MHz V/µs ns ns ns VOUT = 1 V p-p, 10 MHz f ≥ 10 kHz −70 14 dB nV/√Hz Differential Common-mode Differential Common-mode 90 65 32 1.3 to 3.7 ±2.5 5 3 2 3 dB dB dB V V MΩ MΩ pF pF VOUT = ±1 V DC, measured at G = 11 70 0.25 dB % mV µV/°C µA µA nA/°C µA nA/°C DC, VCM = −3.5 V to +3.5 V VCM = 1 V p-p, f = 10 MHz VCM = 1 V p-p, f = 100 MHz V+IN − V−IN = 0 V 76 −3.4 TMIN to TMAX Input Bias Current (+IN, −IN) Input Bias Current (REF, FB) Input Bias Current Drift Input Offset Current Input Offset Current Drift OUTPUT PERFORMANCE Voltage Swing Output Current Short Circuit Current COMPARATOR PERFORMANCE VOH VOL Hysteresis Width Input Bias Current Propagation Delay, tPLH Propagation Delay, tPHL Output Rise Time Output Fall Time POWER-DOWN PERFORMANCE Power-Down VIH Power-Down VIL Power-Down IIH Power-Down IIL Power-Down Assert Time Typ +3.4 15 −3 −4.5 TMIN to TMAX (+IN, −IN, REF, FB) (+IN, −IN, REF, FB) TMIN to TMAX −2.3 RLOAD = 150 Ω 0.88 16 Input driven low 10% to 90% 10% to 90% PD = VCC PD = GND Rev. A | Page 7 of 24 +1.3 ±3 3.58 40 150 Short to GND RL = 10 kΩ RL = 10 kΩ +2.7 +3 3.02 V mA mA 32 3.5 20 15 15 11 V V mV µA ns ns ns ns VS+ − 1.5 VS+ − 2.5 1 230 0.5 V V µA µA µs 0.25 AD8143 Parameter POWER SUPPLY Operating Range Quiescent Current, Positive Supply PSRR, Positive Supply Data Sheet Test Conditions/Comments Min Typ Max Unit 31.5 −86 24 38.8 −76 V mA dB 4.5 DC Rev. A | Page 8 of 24 Data Sheet AD8143 ABSOLUTE MAXIMUM RATINGS Rating 24 V See Figure 2 –65°C to +125°C –40°C to +85°C 300°C 150°C Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability. THERMAL RESISTANCE θJA is specified for the worst-case conditions, that is, θJA is specified for a device soldered in the circuit board with its exposed paddle soldered to a pad on the PCB surface, which is thermally connected to a copper plane. Table 5. Thermal Resistance Package Type 5 mm × 5 mm, 32-Lead LFCSP θJA 45 θJC 7 Unit °C/W Maximum Power Dissipation The maximum safe power dissipation in the AD8143 package is limited by the associated rise in junction temperature (TJ) on the die. At approximately 150°C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8143. Exceeding a junction temperature of 150°C for an extended period can result in changes in the silicon devices potentially causing failure. Airflow increases heat dissipation, effectively reducing θJA. In addition, more metal directly in contact with the package leads from metal traces, through-holes, ground, and power planes reduces the θJA. The exposed paddle on the underside of the package must be soldered to a pad on the PCB surface which is thermally connected to a copper plane to achieve the specified θJA. Figure 2 shows the maximum safe power dissipation in the package vs. the ambient temperature for the 32-lead LFCSP (45°C/W) on a JEDEC standard 4-layer board with the underside paddle soldered to a pad which is thermally connected to a PCB plane. Extra thermal relief is required for operation at high supply voltages. See the Applications Information section for details. θJA values are approximations. 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 –40 05538-056 Parameter Supply Voltage Power Dissipation Storage Temperature Range Operating Temperature Range Lead Temperature Range (Soldering 10 sec) Junction Temperature power dissipated due to all of the loads is equal to the sum of the power dissipation due to each individual load. RMS voltages and currents must be used in these calculations. MAXIMUM POWER DISSIPATION (W) Table 4. –20 0 20 40 60 80 AMBIENT TEMPERATURE (°C) Figure 2. Maximum Power Dissipation vs. Temperature for a 4-Layer Board ESD CAUTION The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). The power dissipated due to the load drive depends upon the particular application. For each output, the power due to load drive is calculated by multiplying the load current by the associated voltage drop across the device. The Rev. A | Page 9 of 24 AD8143 Data Sheet 32 31 30 29 28 27 26 25 GND IN–_B IN+_B FB_B REF_B DIS/PD VS– GND PIN CONFIGURATION AND FUNCTION DESCRIPTIONS 1 2 3 4 5 6 7 8 AD8143 TOP VIEW (Not to Scale) 24 23 22 21 20 19 18 17 GND OUT_B OUT_G OUT_R VS+ COMPB_IN+ COMPB_IN– GND NOTES 1. THE EXPOSED PAD ON THE UNDERSIDE OF THE DEVICE MUST BE CONNECTED TO GROUND. 05538-050 GND IN+_R IN–_R COMPA_IN+ COMPA_IN– COMPA_OUT COMPB_OUT GND 9 10 11 12 13 14 15 16 GND REF_G FB_G IN+_G IN–_G REF_R FB_R GND Figure 3. Pin Configuration Table 6. Pin Function Descriptions Pin No. 1, 8, 9,16, 17, 24, 25, 32 2 3 4 5 6 7 10 11 12 13 14 15 18 19 20 21 22 23 26 27 28 29 30 31 0 Mnemonic GND REF_G FB_G IN+_G IN−_G REF_R FB_R IN+_R IN−_R COMPA_IN+ COMPA_IN− COMPA_OUT COMPB_OUT COMPB_IN− COMPB_IN+ VS+ OUT_R OUT_G OUT_B VS− DIS/PD REF_B FB_B IN+_B IN−_B EPAD Description Signal Ground and Thermal Plane Connection (See the Applications Information Section) Reference Input, Green Channel Feedback Input, Green Channel Noninverting Input, Green Channel Inverting Input, Green Channel Reference Input, Red Channel Feedback Input, Red Channel Noninverting Input, Red Channel Inverting Input, Red Channel Positive Input, Comparator A Negative Input, Comparator A Output, Comparator A Output, Comparator B Negative Input, Comparator B Positive Input, Comparator B Positive Power Supply Output, Red Channel Output, Green Channel Output, Blue Channel Negative Power Supply Disable/Power Down Reference Input, Blue Channel Feedback Input, Blue Channel Noninverting Input, Blue Channel Inverting Input, Blue Channel Exposed Pad. The exposed pad on the underside of the device must be connected to ground (see the Applications Information section). Rev. A | Page 10 of 24 Data Sheet AD8143 TYPICAL PERFORMANCE CHARACTERISTICS Unless otherwise noted, G = 1, RL = 150 Ω, CL = 2 pF, VS = ±5 V, TA = 25°C. Refer to the circuit in Figure 38. 3 3 2 2 VS = ±5 1 1 0 0 –1 GAIN (dB) GAIN (dB) VS = ±12 VS = ±12 –2 –3 –1 –2 VS = ±5 –3 –4 –4 VS = +5 –5 –6 –7 1 VOUT = 2V p-p 05538-002 VOUT = 0.2V p-p 10 VS = +5 05538-005 –5 –6 –7 1 100 10 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 4. Small Signal Frequency Response at Various Power Supplies, G = 1 Figure 7. Large Signal Frequency Response at Various Power Supplies, G = 1 9 9 8 8 VS = +5 VS = +5 7 7 6 6 5 GAIN (dB) VS = ±12 4 4 VS = ±5 5 VS = ±12 4 3 2 2 1 1 –1 1 VOUT = 2V p-p 05538-003 VOUT = 0.2V p-p 0 10 05538-006 GAIN (dB) VS = ±5 0 –1 1 100 10 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 5. Small Signal Frequency Response at Various Power Supplies, G = 2 Figure 8. Large Signal Frequency Response at Various Power Supplies, G = 2 3 3 2 2 RL = 1kΩ 1 1 0 0 GAIN (dB) –1 RL = 150Ω –2 –3 –4 –1 RL = 150Ω –2 –3 –4 –5 –5 –6 –7 1 VOUT = 2V p-p 10 05538-007 VOUT = 0.2V p-p 05538-004 GAIN (dB) RL = 1kΩ –6 –7 100 1 10 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 6. Small Signal Frequency Response at Various Loads Figure 9. Large Signal Frequency Response at Various Loads Rev. A | Page 11 of 24 AD8143 Data Sheet 5 5 4 4 G = 1, CL = 10pF, RSNUB = 40Ω 3 3 G = 1, CL = 2pF 2 1 0 –1 G = 2, CL = 2pF G = 2, CL = 10pF, RSNUB = 40Ω 1 GAIN (dB) 0 G = 2, CL = 2pF –1 G = 1, CL = 2pF –2 –2 –3 –3 –4 05538-013 RL = 1kΩ VOUT = 0.2V p-p –5 1 10 100 RL = 1kΩ VOUT = 2V p-p –4 05538-014 GAIN (dB) G = 1, CL = 10pF, RSNUB = 40Ω G = 2, CL = 10pF, RSNUB = 40Ω 2 –5 10 1 1000 100 1000 FREQUENCY (MHz) FREQUENCY (MHz) Figure 10. Small Signal Frequency Response at Various Gains and 10 pF Capacitive Load Buffered by 40 Ω Resistor Figure 13. Large Signal Frequency Response at Various Gains and 10 pF Capacitive Load Buffered by 40 Ω Resistor 3 3 2 2 G=1 1 1 0 0 –1 –1 GAIN (dB) –2 G=2 –3 –2 –3 –4 –4 –5 –5 –6 –7 1 VOUT = 2V p-p 05538-009 VOUT = 0.2V p-p 10 G=2 05538-012 GAIN (dB) G=1 –6 –7 1 100 10 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 14. Large Signal Frequency Response at Various Gains Figure 11. Small Signal Frequency Response at Various Gains 0.5 0.4 80 0 70 –20 MAGNITUDE OPEN LOOP-GAIN (dB) GAIN (dB) –40 60 RL = 1kΩ, VOUT = 0.2V p-p 0.2 0.1 0 –0.1 RL = 150Ω, VOUT = 0.2V p-p –0.2 50 –60 40 –80 –100 30 PHASE RL = 150Ω, VOUT = 2V p-p 20 –120 10 –140 0 –160 OPEN LOOP-PHASE (Degrees) RL = 1kΩ, VOUT = 2V p-p 0.3 –0.4 –0.5 1 10 100 –10 0.001 0.01 0.1 1 10 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 15. Open-Loop Gain and Phase Responses Figure 12. 0.1 dB Flatness for Various Loads and Output Amplitudes Rev. A | Page 12 of 24 –180 1000 05538-016 05538-010 –0.3 Data Sheet AD8143 100 100 VS = ±12V 80 70 60 +5V 50 40 ±5V 30 20 10 0 0.1 1 10 100 05538-021 INPUT VOLTAGE NOISE (nV/√Hz) ±12V 05538-020 COMMON-MODE REJECTION (dB) 90 10 0.00001 1000 0.0001 0.001 FREQUENCY (MHz) 0.01 0.1 1 10 FREQUENCY (MHz) Figure 16. Common-Mode Rejection Ratio vs. Frequency at Various Supplies Figure 19. Input Referred Voltage Noise vs. Frequency 200 1.5 VOUT = 2V p-p VOUT = 0.2V p-p 150 1.0 0.5 50 VOLTAGE (V) VOLTAGE (mV) 100 G = 1, RL = 150Ω G = 1, RL = 1kΩ G = 2, RL = 150Ω G = 2, RL = 1kΩ 0 –50 G = 1, RL = 150Ω G = 1, RL = 1kΩ G = 2, RL = 150Ω G = 2, RL = 1kΩ 0 –0.5 –100 05538-015 –200 0 10 20 30 40 50 60 70 80 90 05538-018 –1.0 –150 –1.5 0 100 10 20 30 40 Figure 17. Small Signal Transient Response at Various Gains and Loads 60 70 80 90 100 Figure 20. Large Signal Transient Response at Various Gains and Loads 200 1.5 G = 2, CL = 10pF, RSNUB = 40Ω G = 2, CL = 2pF G = 2, CL = 2pF 150 G = 2, CL = 10pF, RSNUB = 40Ω 1.0 100 OUTPUT VOLTAGE (dB) 50 0 –50 –100 G = 1, CL = 2pF G = 1, CL = 10pF, RSNUB = 40Ω –200 0 10 20 30 40 50 60 0 RL = 1kΩ VOUT = 2V p-p –0.5 G = 1, C L = 10pF, RSNUB = 40Ω –1.0 05538-017 RL = 1kΩ VOUT = 0.2V p-p –150 G = 1, CL = 2pF 0.5 70 80 90 05538-019 OUTPUT VOLTAGE (mV) 50 TIME (ns) TIME (ns) –1.5 0 100 TIME (ns) 10 20 30 40 50 60 70 80 90 100 TIME (ns) Figure 18. Small Signal Transient Response at Various Gains and 10 pF Capacitive Load Buffered by 40 Ω Resistor Figure 21. Large Signal Transient Response at Various Gains and 10 pF Capacitive Load Buffered by 40 Ω Resistor Rev. A | Page 13 of 24 AD8143 Data Sheet 1.25 0.5 1.00 0.4 0.75 INPUT 0.3 0.50 0.2 1400 0 0 –0.25 –0.1 –0.50 –0.2 +SR, RL = 150 –SR, RL = 150 +SR, RL = 1k –SR, RL= 1k 600 400 –0.3 OUTPUT –1.00 –1.25 0 800 –0.4 10 20 30 40 50 60 70 80 90 –0.5 100 200 05538-023 –0.75 SLEW RATE (V/s) 0.1 ERROR ERROR (%) 0.25 1000 05538-027 VOLTAGE (V) 1200 0 0 Figure 22. Settling Time (0.1%) at Various Loads 1.0 1.5 2.0 2.5 –50 –30 –55 –40 VS = ±12V –65 4.0 4.5 VS = ±5V –70 –60 –70 –80 Vs = ±12V –75 –100 0.1 100 10 05538-055 05538-047 1 Vs = ±5V –90 1 FREQUENCY (MHz) 10 100 FREQUENCY (MHz) Figure 23. Second Harmonic Distortion vs. Frequency and Power Supplies, VO = 2 V p-p, G = 2 Figure 26. Third Harmonic Distortion vs. Frequency and Power Supplies, VO = 2 V p-p, G = 2 –30 –50 –40 DISTORTION (dBc) –55 DISTORTION (dBc) 3.5 –50 –60 –80 0.1 3.0 Figure 25. Slew Rate vs. Input Voltage Swing at Various Loads DISTORTION (dBc) DISTORTION (dBc) 0.5 OUTPUT VOLTAGE (V p-p) TIME (ns) –60 VS = ±12V –65 –50 –60 VS = ±12V –70 –70 05538-048 –75 0.1 –80 1 10 –90 0.1 100 VS = ±5V 1 10 05538-049 VS = ±5V 100 FREQUENCY (MHz) FREQUENCY (MHz) Figure 24. Second Harmonic Distortion vs. Frequency and Power Supplies, VO = 2 V p-p Figure 27. Third Harmonic Distortion vs. Frequency and Power Supplies, VO = 2 V p-p Rev. A | Page 14 of 24 AD8143 54 4 52 3 OUTPUT VOLTAGE (V) IS+ 50 48 46 44 IS– 42 2 1 0 –1 –2 VS = ±12V RL =  38 –5 –4 –3 –2 –1 0 1 2 3 –3 –4 –5 5 4 05538-026 40 05538-022 –4 –3 DIFFERENTIAL INPUT VOLTAGE (V) Figure 28. Power Supply Current vs. Differential Input Voltage at ±12 V Supplies 60 0 1 2 3 4 5 Figure 31. Differential Input Operating Range IS+, VS = ±12V 50 SUPPLY CURRENT (mA) 45 45 IS–, VS = ±12V 40 35 IS–, VS = ±5V 30 IS+, VS = ±5V 25 20 IS+ 40 IS– 35 05538-031 30 15 10 –50 –40 –30 –20 –10 0 RL =  25 10 20 30 40 50 60 70 80 90 100 5 6 7 TEMPERATURE (C) 9 10 11 12 Figure 32. Power Supply Current vs. Power Supply Voltage 0 0 VS = ±5V –10 –10 VS = +5V –20 VS = ±5V –20 –30 VS = ±12V –30 VS = ±12V –50 –60 –40 –50 –60 –70 –80 –80 05538-046 –70 –90 0.1 1 10 100 05538-045 PSRR (dB) –40 –100 0.01 8 SUPPLY VOLTAGE (VS) Figure 29. Power Supply Current vs. Temperature PSRR (dB) –1 50 RL =  55 SUPPLY CURRENT (mA) –2 DIFFERENTIAL INPUT VOLTAGE (V) 05538-024 SUPPLY CURRENT (mA) Data Sheet –90 –100 0.01 1000 FREQUENCY (MHz) 0.1 1 10 100 1000 FREQUENCY (MHz) Figure 30. Positive Power Supply Rejection Ratio vs. Frequency Figure 33. Negative Power Supply Rejection Ratio vs. Frequency Rev. A | Page 15 of 24 AD8143 15 Data Sheet 10 3.5 VS = ±12V +VSAT_±12V 3.0 VS = ±5V 5 2.5 +VSAT_±5V VOUT (V) OUTPUT VOLTAGE (V) 4.0 G = +2 (RF = RG = 499Ω) AND VS = ±5V G = +5 (RF = 8.06kΩ RG = 2kΩ) AND VS = ±12V 0 2.0 1.5 –5 –VSAT_±5V 1.0 –10 0 100 200 300 400 500 600 700 800 900 0 –25 1000 OUTPUT LOAD (Ω) 2 × VIN G=2 4 3 1 0 –1 OUTPUT –3 –4 05538-029 VOLTAGE (V) 2 –5 –6 0 100 200 300 400 500 600 –15 –10 –5 0 5 10 Figure 36. Comparator Hysteresis 6 –2 –20 VIN (mV) Figure 34. Output Saturation Voltage vs. Output Load 5 05538-032 –15 0.5 05538-025 –VSAT_±12V 700 800 900 1000 TIME (ns) Figure 35. Output Overdrive Recovery Rev. A | Page 16 of 24 15 20 25 Data Sheet AD8143 THEORY OF OPERATION The AD8143 amplifiers use an architecture called active feedback, which differs from that of conventional op amps. The most obvious differentiating feature is the presence of two separate pairs of differential inputs compared to a conventional op amp’s single pair. Typically, for the active-feedback architecture, one of these input pairs is driven by a differential input signal, while the other is used for the feedback. This active stage in the feedback path is where the term active feedback is derived. The active feedback architecture offers several advantages over a conventional op amp in several types of applications. Among these are excellent common-mode rejection, wide input commonmode range, and a pair of inputs that are high impedance and completely balanced in a typical application. In addition, while an external feedback network establishes the gain response as in a conventional op amp, its separate path makes it entirely independent of the signal input. This eliminates any interaction between the feedback and input circuits, which traditionally causes problems with CMRR in conventional differential-input op amp circuits. Another advantage of active feedback is the ability to change the polarity of the gain merely by switching the differential inputs. A high input impedance inverting amplifier can therefore be made. Besides high input impedance, a unity-gain inverter with the AD8143 has noise gain of unity, producing lower output noise and higher bandwidth than op amps that have noise gain equal to 2 for a unity-gain inverter. The two differential input stages of the AD8143 are each transconductance stages that are well-matched. These stages convert the respective differential input voltages to internal currents. The currents are then summed and converted to a voltage, which is buffered to drive the output. The compensation capacitor is included in the summing circuit. When the feedback path is closed around the part, the output drives the feedback input to that voltage which causes the internal currents to sum to zero. This occurs when the two differential inputs are equal and opposite; that is, their algebraic sum is zero. In a closed-loop application, a conventional op amp has its differential input voltage driven to near zero under non-transient conditions. The AD8143 generally has differential input voltages at each of its input pairs, even under equilibrium conditions. As a practical consideration, it is necessary to internally limit the differential input voltage with a clamp circuit. Thus, the input dynamic ranges are limited to about 2.5 V for the AD8143 (see the Specifications section for more detail). For this and other reasons, it is not recommended to reverse the input and feedback stages of the AD8143, even though some apparently normal functionality may be observed under some conditions. Rev. A | Page 17 of 24 AD8143 Data Sheet APPLICATIONS INFORMATION The AD8143 contains three independent active-feedback amplifiers that can be effectively applied as differential line receivers for red-green-blue (RGB) signals or component video, such as YPbPr, signals transmitted over unshielded-twisted-pair (UTP) cable. The AD8143 also contains two general-purpose comparators with hysteresis that can be used to receive digital signals or to extract video synchronization pulses from received common-mode signals that contain encoded synchronization signals. In this configuration, the voltage applied to the REF pin appears at the output with a gain of 1 + RF/RG. To achieve unity gain from VREF to VOUT in this configuration, divide VREF by the same factor used in the feedback loop; the same RF and RG values can be used. Figure 38 illustrates this approach. +5V 0.01µF + VIN An internal linear voltage regulator derives power for the comparators from the positive supply; therefore, the AD8143 must always have a minimum positive supply voltage of 4.5 V. – + RF REF – VREF FB RG The AD8143 includes a power-down feature that can be asserted to reduce the supply current when a particular device is not in use. RF RG 0.01µF –5V BASIC CLOSED-LOOP GAIN CONFIGURATIONS As described in the Theory of Operation section, placing a resistive feedback network between an amplifier output and its respective feedback amplifier input creates a stable negative feedback amplifier. It is important to note that the closed-loop gain of the amplifier used in the signal path is defined as the amplifier’s single-ended output voltage divided by its differential input voltage. Therefore, each amplifier in the AD8143 provides differential-to-single-ended gain. Additionally, the amplifier used for feedback has two high impedance inputs—the FB input, where the negative feedback is applied, and the REF input, which can be used as an independent single-ended input to apply a dc offset to the output signal. Some basic gain configurations implemented with an AD8143 amplifier are shown in Figure 37 through Figure 39. Figure 38. Basic Gain Circuit: VOUT = VIN (1 + RF/RG) + VREF The gain equation for the circuit in Figure 38 is VOUT = VIN (1 + RF/RG) + VREF (2) Another configuration that provides the same gain equation as Equation 2 is shown in Figure 39. In this configuration, it is important to keep the source resistance of VREF much smaller than RG to avoid gain errors. +5V 0.01µF + VIN – + REF +5V – VOUT FB 0.01µF + RG RF – VREF + REF – VOUT FB VREF RG Figure 39. Basic Gain Circuit: VOUT = VIN (1 + RF/RG) + VREF RF 05538-038 0.01µF –5V Figure 37. Basic Gain Circuit: VOUT = (VIN + VREF)(1 + RF/RG) The gain equation for the circuit in Figure 37 is VOUT = (VIN + VREF)(1 + RF/RG) 0.01µF –5V 05538-040 VIN VOUT 05538-039 OVERVIEW (1) For stability reasons, the inductance of the trace connected to the REF pin must be kept to less than 10 nH. The typical inductance of 50 Ω traces on the outer layers of the FR-4 boards is 7 nH/in, and on the inner layers, it is typically 9 nH/in. Vias must be accounted for as well. The inductance of a typical via in a 0.062-inch board is on the order of 1.5 nH. If longer traces are required, a 200 Ω resistor should be placed in series with the trace to reduce the Q-factor of the inductance. Rev. A | Page 18 of 24 Data Sheet AD8143 In many dual-supply applications, VREF can be directly connected to ground right at the device. INPUT CLAMPING TERMINATING THE INPUT One of the key benefits of the active-feedback architecture is the separation that exists between the differential input signal and the feedback network. Because of this separation, the differential input maintains its high CMRR and provides high differential and common-mode input impedances, making line termination a simple task. Most applications that use the AD8143 involve transmitting broadband video signals over 100 Ω UTP cable and use dc-coupled terminations. The two most common types of dc-coupled terminations are differential and common-mode. Differential termination of 100 Ω UTP is implemented by simply connecting a 100 Ω resistor across the amplifier input, as shown in Figure 40. +5V 0.01µF 100Ω UTP 100Ω + VIN – + REF – VOUT FB RG RF 05538-041 0.01µF –5V Figure 40. Differential-Mode Termination Some applications require common-mode terminations for common-mode currents generated at the transmitter. In these cases, the 100 Ω termination resistor is split into two 50 Ω resistors. The required common-mode termination voltage is applied at the tap between the two resistors. In many of these applications, the common-mode tap is connected to ground (VTERM (CM) = 0). This scheme is illustrated in Figure 41. The differential input that is assigned to receive the input signal includes clamping diodes that limit the differential input swing to approximately 5.5 V p-p at 25°C. Because of this, the input and feedback stages should never be interchanged. Figure 31 illustrates the clamping action at the signal input stage. The supply current drawn by the AD8143 has a strong dependence on input signal magnitude because the input transconductance stages operate with differential input signals that can be up to a few volts peak-to-peak. This behavior is distinctly different from that of traditional op-amps, where the differential input signal is driven to essentially 0 V by negative feedback. Figure 28 illustrates the supply current dependence on input voltage. For most applications, including receiving RGB video signals, the input signal magnitudes encountered are well within the safe operating limits of the AD8143 over its full power supply and operating temperature ranges. In some extreme applications where large differential and/or common-mode voltages can be encountered, external clamping may be necessary. Another application where external common-mode clamping is sometimes required is when an unpowered AD8143 receives a signal from an active driver. In this case, external diodes are required when the current drawn by the internal ESD diodes cannot be kept to less than 5 mA. When using ±12 V supplies, the differential input signal must be kept to less than 4 V p-p. In applications that use ±12 V supplies where the input signals are expected to reach or exceed 4 V p-p, external differential clamping at a maximum of 4 V p-p is required. Figure 42 shows a general approach to external differentialmode clamping. POSITIVE CLAMP + VIN +5V 50Ω 100Ω UTP 50Ω 0.01µF – + VIN – NEGATIVE CLAMP RS RT + VOUT – RS – 05538-051 + REF VOUT FB Figure 42. Differential-Mode Clamping VTERM(CM) RF 0.01µF –5V Figure 41. Common-Mode Termination 05538-042 RG The positive and negative clamps are nonlinear devices that exhibit very low impedance when the voltage across them reaches a critical threshold (clamping voltage), thereby limiting the voltage across the AD8143 input. The positive clamp has a positive threshold, and the negative clamp has a negative threshold. Rev. A | Page 19 of 24 AD8143 Data Sheet A simple way to implement a clamp is to use a number of diodes in series. The resultant clamping voltage is then the sum of the clamping voltages of individual diodes. A 1N4448 diode has a forward voltage of approximately 0.70 V to 0.75 V at typical current levels that are seen when it is being used as a clamp, and 2 pF maximum capacitance at 0 V bias. (The capacitance of a diode decreases as its reverse bias voltage is increased.) The series connection of two 1N4448 diodes, therefore, has a clamping voltage of 1.4 V to 1.5 V. Figure 43 shows how to limit the differential input voltage applied to an AD8143 amplifier to ±1.4 V to ±1.5 V (2.8 V p-p to 3.0 V p-p). Note that the resulting capacitance of the two series diodes is half that of one diode. Different numbers of series diodes can be used to obtain different clamping voltages. RT is the differential termination resistor and the series resistances, RS, limit the current into the diodes. The series resistors should be highly matched in value to preserve high frequency CMRR. POSITIVE CLAMP NEGATIVE CLAMP – 3 HBAT-540C 1 V– RT V+ 2 RS HBAT-540C + 3 – VOUT 1 V– Figure 44. External Common-Mode Clamping The series resistances, RS, limit the current in each leg, and the Schottky diodes limit the voltages on each input to approximately 0.3 V to 0.4 V over the positive power supply, V+ and to 0.3 V to 0.4 V below the negative power supply, V−. The maximum value of RS is determined by the required signal bandwidth, the line impedance, and the effective differential capacitance due to the AD8143 inputs and the diodes. The two most important issues with regard to printed circuit board (PCB) layout are minimizing parasitic signal trace reactances in the feedback network and providing sufficient thermal relief. RT + VOUT – Excessive parasitic reactances in the feedback network cause excessive peaking in the amplifier’s frequency response and excessive overshoot in its step response due to a reduction in phase margin. Oscillation occurs when these parasitic reactances are increased to a critical point where the phase margin is reduced to zero. Minimizing these reactances is important to obtain optimal performance from the AD8143. 05538-052 – VIN RS PRINTED CIRCUIT BOARD LAYOUT CONSIDERATIONS + RS + As with the differential clamp, the series resistors should be highly matched in value to preserve high frequency CMRR. RS VIN V+ 2 05538-044 A diode is a simple example of such a clamp. Schottky diodes generally have lower clamping voltages than typical signal diodes. The clamping voltage should be larger than the largest expected signal amplitude, with enough margin to ensure that the received signal passes without being distorted. Figure 43. Using Two 1N4448 Diodes in Series as a Clamp There are many other nonlinear devices that can be used as clamps. The best choice for a particular application depends upon the desired clamping voltage, response time, parasitic capacitance, and other factors. When using external differential-mode clamping, it is important to ensure that the series resistors (RS), the sum of the parasitic capacitance of the clamping devices, and the input capacitance of the AD8143 are small enough to preserve the desired signal bandwidth. Figure 44 shows a specific example of external common-mode clamping. When operating at ±12 V power, it is important to pay special attention to removing heat from the AD8143. Besides the special layout considerations previously mentioned and expounded upon in the following sections, general high speed layout practices must be adhered to when applying the AD8143. Controlled impedance transmission lines are required for incoming and outgoing signals, referenced to a ground plane. Rev. A | Page 20 of 24 Data Sheet AD8143 Typically, the input signals are received over 100 Ω differential transmission lines. A 100 Ω differential transmission line is readily realized on the printed circuit board using two wellmatched, closely-spaced 50 Ω single-ended traces that are coupled through the ground plane. The traces that carry the single-ended output signals are most often 75 Ω for video signals. Output signal connections should include series termination resistors that are matched to the impedance of the line they are driving. Broadband power supply decoupling networks should be placed as close as possible to the supply pins. Small surface-mount ceramic capacitors are recommended for these networks, and tantalum capacitors are recommended for bulk supply decoupling. Minimizing Parasitic Reactances in the Feedback Network Parasitic trace capacitance and inductance are both reduced when the traces that connect the feedback network together are reduced in length. Removing the copper from all planes below the traces reduces trace capacitance, but increases trace inductance because the loop area formed by the trace and ground plane is increased. A reasonable compromise that works well is to void all copper directly under the feedback loop traces and component pads with margins on each side approximately equal to one trace width. Combining this technique with minimizing trace lengths is effective in keeping parasitic trace reactances in the feedback loop to a minimum. Additionally, all components used in the feedback network should be in 0402 surface-mount packages. Figure 45 illustrates the magnified view of a proven feedback network layout that provides excellent performance. Note that the internal layers are not shown. A conservative estimate for feedback-loop trace capacitance in each loop of the layout shown in Figure 45 is 2 pF. This value is viewed as the minimum load capacitance and is reflected in the frequency response and transient response plots. Maximizing Heat Removal The AD8143 pinout includes ground connections on its corner pins to facilitate heat removal. These pins should be connected to the exposed paddle on the underside of the AD8143 and to a ground plane on the component side of the board. Additionally, a 5 × 5 array of thermal vias connecting the exposed pad to internal ground planes should be placed inside the PCB pad that is soldered to the exposed pad. Using these techniques is highly recommended in all applications, and is required in ±12 V applications where power dissipation is the greatest. Figure 45 illustrates how to optimize the circuit board layout for heat removal. Designs must often conform to design-for-manufacturing (DFM) rules that stipulate how to lay out PCBs in such a way as to facilitate the manufacturing process. Some of these rules require thermal relief on pads that connect to planes, and the rules may preclude the use of the technique illustrated in Figure 45. In these cases, the ground pins should be connected to the exposed paddle and component-side ground plane using techniques that conform to the DFM requirements. GND CFG GND RGB CFB RGG RFB GND RFG It is strongly recommended that the layout shown in Figure 45, or something very similar, be used for the three AD8143 feedback networks. GND = COMPONENT SIDE = CIRCUIT SIDE GND RFR GND CFR 05538-043 RGR GND Figure 45. Recommended Layout for Feedback Loops and Grounding Rev. A | Page 21 of 24 AD8143 Data Sheet DRIVING A CAPACITIVE LOAD The AD8143 typically drives either high impedance loads, such as crosspoint switch inputs, or doubly terminated coaxial cables. A gain of 1 is commonly used in the high impedance case because the 6 dB transmission line termination loss is not incurred. A gain of 2 is required when driving cables to compensate for the 6 dB termination loss. In all cases, the output must drive the parasitic capacitance of the feedback loop, conservatively estimated to be 2 pF, in addition to the capacitance presented by the actual load. When driving a high impedance input, it is recommended that a small series resistor be used to buffer the input capacitance of the device being driven. Clearly, the resistor value must be small enough to preserve the required bandwidth. In the ideal doubly terminated cable case, the AD8143 output sees a purely resistive load. In reality, there is some residual capacitance, and this is buffered by the series termination resistor. Figure 46 illustrates the high impedance case, and Figure 47 illustrates the cabledriving case. +5V 0.01µF + RS REF CIN FB RF RG –5V +5V 0.01µF + VIN RS REF CS RL FB 0.01µF –5V In addition to general-purpose applications, the two on-chip comparators can be used to receive differential digital information or to decode video sync pulses from received common-mode voltages. Built-in hysteresis helps to eliminate false triggers from noise. The AD8143 is particularly useful in keyboard video mouse (KVM) applications. KVM networks transmit and receive computer video signals, which are typically comprised of red, green, and blue (RGB) video signals and separate horizontal and vertical sync signals. Because the sync signals are separate and not embedded in the color signals, it is advantageous to transmit them using a simple scheme that encodes them among the three common-mode voltages of the RGB signals. The AD8134 triple differential driver is a natural complement to the AD8143 and performs the sync pulse encoding with the necessary circuitry on-chip. 05538-054 RF RG COMPARATORS SYNC PULSE EXTRACTION USING COMPARATORS Figure 46. Buffering the Input Capacitance of a High-Z Load – The power-down feature is intended to be used to reduce power consumption when a particular device is not in use, and does not place the output in a High-Z state when asserted. The power-down feature is asserted when the voltage applied to the power-down pin drops to approximately 2 V below the positive supply. The AD8143 is enabled by pulling the power-down pin to the positive supply. An internal linear voltage regulator derives power for the comparators from the positive supply; therefore, the AD8143 must always have a minimum positive supply voltage of 4.5 V. 05538-053 0.01µF POWER-DOWN The comparator outputs are not designed to drive transmission lines. When the signals detected by the comparators are driven over cables or controlled impedance printed circuit board traces, the comparator outputs must be fed to a spare logic gate, FPGA, or other device that is capable of driving signals over transmission lines. VIN – Small and large signal frequency responses for the High-Z case with a 40 Ω series resistor and 10 pF load capacitance are shown in Figure 10 and Figure 13; transient responses for the same conditions are shown in Figure 18 and Figure 21. In the cable driving case shown in Figure 47, CS