AD8339

Quad I/Q Demodulator And Phase Shifter Preliminary Technical Data FEATURES Quad Integrated I/Q Demodulator 16 Phase Select on each Output (22.5° per step) Quadrature Demodulation Accuracy Phase Accuracy ±1° Amplitude Balance ±0.25 dB Bandwidth 4LO: LF – 100 MHz; RF: LF - 25 MHz Baseband: determined by external filtering Output Dynamic Range 158 dB (1 Hz Bandwidth) LO Drive > –10 dBm (50 Ω); 200 mVpp Supply: ±5 V Power Consumption 73 mW/channel (290 mW total) Power Down via SPI (Each Channel and Complete Chip) AD8339 FUNCTIONAL BLOCK DIAGRAM RF1 2 0 φ φ φ φ 4 BIAS I1 Q1 Q2 I2 SCLK SDI SDO CSB I3 Q3 Q4 I4 90 90 RSET RF2 2 0 4xLO RF3 2 /4 Serial Interface 4 0 2 φ φ φ φ APPLICATIONS Medical Imaging (CW Ultrasound Beamforming) Phased Array Systems Radar Adaptive Antennas Communication Receivers VPOS COMM VNEG 90 90 0 2 RF4 Figure 1. Functional Block Diagram GENERAL DESCRIPTION The AD8339 is a Quad I/Q demodulator intended to be driven by a low noise preamplifier with differential outputs; it is optimized for the LNA in the AD8332/4/5 family of VGAs. The part consists of four identical I/Q demodulators with a 4x local oscillator (LO) input that divides this signal and generates the necessary 0° and 90° phases of the internal LO that drive the mixers. The four I/Q demodulators can be used independently of each other (assuming that a common LO is acceptable) since each has a separate RF input. The major application is continuous wave (CW) analog beamforming in ultrasound. Since in a beamforming application the outputs of many channels are summed coherently, the signals need to be phase aligned. A reset pin for the LO divider that synchronizes multiple ICs to start in the same quadrant is provided. Sixteen discrete phase rotations in 22.5° increments can be selected independently for each channel. For example, if CH1 is used as a reference and CH2 has an I/Q phase lead of 45°, then by choosing the correct code one can phase align CH2 with CH1. The mixer outputs are provided in current form so that they can be easily summed. The summed current outputs, one each for the I and Q signals, will need to be converted to a voltage by a high dynamic range current-to-voltage (I-V) converter. A good choice for this transimpedance amplifier is the AD8021 because of its low noise. Following the current summation the combined signal is presented to a high resolution AD converter (ADC) like the AD7665 (16b/570 ksps). An SPI compatible serial interface is provided for ease of programming the phase of each channel; the interface allows daisychaining by shifting the data through each chip from SDI to SDO. The SPI also allows for power down of each individual channel and the complete chip. During power down the serial interface remains active so that the device can be programmed again. The dynamic range is >158 dB (1 Hz BW) at the I and Q outputs. Note that the following transimpedance amplifier is an important element in maintaining this dynamic range and attention needs to be paid to component selection. The AD8339 will be available in a 6x6 mm 40 pin LFCSP for the industrial temperature range of -40°C to +85°C. Rev. PrA - 12/19/06 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective companies. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. www.analog.com Tel: 781.329.4700 Fax: 781.326.8703 © 2007 Analog Devices, Inc. All rights reserved. AD8339 Preliminary Technical Data TABLE OF CONTENTS AD8339 Specifications.................................................................... 3 Absolute Maximum Ratings............................................................ 5 ESD Caution.................................................................................. 5 Pin Configuration and Function Descriptions............................. 6 Equivalent Input Circuits ................................................................ 7 Typical Performance Characteristics ............................................. 8 Test Circuits....................................................................................... 9 Theory of Operation ...................................................................... 10 Quadrature Generation ............................................................. 10 I/Q Demodulator and Phase Shifter ........................................ 10 Dynamic Range and Noise........................................................ 11 Summation of Multiple Channels (Analog Beamforming).. 12 Phase Compensation and Analog Beamforming................... 12 Serial Interface ............................................................................ 13 ENBL Bits ................................................................................ 13 Applications..................................................................................... 14 Logic Inputs and Interfaces....................................................... 14 Reset Input .................................................................................. 14 Outline Dimensions ....................................................................... 15 REVISION HISTORY 12/19/06 – Rev. Prelim A1 Rev. PrA | Page 2 of 15 Preliminary Technical Data AD8339 SPECIFICATIONS AD8339 Table 1. VS = ±5 V, TA = 25°C, 4 fLO = 20 MHz, fRF = 5.01 MHz, fBB = 10 kHz, PLO ≥ 0 dBm, per channel performance, dBm (50 Ω) unless otherwise noted. Single channel AD8021 LPF values RFILT = 1.58 kΩ and CFILT = 1 nF (see Figure 2). Parameter OPERATING CONDITIONS LO Frequency Range Conditions 4x internal LO at pins 4LOP and 4LON Square Wave Sine Wave Mixing Limited by external filtering Min Typ Max Unit RF Frequency Range Baseband Bandwidth LO Input Level VSUPPLY (VS) Temperature Range DEMODULATOR PERFORMANCE Input Impedance Transconductance LF TBD LF LF ±4.5 -40 0 ±5 100 100 25 25 13 ±5.5 +85 MHz MHz MHz MHz dBm V °C kΩ||pF kΩ||pF Dynamic Range Max Input Swing Peak Output Current (No Filtering) Input P1dB Third Order Intermodulation (IM3) Equal Input Levels Unequal Input Levels Third Order Input Intercept (IIP3) LO Leakage RF - Differential LO – Differential Demodulated IOUT/VIN; Each Ix or Qx output after low pass filtering measured from RF inputs All Phases IP1dB minus Input referred noise (dBm) Differential; Inputs biased at 2.5V; Pins RFxP, RFxN 0° Phase Shift 45° Phase Shift Ref = 50 Ω Ref = 1VRMS fRF1 = 5.010 MHz, fRF2 = 5.015 MHz, fLO = 5.023 MHz Baseband tones: -7 dBm @ 8 kHz and 13 kHz Baseband tones: -1 dBm @ 8 kHz and -31 dBm @13 kHz Same conditions as IM3 Measured at RF inputs, worst phase, measured into 50 Ω Measured at baseband outputs, worst phase, AD8021 disabled, measured into 50 Ω All codes, see Figure XX Output Noise ÷ Conversion Gain (see Figure XX) Output noise ÷ 1.58 kΩ With AD8332 LNA RS = 50 Ω, RFB = ∞ RS = 50 Ω, RFB = 1.1k Ω RS = 50 Ω, RFB = 274 Ω Pins 4LOP and 4LON Pins RFxP and RFxN Pins 4LOP and 4LON (each pin) For maximum differential swing; Pins RFxP and RFxN (DC-coupled to AD8332 output) Pins IxOP and QxOP 7||7 100||1 1.1 158 2.7 ±2.4 ±3.3 14.5 1.5 mS dB (1Hz BW) Vpp mA mA dBm dBV -75 TBD 30 TBD TBD 4.7 TBD TBD TBD TBD TBD -2 -35 0.2 2.5 -1.5 0.7 3.8 dBc dBc dBm dBm dBm dB nV/√Hz pA/√Hz dB dB dB μA μA V V V Conversion Gain Input Referred Noise Output Current Noise Noise Figure Bias Current LO Common Mode Range Range RF Common Mode Voltage Output Compliance Range Rev. PrA | Page 3 of 15 AD8339 PHASE ROTATION PERFORMANCE Phase Increment Quadrature Phase Error I/Q Amplitude Imbalance Channel-to-Channel Matching LOGIC INTERFACES Logic Level High Logic Level Low Bias Current Input Resistance LO Divider RSET Setup Time LO Divider RSET High Pulse Width LO Divider RSET Setup Time Phase Response Time Enable Response Time Output Logic Level High Logic Level Low SPI TIMING CHARACTERISTICS SCLK Frequency CSB to SCLK Setup Time SCLK High Pulse Width SCLK Low Pulse Width Data Access Time after SCLK Falling Edge Data Setup Time Prior to SCLK Rising Edge Data Hold Time after SCLK Rising Edge CSB High Pulse Width SCLK Fall to CSB Fall Hold Time SCLK Fall to CSB Rise Hold Time POWER SUPPLY Supply Voltage Quiescent Current Over Temperature Quiescent Power Disable Current PSRR One CH is reference, others are stepped 16 Phase Steps per Channel Preliminary Technical Data 22.5 ±1 0.25 ±1 ±0.5 1.5 0.9 Logic High (pulled to +5V) Logic Low (pulled to GND) RSET rising edge to 4LOP-4LON (Differential) rising edge RSET falling edge prior to 4LOP-4LON (Differential) rising edge Measured from CSB going high Measured from CSB going high (with 0.1 μF cap on pin LODC) Pin SDO Loaded with 5 pF and next SDI input Loaded with 5 pF and next SDI input Pins SDI,SDO,CSB,SCLK, RSTS fCLK T1 T2 T3 T4 T5 T6 T7 T8 T9 Pins VPOS,VNEG VPOS, all phase bits = 0 VNEG, all phase bits = 0 -40°C < TA < 85°C Per Channel, all phase bits = 0 Per Channel max (depends on phase bits) All Channels Disabled; SPI stays on VPOS to Ix/Qx outputs (meas. @ AD8021 output) VNEG to Ix/Qx outputs (meas. @ AD8021 output) 5 20 5 TBD 15 0.5 0 4 Ix to Qx; all phases, 1σ Ix to Qx; all phases, 1σ Phase Match I-to-I and Q-to-Q; -40°C < TA < 85°C Ampl. Match I-to-I and Q-to-Q; -40°C < TA < 85°C Pins SDI,CSB,SCLK, RSTS,RSET ° ° dB ° dB V V μA μA MΩ ns ns ns μs μs 1.7 1.9 0.2 0.5 10 V V MHz ns ns ns ns ns ns ns ns TBD TBD TBD TBD TBD TBD TBD TBD TBD ±4.5 ±5 37.5 -21 73 TBD 2.6 TBD TBD ±5.5 TBD TBD V mA mA mA mW mW mA dB dB Rev. PrA | Page 4 of 15 Preliminary Technical Data ABSOLUTE MAXIMUM RATINGS Table 2. Parameter Voltages Supply Voltage VS RF Inputs 4LO Inputs Outputs (IxOP, QxOP) Digital Inputs SDO Output LODC Pin Thermal Data —4 Layer Jedec Board No Air Flow (Exposed Pad Soldered to PC Board) θJA θJB θJC ΨJT ΨJB Maximum Junction Temperature Maximum Power Dissipation (Exposed Pad Soldered to PC Board) Operating Temperature Range Storage Temperature Range Lead Temperature Range (Soldering 60 sec) Rating ±6 V +6 V, GND +6 V, GND +1 V, -6 V +6 V, GND +6 V, GND +6 V (max) VPOS –1.5 V (min) AD8339 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. TBD°C/W TBD°C/W TBD°C/W TBD°C/W TBD°C/W 150°C TBD W –40°C to +85°C –65°C to +150°C 300°C ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. PrA | Page 5 of 15 AD8339 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS 40 RSTS 38 RF1P 37 RF1N 36 COMM 35 VPOS 34 RSET 33 I1OP 32 Q1OP 31 VNEG 39 SDI Preliminary Technical Data RF2N RF2P COMM COMM SCLK CSB VPOS VPOS RF3P 1 2 3 4 5 6 7 8 9 Pin 1 Identifier 30 Q2OP 29 I2OP 28 VPOS 27 VPOS AD8339 Top View (not to scale) 26 4LOP 25 4LON 24 VNEG 23 VNEG 22 I3OP 21 Q3OP RF3N 10 VPOS 11 SDO 12 RF4P 13 RF4N 14 COMM 15 VPOS 16 LODC 17 I4OP 18 Q4OP 19 VNEG 20 Figure 2. 40-Lead LFCSP Table 3. Pin Function Descriptions Pin No. 1, 2, 9, 10, 13, 14, 37, 38 3, 4, 15, 36 5 6 7, 8, 11, 16, 27, 28, 35 12 17 18, 19, 21, 22, 29, 30, 32, 33 20, 23, 24, 31 25, 26 34 39 40 Mnemonic RF1P-RF4P RF1N-RF4N COMM SCLK CSB VPOS Description RF Inputs. No internal bias. The optimum common mode voltage for maximum symmetrical input differential swing is 2.5 V if ±5 V supplies are used. Ground Serial Interface – Clock Serial Interface – Chip Select Bar. Active Low. Positive Supply. These pins should be decoupled with a ferrite bead in series with the supply, plus a 0.1 μF and 1 nF capacitor between the VPOS pins and ground. Since the VPOS pins are internally connected, one set of supply decoupling components on each side of the chip should be sufficient. Serial Interface – Data Output. Normally connected to SDI of next chip or left open. Decoupling Pin for LO. A 0.1 μF capacitor should be connected between this pin and ground. Value of cap does influence chip enable/disable times. I/Q Outputs. These outputs provide a bidirectional current that can be converted back to a voltage via a transimpedance amplifier. Multiple outputs can be summed together through simply connecting them (Wire-OR). The bias voltage should be set to 0 V or less by the transimpedance amplifier, see Figure 7. Negative Supply. These pin should be decoupled with a ferrite bead in series with the supply, plus a 0.1 μF and 1 nF capacitor between the pin and ground. Since the VNEG pins are internally connected, one set of supply decoupling components should be sufficient. LO Inputs. No internal bias; optimally biased by an LVDS driver. For best performance, these inputs should be driven differentially. LO Interface - Reset. Logic threshold is at about 1.1 V and therefore can be driven by >1.8 V CMOS logic. Serial Interface – Data Input. Logic threshold is at about 1.1 V and therefore can be driven by >1.8 V CMOS logic. Reset for SPI Interface. Logic threshold is at about 1.1 V and therefore can be driven by >1.8 V CMOS logic. For quick testing without the need to program the SPI, the voltage on the RSTS pin should be pulled to -1.4 V; this enables all four channels in the Phase (I=1,Q=0) state. SDO LODC I1OP-I4OP, Q1OP-Q4OP VNEG 4LOP, 4LON RSET SDI RSTS Rev. PrA | Page 6 of 15 Preliminary Technical Data EQUIVALENT INPUT CIRCUITS VPOS VPOS RSTS SCLK SDI SDO CSB AD8339 LOGIC INTERFACE RFxP COMM RFxN COMM Figure 3. Logic Inputs Figure 6. RF Inputs VPOS COMM 4LOP Ix Qx 4LON COMM VNEG Figure 4. Local Oscillator Inputs Figure 7. Output Drivers VPOS LODC COMM Figure 5. LO Decoupling Pin Rev. PrA | Page 7 of 15 AD8339 TYPICAL PERFORMANCE CHARACTERISTICS Preliminary Technical Data VS = ±5 V, TA = 25°C, 4fLO = 20 MHz, fLO = 5 MHz, fRF= 5.01 MHz, fBB = 10 kHz, PLO ≥ 0 dBm (50Ω); single-ended sine wave; per channel performance, differential voltages, dBm (50Ω), phase select code = 0000, unless otherwise noted (see Error! Reference source not found.). Rev. PrA | Page 8 of 15 Preliminary Technical Data TEST CIRCUITS AD8339 Rev. PrA | Page 9 of 15 AD8339 THEORY OF OPERATION The AD8339 is a quad I/Q demodulator with a programmable phase shifter for each channel. The primary application is phased array beamforming in medical ultrasound. Other potential applications might be phased array radar, and smart antennas for mobile communications. The AD8339 can also be used in applications that require multiple well-matched I/Q demodulators. The AD8339 is architecturally very similar to its predecessor – the AD8333. The major differences are: Preliminary Technical Data divide-by-four logic circuit. The divider is dc-coupled and inherently broadband; the maximum LO frequency is limited only by its switching speed. The duty cycle of the quadrature LO signals is intrinsically 50% and is unaffected by the asymmetry of the externally connected 4xLO input. Furthermore, the divider is implemented such that the 4xLO signal re-clocks the final flip-flops that generate the internal LO signals and thereby minimizes noise introduced by the divide circuitry. For optimum performance, the 4xLO input is driven differentially, but can also be driven single-ended. A good choice for a drive is an LVDS device. The common-mode range on each pin is approximately 0.2 V to 3.8 V with the nominal ±5 V supplies. The minimum 4xLO level is frequency dependent. For optimum noise performance it is important to ensure that the LO source has very low phase noise (jitter) and adequate input level to assure stable mixer-core switching. The gain through the divider determines the LO signal level vs. RF frequency. The AD8339 can be operated to very low frequencies at the LO inputs if a square wave is used to drive the LO. Beamforming applications require a precise channel-to-channel phase relationship for coherence among multiple channels. A reset pin is provided to synchronize the LO divider circuits in different AD8339s when they are used in arrays. The RSET pin resets the dividers to a known state after power is applied to multiple AD8339s. A logic input must be provided to the RSET pin when using more than one AD8339. Note that at least one channel must be enabled for the LO interface to also be enabled and the LO reset to work. See the Reset Input section in the applications section for more detail. 1. 2. the addition of a serial (SPI) interface that allows daisychaining of multiple devices reduced power per channel at the expense of a slight decrease in dynamic range Figure 1 shows the block diagram and pinout of the AD8339. Four RF inputs accept signals from the RF sources, and a local oscillator (applied to differential input pins marked 4LOP and 4 LON) common to all channels, comprise the analog inputs. Each channel has the option to program 16 delay states/360° (or 22.5°/step) selectable via the SPI port. The part has two reset inputs: RSET is used to synchronize the LO dividers in multiple AD8339s used in arrays; RSTS is used to set the SPI port bits to all zeros. This can be useful in testing or when one quickly wants to turn off the device without first programming the SPI port. RSTS RF1P RF1N COMM VPOS RSET I1OP Q1OP VNEG 31 SDI 39 40 38 37 36 35 34 33 RF2N 1 BIAS V/I φ 32 I/V 30 Q2OP CHANNEL 1 φ I/V 29 I2OP RF2P 2 COMM 3 RSTS SDI φ V/I I/V 28 VPOS CHANNEL 2 φ I/V 27 VPOS COMM 4 SCLK 5 SCLK 0 90 26 LO Divide-by-4 25 4LOP I/Q DEMODULATOR AND PHASE SHIFTER The I/Q demodulators consist of double-balanced Gilbert cell mixers. The RF input signals are converted into currents by transconductance stages that have a maximum differential input signal capability of 2.7 V p-p. These currents are then presented to the mixers, which convert them to baseband: RF − LO and RF + LO. The signals are phase shifted according to the codes programmed into the SPI latch (see Table 4); the phase bits are labeled PHx0 through PHx3 where ‘0’ indicates LSB and ‘3’ indicates MSB. The phase shift function is an integral part of the overall circuit (patent pending). The phase shift listed in Column 1 of Table 4 is defined as being between the baseband I or Q channel outputs. As an example, for a common signal applied to a pair of RF-inputs to an AD8339, the baseband outputs are in phase for matching phase codes. However, if the phase code for Channel 1 is 0000 and that of Channel 2 is 0001, then Channel 2 leads Channel 1 by 22.5°. Following the phase shift circuitry, the differential current signal is converted from differential to single-ended via a CSB 6 CSB 4LON VPOS 7 Serial Interface (SPI) V/I φ I/V 24 VNEG CHANNEL 3 φ I/V 23 VNEG VPOS 8 RF3P 9 V/I φ I/V 22 I3OP CHANNEL 4 φ I/V 21 Q3OP RF3N 10 SDO 11 12 13 14 15 16 17 18 19 VPOS SDO RF4P RF4N COMM VPOS LODC I4OP Q4OP Figure 1. Block Diagram and Pinout Each of the current formatted I and Q outputs sum together for beamforming applications. Multiple channels are summed and converted to a voltage using a transimpedance amplifier. If desired, channels can also be used individually. QUADRATURE GENERATION The internal 0° and 90° LO phases are digitally generated by a VNEG 20 Rev. PrA | Page 10 of 15 Preliminary Technical Data current mirror. An external transimpedance amplifier is needed to convert the I and Q outputs to voltages. Table 4. Phase Select Code for Channel-to-Channel Phase Shift φ-Shift 0º 22.5º 45º 67.5º 90º 112.5º 135º 157.5º 180º 202.5º 225º 247.5º 270º 292.5º 315º 337.5º PHx3 (MSB) 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 PHx2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 PHx1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 PHx0 (LSB) 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 AD8339 voltage noise density (en) of the AD8339 is nominally about TBD nV/√Hz. For the noise of the AD8339 to degrade the system noise figure (NF) by 1 dB, the combined noise of the source and the LNA should be about twice that of the AD8339 or TBD nV/√Hz. If the noise of the circuitry before the AD8339 is less than TBD nV/√Hz then the system NF degrades more than 1 dB. For example, if the noise contribution of the LNA and source is equal to the AD8339, or TBD nV/√Hz, the degradation is 3 dB. If the circuit noise preceding the AD8339 is 1.3× as large as that of the AD8339 (or about TBD nV/√Hz) the degradation is 2 dB. For a circuit noise 1.45× that of the AD8339 (TBD nV/√Hz) the degradation is 1.5 dB. To determine the input referred noise it is important to know the active low pass filter (LPF) values RFILT and CFILT, shown in Figure 2. Typical filter values for a single channel are 1.58 kΩ and 1 nF, and implement a 100 kHz single-pole LPF. In the case that two channels are summed as is done on the evaluation board, the values would be the same as for a single channel of the AD8333, namely 787 Ω and 2.2 nF. If the RF and LO are offset by 10 kHz, the demodulated signal is 10 kHz and is passed by the LPF. The single-channel mixing gain, from the RF input to the AD8021 output (for example, I1´, Q1´) is approximately 1.7 (4.7 dB). This together with the TBD nV/√Hz of AD8339 noise results in about TBD nV/√Hz at the AD8021 output. Since the AD8021, including the 1.58 kΩ feedback resistor, contributes another 6.3 nV/√Hz, the total output referred noise is about TBD nV/√Hz. This value can be adjusted by increasing the filter resistor while maintaining the corner frequency, thereby increasing the gain. The factor limiting the magnitude of the gain is the output swing and drive capability of the op-amp selected for the I-to-V converter, in this instance the AD8021. Because any amplifier has limited drive capability there will be a finite number of channels that can be summed. This is explained in great detail in the section below called – Channel Summing. DYNAMIC RANGE AND NOISE Figure 2 is an interconnection block diagram of two channels (1/2 of the AD8339), more channels are easily added to the summation (up to 16 when using an AD8021 as the summation amplifier) by wire-or connecting the outputs as shown for two channels. For optimum system noise performance, the RF input signal is provided by a very low noise amplifier such as the LNA of the AD8332/AD8334 or the preamplifier of the AD8335. In beamformer applications, the I and Q outputs of a number of receiver channels are summed (for example, the two channels illustrated in Figure 2). The dynamic range of the system increases by the factor 10log10(N), where N is the number of channels (assuming random uncorrelated noise.) The noise in the two channel example of Figure 2 is increased by 3 dB while the signal doubles (+6 dB), yielding an aggregate SNR improvement of (+6 − 3) = +3 dB. For four channels the dynamic range will increase by +6 dB and so on. Judicious selection of the RF amplifier ensures the least degradation in dynamic range. The input referred spectral Rev. PrA | Page 11 of 15 AD8339 TRANSMITTER T/R SW TRANSDUCER RFB Preliminary Technical Data AD8332 LNA OR AD8335 PREAMP CH1 RF CH1 PHASE SELECT 2 4 2 Φ 2 2 2 I1 * CFILT RFILT ΣI AD8333 ½ AD8339 0° 90° CLOCK GENERATOR ÷4 90° Φ Q1 AD8021 CFILT * RFILT ΣQ ADC 16-BIT I DATA 570kSPS 2 Φ 2 Q2 AD7665 OR AD7686 ADC 16-BIT 570kSPS Q DATA 0° 2 CH2 RF TRANSMITTER T/R SW TRANSDUCER RFB 2 4 Φ 2 I2 AD8021 AD8332 LNA OR AD8335 PREAMP CH2 PHASE SELECT **UP to 16 channels of Up TO 8 CHANNELS PER AD8021 AD8339 per AD8021 Figure 2. Interconnection Block Diagram for ½ of AD8339 SUMMATION OF MULTIPLE CHANNELS (ANALOG BEAMFORMING) Beamforming, as applied to medical ultrasound, is defined as the phase alignment and summation of signals generated from a common source, but received at different times by a multielement ultrasound transducer. Beamforming has two functions: it imparts directivity to the transducer, enhancing its gain and it defines a focal point within the body from which the location of the returning echo is derived. The primary application for the AD8339 is in analog beamforming circuits for ultrasound. converted by a very large dynamic range I/Q demodulator. The resultant I and Q signals are filtered and then sampled by two high resolution AD converters. The sampled signals are processed to extract the relevant Doppler information. Alternatively, the RF signal can be processed by downconversion on each channel individually, phase shifting the down-converted signal, and then combining all channels. The AD8333 and the AD8339 provide the means to implement this architecture. The down-conversion is done by an I/Q demodulator on each channel, and the summed current output is the same as in the delay line approach. The subsequent filters after the I-to-V conversion and the AD converters are similar. The AD8339 integrates the phase shifter, frequency conversion, and I/Q demodulation into a single package, and directly yields the baseband signal. Figure 3 is a simplified diagram showing the idea for two channels. The ultrasound wave USW is received by two transducer elements, TE1 and TE2, in an ultrasound probe and generates signals E1 and E2. In this example, the phase at TE1 leads the phase at TE2 by 45°. TRANSDUCER ELEMENTS TE1 AND TE2 CONVERT USW TO ELECTRICAL AD8332 USW AT TE1 SIGNALS LEADS USW ES1 LEADS AT TE2 BY ES2 BY 45° 45° 19dB 45° LNA E1 E2 19dB LNA CH 2 PHASE LEAD 45° PHASE COMPENSATION AND ANALOG BEAMFORMING Modern ultrasound machines used for medical applications employ an array of receivers for beamforming, with typical CW Doppler array sizes up to 64 receiver channels phase-shifted and summed together to extract coherent information. When used in multiples, the desired signals from each of the channels can be summed to yield a larger signal (increased by a factor N, where N is the number of channels), while the noise is increased by the square root of the number of channels. This technique enhances the signal to noise performance of the machine. The critical elements in a beamformer design are the means to align the incoming signals in the time domain, and the means to sum the individual signals into a composite whole. In traditional analog beamformers incorporating Doppler, a V-to-I converter per channel and a cross-point switch precede passive delay lines used as a combined phase shifter and summing circuit. The system operates at the receive frequency (RF) through the delay line which also sums the signals from the various channels, and then the combined signal is down- ½AD8333 AD8339 PHASE BIT SETTINGS CH 1 REF (NO PHASE LEAD) S1 AND S2 ARE NOW IN PHASE S1 S2 05543-063 05543-038 SUMMED OUTPUT S1 + S2 Figure 3. Simplified Example of the AD8339 Phase Shifter Rev. PrA | Page 12 of 15 Preliminary Technical Data In a real application, the phase difference depends on the element spacing, λ (wavelength), speed of sound, angle of incidence, and other factors. The signals ES1 and ES2 are amplified 19 dB by the low-noise amplifiers in the AD8332; for lower performance portable ultrasound applications, the combination of the AD8335 and the AD8339 result in the lowest power per channel. For optimum signal-to-noise performance, the output of the LNA is applied directly to the input of the AD8339. In order to sum the signals ES1 and ES2, ES2 is shifted 45° relative to ES1 by setting the phase code in Channel 2 to 0010. The phase aligned current signals at the output of the AD8333 are summed in an I-to-V converter to provide the combined output signal with a theoretical improvement in dynamic range of 3 dB for the sum of two channels. AD8339 transferred to the latch. Depending on the data loaded the corresponding channels will be enabled, and the phases on each channel will be set. Figure XYZ shows how the timing might look when two AD8339s have their data loaded. ENBL Bits If all four ENBL bits are set to ‘0’, then only the SPI port is powered up. This feature allows for very low power consumption (about 13 mW per AD8339 or 3.25 mW per channel) when the CW Doppler function is not needed. Since the SPI port stays alive even when the rest of the chip is powered down, the part can be awakened again by simply programming the port. As soon as the CSB signal goes high, the part turns on again. It should be pointed out that this will take a fair amount of time because of the external capacitor on the LODC pin. It will take about 10-20 μs with the recommended 0.1 μF decoupling cap. The decoupling cap on this pin is intended to reduce bias noise contribution in the LO divider chain. The user can experiment with the value of this decoupling capacitor to see what the smallest value can be without any dynamic range degradation within the frequency band of interest. The SPI also has an additional pin that can be used in a test mode, or as a quick way to reset the SPI and de-power the chip. All bits in both the shift register and the latch can be reset to ‘0’ when pin RSTS is pulled above about 1.2 V. For quick testing without the need to program the SPI, the voltage on the RSTS pin should be first pulled high and then pulled to -1.4 V; this enables all four channels in the (I=1,Q=0) state (all phase bits are 0000). SERIAL INTERFACE The AD8339 contains a 4-wire SPI compatible digital interface (SDI, SCLK, CSB, and SDO). The interface is comprised of a 20bit shift register plus a latch. The shift register needs to be loaded MSB first. The data allows control over each channel’s phases, plus the last four bits shifted into the register determine the enable state of the individual channels. Figure XYZ shows a block and timing diagram of the serial interface. The shift direction is to the “right” with MSB first. As soon CSB goes low, the data in the latch is protected and new data can be loaded into the shift register. If only one AD8339 needs to be programmed, then only 20 bits need to be shifted into the part before CSB goes high. As soon as CSB goes high, the data loaded into the shift register will be Figure XYZ. SPI - Block and Timing Diagram Rev. PrA | Page 13 of 15 AD8339 APPLICATIONS The AD8339 is the key component of a phase-shifter system that aligns time-skewed information contained in RF signals. Combined with a variable gain amplifier (VGA) and low noise amplifier (LNA) as in the AD8334/5 VGA family, the AD8339 forms a complete analog receiver for a high-performance ultrasound system. Preliminary Technical Data least the tSET-UP should be ≥ 5 ns. An optimal timing set-up would be for the RSET pulse to go high on a 4 x LO falling edge and go low on a 4 x LO falling edge; this gives 10 ns of set-up time even at a 4 x LO frequency of 50 MHz (12.5 MHz internal LO). Synchronization of multiple AD8339s can be checked as follows: LOGIC INPUTS AND INTERFACES All logic inputs of the AD8339 including the SPI and RSET pins are CMOS compatible down to 1.8 V. Each logic input pin has a Schmitt trigger activated input that contains a threshold that is centered at about 1.1 V with a hysteresis of ±0.1 V around this value. The LO divider RSET pin has a slightly higher threshold at about 1.3 V and a hysteresis of about ±0.1 V. This input also can still be driven by 1.8 V CMOS logic. The only logic output, SDO, generates a signal that has a logic low level of about 0.2 V and a logic high level of about 1.9V to allow for easy interfacing to the next AD8339 SDI input. 1. Activate at least one channel per AD8339 by setting the appropriate channel enable bit in the serial interface. Set the phase code of all AD8339 channels the same, for example, 0000. Apply the same test signal to all devices that generates a sine wave in the baseband output and measure the output of one channel per device. Apply a RSET pulse to all AD8339s. Since all the phase codes of the AD8339s should be the same, the combined signal of multiple devices should be N times bigger than a single channel. If the combined signal is less than N times one channel, then the LO phases of the individual AD8339s are most likely in error. 2. 3. 4. 5. RESET INPUT The RSET pin is used to synchronize the LO dividers in AD8339 arrays. Because they are driven by the same internal LO, the four channels in any AD8339 are inherently synchronous. However, when multiple AD8339s are used it is possible that their dividers wake up in different phase states. The function of the RSET pin is to phase align all the LO signals in multiple AD8339s. The 4 × LO divider of each AD8339 can initiate in one of four possible states - 0°, 90°, 180°, and 270° relative to other AD8339s. The internally generated I/Q signals of each AD8339 LO are always at a 90° angle relative to each other, but a phase shift can occur during power up between the internal LOs of the different AD8339s. The LO divider reset function has been improved in the AD8339 over the AD8333. The RSET pin still provides an asynchronous reset of the LO dividers by forcing the internal LO to "hang", however, now the LO reset function is fast and does not require a shut-down of the 4 x LO input signal. The RSET mechanism also allows the measurement of nonmixing gain from the RF input to the output. The rising edge of the active high RSET pulse can occur at any time; however, the duration should be ≥ 20 ns minimum. When the RSET pulse transitions from high to low, the LO dividers are reactivated on the next rising edge of the 4 x LO clock. To guarantee synchronous operation of an array of AD8339s the RSET pulse needs to go low on all devices before the next rising edge of the 4 x LO clock. Therefore it is best to have the RSET pulse go low on the falling edge of the 4 x LO clock; at the very Rev. PrA | Page 14 of 15 Preliminary Technical Data OUTLINE DIMENSIONS AD8339 Figure 8. 40-Lead Chip Scale Package Rev. PrA | Page 15 of 15 PR06587-0-1/07(PrA)