Data Sheet
50 Hz to 3.8 GHz
65 dB TruPwr™ Detector
AD8362
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Complete, fully calibrated measurement/control system
Accurate rms-to-dc conversion from 50 Hz to 3.8 GHz
Input dynamic range of >65 dB: −52 dBm to +8 dBm in 50 Ω
Waveform and modulation independent, such as
GSM/CDMA/TDMA
Linear-in-decibels output, scaled 50 mV/dB
Law conformance error of 0.5 dB
All functions temperature and supply stable
Operates from 4.5 V to 5.5 V at 24 mA
Power-down capability to 1.3 mW
DECL
CHPF
INHI
x2
CLPF
INLO
VOUT
x2
VTGT
ACOM
VSET
APPLICATIONS
AD8362
BIAS
VREF
COMM
PWDN
VPOS
02923-001
Power amplifier linearization/control loops
Transmitter power controls
Transmitter signal strength indication (TSSI)
Radio frequency (RF) instrumentation
Figure 1.
GENERAL DESCRIPTION
Table 1. Next Generation Upgrades for the AD8362
The AD8362 is a true rms-responding power detector that has
a 65 dB measurement range. It is intended for use in a variety of
high frequency communication systems and in instrumentation
requiring an accurate response to signal power. It is easy to use,
requiring only a single supply of 5 V and a few capacitors. It
operates from arbitrarily low frequencies to over 3.8 GHz and
accepts inputs from −52 dBm to +8 dBm with crest factors that
are typical of quadrature amplitude modulation (QAM) and
orthogonal frequency division multiplexing (OFDM) modulation
schemes.
Device No.
ADL5902
The AD8362 has a 1.3 mW power consumption when powered
down by a logic high applied to the PWDN pin. It powers up
within about 20 μs to its nominal operating current of 20 mA at
25°C. The AD8362 is supplied in a 16-lead TSSOP for operation
over the temperature range of −40°C to +85°C.
Rev. F
HMC1120
ADL5906
AD8363
ADL5903
ADL5501
Product Description
Improved temperature stability, single-ended
input drive.
Improved sensitivity and range, includes envelope
tracking and peak-hold function, single-ended
input drive.
Improved temperature stability, single-ended
input drive. Slightly lower input range.
Improved temperature stability, single-ended
input drive. Lower input range.
Lower range, significantly lower power consumption,
single-ended input drive.
Lower range, significantly lower power consumption,
linear in V/V output, single-ended input drive.
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AD8362
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Operation in RF Measurement Mode.......................................... 20
Applications ....................................................................................... 1
Basic Connections ...................................................................... 20
Functional Block Diagram .............................................................. 1
Device Disable ............................................................................ 20
General Description ......................................................................... 1
Recommended Input Coupling ................................................ 20
Revision History ............................................................................... 3
Operation at Low Frequencies.................................................. 21
Specifications..................................................................................... 4
Choosing a Value for CHPF...................................................... 22
Absolute Maximum Ratings............................................................ 7
Choosing a Value for CLPF....................................................... 22
ESD Caution .................................................................................. 7
Pin Configuration and Function Descriptions ............................. 8
Adjusting VTGT to Accommodate Signals with Very High
Crest Factors ............................................................................... 23
Equivalent Circuits ........................................................................... 9
Altering the Slope ....................................................................... 23
Typical Performance Characteristics ........................................... 10
Temperature Compensation and Reduction of Transfer
Function Ripple .......................................................................... 24
Characterization Setup .................................................................. 16
Equipment ................................................................................... 16
Analysis ........................................................................................ 16
Circuit Description ......................................................................... 17
Square Law Detection ................................................................ 17
Voltage vs. Power Calibration ................................................... 18
Offset Elimination ...................................................................... 19
Time-Domain Response of the Closed Loop ......................... 19
Temperature Compensation at Various WiMAX Frequencies up
to 3.8 GHz........................................................................................ 25
Operation in Controller Mode ................................................. 27
RMS Voltmeter with 90 dB Dynamic Range .......................... 28
AD8362 Evaluation Board ............................................................ 29
Outline Dimensions ....................................................................... 32
Ordering Guide .......................................................................... 32
Rev. F | Page 2 of 33
Data Sheet
AD8362
REVISION HISTORY
9/15—Rev. E to Rev. F
Changes to General Description Section ....................................... 1
Added Table 1; Renumbered Sequentially .................................... 1
Change to Equation 3 .....................................................................17
Changes to Ordering Guide ...........................................................32
12/12—Rev. D to Rev. E
Changes to Time Domain Response of the Closed Loop Section.. 18
Changes to Equation 13, Choosing a Value for CLPF Section
and Equation 14 ...............................................................................21
Changes to Ordering Guide ...........................................................31
6/07—Rev. C to Rev. D
Changes to Features, General Description .................................... 1
Changes to Table 1 ............................................................................ 3
Changes to Table 2 ............................................................................ 6
Added Figure 21 to Figure 25 ........................................................11
Changes to Equipment Section .....................................................15
Changes to Circuit Description Section .......................................16
Changes to Single-Ended Input Drive Section ............................19
Changes to Choosing a Value for CHPF section ........................21
Changes to Choosing a Value for CLPF section .........................21
Changes to Figure 57 ......................................................................23
Changes to Figure 58 ......................................................................24
Added Temperature Compensation at Various WiMAX
Frequencies up to 3.8 GHz Section ...............................................24
Changes to Ordering Guide ...........................................................31
9/05—Rev. B to Rev. C
Changes to Specifications ................................................................ 3
Changes to Table 3 ............................................................................ 7
Deleted Figure 16 to Figure 18; Renumbered Sequentially ....... 10
Changes to Figure 32 and Figure 33 ............................................. 13
Replaced Circuit Description Section .......................................... 15
Changes to Operation in RF Measurement Mode Section........ 18
Deleted Using the AD8362 Section .............................................. 20
Deleted Main Modes of Operation Section ................................. 22
Changes to Operation in Controller Mode Section ................... 23
Changes to AD8362 Evaluation Board Section .......................... 25
Deleted General Applications Section.......................................... 29
3/04—Rev. A to Rev. B
Updated Format ................................................................. Universal
Changes to Specifications ................................................................ 3
Changes to the Offset Elimination Section ................................. 16
Changes to the Operation at Low Frequencies Section ............. 17
Changes to the Time-Domain Response of the Closed Loop
Section .............................................................................................. 17
Changes to Equation 13 ................................................................. 24
Changes to Table 5 .......................................................................... 31
6/03—Rev. 0 to Rev. A
Updated Ordering Guide ................................................................. 5
Change to Analysis Section ........................................................... 12
Updated AD8362 Evaluation Board Section ............................... 26
2/03—Revision 0: Initial Version
Rev. F | Page 3 of 33
AD8362
Data Sheet
SPECIFICATIONS
VS = 5 V, T = 25°C, ZO = 50 Ω, differential input drive via balun 1, VTGT connected to VREF, VOUT tied to VSET, unless otherwise noted.
Table 2.
Parameter
OVERALL FUNCTION
Maximum Input Frequency
Input Power Range (Differential)
Nominal Low End of Range
Nominal High End of Range
Input Voltage Range (Differential)
Nominal Low End of Range
Nominal High End of Range
Input Power Range (S-Sided)
Nominal Low End of Range
Nominal High End of Range
Input Voltage Range (S-Sided)
Nominal Low End of Range
Nominal High End of Range
Input Power Range (S-Sided)
Nominal Low End of Range
Nominal High End of Range
Output Voltage Range
Nominal Low End of Range
Nominal High End of Range
Output Scaling (Log Slope)
Law Conformance Error
RF INPUT INTERFACE
Input Resistance
OUTPUT INTERFACE
Available Output Range
Absolute Voltage Range
Nominal Low End of Range
Nominal High End of Range
Source/Sink Current
Slew Rate Rising
Slew Rate Falling
Rise Time, 10% to 90%
Fall Time, 90% to 10%
Wideband Noise
VSET INTERFACE
Nominal Input Voltage Range
Input Resistance
Scaling (Log Slope)
Scaling (Log Intercept)
VOLTAGE REFERENCE
Output Voltage
Temperature Sensitivity
Output Resistance
Test Conditions/Comments
Min
Typ
Max
Unit
3.8
GHz
−52
8
dBm
dBm
1.12
1.12
mV rms
V rms
−40
0
dBm
dBm
2.23
2.23
mV rms
V rms
−35
12 4
dBm
dBm
100
4.9
50
±0.5
mV
V
mV/dB
dB
100
200
Ω
Ω
dB referred to 50 Ω impedance level, f ≤ 2.7 GHz, into 1:4 balun1
RMS voltage at input terminals, f ≤ 2.7 GHz, into input of the device
Single-ended drive, CW input, f ≤ 2.7 GHz, into input resistive network 2
RMS voltage at input terminals, f ≤ 2.7 GHz
Single-ended drive, CW input, f ≥ 2.7 GHz, into matched input network 3
RL ≥ 200 Ω to ground
In general, VS − 0.1 V
Over central 60 dB range, f ≤ 2.7 GHz
Pin INHI, Pin INLO, ac-coupled, at low frequencies
Single-ended drive, with respect to DECL
Differential drive
Pin VOUT
RL ≥ 200 Ω to ground
Measurement mode, f = 900 MHz, PIN = −52 dBm
Measurement mode, f = 900 MHz, PIN = +8 dBm
VOUT held at VS/2, to 1% change
CL = open
CL = open
0.2 V to 1.8 V, CLPF = Open
1.8 V to 0.2 V, CLPF = Open
CLPF = 1000 pF, fSPOT ≤ 100 kHz
Pin VSET
To ±1 dB error
f = 900 MHz
f = 900 MHz, into 1:4 balun
Pin VREF
25°C
−40°C ≤ TA ≤ +85°C
0.1
4.9
V
0.32
3.44
0.48
3.52
V
V
mA
V/µs
V/µs
ns
µs
nV/√Hz
3.75
V
kΩ
mV/dB
dBm
dBV
48
60
5
45
0.4
70
0.5
46
−64
−77
1.225
Rev. F | Page 4 of 33
68
50
−60
−73
1.25
0.08
8
54
−56
−69
1.275
V
mV/°C
Ω
Data Sheet
Parameter
RMS TARGET INTERFACE
Nominal Input Voltage Range
Input Bias Current
Incremental Input Resistance
POWER-DOWN INTERFACE
Logic Level to Enable
Logic Level to Disable
Input Current
Enable Time
Disable Time
POWER SUPPLY INTERFACE
Supply Voltage
Quiescent Current
Supply Current
900 MHz
Dynamic Range
Deviation vs. Temperature
Logarithmic Slope
Logarithmic Intercept
Deviation from CW Response
1.9 GHz
Dynamic Range
Deviation vs. Temperature
Logarithmic Slope
Logarithmic Intercept
Deviation from CW Response
2.2 GHz
Dynamic Range
Deviation vs. Temperature
Logarithmic Slope
Logarithmic Intercept
Deviation from CW Response
AD8362
Test Conditions/Comments
Pin VTGT
Measurement range = 60 dB, to ±1 dB error
VTGT = 1.25 V
VTGT = 0 V
Pin PWDN
Logic low enables
Logic high disables
Logic high
Logic low
From PWDN low to VOUT within 10% of final value, CLPF = 1000 pF
From PWDN high to VOUT within 10% of final value, CLPF = 1000 pF
Pin VPOS
Min
0.625
46
−64
Error referred to best-fit line (linear regression)
±1.0 dB linearity, CW input
±0.5 dB linearity, CW input
Deviation from output at 25°C
−40°C < TA < +85°C, PIN = −45 dBm
−40°C < TA < +85°C, PIN = −20 dBm
−40°C < TA < +85°C, PIN = +5 dBm
5.5 dB peak-to-rms ratio (IS95 reverse link)
12.0 dB peak-to-rms ratio (W-CDMA 4 channels)
18.0 dB peak-to-rms ratio (W-CDMA 15 channels)
Rev. F | Page 5 of 33
2.5
V
μA
μA
kΩ
1
V
V
μA
μA
ns
μs
5.5
22
V
mA
mA
230
5
14.5
2.5
Error referred to best-fit line (linear regression)
±1.0 dB linearity, CW input
±0.5 dB linearity, CW input
Deviation from output at 25°C
−40°C < TA < +85°C, PIN = −45 dBm
−40°C < TA < +85°C, PIN = −20 dBm
−40°C < TA < +85°C, PIN = +5 dBm
5.5 dB peak-to-rms ratio (IS95 reverse link)
12.0 dB peak-to-rms ratio (W-CDMA 4 channels)
18.0 dB peak-to-rms ratio (W-CDMA 15 channels)
Unit
3
When disabled
Error referred to best-fit line (linear regression)
±1 dB linearity, CW input
±0.5 dB linearity, CW input
Deviation from output at 25°C
−40°C < TA < +85°C, PIN = −45 dBm
−40°C < TA < +85°C, PIN = −20 dBm
−40°C < TA < +85°C, PIN = +5 dBm
Max
−28
−52
52
4.5
5.5 dB peak-to-rms ratio (IS95 reverse link)
12.0 dB peak-to-rms ratio (W-CDMA 4 channels)
18.0 dB peak-to-rms ratio (W-CDMA 15 channels)
Typ
5
20
0.2
65
62
dB
dB
−1.7
−1.4
−1.0
50
−60
0.2
0.2
0.5
dB
dB
dB
mV/dB
dBm
dB
dB
dB
54
−56
65
62
dB
dB
−0.6
−0.5
−0.3
51
−59
0.2
0.2
0.5
dB
dB
dB
mV/dB
dBm
dB
dB
dB
65
65
dB
dB
−1.8
−1.6
−1.3
50.5
−61
0.2
0.2
0.5
dB
dB
dB
mV/dB
dBm
dB
dB
dB
AD8362
Parameter
2.7 GHz
Dynamic Range
Deviation vs. Temperature
Logarithmic Slope
Logarithmic Intercept
Deviation from CW Response
3.65 GHz
Dynamic Range
Deviation vs. Temperature
Data Sheet
Test Conditions/Comments
Error referred to best-fit line (linear regression)
±1.0 dB linearity, CW input
±0.5 dB linearity, CW input
Deviation from output at 25°C
−40°C < TA < +85°C, PIN = −40 dBm
−40°C < TA < +85°C, PIN = −15 dBm
−40°C < TA < +85°C, PIN = +5 dBm
5.5 dB peak-to-rms ratio (IS95 reverse link)
12.0 dB peak-to-rms ratio (W-CDMA 4 channels)
18.0 dB peak-to-rms ratio (W-CDMA 15 channels)
Single-ended drive3
Error referred to best-fit line (linear regression)
±1.0 dB linearity, CW input
±0.5 dB linearity, CW input
Deviation from output at 25°C
−40°C < TA < +85°C, PIN = −35 dBm
−40°C < TA < +85°C, PIN = −15 dBm
−40°C < TA < +85°C, PIN = +10 dBm
Logarithmic Slope
Logarithmic Intercept
1
1:4 balun transformer, M/A-COM ETC 1.6-4-2-3.
See Figure 48.
3
See Figure 50.
4
The limitation of the high end of the power range is due to the test equipment not the device under test.
2
Rev. F | Page 6 of 33
Min
Typ
Max
Unit
63
62
dB
dB
−5.3
−5.5
−4.8
50.5
−58
0.2
0.2
0.4
dB
dB
dB
mV/dB
dBm
dB
dB
dB
51
50
dB
dB
−3
−3.5
−3.5
51.7
−45
dB
dB
dB
mV/dB
dBm
Data Sheet
AD8362
ABSOLUTE MAXIMUM RATINGS
Table 3.
Parameter
Supply Voltage VPOS
Input Power (Into Input of Device)
Equivalent Voltage
Internal Power Dissipation
θJA
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Lead Temperature (Soldering, 60 sec)
Rating
5.5 V
15 dBm
2 V rms
500 mW
125°C/W
125°C
−40°C to +85°C
−65°C to +150°C
300°C
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
ESD CAUTION
Rev. F | Page 7 of 33
AD8362
Data Sheet
COMM
1
16
ACOM
CHPF
2
15
VREF
DECL
3
AD8362
14
VTGT
INHI
4
13
VPOS
INLO
TOP VIEW
(Not to Scale)
5
12
VOUT
DECL
6
11
VSET
PWDN
7
10
ACOM
COMM
8
9
CLPF
02923-002
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 2. Pin Configuration
Table 4. Pin Function Descriptions
Pin
No.
1, 8
2
3, 6
Mnemonic
COMM
CHPF
DECL
4, 5
INHI, INLO
7
9
10, 16
11
PWDN
CLPF
ACOM
VSET
12
13
14
VOUT
VPOS
VTGT
15
VREF
Description
Common Connection. Connect via low impedance to system common.
Input HPF. Connect to common via a capacitor to determine 3 dB point of input signal high-pass filter.
Decoupling Terminals for INHI and INLO. Connect to common via a large capacitance to complete
input circuit.
Differential Signal Input Terminals. Input Impedance = 200 Ω. Can also be driven single-ended, in
which case, the input impedance reduces to 100 Ω.
Disable/Enable Control Input. Apply logic high voltage to shut down the AD8362.
Connection for Ground Referenced Loop Filter Integration (Averaging) Capacitor.
Analog Common Connection for Output Amplifier.
Setpoint Input. Connect directly to VOUT for measurement mode. Apply setpoint input to this pin for
controller mode.
RMS Output. In measurement mode, VOUT is normally connected directly to VSET.
Connect to 5 V Power Supply.
The logarithmic intercept voltage is proportional to the voltage applied to this pin. The use of a lower
target voltage increases the crest factor capacity. Normally connected to VREF.
General-Purpose Reference Voltage Output of 1.25 V. Usually connected only to VTGT.
Rev. F | Page 8 of 33
Equivalent
Circuit
Circuit A
Circuit B
Circuit C
Circuit D
Circuit E
Data Sheet
AD8362
EQUIVALENT CIRCUITS
VPOS
VPOS
DECL
50kΩ
COMM
VTGT
50kΩ
ACOM
VTGT
INTERFACE
GAIN = 0.12
100Ω
02923-005
INHI
COMM
VGA
Figure 5. Circuit C
RAIL-TO-RAIL
OUTPUT
INLO
0.7V
VPOS
02923-003
COMM
COMM
Figure 6. Circuit D
Figure 3. Circuit A
VPOS
SOURCE ONLY
REF BUF
~35kΩ
VSET
~35kΩ
ACOM
500Ω
~0.35V
VSET
INTERFACE
VPOS
VOUT
13kΩ
ACOM
5kΩ
02923-004
ACOM
COMM
COMM
Figure 7. Circuit E
Figure 4. Circuit B
Rev. F | Page 9 of 33
02923-007
DECL
VOUT
2kΩ
CLPF
VPOS
02923-006
100Ω
AD8362
Data Sheet
100MHz
2200MHz
VOUT (V)
2.0
2700MHz
1.5
5
10 15
2.5
VOUT (V)
ERROR IN VOUT (dB)
100MHz
0.5
0
2200MHz
2.4
1.8
–40°C
0
2.0
1.6
10 15
3.0
3.6
2.4
3.2
1.8
–40°C
1.2
0.6
+25°C
0
+85°C
–0.6
–40°C
–1.2
+85°C
INPUT AMPLITUDE (dBm)
0
5
–3.0
10 15
2.5
2.0
1.5
0
–3.0
10 15
5
IS95 REVERSE LINK
W-CDMA 8-CHANNEL
W-CDMA 15-CHANNEL
1.0
0.5
0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
02923-010
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
–2.4
CW
3.0
–2.4
+25°C
–1.8
+85°C
3.5
–1.8
0.8
+25°C
4.0
ERROR IN VOUT (dB)
4.0
–1.2
+85°C
–40°C
Figure 12. VOUT and Law Conformance vs. Input Amplitude,
Frequency 2200 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C
VOUT (V)
5
–0.6
+25°C
0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
02923-009
0
1.2
0.6
2.4
2700MHz
Figure 9. Logarithmic Law Conformance vs. Input Amplitude,
Frequencies: 100 MHz, 900 MHz, 1900 MHz, 2200 MHz, and 2700 MHz;
Sine Wave, Differential Drive
VOUT (V)
3.0
0.4
INPUT AMPLITUDE (dBm)
0
–3.0
10 15
5
3.6
0.8
900MHz
–3.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0.4
0
4.0
1.2
1900MHz
–2.0
1.2
–1.8
–2.4
2.8
1.5
1.6
–1.2
+25°C
+85°C
3.2
2.0
2.0
–0.6
+85°C
Figure 11. VOUT and Law Conformance vs. Input Amplitude,
Frequency 1900 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C
3.0
2.4
+25°C
–40°C
0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
Figure 8. Output Voltage (VOUT) vs. Input Amplitude (dBm),
Frequencies: 100 MHz, 900 MHz, 1900 MHz, 2200 MHz, and 2700 MHz;
Sine Wave, Differential Drive
2.8
0
1.6
02923-008
0
INPUT AMPLITUDE (dBm)
–2.5
0.6
2.0
0.4
1900MHz
0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
–1.5
2.4
0.8
900MHz
0.5
1.2
–40°C
1.2
1.0
–1.0
1.8
ERROR IN VOUT (dB)
VOUT (V)
2.5
–0.5
2.4
3.2
2.8
3.0
1.0
3.6
02923-012
3.5
3.0
Figure 10. VOUT and Law Conformance vs. Input Amplitude,
Frequency 900 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C
0
5
10 15
02923-013
4.0
4.0
02923-011
4.5
ERROR IN VOUT (dB)
TYPICAL PERFORMANCE CHARACTERISTICS
Figure 13. VOUT vs. Input Amplitude with Different Waveforms, CW, IS95
Reverse Link, W-CDMA 8-Channel, W-CDMA 15-Channel, Frequency 900 MHz
Rev. F | Page 10 of 33
Data Sheet
AD8362
4.0
3.0
2.5
3.5
3.0
1.5
1.0
IS95 REVERSE LINK
CW
0
–0.5
W-CDMA 15-CHANNEL
–1.5
1.0
–2.0
0
10 15
5
0
02923-014
3.0
2.5
2.5
2.0
2.0
W-CDMA
8-CHANNEL
ERROR IN VOUT (dB)
0.5
W-CDMA
4-CHANNEL
CW
0
–0.5
–1.0
–1.5
W-CDMA 15-CHANNEL
0.5
0
–0.5
–1.0
–1.5
–2.5
–2.5
5
10
+25°C
+85°C
–3.0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
02923-015
0
Figure 15. Output Error from CW Linear Reference vs. Input Amplitude
with Different W-CDMA Channel Loading, 4-Channel, 8-Channel,
15-Channel, Frequency 2200 MHz, VTGT = 1.25 V
0
5
10
Figure 18. Logarithmic Law Conformance vs. Input Amplitude,
3 Sigma to Either Side of Mean, Sine Wave, Frequency 900 MHz,
Temperatures: −40°C, +25°C, and +85°C
3.0
4.0
2.5
3.5
–45°C
2.0
ERROR IN VOUT (dB)
3.0
2.5
2.0
1.5
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
1.0
–2.0
0.5
–2.5
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
0
5
10
+85°C
–3.0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
02923-016
0
–40°C
1.0
–2.0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
10
1.5
–2.0
–3.0
5
Figure 17. VOUT vs. Input Amplitude, 3 Sigma to Either Side of Mean,
Sine Wave, Frequency 1900 MHz, Part-to-Part Variation
3.0
1.0
0
INPUT AMPLITUDE (dBm)
Figure 14. Output Error from CW Linear Reference vs. Input Amplitude
with Different Waveforms, CW, IS95 Reverse Link, W-CDMA 8-Channel,
W-CDMA 15-Channel, Frequency 900 MHz, VTGT = 1.25 V
1.5
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
02923-018
–3.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
02923-017
0.5
–2.5
ERROR IN VOUT (dB)
2.0
1.5
–1.0
VOUT (V)
2.5
Figure 16. VOUT vs. Input Amplitude, 3 Sigma to Either Side of Mean,
Sine Wave, Frequency 900 MHz, Part-to-Part Variation
Rev. F | Page 11 of 33
+25°C
0
5
10
Figure 19. Logarithmic Law Conformance vs. Input Amplitude,
3 Sigma to Either Side of Mean, Sine Wave, Frequency 1900 MHz,
Temperatures: −40°C, +25°C, and +85°C
02923-019
0.5
W-CDMA 8-CHANNEL
VOUT (V)
ERROR IN VOUT (dB)
2.0
AD8362
Data Sheet
–40°C
3.5
1.5
1.0
0.5
VOUT (V)
0
–0.5
–1.0
–1.5
–2.0
2
2.0
0
1.5
–2
1.0
–4
0.5
–6
0
–60
–50
–40
–30
–20
–10
0
10
–8
20
INPUT AMPLITUDE (dBm)
Figure 23. VOUT and Law Conformance vs. Input Amplitude for 15 Devices,
Frequency 2800 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C,
No Temperature Compensation, Single-Ended Drive, See Figure 50
6
3.5
8
+85°C
+25°C
–40°C 6
3.0
4
3.0
4
2.5
2
2.5
2
2.0
0
2.0
0
1.5
–2
1.5
–2
1.0
–4
1.0
–4
0.5
–6
0.5
–6
–50
–40
–30
–20
–10
0
10
–8
20
VOUT (V)
0
–60
02923-021
0
–60
ERROR (dB)
4.0
+85°C
+25°C
–40°C
INPUT AMPLITUDE (dBm)
Figure 21. VOUT and Law Conformance vs. Input Amplitude for 15 Devices,
Frequency 2350 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C,
No Temperature Compensation, Single-Ended Drive, See Figure 50
–50
–40
–30
–20
–10
0
10
–8
20
INPUT AMPLITUDE (dBm)
Figure 24. VOUT and Law Conformance vs. Input Amplitude for 15 Devices,
Frequency 3450 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C,
No Temperature Compensation, Single-Ended Drive, See Figure 50
4.0
8
6
3.5
6
3.0
4
3.0
4
2.5
2
2.5
2
2.0
0
2.0
0
1.5
–2
1.5
–2
1.0
–4
1.0
–4
0.5
–6
0.5
–6
0
–60
–50
–40
–30
–20
–10
INPUT AMPLITUDE (dBm)
0
10
–8
20
VOUT (V)
0
–60
02923-022
3.5
+85°C
+25°C
–40°C
ERROR (dB)
8
4.0
02923-023
10
Figure 22. VOUT and Law Conformance vs. Input Amplitude for 15 Devices,
Frequency 2600 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C,
No Temperature Compensation, Single-Ended Drive, See Figure 50
ERROR (dB)
5
8
3.5
VOUT (V)
2.5
02923-024
0
Figure 20. Logarithmic Law Conformance vs. Input Amplitude,
3 Sigma to Either Side of Mean, Sine Wave, Frequency 2200 MHz,
Temperatures: −40°C, +25°C, and +85°C
VOUT (V)
4
+25°C
–3.0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
4.0
3.0
–50
–40
–30
–20
–10
INPUT AMPLITUDE (dBm)
0
10
–8
20
ERROR (dB)
+85°C
02923-020
–2.5
6
02923-025
ERROR IN VOUT (dB)
2.0
8
+85°C
+25°C
–40°C
ERROR (dB)
4.0
3.0
2.5
Figure 25. VOUT and Law Conformance vs. Input Amplitude for 15 Devices,
Frequency 3650 MHz, Sine Wave, Temperatures: −40°C, +25°C, and +85°C,
No Temperature Compensation, Single-Ended Drive, See Figure 50
Rev. F | Page 12 of 33
Data Sheet
AD8362
2.0
51.5
+85°C
51.0
+25°C
50.5
–40°C
1.5
CHANGE IN INTERCEPT (dB)
50.0
49.5
900MHz
0
–0.5
–1.0
2700
FREQUENCY (MHz)
2200MHz
–2.0
–40 –30 –20 –10
02923-026
2600
2500
2400
2300
2200
2100
2000
1900
1800
1700
1600
1500
1400
1300
1200
1100
900
1900MHz
0.5
–1.5
1000
49.0
1.0
0
10
20
30
40
50
60
70
80
90
TEMPERATURE (°C)
02923-029
SLOPE (mV)
52.0
Figure 29. Change in Logarithmic Intercept vs. Temperature, 3 Sigma to
Either Side of Mean, Frequencies: 900 MHz, 1900 MHz, and 2200 MHz
Figure 26. Logarithmic Slope vs. Frequency,
Temperatures: −40°C, +25°C, and +85°C
–53
–54
100
–55
+85°C
80
–57
HITS
INTERCEPT (dBm)
–56
+25°C
–58
60
–59
40
–40°C
–60
–61
20
3.0
52
53
80
2.5
900MHz
70
2.0
1.5
60
1900MHz
50
HITS
0.5
0
40
–0.5
30
–1.0
–1.5
–2.5
20
2200MHz
–3.0
–40 –30 –20 –10
0
10 20 30 40 50
TEMPERATURE (°C)
10
60
70
80
90
0
–61.0
Figure 28. Change in Logarithmic Slope vs. Temperature, 3 Sigma to Either
Side of Mean, Frequencies: 900 MHz, 1900 MHz, and 2200 MHz
Rev. F | Page 13 of 33
–60.5
–60.0
–59.5
–59.0
INTERCEPT (dBm)
–58.5
–58.0
02923-031
–2.0
02923-028
CHANGE IN SLOPE (mV)
50
51
SLOPE (mV/dB)
Figure 30. Slope Distribution, Frequency 900 MHz
Figure 27. Logarithmic Intercept vs. Frequency,
Temperatures: −40°C, +25°C, and +85°C
1.0
49
02923-030
FREQUENCY (MHz)
0
48
02923-027
2700
2600
2500
2400
2300
2200
2100
2000
1900
1800
1700
1600
1500
1400
1300
1200
1100
900
–63
1000
–62
Figure 31. Logarithmic Intercept Distribution, Frequency 900 MHz
AD8362
Data Sheet
–4
2.0
–6
–30dBm
1.5
–8
0.5V/DIV
1.0
0
2
4
6
8
12
10
TIME (µs)
14
16
18
–10
–12
–14
20
0
3.0
VOUT
2.0
1.0
0
0
2
4
6
8
10
12
TIME (ms)
14
16
18
0
4.0
0
3.5
–2
3.0
–4
1.5
1.0
–14
20
0
2V/DIV
VPOS
2.5
–12
–6
+2dBm
–10dBm
–20dBm
–30dBm
1V/DIV
4
–8
–10
–12
0
2
4
6
8
12
10
TIME (ms)
14
16
18
–14
20
Figure 36. Output Response to Gating on Power Supply for Various RF Input
Levels, Carrier Frequency 900 MHz, CLPF = 0
2V/DIV
3.0
–2
–10dBm
2.5
–4
–20dBm
–6
–30dBm
0.5V/DIV
2
0
+2dBm
VOUT
4
–8
1.0
–10
0.5
–12
0
2
4
6
8
12
10
TIME (µs)
14
16
18
–14
20
POWER-DOWN PIN (V)
3.5
VOUT (V)
6
2.0
100MHz
3GHz
02923-034
POWERDOWN
4.0
PIN
0
–14
20
6
4.5
1.5
18
16
2
Figure 33. Output Response to RF Burst Input for Various RF Input Levels,
Carrier Frequency 900 MHz, CLPF = 0.1 µF
2.0
14
4.5
–10
5.0
12
10
TIME (ms)
5.0
–8
0.5V/DIV
0.5
8
2
–6
–30dBm
1.5
6
4
–4
–20dBm
4
5.5
–2
–10dBm
2.5
2
6
VOUT (V)
VOUT (V)
+2dBm
0
Figure 35. Output Response Using Power-Down Mode for Various RF Input
Levels, Carrier Frequency 900 MHz, CLPF = 0.1 µF
RF BURST ENABLE (V)
2V/DIV
3.5
–8
0.5V/DIV
0.5
02923-033
RF BURST
ENABLE
–6
–30dBm
1.0
5.0
4.0
2.0
–12
Figure 32. Output Response to RF Burst Input for Various
RF Input Levels, Carrier Frequency 900 MHz, CLPF = Open
4.5
–4
–20dBm
–10
0.5
0
–2
–10dBm
2.5
1.5
0
+2dBm
POWER-DOWN PIN (V)
–10dBm
–20dBm
3.5
3.0
4
2
02923-035
–2
2V/DIV
POWER-DOWN PIN (V)
VOUT
2.5
4.0
0
+2dBm
3.0
4.5
2
Figure 34. Output Response Using Power-Down Mode for Various RF Input
Levels, Carrier Frequency 900 MHz, CLPF = 0
02923-037
VOUT (V)
3.5
4
6
02923-036
2V/DIV
VOUT (V)
RF BURST
ENABLE
4.0
5.0
02923-032
4.5
6
RF BURST ENABLE (V)
5.0
Figure 37. INHI, INLO Differential Input Impedance, 100 MHz to 3 GHz
Rev. F | Page 14 of 33
Data Sheet
AD8362
5
300
250
–5
200
–10
HITS
–15
100
–20
50
–40 –30 –20 –10
0
10
20
30
40
TEMPERATURE (°C)
50
60
70
80
90
02923-038
–25
–30
150
Figure 38. Change in VREF vs. Temperature, 3 Sigma to Either Side of Mean
Rev. F | Page 15 of 33
0
1.230
1.235 1.240
1.245 1.250
1.255 1.260
VREF (V)
Figure 39. VREF Distribution
1.265 1.270
02923-039
CHANGE IN VREF (mV)
0
AD8362
Data Sheet
CHARACTERIZATION SETUP
TEK TDS5104
SCOPE
The general hardware configuration used for most of the AD8362
characterization is shown in Figure 40. The signal source is a
Rohde & Schwarz SMIQ03B. A 1:4 balun transformer is used to
transform the single-ended RF signal to differential form. For
frequencies above 3.0 GHz, an Agilent 8521A signal source was
used. For the response measurements in Figure 32 and Figure 33,
the configuration shown in Figure 41 is used. For Figure 34 and
Figure 35, the configuration shown in Figure 42 is used. For
Figure 36, the configuration shown in Figure 43 is used.
AD8362
TEK P5050
VOLTAGE PROBE
C1
SMT03
SIGNAL
GENERATOR
BALUN
3dB
C2
C3
RF 50Ω
COMM
ACOM
CHPF
VREF
DECL
VTGT
INHI
VPOS
INLO
VOUT
DECL
VSET
PWDN
ACOM
COMM
CLPF
HPE3631A
POWER
SUPPLY
C4
AD8362
3dB RFIN
VOUT
MULTIMETER
HP34401A
Figure 41. Response Measurement Setup for Modulated Pulse
TEK TDS5104
SCOPE
02923-040
PC
CONTROLLER
AD8362
TEK P5050
VOLTAGE PROBE
C1
Figure 40. Primary Characterization Setup
SMT03
SIGNAL
GENERATOR
ANALYSIS
The slope and intercept are derived using the coefficients of
a linear regression performed on data collected in its central
operating range. Error is stated in two forms: error from the
linear response to the CW waveform and output delta from
25°C performance.
Error dB
VOUT Slope PIN PZ
Slope
ACOM
CHPF
VREF
DECL
VTGT
INHI
VPOS
INLO
VOUT
DECL
VSET
PWDN
ACOM
COMM
CLPF
BALUN
3dB
C2
RF 50Ω
C3
HP8112A
PULSE
GENERATOR
The error from linear response to the CW waveform is the
decibel difference in output from the ideal output defined by
the conversion gain and output reference. This is a measure of
the linearity of the device response to both CW and modulated
waveforms. The error in dB is calculated by
COMM
HPE3631A
POWER
SUPPLY
C4
02923-042
CHARACTERIZATION
BOARD
Figure 42. Response Measurement Setup for Power-Down Step
HP8112A
PULSE
GENERATOR
AD811
50Ω
732Ω
(1)
AD8362
where PZ is the x intercept, expressed in dBm.
Error from the linear response to the CW waveform is not a
measure of absolute accuracy because it is calculated using
the slope and intercept of each device. However, it verifies the
linearity and the effect of modulation on the device response.
Error from the 25°C performance uses the performance of a
given device and waveform type as the reference; it is predominantly a measurement of output variation with temperature.
C1
3dB
COMM
ACOM
CHPF
VREF
DECL
VTGT
INHI
VPOS
INLO
VOUT
DECL
VSET
PWDN
ACOM
COMM
CLPF
BALUN
C2
C3
SMT03
SIGNAL
GENERATOR
TEK TDS5104
SCOPE
TEK P5050
VOLTAGE
PROBE
0.01µF
100pF
C4
RF 50Ω
Figure 43. Response Measurement Setup for Gated Supply
Rev. F | Page 16 of 33
02923-043
SMIQ03B
RF SOURCE
02923-041
EQUIPMENT
Data Sheet
AD8362
CIRCUIT DESCRIPTION
The AD8362 is a fully calibrated, high accuracy, rms-to-dc
converter providing a measurement range of over 65 dB. It is
capable of operating from signals as low in frequency as a few
hertz to at least 3.8 GHz. Unlike earlier rms-to-dc converters,
the response bandwidth is completely independent of the signal
magnitude. The −3 dB point occurs at about 3.5 GHz. The
capacity of this device to accurately measure waveforms having a
high peak-to-rms ratio (crest factor) is independent of either
the signal frequency or its absolute magnitude, over a wide
range of conditions.
This unique combination allows the AD8362 to be used as a
calibrated RF wattmeter covering a power ratio of >1,000,000:1,
a power controller in closed-loop systems, a general-purpose
rms-responding voltmeter, and in many other low frequency
applications.
The device comprises the core elements of a high performance
AGC loop (see Figure 44), laser-trimmed during manufacturing
to close tolerances while fully operational at a test frequency of
100 MHz. Its linear, wideband VGA provides a general voltage
gain, GSET; this can be controlled in a precisely exponential (linearin-dB) manner over the full 68 dB range from −25 dB to +43 dB
by a voltage, VSET. However, to provide adequate guardbanding,
only the central 60 dB of this range, from −21 dB to +39 dB, is
normally used. The Adjusting VTGT to Accommodate Signals
with Very High Crest Factors section shows how this basic
range can be shifted up or down.
AMPLITUDE TARGET
FOR VSIG
–25dB TO +43dB
MATCH WIDEBAND SQUARERS
VTGT
INHI
VGA
X2
X2
ISQU
ITGT
VSIG
INLO
× 0.06
VATG
ACOM
OFFSET
NULLING
CF
OUTPUT
FILTER
VOUT
SETPOINT
INTERFACE
VSET
BAND GAP
REFERENCE
INTERNAL
RESISTORS
SET BUFFER
GAIN TO 5
where:
GO is a basic fixed gain.
VGNS is a scaling voltage that defines the gain slope (the dB
change per volt). Note that the gain decreases with VSET.
The VGA output is
VSIG = GSETVIN = GOVIN exp(−VSET/VGNS)
(3)
where VIN is the ac voltage applied to the input terminals
of the AD8362.
As explained in the Recommended Input Coupling section, the
input drive can either be single-sided or differential, although
dynamic range is maximized with a differential input drive. The
effect of high frequency imbalances when using a single-sided
drive is less apparent at low frequencies (from 50 Hz to 500 MHz),
but the peak input voltage capacity is always halved relative to
differential operation.
SQUARE LAW DETECTION
The output of the variable gain amplifier (VSIG) is applied to
a wideband square law detector, which provides a true rms
response to this alternating signal that is essentially independent
of waveform. Its output is a fluctuating current (ISQU) that has
a positive mean value. This current is integrated by an on-chip
capacitance (CF), which is usually augmented by an external
capacitance (CLPF) to extend the averaging time. The resulting
voltage is buffered by a gain of 5, dc-coupled amplifier whose
rail-to-rail output (VOUT) can be used for either measurement
or control purposes.
CLPF
EXTERNAL
ACOM
02923-044
1.25V
Figure 44. Basic Structure of the AD8362
(2)
The fluctuating current (ISQU) is balanced against a fixed
setpoint target current (ITGT) using current mode subtraction.
With the exact integration provided by the capacitor(s), the
AGC loop equilibrates when
CLPF
VREF
GSET = GO exp(−VSET/VGNS)
In most applications, the AGC loop is closed via the setpoint
interface pin, VSET, to which the VGA gain control voltage on
VOUT is applied. In measurement modes, the closure is direct
and local by a simple connection from the output of the VOUT
pin to the VSET pin. In controller modes, the feedback path is
around some larger system, but the operation is the same.
GSET
CHPF
The VGA gain has the form
MEAN(ISQU) = ITGT
(4)
The current, ITGT, is provided by a second-reference squaring
cell whose input is the amplitude-target voltage VATG. This is
a fraction of the voltage VTGT applied to a special interface,
which accepts this input at the VTGT pin. Because the two
squaring cells are electrically identical and are carefully implemented in the IC, process and temperature-dependent variations
in the detailed behavior of the two square-law functions cancel.
Rev. F | Page 17 of 33
AD8362
Data Sheet
(5)
In a formal solution, extract the square root of both sides to provide
an explicit value for the root-mean-square (rms) value. However, it
is apparent that by forcing this identity through varying the VGA
gain and extracting the mean value by the filter provided by the
capacitor(s), the system inherently establishes the relationship
rms(VSIG) = VATG
1.5
2.9
rms[GOVIN/VATG] = exp(VSET/VGNS)
VOUT (V)
(7)
As a measurement device, VIN is the unknown quantity and all
other parameters can be fixed by design. To solve Equation 7,
1.0
1.7
0.8
0.5
VSET = VGNS log[rms(VIN)/VZ]
0.2
(9)
0
2.0
1.1
(8)
0.5
2.3
1.4
therefore,
+25°C
–40°C
–0.5
+85°C
–1.0
–1.5
+25°C
–2.0
+85°C
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT AMPLITUDE (dBm)
The quantity VZ = VATG/GO is defined as the intercept voltage
because VSET must be 0 when rms (VIN) = VZ.
–2.5
0
5
–3.0
10 15
Figure 45. Output Voltage and Law Conformance Error
at TA = −40°C, +25°C, and +85°C
When connected as a measurement device, the output of the
buffer is tied directly to VSET, which closes the AGC loop.
Making the substitution VOUT = VSET and changing the
log base to 10, as needed in a decibel conversion,
(10)
where VSLP is the slope voltage, that is, the change in output
voltage for each decade of change in the input amplitude.
Note that VSLP = VGNS log (10) = 2.303 VGNS.
In the AD8362, VSLP is laser-trimmed to 1 V using a 100 MHz
test signal. Because a decade corresponds to 20 dB, this slope
can also be stated as 50 mV/dB. The Altering the Slope section
explains how the effective value of VSLP can be altered by the
user. The intercept, VZ, is also laser-trimmed to 224 µV (−60 dBm
relative to 50 Ω). In an ideal system, VOUT would cross zero
for an rms input of that value. In a single-supply realization of
the function, VOUT cannot run fully down to ground; here, VZ
is the extrapolated value.
VOLTAGE vs. POWER CALIBRATION
The AD8362 can be used as an accurate rms voltmeter from
arbitrarily low frequencies to microwave frequencies. For low
frequency operation, the input is usually specified either in
volts rms or in dBV (decibels relative to 1 V rms).
Figure 45 shows the output of the circuit of Figure 47 over the
full input range. The agreement with the ideal function (law
conformance) is also shown. This was determined by linear
regression on the data points over the central portion of the
transfer function for the +25°C data.
The error at −40°C, +25°C, and +85°C was then calculated by
subtracting the ideal output voltage at each input signal level
from the actual output and dividing this quantity by the mean
slope of the regression equation to provide a measurement of
the error in decibels (scaled on the right-hand axis of Figure 45).
The error curves generated in this way reveal not only the deviations from the ideal transfer function at a nominal temperature,
but also the additional errors caused by temperature changes.
Notice that there is a small temperature dependence in the
intercept (the vertical position of the error plots).
Figure 45 further reveals a periodic ripple in the conformance
curves. This is due to the interpolation technique used to select
the signals from the attenuator, not only at discrete tap points,
but also anywhere in between, thus providing continuous
attenuation values. The selected signal is then applied to the
3.5 GHz, 40 dB fixed gain amplifier in the remaining stages of
the VGA of the AD8362.
At high frequencies, signal levels are commonly specified in
power terms. In these circumstances, the source and termination impedances are an essential part of the overall scaling. For
this condition, the output voltage can be expressed as
VOUT = SLOPE × (PIN − PZ)
2.0
–40°C
3.2
2.6
rms[GOVIN exp(−VSET/VGNS)] = VATG
VOUT = VSLP log10[rms(VIN)/VZ]
2.5
3.5
(6)
Substituting the value of VSIG from Equation 3,
3.0
3.8
ERROR IN VOUT (dB)
MEAN(VSIG2) = VATG2
In practice, the response deviates slightly from the ideal straight
line suggested by Equation 11. This deviation is called the law
conformance error. In defining the performance of high accuracy
measurement devices, it is customary to provide plots of this
error. In general terms, it is computed by extracting the best
straight line to the measured data using linear regression over
a substantial region of the dynamic range and under clearly
specified conditions.
02923-045
Accordingly, VTGT (and its fractional part VATG) determines
the output that must be provided by the VGA for the AGC loop
to settle. Because the scaling parameters of the two squarers are
accurately matched, it follows that Equation 4 is satisfied only when
(11)
where PIN and the intercept PZ are expressed in dBm.
Rev. F | Page 18 of 33
Data Sheet
AD8362
An approximate schematic of the signal input section of the
AD8362 is shown in Figure 46. The ladder attenuator is composed of 11 sections (12 taps), each of which progressively
attenuates the input signal by 6.33 dB. Each tap is connected to
a variable transconductance cell whose bias current determines
the signal weighting given to that tap. The interpolator determines
which stages are active by generating a discrete set of bias currents,
each having a Gaussian profile. These are arranged to move
from left to right, thereby determining the attenuation applied
to the input signal as the gain is progressively lowered over the
69.3 dB range under control of the VSET input. The detailed
manner in which the transconductance of adjacent stages varies
as the virtual tap point slides along the attenuator accounts for
the ripple observed in the conformance curves. Its magnitude is
slightly temperature dependent and varies with frequency (see
Figure 10, Figure 11, and Figure 12). Notice that the system
responses to signal inputs at INHI and INLO are not completely
independent; these pins do not constitute a fully floating
differential input.
ATTENUATION
CONTROL
GAUSSIAN INTERPOLATOR
gm
gm
gm
gm
TO FIXED
GAIN STAGE
INHI
DECL
STAGE 1
6.33dB
STAGE 2
6.33dB
02923-046
INLO
STAGE 11
6.33dB
Figure 46. Simplified Input Circuit
OFFSET ELIMINATION
To address the small dc offsets that arise in the VGA, an offsetnulling loop is used. The high-pass corner frequency of this
loop is internally preset to 1 MHz, which is sufficiently low for
most high frequency applications.
When using the AD8362 in low frequency applications, the
corner frequency can be reduced as needed by the addition of a
capacitor from the CHPF pin to ground having a nominal value
of 200 μF/Hz.
For example, to lower the high-pass corner frequency to
150 Hz, a capacitance of 1.33 μF is required. The offset voltage
varies depending on the actual gain at which the VGA is
operating, and thus on the input signal amplitude.
Baseline variations of this sort are a common aspect of all
VGAs, but they are more evident in the AD8362 because of the
method of its implementation, which causes the offsets to ripple
along the gain axis with a period of 6.33 dB. When an excessively
large value of CHPF is used, the offset correction process can
lag the more rapid changes in the gain of the VGA, which in
turn can increase the time required for the loop to fully settle
for a given steady input amplitude.
TIME-DOMAIN RESPONSE OF THE CLOSED LOOP
The external low-pass averaging capacitance (CLPF) added at
the output of the squaring cell is chosen to provide adequate
filtering of the fluctuating detected signal. The optimum value
depends on the application; as a guideline, a value of roughly
900 μF/Hz should be used. For example, a capacitance of 5 μF
provides adequate filtering down to 180 Hz. Note that the
fluctuation in the quasi-dc output of a squaring cell operating
on a sine wave input is a raised cosine at twice the signal
frequency, easing this filtering function.
In the standard connections for the measurement mode, the
VSET pin is tied to VOUT. For small changes in input amplitude (a few decibels), the time-domain response of this loop
is essentially linear, with a 3 dB low-pass corner frequency of
nominally fLP = 1/(2π × CLPF × 1100). Internal time delays
around this local loop set the minimum recommended value
of this capacitor to about 300 pF, resulting in an fLP of
approximately 500 kHz.
When large and abrupt changes of input amplitude occur,
the loop response becomes nonlinear and exhibits slew rate
limitations.
Rev. F | Page 19 of 33
AD8362
Data Sheet
OPERATION IN RF MEASUREMENT MODE
The balun outputs must be ac-coupled to the input of the
AD8362. The balun used in this example (M/A-COM ETC
1.6-4-2-3) is specified for operation from 0.5 GHz to 2.5 GHz.
BASIC CONNECTIONS
Basic connections for operating the AD8362 in measurement
mode are shown in Figure 47. While the AD8362 requires a
single supply of nominally 5 V, its performance is essentially
unaffected by variations of up to ±10%.
The supply is connected to the VPOS pin using the decoupling
network also displayed in Figure 47. The capacitors used in this
network must provide a low impedance over the full frequency
range of the input and should be placed as close as possible to
the VPOS pin. Two different capacitors are used in parallel to
reduce the overall impedance because these have different resonant frequencies. The measurement accuracy is not critically
dependent on supply decoupling because the high frequency
signal path is confined to the relevant input pins. Lead lengths
from both DECL pins to ground and from INHI/INLO to the
input coupling capacitors should be as short as possible. All
COMM pins should also connect directly to the ground plane.
To place the device in measurement mode, connect VOUT to
VSET and connect VTGT directly to VREF.
If a center-tapped, flux-coupled transformer is used, connect
the center tap to the DECL pins, which are biased to the same
potential as the inputs (approximately 3.6 V).
At lower frequencies where impedance matching is not necessary, the AD8362 can be driven from a low impedance differential
source, remembering the inputs must be ac-coupled.
Choosing Input Coupling Capacitors
As noted, the inputs must be ac-coupled. The input coupling
capacitors combine with the 200 Ω input impedance to create
an input high pass corner frequency equal to
fHP = 1/(200 × π × CC)
Typically, fHP should be set to at least one tenth the lowest input
frequency of interest.
Single-Ended Input Drive
As previously noted, the input stages of the AD8362 are optimally
driven from a fully balanced source, which should be provided
wherever possible. In many cases, unbalanced sources can be
applied directly to one or the other of the two input pins. The
chief disadvantage of this driving method is a 10 dB to 15 dB
reduction in dynamic range at frequencies above 500 MHz.
The AD8362 is disabled by a logic high on the PWDN pin,
which can be directly grounded for continuous operation.
When enabled, the supply current is nominally 20 mA and
essentially independent of supply voltage and input signal
strength. When powered down by a logic low on PWDN,
the supply current is reduced to 230 μA.
RECOMMENDED INPUT COUPLING
The full dynamic range of the AD8362, particularly at very
high frequencies (above 500 MHz), is realized only when the
input is presented to it in differential (balanced) form. In Figure 47,
a transmission line balun is used at the input. Having a 1:4
impedance ratio (1:2 turns ratio), the 200 Ω differential input
resistance of the AD8362 becomes 50 Ω at the input to the balun.
Figure 48 illustrates one of many ways of coupling the signal
source to the AD8362. Because the input pins are biased to
about 3.6 V (for VS = 5 V), dc-blocking capacitors are required
when driving from a grounded source. For signal frequencies
>5 MHz, a value of 1 nF is adequate. While either INHI or
INLO can be used, INHI is chosen here.
AD8362
C10
1000pF
T1
ETC1.6-4-2-3
C4
C6 1nF
100pF
C5
100pF
C7
1nF
COMM
ACOM 16
2
CHPF
VREF 15
3
DECL
VTGT 14
4
INHI
VPOS 13
5
INLO
VOUT 12
6
DECL
VSET 11
7
PWDN
ACOM 10
8
COMM
CLPF 9
C3
0.1µF
1nF
1nF
1nF
1nF
C1
0.1µF
C2
1nF
VOUT
ACOM 16
2 CHPF
VREF 15
3 DECL
VTGT 14
4 INHI
VPOS 13
5 INLO
VOUT 12
6 DECL
VSET 11
7 PWDN
ACOM 10
8 COMM
CLPF 9
Figure 48. Input Coupling from a Single-Ended 50 Ω Source
02923-047
SIGNAL
INPUT
Z = 50Ω
1
0.01µF
RF INPUT
100Ω
VS
5V @ 24mA
C8
1000pF
AD8362
1 COMM
02923-048
DEVICE DISABLE
1:4 Z-RATIO
(12)
Figure 47. Basic Connections for RF Power Measurement
Rev. F | Page 20 of 33
Data Sheet
AD8362
3.5
1.5
1.0
2.5
2.0
2.25
1.5
2.00
1.0
1.75
0.5
1.50
0
1.25
–0.5
1.00
–1.0
0.75
–1.5
0.50
–2.0
0.25
–2.5
0.5
2.0
0
1.5
–0.5
1.0
–1.0
OPERATION AT LOW FREQUENCIES
0.5
–1.5
In conventional rms-to-dc converters based on junction techniques, the effective signal bandwidth is proportional to the
signal amplitude. In contrast, the 3.5 GHz VGA bandwidth in
the AD8362 is independent of its gain. Because this amplifier is
internally dc-coupled, the system is also used as a high accuracy
rms voltmeter at low frequencies, retaining its temperaturestable, decibel-scaled output (for example, in seismic, audio,
and sonar instrumentation).
0
–55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN (dBm)
0
5
ERROR (dB)
–2.0
10
Figure 49. Transfer Function at Various Frequencies when the
RF Input is Driven Single-Ended
RF INPUT
1nF
2.7nH
4.7nH
1nF
1nF
1nF
1 COMM
ACOM 16
2 CHPF
VREF 15
3 DECL
VTGT 14
4 INHI
VPOS 13
5 INLO
VOUT 12
6 DECL
VSET 11
7 PWDN
ACOM 10
8 COMM
CLPF 9
02923-050
AD8362
0.01µF
0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10
Figure 50. Input Matching for Operation at Frequencies ≥2.7 GHz
For operation at frequencies ≥2.7 GHz, some additional
components are required to match the AD8362 input to 50 Ω
(see Figure 50). As the operating frequency increases, there is
also corresponding shifting in the operating power range (see
Figure 51).
–5
0
5
10
15
–3.0
INPUT AMPLITUDE (dBm)
02923-051
2.5
Figure 51. Transfer Function at Various Frequencies ≥2.7 GHz when
the RF Input is Driven Single-Ended
02923-049
VOUT (V)
3.0
2.0
450MHz
1900MHz
2500MHz
900MHz
2140MHz
2.50
3.0
2.8GHz
3.45GHz
3.65GHz
ERROR (dB)
4.0
3.00
2.75
VOUT (V)
An external 100 Ω shunt resistor combines with the internal
100 Ω single-ended input impedance to provide a broadband
50 Ω match. The unused input (in this case, INLO) is ac-coupled
to ground. Figure 49 shows the transfer function of the AD8362
at various frequencies when the RF input is driven singleended. The results show that transfer function linearity at the
top end of the range is degraded by the single-ended drive.
While the AD8362 can be operated at arbitrarily low frequencies,
an ac-coupled input interface must be maintained. In such cases,
the input coupling capacitors should be large enough so that the
lowest frequency components of the signal to be included in the
measurement are minimally attenuated. For example, for a 3 dB
reduction at 1.5 kHz, capacitances of 1 µF are needed because the
input resistance is 100 Ω at each input pin (200 Ω differentially),
and the calculation is 1/(2π × 1.5 kΩ × 100) = 1 μF. In addition, to
lower the high-pass corner frequency of the VGA, a large capacitor must be connected between the CHPF pin and ground (see
the Choosing a Value for CHPF section).
More information on the operation of the AD8362 and other
RF power detectors at low frequency is available in AN-691
Application Note, Operation of RF Detector Products at Low
Frequency.
Rev. F | Page 21 of 33
AD8362
Data Sheet
CHPF = 1/(2π × 800 × fHP)
(13)
For operation at frequencies as low as 100 kHz, set fHP to
approximately 25 kHz (CHPF = 8 nF). For frequencies above
approximately 2 MHz, no external capacitance is required
because there is adequate internal capacitance on this node.
CHOOSING A VALUE FOR CLPF
Table 5 shows recommended values of CLPF for popular
modulation schemes. In each case, CLPF is increased until
residual output noise falls below 50 mV. A 10% to 90% step
response to an input step is also listed. Where the increased
response time is unacceptably high, CLPF must be reduced. If
the output of the AD8362 is sampled by an ADC, averaging in
the digital domain can further reduce the residual noise.
Figure 52 shows how residual ripple and rise/fall time vary with
filter capacitance when the AD8362 is driven by a single carrier
W-CDMA signal (Test Model 1-64) at 2140 MHz.
RESIDUAL RIPPLE (mV p-p)
In the standard connections for the measurement mode, the
VSET pin is tied to VOUT. For small changes in input amplitude such as a few decibels, the time-domain response of this
loop is essentially linear with a 3 dB low-pass corner frequency
of nominally fLP = 1/(2π × CLPF × 1100). Internal time delays
around this local loop set the minimum recommended value
of this capacitor to about 300 pF, making fLP = 500 kHz.
For operation at lower signal frequencies, or whenever the
averaging time needs to be longer, use
CLPF = 1/(2π × fLP × 1100)
(14)
When the input signal exhibits large crest factors, such as a
CDMA or W-CDMA signal, CLPF must be much larger than
might seem necessary.
180
170
160
150
140
130
120
110
100
90
80
70
60
50
40
30
20
10
0
RESIDUAL RIPPLE (mV p-p)
FALL TIME (ms)
RISE TIME (ms)
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
FILTER CAPACITANCE (µF)
0.8
0.9
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
1.0
RISE/FALL TIME (ms)
The 3.5 GHz VGA of the AD8362 includes an offset cancellation loop, which introduces a high-pass filter effect in its
transfer function. To measure the amplitude of the input signal
properly, the corner frequency (fHP) of this filter must be well
below that of the lowest input signal in the desired measurement
bandwidth frequency. The required value of the external
capacitor is given by
This is due to the presence of significant low frequency
components in the complex, pseudorandom modulation, which
generates fluctuations in the output of the AD8362. Increasing
CLPF also increases the step response of the AD8362 to a
change at its input.
02923-052
CHOOSING A VALUE FOR CHPF
Figure 52. Residual Ripple, Rise and Fall Time vs. Filter Capacitance,
Single Carrier W-CDMA Input Signal, Test Model 1-64
Table 5. Recommended CLPF Values for Various Modulation Schemes
Modulation Scheme/Standard
W-CDMA , Single-Carrier, Test Model 1-64
W-CDMA, 4-Carrier, Test Model 1-64
CDMA2000, Single-Carrier, 9CH Test Model
CDMA2000, 3-Carrier, 9CH Test Model
WiMAX 802.16 (64QAM, 256 Subcarriers, 10 MHz Bandwidth)
Crest Factor
12.0 dB
11.0 dB
9.1 dB
11.0 dB
14.0 dB
Rev. F | Page 22 of 33
CLPF
0.1 µF
0.1 µF
0.1 µF
0.1 µF
0.1 µF
Residual Ripple
28 mV p-p
20 mV p-p
38 mV p-p
29 mV p-p
30 mV p-p
Response Time (Rise/Fall)
10% to 90%
171 µs/1.57 ms
162 µs/1.55 ms
179 µs /1.55 ms
171 µs/1.55 ms
157 µs/1.47 ms
Data Sheet
AD8362
For example, if this pin is supplied from VREF via a simple
resistive attenuator of 1 kΩ:1 kΩ, the output required from the
VGA is halved to 37.5 mV rms. Under these conditions, the
effective headroom in the signal path that drives the squaring
cell is doubled. In principle, this doubles the peak crest factor
that can be handled by the system.
Figure 53 and Figure 54 show the effect of varying VTGT on
measurement accuracy when the AD8362 is swept with a series
of signals with different crest factors, varying from CW with a
crest factor of 3 dB, to a W-CDMA carrier (Test Model 1-64)
with a crest factor of 10.6 dB. The crest factors of each signal are
listed in the plots. In Figure 53, VTGT is set to its nominal value
of 1.25 V, while in Figure 54, it is reduced to 0.625 V.
2.0
1.5
–0.5
0
5
1.5
–0.5
1.0
ERROR QPSK 4dB CF
ERROR 256QAM 8.2dB CF
ERROR CW
ERROR 64QAM 7.7dB CF
ERROR WCDMA TM1-64 10.6dB CF
0.5
0
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN (dBm)
0
5
ERROR (dB)
0
–1.0
–1.5
–2.0
10
Figure 54. Transfer Function and Law Conformance for Signals with
Varying Crest Factors, VTGT = 0.625 V, CLPF = 0.1 μF
Reducing VTGT also reduces the intercept. More significant in
this case, however, is the behavior of the error curves. Note that
in Figure 54 all of the error curves sit on one another, while in
Figure 53, there is some vertical spreading. This suggests that
VTGT should be reduced in those applications where a wide
range of input crest factors is expected. As noted, VTGT can
also be increased above its nominal level of 1.25 V. While this
can be used to increase the intercept, it would have the undesirable effect of degrading measurement accuracy in situations
where the crest factor of the signal being measured varies
significantly.
–1.0
–1.5
–2.0
10
AD8362
Figure 53. Transfer Function and Law Conformance for Signals with
Varying Crest Factors, VTGT = 1.25 V
1
COMM
ACOM 16
2
CHPF
VREF 15
3
DECL
VTGT 14
4
INHI
VPOS 13
5
INLO
VOUT 12
6
DECL
VSET 11
7
PWDN
ACOM 10
8
COMM
CLPF 9
R1
VOUT
R2
02923-055
0
0
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN (dBm)
2.0
None of the changes in operating conditions discussed so far
affects the logarithmic slope (VSLP) in Equation 10. This can
readily be altered by controlling the fraction of VOUT that is
fed back to the setpoint interface at the VSET pin. When the full
signal from VOUT is applied to VSET, the slope assumes its
nominal value of 50 mV/dB. It can be increased by including a
voltage divider between these pins, as shown in Figure 55.
2.0
0.5
0.5
1.0
0.5
ERROR QPSK 4dB CF
ERROR 256QAM 8.2dB CF
ERROR CW
ERROR 64QAM 7.7dB CF
ERROR WCDMA TM1-64 10.6dB CF
1.0
ALTERING THE SLOPE
2.5
1.0
1.5
1.5
ERROR (dB)
VOUT (V)
3.0
CW
64QAM
WCDMA TM1-64
QPSK
256QAM
2.5
02923-053
3.5
VOUT CW
VOUT 64QAM
VOUT WCDMA TM1-64
VOUT QPSK
VOUT 256QAM
VOUT (V)
In the default scheme, the VREF of 1.25 V positions this target
to 0.06 × 1.25 V = 75 mV. In principle, however, VTGT can be
driven by voltages that are larger or smaller than 75 mV. This
technique can be used to move the intercept, which increases or
decreases the input sensitivity of the device, or to improve the
accuracy when measuring signals with large crest factors.
3.0
VOUT
VOUT
VOUT
VOUT
VOUT
02923-054
3.5
An external direct connection between VREF (1.25 V) and VTGT
sets up the internal target voltage, which is the rms voltage that
must be provided by the VGA to balance the AGC feedback loop.
4.0
2.0
4.0
ADJUSTING VTGT TO ACCOMMODATE SIGNALS
WITH VERY HIGH CREST FACTORS
Figure 55. External Network to Raise Slope
Rev. F | Page 23 of 33
AD8362
Data Sheet
Moderately low resistance values should be used to minimize
scaling errors due to the 70 kΩ input resistance at the VSET pin.
This resistor string also loads the output, and it eventually reduces
the load-driving capabilities if very low values are used. To
calculate the resistor values, use
3.5
2.5
where:
SD is the desired slope, expressed in mV/dB.
R2' is the value of R2 in parallel with 70 kΩ.
ERROR (dB –40°C)
0
2.0
1.5
ERROR (dB +25°C)
1.0
For example, using R1 = 1.65 kΩ and R2 = 1.69 kΩ (R2' =
1.649 kΩ), the nominal slope is increased to 100 mV/dB.
Note, however, that doubling the slope in this manner reduces
the maximum input signal to approximately −10 dBm because
of the limited swing of VOUT (4.9 V with a 5 V power supply).
0
–60
–50
–40
–20
–10
0
10
–2
Figure 56. Transfer Function and Linearity with Combined Ripple Reduction
and Temperature Compensation Circuits, Frequency = 2.14 GHz,
Single-Carrier W-CDMA, Test Model 1-64
Because of the reduced filter capacitor, the rms voltage appearing
at the output of the error amplifier now contains significant
peak-to-peak noise. While it is critical to feed this signal back to
the VGA gain control input with the noise intact, the rms voltage
going to the external measurement node can be filtered using a
simple filter to yield a largely noise-free rms voltage.
The transfer function ripple and intercept drift of the AD8362
can be reduced using two techniques detailed in Figure 57.
CLPF is reduced from its nominal value. For broadbandmodulated input signals, this results in increased noise at the
output that is fed back to the VSET pin.
The noise contained in this signal causes the gain of the VGA to
fluctuate around a central point, moving the wiper of the
Gaussian Interpolator back and forth on the R-2R ladder.
The circuit shown in Figure 57 also incorporates a temperature
sensor that compensates temperature drift of the intercept. Because
the temperature drift varies with frequency, the amount of
compensation required must also be varied using R1 and R2.
Because the gain-control voltage is constantly moving across at
least one of taps of the Gaussian Interpolator, the relationship
between the rms signal strength of the VGA output and the
VGA control voltage becomes independent of the VGA gain
control ripple (see Figure 56). The signal being applied to the
squaring cell is now lightly AM modulated. However, this does
not change the peak-to-average ratio of the signal.
These compensation techniques are discussed in more detail in
AN-653 Application Note, Improving Temperature, Stability, and
Linearity of High Dynamic Range RMS RF Power Detectors.
5V
1nF
0.1µF
VPOS
0.1µF
1
1kΩ
VOUT
AD83621
3
7
AD8031
VSET
1µF
VREF
6
VOUT_COMP
5
2
R1
4
VTGT
COMM
–30
PIN (dBm)
TEMPERATURE COMPENSATION AND REDUCTION
OF TRANSFER FUNCTION RIPPLE
5V
VOUT (+25°C)
VOUT (–40°C)
VOUT (+85°C)
0.5
–1
ERROR (dB +85°C)
02923-056
(15)
ERROR (dB)
1
3.0
VOUT (V)
R1 = R2' (SD/50 − 1)
2
4.0
ACOM
CLPF
440pF
R2
FREQUENCY (MHz)
900
1900
2200
5V
R1 (kΩ)
1.02
1
1
R2 (kΩ)
25.5
82.5
19.1
0.1µF
2
PINS
OMITTED FOR CLARITY.
1
VTEMP
TMP36F
5
Figure 57. Temperature Compensation and Reduction of Transfer Function Ripple
Rev. F | Page 24 of 33
02923-057
1ADDITIONAL
Data Sheet
AD8362
TEMPERATURE COMPENSATION AT VARIOUS WiMAX FREQUENCIES UP TO 3.8 GHz
Figure 59 through Figure 63 show these results. The compensation is simple and relies on the TMP36 precision temperature
sensor driving one side of the resistor divider as the AD8362
drives the other side. The output is at the junction of the two
resistors (see Figure 58). At 25°C, TMP36 has an output voltage
of 750 mV and a temperature coefficient of 10 mV/°C. As the
temperature increases, the voltage from the AD8362 drops and
the voltage from the TMP36 rises. R1 and R2 are chosen so the
voltage at the center of the resistor divider remains steady over
temperature. In practice, R2 is much larger than R1 so that the
output voltage from the circuit is close to the voltage of the VOUT
pin. The resistor ratio R2/R1 is determined by the temperature
drift of the AD8362 at the frequency of interest. To calculate the
values of R1 and R2, first calculate the drift at a particular input
level, −15 dBm in this case. To do this, calculate the average
drift over the temperature range from 25°C to 85°C. Using the
following equation, the average drift in dB/°C is obtained.
dB/°C =
dBError
ΔTemperature
Table 6 shows the resultant values for R2 and R1 for frequencies
ranging from 2350 MHz to 3650 MHz. Figure 59 through Figure 63
show the performance over temperature for the AD8362 with
temperature compensation at frequencies across the WiMAX
band. The compensation factor chosen optimizes temperature
drift in the 25°C to 85°C range. This can be altered depending
on the temperature requirements for the application.
Table 6. Recommended Resistor Values for Temperature
Compensation at Various Frequencies
Freq.
(MHz)
2350
2600
2800
3450
3650
Average
Drift at
−15 dBm
(mV/°C)
−1.7600
−2.2639
−2.5102
−2.7402
−2.9544
Slope
(mV/dB)
51
51.45
51.68
51.61
51.73
R1
(kΩ)
4.99
4.99
4.99
4.99
4.99
R2
(kΩ)
28
22.1
20
18.2
16.9
5V
0.1µF
2.7nH
4.7nH
(16)
In this example, the drift of the AD8362 from 25°C to 85°C is
−2.07 dB and the temperature delta is 60°C, which results in
−0.0345 dB/°C drift. This temperature drift in dB/°C is converted to mV/°C through multiplication by the logarithmic slope
(51 mV/dB at 2350 MHz). The result is −1.76 mV/°C. The
following equation calculates the values of R1 and R2:
10 mV/°C
R2
=
R1 AD8362 Drift(mV/°C)
Average
Drift at
−15 dBm
(dB/°C)
−0.0345
−0.0440
−0.0486
−0.0531
−0.0571
(17)
Rev. F | Page 25 of 33
1nF
1nF
AD8362
INHI
INLO
VTGT
VOUT
VSET
CLPF
VREF
VTEMP
R1
R2
2
1
TMP36F
5
0.1µF
VOUT
Figure 58. AD8362 with Temperature Compensation Circuit
02923-058
The AD8362 is ideally suited for measuring WiMAX type
signals because crest factor changes in the modulation scheme
have very little effect on the accuracy of the measurement.
However, at higher frequencies, the AD8362 drifts more over
temperature often making temperature compensation necessary.
Temperature compensation is possible because the part-to-part
variation over temperature is small, and temperature change
only causes a shift in the AD8362 intercept. Typically, users
choose to compensate for temperature changes digitally. However, temperature compensation is possible using an analog
temperature sensor. Because the drift of the output voltage is
due mainly to intercept shift, the whole transfer function tends
to drop with increasing temperature, while the slope remains
quite stable. This makes the temperature drift independent of
input level. Compensating the drift based on a particular
input level (for example, −15 dBm), holds up well over the
dynamic range.
3.5
3.0
4
3.0
4
2.5
2
2.5
2
2.0
0
2.0
0
1.5
–2
1.5
–2
1.0
–4
1.0
–4
0.5
–6
0.5
–6
–40
–30
–20
–10
0
10
–8
20
VOUT (V)
0
–60
INPUT AMPLITUDE (dBm)
4
2.5
2
2.0
0
1.5
–2
1.0
–4
0.5
–6
–20
–10
0
10
–8
20
INPUT AMPLITUDE (dBm)
Figure 60. AD8362 VOUT and Error with Linear Temperature
Compensation at 2600 MHz
3.0
4
2.5
2
2.0
0
1.5
–2
1.0
–4
0.5
–6
–50
–40
–30
–20
–10
0
10
–8
20
INPUT AMPLITUDE (dBm)
+125°C
+105°C
+85°C
+25°C
–40°C
–8
20
8
6
4
2
2.0
0
1.5
–2
1.0
–4
0.5
–6
–50
–40
–30
–20
–10
INPUT AMPLITUDE (dBm)
0
10
–8
20
Figure 63. AD8362 VOUT and Error with Linear Temperature Compensation
at 3650 MHz, Temperature Compensation is Optimized for 85°C
02923-061
3.5
8
+85°C
+25°C
–40°C 6
0
–60
10
2.5
0
–60
ERROR (dB)
VOUT (V)
4.0
0
3.0
02923-060
–30
–10
3.5
VOUT (V)
3.0
–40
–20
4.0
ERROR (dB)
VOUT (V)
3.5
–50
–30
Figure 62. AD8362 VOUT and Error with Linear Temperature
Compensation at 3450 MHz
8
+85°C
+25°C
–40°C 6
0
–60
–40
INPUT AMPLITUDE (dBm)
Figure 59. AD8362 VOUT and Error with Linear Temperature
Compensation at 2350 MHz
4.0
–50
ERROR (dB)
–50
02923-063
0
–60
ERROR (dB)
3.5
8
+85°C
+25°C
–40°C 6
02923-059
VOUT (V)
4.0
8
+85°C
+25°C
–40°C 6
4.0
ERROR (dB)
Data Sheet
02923-062
AD8362
Figure 61. AD8362 VOUT and Error with Linear Temperature
Compensation at 2800 MHz
Rev. F | Page 26 of 33
Data Sheet
AD8362
To operate in controller mode, the link between VSET and
VOUT is broken. A setpoint voltage is applied to the VSET
input, while VOUT is connected to the gain control terminal
of the VGA, and the AD8362 RF input is connected to the output of the VGA (generally using a directional coupler or power
splitter and some additional attenuation). Based on the defined
relationship between VOUT and the RF input signal when the
device is in measurement mode, the AD8362 adjusts the voltage
on VOUT (VOUT is now an error amplifier output) until the
level at the RF input corresponds to the applied VSET. For
example, in a closed loop system, if VSET is set to 3 V, VOUT
increases or decreases until the input signal is equal to 0 dBm.
This relationship follows directly from the measurement mode
transfer function (see Figure 10, Figure 11, and Figure 12).
Therefore, when the AD8362 operates in controller mode, there
is no defined relationship between VSET and VOUT. VOUT
settles to a value that results in balance between the input signal
levels appearing at INHI/INLO and VSET.
OPERATION IN CONTROLLER MODE
The AD8362 provides a controller mode feature at the VOUT
pin. Using VSET for the setpoint voltage, it is possible for the
AD8362 to control subsystems such as power amplifiers (PAs),
VGAs, or variable voltage attenuators (VVAs), which have
output power that decreases monotonically with respect to
their (increasing) gain control signal.
CONTROLLED SYSTEM
(OUTPUT POWER
DECREASES AS
VAPC INCREASES)
POUT
OUTPUT
VAPC
PIN
INPUT
OUTPUT CONTROL VOLTAGE
0.1V TO 4.9V
VS
AD8362
1:4 Z-RATIO
C8
1000pF
C4
C6
100pF 1nF
C10
1000pF
T1
ETC1.6-4-2-3
C5
100pF
C7
1nF
1 COMM
ACOM 16
2 CHPF
VREF 15
3 DECL
VTGT 14
4 INHI
VPOS 13
5 INLO
VOUT 12
6 DECL
VSET 11
7 PWDN
ACOM 10
8 COMM
CLPF 9
C1
0.1µF
C2
1nF
SETPOINT
VOLTAGE
INPUT
0V TO 3.5V
C3
(SEE TEXT)
Figure 64. Basic Connections for Controller Mode Operation
02923-064
ATTN
For this output power control loop to be stable, a groundreferenced capacitor must be connected to the CLPF pin.
This capacitor integrates the internal error current that is
present when the loop is not balanced.
Increasing VSET, which corresponds to demanding a higher
signal from the VGA, tends to decrease VOUT. The VGA or VVA
therefore must have a negative sense. In other words, increasing
the gain control voltage decreases gain. If this is not the case, an
op amp, configured as an inverter with suitable level shifting, can
be used to correct the sense of the VOUT signal.
Rev. F | Page 27 of 33
AD8362
Data Sheet
Due to the noise generated from the AD8330, the dynamic
range is limited to approximately 90 dB. This can be achieved
only when a band-pass filter is used at the operating frequency
between the AD8330 and AD8362.
RMS VOLTMETER WITH 90 dB DYNAMIC RANGE
The 65 dB range of the AD8362 can be extended by adding a
standalone VGA as a preamplifier whose gain control input is
derived directly from VOUT. This extends the dynamic range
by the gain control range of this second amplifier. When this
VGA also provides a linear-in-dB (exponential) gain control
function, the overall measurement remains linearly scaled in
decibels. The VGA gain must decrease with an increase in its
gain bias in the same way as the AD8362. Alternatively, an
inverting op amp with suitable level shifting can be used. It is
convenient to select a VGA needing only a single 5 V supply
and capable of generating a fully balanced differential output.
All of these conditions are met by the AD8330. Figure 66 shows
the schematic. Also, note that the AD8131 is used to convert a
single-ended input into the differential-ended input needed by
the AD8330. The AD8131 gain of 2 does create a dc offset on
the output of the AD8362, but this is removed by connecting
0.5 V to the VMAG on AD8330.
Figure 65 shows data results of the extended dynamic range at
70 MHz with error in VOUT.
Using the inverse gain mode (MODE pin low) of the AD8330,
its gain decreases on a slope of 30 mV/dB to a minimum value
of 3 dB for a gain voltage (VDBS) of 1.5 V. VDBS is 40% of the
output of the AD8362. Over the 3 V range from 0.5 V to 3.5 V,
the gain of the AD8330 varies by (0.4 × 3 V)/(30 mV/dB), or
40 dB. Combined with the 65 dB gain span of the AD8362, this
results in a 100 dB variation for a 3 V change in VOUT.
–73
–63
INPUT (dBV)
–53 –43 –33
–23
–13
–3
7
6
2.5
4
2.0
2
1.5
0
1.0
–2
0.5
–4
0
–90
–80
–70
–60
–50
–40 –30 –20
INPUT (dBm)
–10
0
10
–6
20
Figure 65. Output and Conformance for the AD8330/AD8362
Extended Dynamic Range Circuit
+5V
0.1µF
GAIN OF 2
0.1µF
AD8362
0.1µF
1
COMM
ACOM 16
2
CHPF
VREF 15
3
DECL
VTGT 14
OPHI
4
INHI
VPOS 13
INLO
OPLO
5
INLO
VOUT 12
MODE
CMOP
6
DECL
VSET 11
7
PWDN
ACOM 10
8
COMM
CLPF 9
0.1µF
10µF
0.01µF
ENBL
OFST
VPOS
VPS1
AD8131
8
INPUT
3
4
INHI
0.1µF
–5V
BAND-PASS
@ 70MHz
AD8330
1
29.9Ω
VPSO
0.1µF
2
49.9Ω
CNTR
6
5
0.1µF
0.01µF
VDBS
CMGN
COMM
VMAG
+0.5V
VOUT
0.1µF
10µF
2kΩ
2kΩ
Figure 66. RMS Voltmeter with 90 dB Dynamic Range
Rev. F | Page 28 of 33
02923-066
0.1µF
ERROR IN VOUT (dB)
–83
02923-065
OUTPUT (V)
–103 –93
3.0
Data Sheet
AD8362
AD8362 EVALUATION BOARD
The AD8362 evaluation board provides for a number of different operating modes and configurations, including many
described in this data sheet. The measurement mode is set up
by positioning SW2 as shown in Figure 67. The AD8362 can be
operated in controller mode by applying the setpoint voltage to
the VSET connector, and flipping SW2 to its alternate position.
with SW1 switched to its alternate position, LK1 removed, and
the external target voltage applied to the VTGT connector.
The internal voltage reference is used for the target voltage when
SW1 is in the position shown in Figure 67. This voltage may be
reduced optionally via a voltage divider implemented with
R4 and R5, with LK1 in place, and SW1 switched to its alternate
position. Alternatively, an external target voltage may be used
The AD8362 is powered up with SW3 in the position shown in
Figure 67 and connector PWDN open. The device can be powered
down by either connecting a logic high voltage to a connector,
PWDN, with SW3 in the position, or by switching SW3 to its
alternate position.
In measurement mode, the slope of the response at VOUT may
be increased by using a voltage divider implemented with resistors in Position R17 and Position R9, and with SW2 switched to
its alternate position.
R1
0Ω
AGND
VPOS
C1
0.1µF
C8
1000pF
C10
1000pF
R15
0Ω
AD8362
C7
1000pF
C6
100pF
T1
RFIN
R16
OPEN
C5
100pF
C4
1000pF
1
COMM
ACOM 16
2
CHPF
VREF 15
3
DECL
VTGT 14
4
INHI
VPOS 13
5
INLO
VOUT
6
DECL
VSET 11
7
PWDN
ACOM 10
8
COMM
SW3
PWDN
CLPF
R13
10kΩ
12
R4
0Ω
SW1
R6
0Ω
C3
0.1µF
SW2
R7
0Ω
VOUT
VSET
9
R10
0Ω
C9
OPEN
Figure 67. Evaluation Board Schematic
Rev. F | Page 29 of 33
LK1
VTGT
R8
R17
OPEN 0Ω
R9
10kΩ
VREF
R5
10kΩ
02923-067
R14
OPEN
C2
100pF
Data Sheet
02923-068
AD8362
02923-069
Figure 68. Component Side Metal of Evaluation Board
Figure 69. Component Side Silkscreen of Evaluation Board
Rev. F | Page 30 of 33
Data Sheet
AD8362
Table 7. Bill of Materials
Designator
T1
Description
Part Number
ETC 1.6-4-2-3
(M/A-COM)
C1
C2
C3, C9
C4, C7, C10
C5, C6
C8
DUT
LK1
R1, R6, R7, R8, R10, R15
R4, R5
R9, R17
R13
R16
SW1
SW2
SW3
Supply filtering/decoupling capacitor
Supply filtering/decoupling capacitor
Output low-pass filter capacitor
Input bias-point decoupling capacitors
Input signal coupling capacitors
Input high-pass filter capacitor
AD8362
Use to reduce VTGT or to externally apply a voltage to VTGT
Jumpers
Use to reduce VTGT or to externally apply a voltage to VTGT
Slope adjustment resistors (see the Altering the Slope section)
Power-up terminating resistor
Not installed
Use to reduce VTGT or to externally apply a voltage to VTGT
Measurement mode/controller mode selector
Power-down/power-up or external power-down selector
Rev. F | Page 31 of 33
Default Value
0.1 µF
100 pF
C3 = 0.1 µF, C9 = open
1000 pF
100 pF
1000 pF
AD8362ARU
LK1 = open
0Ω
R4 = 0 Ω, R5 = 10 kΩ
R9 = 10 kΩ, R17 = open
R13 = 10 kΩ
Open
SW1 connects VREF to VTGT
SW2 connects VSET to VOUT
SW3 connects PWDN to R13
AD8362
Data Sheet
OUTLINE DIMENSIONS
5.10
5.00
4.90
16
9
4.50
4.40
4.30
6.40
BSC
1
8
PIN 1
1.20
MAX
0.15
0.05
0.65
BSC
0.30
0.19
COPLANARITY
0.10
0.20
0.09
SEATING
PLANE
8°
0°
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153-AB
Figure 70. 16-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-16)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
AD8362ARU
AD8362ARUZ
AD8362ARUZ-REEL7
AD8362-EVALZ
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
16-Lead Thin Shrink Small Outline Package [TSSOP], Tube
16-Lead Thin Shrink Small Outline Package [TSSOP], Tube
16-Lead Thin Shrink Small Outline Package [TSSOP], 7" Tape and Reel
Evaluation Board
Z = RoHS Compliant Part.
Rev. F | Page 32 of 33
Package Option
RU-16
RU-16
RU-16
Data Sheet
AD8362
NOTES
©2003–2015 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D02923-0-9/15(F)
Rev. F | Page 33 of 33
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