ADP1110AR

a Micropower, Step-Up/Step-Down Switching Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V ADP1110 FUNCTIONAL BLOCK DIAGRAMS SET FEATURES Operates at Supply Voltages From 1.0 V to 30 V Step-Up or Step-Down Mode Minimal External Components Required Low-Battery Detector User-Adjustable Current Limiting Fixed or Adjustable Output Voltage Versions 8-Pin DIP or SO-8 Package APPLICATIONS Cellular Telephones Single-Cell to 5 V Converters Laptop and Palmtop Computers Pagers Cameras Battery Backup Supplies Portable Instruments Laser Diode Drivers Hand-Held Inventory Computers ADP1110 A2 VIN GAIN BLOCK/ ERROR AMP ILIM SW1 220mV REFERENCE A0 A1 OSCILLATOR DRIVER Q1 COMPARATOR R1 R2 300kΩ GND SENSE SW2 ADP1110 Block Diagram—Fixed Output Version SET ADP1110 A2 VIN GAIN BLOCK/ ERROR AMP A0 ILIM SW1 220mV REFERENCE OSCILLATOR DRIVER GENERAL DESCRIPTION The ADP1110 is part of a family of step-up/step-down switching regulators that operate from an input voltage supply as little as 1.0 V. This very low input voltage allows the ADP1110 to be used in applications that use a single cell as the primary power source. The ADP1110 can be configured to operate in either step-up or step-down mode, but for input voltages greater than 3 V, the ADP1111 would be a more effective solution. An auxiliary gain amplifier can serve as a low battery detector or as a linear regulator. The quiescent current of 300 µA makes the ADP1110 useful in remote or battery powered applications. A1 Q1 COMPARATOR GND FB SW2 ADP1110 Block Diagram—Adjustable Output Version The 70 kHz frequency operation also allows for the use of surface-mount external capacitors and inductors. Battery protection circuitry limits the effect of reverse current to safe levels at reverse voltages up to 1.6 V. R EV. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 World Wide Web Site: http://www.analog.com Fax: 617/326-8703 © Analog Devices, Inc., 1996 ADP1110–SPECIFICATIONS (0 C to +70 C, V = 1.5 V unless otherwise noted) IN Parameter QUIESCENT CURRENT INPUT VOLTAGE COMPARATOR TRIP POINT VOLTAGE OUTPUT SENSE VOLTAGE Conditions Switch Off Step-Up Mode Step-Down Mode ADP11101 ADP1110-3.3 ADP1110-52 ADP1110-122 ADP1110 ADP1110-3.3 ADP1110-5 ADP1110-12 2 VS IQ VIN Min Typ 300 Max Units µA 1.15 210 220 3.30 5.00 12.00 4 66 90 200 12.6 30 230 3.47 5.25 12.6 8 130 180 400 90 78 12.5 240 500 0.4 V V mV V V V mV mV mV mV kHz % µs nA nA V %/V %/V mV mV mV mV mV V/V mA %/°C VOUT 3.13 4.75 11.4 COMPARATOR HYSTERESIS OUTPUT HYSTERESIS OSCILLATOR FREQUENCY DUTY CYCLE SWITCH ON TIME FEEDBACK PIN BIAS CURRENT SET PIN BIAS CURRENT A0 OUTPUT LOW REFERENCE LINE REGULATION SWITCH SATURATION VOLTAGE STEP-UP MODE ADP1110 VFB = 0 V VSET = VREF IAO = 300 µA VSET = 150 mV 1.0 V ≤ VIN ≤ 1.5 V 1.5 V ≤ VIN ≤ 12 V VIN = 1.5 V, ISW = 400 mA, +25°C TMIN to TMAX VIN = 1.5 V, ISW = 500 mA, +25°C TMIN to TMAX VIN = 5 V, ISW = 1 A, +25°C RL = 100 kΩ3 TA = +25°C 4 fOSC Full Load (VFB < VREF) DC tON IFB ISET VAO 52 62 7.5 70 69 10 150 300 0.15 0.35 0.05 0.1 500 600 650 750 1000 VCESAT 300 400 700 A2 ERROR AMP GAIN REVERSE BATTERY CURRENT CURRENT LIMIT TEMPERATURE COEFFICIENT SWITCH OFF LEAKAGE CURRENT MAXIMUM EXCURSION BELOW GND AV IREV 1000 5000 750 –0.3 VIN, TA = +25°C Measured at SW1 Pin, TA = +25°C ISW1 ≤ 10 µA, Switch Off TA = +25°C ILEAK VSW2 1 –400 10 –350 µA mV NOTES 1 This specification guarantees that both the high and low trip point of the comparator fall within the 210 mV to 230 mV range. 2 This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis. 3 100 kΩ resistor connected between a 5 V source and the AO pin. 4 The ADP1110 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while V IN, ILIM, and SW1 pins are grounded. 5 All limits at temperature extremes are guaranteed via correlation using standard statistical quality control methods. Specifications subject to change without notice. – 2– REV. 0 ADP1110 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATIONS 8-Lead Plastic DIP (N-8) ILIM 1 VIN 2 8 FB (SENSE)* Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . –0.5 V to VIN Feedback Pin Voltage (ADP1110) . . . . . . . . . . . . . . . . . . 5.5 V Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW Operating Temperature Range . . . . . . . . . . . . . 0°C to +70°C Storage Temperature Range . . . . . . . . . . . . . –65°C to 150°C Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C TYPICAL APPLICATION 47µH 5V 8-Lead SOIC (SO-8) ILIM VIN 1 2 8 FB (SENSE)* ADP1110 7 SET A0 ADP1110 TOP VIEW (Not to Scale) 7 SET 6 A0 5 GND TOP VIEW SW1 3 (Not to Scale) 6 SW2 4 TJMAX = 90o, θJA = SW1 3 SW2 4 5 GND 130oC/W TJMAX = 90o, θJA = 150oC/W *FIXED VERSIONS *FIXED VERSIONS PIN DESCRIPTION 1 ILIM 1.5V AA CELL* 2 VIN Mnemonic ILIM Function For normal conditions this pin is connected to VIN. When lower current is required, a resistor should be connected between ILIM and VIN. Limiting the switch current to 400 mA is achieved by connecting a 220 Ω resistor. Input Voltage. Collector Node of Power Transistor. For stepdown configuration, connect to VIN. For stepup configuration, connect to an inductor/diode. Emitter Node of Power Transistor. For stepdown configuration, connect to inductor/diode. For step-up configuration, connect to ground. Do not allow this pin to go more than a diode drop below ground. Ground. Auxiliary Gain (GB) Output. The open collector can sink 300 µA. It can be left open if unused. Gain Amplifier Input. The amplifier has positive input connected to SET pin and negative input connected to 220 mV reference. It can be left open if unused. On the ADP1110 (adjustable) version this pin is connected to the comparator input. On the ADP1110-3.3, ADP1110-5 and ADP1110-12, the pin goes directly to the internal application resistor that set output voltage. SW1 3 ADP1110-5 SENSE 8 GND 5 SW2 4 15µF TANTALUM OPERATES WITH CELL VOLTAGE ≥1.0V *ADD 10µF DECOUPLING CAPACITOR IF BATTERY IS *MORE THAN 2' AWAY FROM ADP1110. VIN SW1 Figure 1. 1.5 V to 5 V Converter SW2 ORDERING GUIDE Model ADP1110AN ADP1110AR ADP1110AN-3.3 ADP1110AR-3.3 ADP1110AN-5 ADP1110AR-5 ADP1110AN-12 ADP1110AR-12 Output Voltage ADJ ADJ 3.3 V 3.3 V 5V 5V 12 V 12 V Package N-8 SO-8 N-8 SO-8 N-8 SO-8 N-8 SO-8 GND AO SET FB/SENSE CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the ADP1110 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE REV. 0 –3– ADP1110-Typical Characteristics 76 1.4 1.2 1 0.8 0.6 0.4 0.2 0 0.1 VIN = +1.2V VIN = +3V VIN = +1V VIN = +5V VIN = +1.5V VIN = +2V 74 OSCILLATOR FREQUENCY – kHz SATURATION VOLTAGE – V 72 OSCILLATOR FREQUENCY 70 68 66 64 62 0.2 0.4 0.5 0.6 0.8 1 ISWITCH CURRENT – A 1.2 1.25 1.4 60 2 4 6 8 10 12 15 18 INPUT VOLTAGE – V 21 24 27 30 Figure 2. Saturation Voltage vs. ISWITCH Current in Step-Up Mode Figure 5. Oscillator Frequency vs. Input Voltage 2 1.8 1.6 1.4 ON VOLTAGE – V 1.9 1.7 1.5 SWITCH CURRENT – A VIN = +12V 1.3 1.1 0.9 0.7 0.5 0.3 0.1 1 1.2 1 0.8 0.6 0.4 0.2 0 0.1 0.2 0.6 ISWITCH CURRENT – A 0.4 0.8 0.9 STEP-DOWN WITH V = +12V 10 RLIM – Ω 100 1000 Figure 3. Switch ON Voltage vs. ISWITCH Current In StepDown Mode Figure 6. Maximum Switch Current vs. RLIM 1800 1600 QUIESCENT CURRENT – µA 1400 1200 QUIESCENT CURRENT 1000 800 600 400 200 0 1 3 6 9 12 15 18 21 24 27 30 INPUT VOLTAGE – V SWITCH CURRENT – A 1.5 1.3 1.1 0.9 0.7 STEP-UP MODE WITH V ≤ +5V 0.5 0.3 0.1 1 10 RLIM – Ω 100 1000 Figure 4. Quiescent Current vs. Input Voltage Figure 7. Maximum Switch Current vs. RLIM –4– REV. 0 ADP1110 78 76 OSCILLATOR FREQUENCY – kHz 0.54 0.53 0.52 74 OSCILLATOR FREQUENCY 70 68 VCE(SAT) – V 72 0.51 VIN = +1.5 @ ISWITCH = +0.5A 0.5 0.49 66 64 0.48 0.47 0 25 TEMPERATURE – C 70 0 25 TEMPERATURE – C 70 Figure 8. Oscillator Frequency vs. Temperature Figure 11. Switch ON Voltage Step-Down vs. Temperature 9.0 8.9 8.8 ON TIME – µs 350 300 QUIESCENT CURRENT 250 8.7 8.6 SWITCH ON TIME 8.5 8.4 8.3 8.2 QUIESCENT CURRENT – µA 200 150 100 50 0 0 25 TEMPERATURE – C 70 0 25 TEMPERATURE – C 70 Figure 9. Switch ON Time vs. Temperature Figure 12. Quiescent Current vs. Temperature 66 65 64 160 155 150 BIAS CURRENT – nA 145 140 BIAS CURRENT 135 130 DUTY CYCLE – % 63 DUTY CYCLE 62 61 60 59 125 120 0 25 TEMPERATURE – C 70 0 25 TEMPERATURE – C 70 Figure 10. Duty Cycle vs. Temperature Figure 13. FB Pin Bias Current vs. Temperature REV. 0 –5– ADP1110 400 350 BIAS CURRENT 300 BIAS CURRENT – nA 250 VREF – mV 220 219 218 217 REFERENCE VOLTAGE 216 215 214 200 150 100 50 0 0 25 TEMPERATURE – C 70 213 212 211 0 25 TEMPERATURE – C 70 Figure 14. Set Pin Bias Current vs. Temperature THEORY OF OPERATION Figure 15. Reference Voltage vs. Temperature The ADP1110 is a flexible, low-power, switch-mode power supply (SMPS) controller. The regulated output voltage can be greater than the input voltage (boost or step-up mode) or less than the input (buck or step-down mode). This device uses a gated-oscillator technique to provide very high performance with low quiescent current. A functional block diagram of the ADP1110 is shown on the first page. The internal 220 mV reference is connected to one input of the comparator, while the other input is externally connected (via the FB pin) to a feedback network connected to the regulated output. When the voltage at the FB pin falls below 220 mV, the 70 kHz oscillator turns on. A driver amplifier provides base drive to the internal power switch, and the switching action raises the output voltage. When the voltage at the FB pin exceeds 220 mV, the oscillator is shut off. While the oscillator is off, the ADP1110 quiescent current is only 300 µA. The comparator includes a small amount of hysteresis, which ensures loop stability without requiring external components for frequency compensation. The maximum current in the internal power switch can be set by connecting a resistor between VIN and the ILIM pin. When the maximum current is exceeded, the switch is turned OFF. The current limit circuitry has a time delay of about 800 ns. If an external resistor is not used, connect ILIM to VIN. Further information on ILIM is included in the “Applications” section of this data sheet. The ADP1110 internal oscillator provides 10 µs ON and 5 µs OFF times, which is ideal for applications where the ratio between VIN and VOUT is roughly a factor of three (such as generating +5 V from a single 1.5 V cell). Wider range conversions, as well as step-down converters, can also be accomplished with a slight loss in the maximum output power that can be obtained. An uncommitted gain block on the ADP1110 can be connected as a low–battery detector. The inverting input of the gain block is internally connected to the 220 mV reference. The noninverting input is available at the SET pin. A resistor divider, connected between VIN and GND with the junction connected to the SET pin, causes the AO output to go LOW when the low battery set point is exceeded. The AO output is an open collector NPN transistor that can sink 300 µA. The ADP1110 provides external connections for both the collector and emitter of its internal power switch, which permits –6– both step-up and step-down modes of operation. For the stepup mode, the emitter (Pin SW2) is connected to GND and the collector (Pin SW1) drives the inductor. For step-down mode, the emitter drives the inductor while the collector is connected to VIN. The output voltage of the ADP1110 is set with two external resistors. Three fixed-voltage models are also available: ADP1110–3.3 (+3.3 V), ADP1110–5 (+5 V) and ADP1110-12 (+12 V). The fixed-voltage models are identical to the ADP1110 except that laser-trimmed voltage-setting resistors are included on the chip. Only three external components are required to form a +3.3 V, +5 V or +12 V converter. On the fixed-voltage models of the ADP1110, simply connect the SENSE pin (Pin 8) directly to the output voltage. COMPONENT SELECTION General Notes on Inductor Selection When the ADP1110 internal power switch turns on, current begins to flow in the inductor. Energy is stored in the inductor core while the switch is on, and this stored energy is then transferred to the load when the switch turns off. Because both the collector and the emitter of the switch transistor are accessible on the ADP1110, the output voltage can be higher, lower, or of opposite polarity than the input voltage. To specify an inductor for the ADP1110, the proper values of inductance, saturation current, and DC resistance must be determined. This process is not difficult, and specific equations for each circuit configuration are provided in this data sheet. In general terms, however, the inductance value must be low enough to store the required amount of energy (when both input voltage and switch ON time are at a minimum) but high enough that the inductor will not saturate when both VIN and switch ON time are at their maximum values. The inductor must also store enough energy to supply the load without saturating. Finally, the dc resistance of the inductor should be low so that excessive power will not be wasted by heating the windings. For most ADP1110 applications, an inductor of 15 µH to 100 µH with a saturation current rating of 300 mA to 1A and dc resistance 6.2 V If the input voltage to the ADP1110 varies over a wide range, a current limiting resistor at Pin 1 may be required. If a particular circuit requires high peak inductor current with minimum input supply voltage, the peak current may exceed the switch maximum rating and/or saturate the inductor when the supply voltage is at the maximum value. See the “Limiting the Switch Current” section of this data sheet for specific recommendations. INCREASING OUTPUT CURRENT IN THE STEP-DOWN REGULATOR where 10 µs is the ADP1110 switch’s “on” time. VIN C2 RLIM 100Ω 1 ILIM 2 3 VIN SW1 FB 8 L1 VOUT D1 1N5818 C1 R1 R2 SW2 4 AO SET GND 6 NC 7 5 ADP1110 NC Figure 20. Step-Down Mode Operation When the switch turns off, the magnetic field collapses. The polarity across the inductor changes, and the switch side of the inductor is driven below ground. Schottky diode D1 then turns on, and current flows into the load. Notice that the Absolute Maximum Rating for the ADP1110’s SW2 pin is 0.5 V below ground. To avoid exceeding this limit, D1 must be a Schottky diode. Using a silicon diode in this application will generate forward voltages above 0.5 V that will cause potentially damaging power dissipation within the ADP1110. The output voltage of the buck regulator is fed back to the ADP1110’s FB pin by resistors R1 and R2. When the voltage at pin FB falls below 220 mV, the internal power switch turns “on” again and the cycle repeats. The output voltage is set by the formula:  R1 V OUT = 220 mV • 1 +   R2 Unlike the boost configuration, the ADP1110’s internal power switch is not saturated when operating in step-down mode. A conservative value for the voltage across the switch in step-down mode is 1.5 V. This results in high power dissipation within the ADP1110 when high peak current is required. To increase the output current, an external PNP switch can be added (Figure 22). In this circuit, the ADP1110 provides base drive to Q1 through R3, while R4 ensures that Q1 turns off rapidly. Because the ADP1110’s internal current limiting function will not work in this circuit, R5 is provided for this purpose. With the value shown, R5 limits current to 2 A. In addition to reducing power dissipation on the ADP1110, this circuit also reduces the switch voltage. When selecting an inductor value for the circuit of Figure 22, the switch voltage can be calculated from the formula: V SW = V R5 + V Q1(SAT ) ≅ 0.6 V + 0.4 V ≅ 1V 0.3Ω INPUT CINPUT RLIM 1 R5 2 R4 220Ω MJE210 R3 330Ω SW1 3 R1 CL L1 OUTPUT ILIM VIN ADP1110 FB 8 AO SET GND SW2 6 7 5 4 D1 1N5821 R2 NC NC When operating the ADP1110 in step-down mode, the output voltage is impressed across the internal power switch’s emitter- Figure 22. High Current Step-Down Operation –10– REV. 0 ADP1110 POSITIVE-TO-NEGATIVE CONVERSION LIMITING THE SWITCH CURRENT The ADP1110 can convert a positive input voltage to a negative output voltage as shown in Figure 23. This circuit is essentially identical to the step-down application of Figure 19, except that the “output” side of the inductor is connected to power ground. When the ADP1110’s internal power switch turns off, current flowing in the inductor forces the output (–VOUT) to a negative potential. The ADP1110 will continue to turn the switch on until its FB pin is 220 mV above its GND pin, so the output voltage is determined by the formula:  R1 V OUT = 220 mV • 1 +   R2 INPUT RLIM 1 2 3 The ADP1110’s RLIM pin permits the switch current to be limited with a single resistor. This current limiting action occurs on a pulse by pulse basis. This feature allows the input voltage to vary over a wide range without saturating the inductor or exceeding the maximum switch rating. For example, a particular design may require peak switch current of 800 mA with a 2.0 V input. If VIN rises to 4 V, however, the switch current will exceed 1.6 A. The ADP1110 limits switch current to 1.5 A and thereby protects the switch, but the output ripple will increase. Selecting the proper resistor will limit the switch current to 800 mA, even if VIN increases. The relationship between RLIM and maximum switch current is shown in Figure 6. The ILIM feature is also valuable for controlling inductor current when the ADP1110 goes into continuous-conduction mode. This occurs in the step-up mode when the following condition is met: CINPUT VIN ILIM SW1 SW2 4 FB 8 GND 5 L1 OUTPUT R1 CL NEGATIVE OUTPUT ADP1110 AO 6 SET 7 V OUT + VD IODE  1  V –V  < 1– DC   IN SW D1 1N5818 R2 NC NC Figure 23. A Positive-to-Negative Converter The design criteria for the step-down application also apply to the positive-to-negative converter. The output voltage should be limited to |6.2 V| unless a diode is inserted in series with the SW2 pin (see Figure 21.) Also, D1 must again be a Schottky diode to prevent excessive power dissipation in the ADP1110. NEGATIVE-TO-POSITIVE CONVERSION where DC is the ADP1110’s duty cycle. When this relationship exists, the inductor current does not go all the way to zero during the time that the switch is OFF. When the switch turns on for the next cycle, the inductor current begins to ramp up from the residual level. If the switch ON time remains constant, the inductor current will increase to a high level (see Figure 25). This increases output ripple and can require a larger inductor and capacitor. By controlling switch current with the ILIM resistor, output ripple current can be maintained at the design values. Figure 26 illustrates the action of the ILIM circuit. The circuit of Figure 24 converts a negative input voltage to a positive output voltage. Operation of this circuit configuration is similar to the step-up topology of Figure 19, except the current through feedback resistor R1 is level-shifted below ground by a PNP transistor. The voltage across R1 is VOUT – VBEQ1. However, diode D2 level-shifts the base of Q1 about 0.6 V below ground thereby cancelling the VBE of Q1. The addition of D2 also reduces the circuit’s output voltage sensitivity to temperature, which otherwise would be dominated by the –2 mV VBE contribution of Q1. The output voltage for this circuit is determined by the formula:  R1 V OUT = 220 mV •    R2 100 90 200mA/div. 10 0% 10mV 10µs Figure 25. ILIM Operation—IL Characteristic Unlike the positive step-up converter, the negative-to-positive converter’s output voltage can be either higher or lower than the input voltage. L1 RLIM 1 2 100 90 D1 R1 Q1 2N3906 10K CL D2 1N4148 POSITIVE OUTPUT 200mA/div. 10 0% ILIM CINPUT VIN SW1 3 10mV 10µs ADP1110 FB 8 AO SET GND SW2 6 7 5 4 R2 Figure 26. ILIM Operation—IL Characteristic NEGATIVE INPUT NC NC Figure 24. A Negative-to-Positive Converter REV. 0 –11– ADP1110 The internal structure of the ILIM circuit is shown in Figure 27. Q1 is the ADP1110’s internal power switch, that is paralleled by sense transistor Q2. The relative sizes of Q1 and Q2 are scaled so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an internal 80 Ω resistor and through the RLIM resistor. These two resistors parallel the base-emitter junction of the oscillatordisable transistor, Q3. When the voltage across R1 and RLIM exceeds 0.6 V, Q3 turns on and terminates the output pulse. If only the 80 Ω internal resistor is used (i.e., the ILIM pin is connected directly to VIN), the maximum switch current will be 1.5 A. Figure 6 gives RLIM values for lower current-limit values. RLIM (EXTERNAL) VIN VIN R1 Q3 ILIM 80Ω (INTERNAL) IQ1 DRIVER 72kHz OSC 200 Q2 VLOGIC RL ADP1110 VBAT R1 220V VREF SET R2 33k Ω GND AO RHYS Figure 28. Setting the Low Battery Detector Trip Point ADP1110 SW1 Q1 POWER SWITCH The circuit of Figure 28 may produce multiple pulses when approaching the trip point due to noise coupled into the SET input. To prevent multiple interrupts to the digital logic, hysteresis can be added to the circuit. Resistor RHYS, with a value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition of RHYS will change the trip point slightly, so the new value for R1 will be: R1 = V LOBATT – 220 mV 220 mV V L – 220 mV  –  R2  RL + RHYS  SW2 Figure 27. ADP1110 Current Limit Operation The delay through the current limiting circuit is approximately 800 ns. If the switch ON time is reduced to less than 3 µs, accuracy of the current trip-point is reduced. Attempting to program a switch ON time of 800 ns or less will produce spurious responses in the switch ON time; however, the ADP1110 will still provide a properly-regulated output voltage. PROGRAMMING THE GAIN BLOCK where VL is the logic power supply voltage, RL is the pull-up resistor, and RHYS creates the hysteresis. The gain block can also be used as a control element to reduce output ripple. The ADP3000 is normally recommended for lowripple applications, but its minimum input voltage is 2 V. The gain-block technique using the ADP1110 can be useful for stepup converters operating down to 1 V. A step-up converter using this technique is shown in Figure 29. This configuration uses the gain block to sense the output voltage and control the comparator. The result is that the comparator hysteresis is reduced by the open loop gain of the gain block. Output ripple can be reduced to only a few millivolts with this technique, versus a typical value of 90 mV for a +5 V converter using just the comparator. For best results, a large output capacitor (1000 µF or more) should be specified. This technique can also be used for step-down or inverting applications, but the ADP3000 is usually a more appropriate choice. See the ADP3000 data sheet for further details. L1 INPUT CINPUT 10µF 270kΩ 1 2 The gain block of the ADP1110 can be used as a low-battery detector, error amplifier or linear post regulator. The gain block consists of an op amp with PNP inputs and an open-collector NPN output. The inverting input is internally connected to the ADP1110’s 220 mV reference, while the noninverting input is available at the SET pin. The NPN output transistor will sink about 300 µA. Figure 28 shows the gain block configured as a low-battery monitor. Resistors R1 and R2 should be set to high values to reduce quiescent current, but not so high that bias current in the SET input causes large errors. A value of 33 kΩ for R2 is a good compromise. The value for R1 is then calculated from the formula: – 220 mV V R1 = LOBATT 220 mV R2 D1 OUTPUT 1N5818 R1 300kΩ 15µH CTX15-4 ILIM VIN SW1 3 ADP1110 SET 7 AO FB GND SW2 6 8 5 4 where VLOBATT is the desired low battery trip point. Since the gain block output is an open-collector NPN, a pull-up resistor should be connected to the positive logic power supply. R2 13.8kΩ CL 1000µF Figure 29. Using the Gain Block to Reduce Output Ripple –12– REV. 0 ADP1110 APPLICATION CIRCUITS All-Surface-Mount, Single-Cell to 5 V Converter 1.5 V to 5 V Dual-Output Step-Up Converter This is a very simple, compact, low-part-count circuit that takes a single alkaline 1.5 V cell input and produces a 5 V output. The output current should be kept to 10 mA or less to conserve battery life. L1 50µH CTX50-4 1 2 This circuit works from a single 1.5 V cell and provides simultaneous outputs of +5 V and –5 V. The accuracy of the negative output suffers slightly because of the extra diode drop of around 0.4 V. L16 8µH D1 1N5817 +5V 10mA ONE ALKALINE CELL 1.5V 1 2 CTX68-4 4.7µF ONE ALKALINE CELL 1.5V ILIM VIN SW1 3 +5V 3mA –5V 3mA 4.7µF ILIM VIN SW1 3 ADP1110-5 SENSE 8 AO SET GND SW2 6 7 5 4 ADP1110-5 SENSE 8 AO SET GND SW2 6 7 5 4 CL 15µF 4.7µF NC NC NOTE: ALL DIODES 1N5818 NC NC Figure 30. All-Surface-Mount, Single-Cell to 5 V Converter All-Surface-Mount, 3 V to 5 V Step-Up Converter Figure 33. 1.5 V to ± 5 V Dual-Output Step-Up Converter All-Surface-Mount Flash Memory VPP Generator Similar to the previous circuit, this circuit takes a 3-volt input and provides a 5 V output at 40 mA. As in the single-cell version, the circuit is compact and uses only four external components. L1 220Ω 1 2 D1 1N5817 50µH CTX50-4 VIN +5V 40mA Figure 34 shows a circuit that can generate the programming voltage, VPP to program flash memory. The key components are the MOSFET and the bipolar transistor. These two devices form a switch that, when ON, allows the ADP1110 to power-up and function as a step-up converter. The output is +12 V at 120 mA. When the MOSFET switch is OFF, the output of the circuit drops to just under +5 V thereby disabling the programming capability. Care should be taken so there is no short-circuit-current limiting in the circuit in either operating mode. TWO ALKALINE CELLS ILIM 3V SW1 3 ADP1110-5 SENSE 8 AO SET GND SW2 6 7 5 4 CL 10µF +5V MMBT4403 10kΩ 1 2 L1 50µH CTX50-4 D1 1N5818 NC NC VPP +12V 120mA Figure 31. All-Surface-Mount, 3 V to 5 V Step-Up Converter All-Surface-Mount, 9 V to 5 V Step-Down Converter ILIM 1kΩ VIN SW1 3 ADP1110-12 SENSE 8 AO SET GND SW2 6 7 5 4 Featuring the same low parts count of the step-up design, this circuit is the complement to the preceding one. The 220 Ω resistor programs the current limit to around 600 mA. RLIM 220Ω 9V BATTERY 1 2 3 MMBF170 LOGIC1 = PROGRAM LOGIC0 = SHUTDOWN NC NC CL 10µF ILIM VIN SW1 SW2 4 L1 50µH CTX50-4 Figure 34. All Surface-Mount Flash Memory VPP Generator +5V 40mA ADP1110-5 SENSE 8 AO SET GND 6 7 5 D1 1N5817 CL 10µF NC NC Figure 32. All-Surface-Mount, 9 V to 5 V Step-Down Converter REV. 0 –13– ADP1110 1.5 V to +5 V, +10 V Dual Output Step-Up Converter 1.5 V-Powered Laser Diode Driver The circuit of Figure 35 illustrates a way to get outputs of +10 V and +5 V from the same converter. The main 5 V output is derived from the feedback provided by the 487 kΩ and 11 kΩ resistors. Capacitor C1 should be a multilayer ceramic variety for best performance, but a good quality tantalum capacitor will also give good performance at lower cost. L1 50µH 220Ω 1 2 Figure 36 shows a circuit suitable for driving many laser diodes that incorporate a photodiode to monitor the laser diode current, this circuit makes use of the gain block and currentlimit functions to provide a feedback system based on the average laser diode current. This current must be controlled very closely or permanent damage to the laser diode is likely to be the result. To ensure that the laser is operating at the proper power level, the actual optical power from the laser should be monitored with a calibrated photodiode or optical power meter. In addition, the actual diode current should also be monitored, and R1 can be adjusted to give the correct output power. NOTES 1. All inductors referenced are Coiltronics CTX-series except where noted. 2. If the source of power is more than an inch or so from the converter, the input to the converter should be bypassed with approximately 10 µF of capacitance. This capacitor should be a good quality tantalum or aluminum electrolytic. TOSHIBA TOLD-9321 D1 +10V 3mA 487kΩ CTX50-4 ONE ALKALINE CELL 1.5V ILIM VIN SW1 3 D3 D2 ADP1110 FB 8 AO SET GND SW2 6 7 5 4 +5V 3mA 4.7µF 11kΩ 4.7µF NC NC NOTE: ALL DIODES 1N5818 Figure 35. 1.5 V to +5 V, +10 V Dual Output Step-Up Converter 0.022µF 5.1kΩ 2N3906 1 2 220Ω 1µF 100µF OS-Con MJE210 1N5818 2Ω 1kΩ 1N4148 1.5V ILIM VIN 10Ω SW1 3 ADP1110 AO 6 FB 8 SET 7 GND SW2 5 4 1kΩ 2.2µH Figure 36. 1.5 V-Powered Laser Diode Driver –14– REV. 0 ADP1110 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 8-Lead Plastic DIP (N-8) 0.430 (10.92) 0.348 (8.84) 8 5 0.280 (7.11) 0.240 (6.10) 1 4 PIN 1 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.93) 0.060 (1.52) 0.015 (0.38) 0.130 (3.30) MIN SEATING PLANE 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 0.022 (0.558) 0.100 0.070 (1.77) 0.014 (0.356) (2.54) 0.045 (1.15) BSC 0.015 (0.381) 0.008 (0.204) 8-Lead SOIC (SO-8) 0.1968 (5.00) 0.1890 (4.80) 8 1 5 4 0.1574 (4.00) 0.1497 (3.80) 0.2440 (6.20) 0.2284 (5.80) PIN 1 0.0098 (0.25) 0.0040 (0.10) 0.0688 (1.75) 0.0532 (1.35) 0.0196 (0.50) x 45° 0.0099 (0.25) SEATING PLANE 0.0500 0.0192 (0.49) (1.27) 0.0138 (0.35) BSC 0.0098 (0.25) 0.0075 (0.19) 8° 0° 0.0500 (1.27) 0.0160 (0.41) REV. 0 –15– – 16– C2212–12–10/96 PRINTED IN U.S.A.