ADP2106-EVAL

1 Amp/1.5 Amp/2 Amp Synchronous, Step-Down DC-to-DC Converters ADP2105/ADP2106/ADP2107 FEATURES Extremely high 97% efficiency Ultralow quiescent current: 20 μA 1.2 MHz switching frequency 0.1 μA shutdown supply current Maximum load current: ADP2105: 1 A ADP2106: 1.5 A ADP2107: 2 A Input voltage: 2.7 V to 5.5 V Output voltage: 0.8 V to VIN Maximum duty cycle: 100% Smoothly transitions into low dropout (LDO) mode Internal synchronous rectifier Small 16-lead 4 mm × 4 mm LFCSP_VQ package Optimized for small ceramic output capacitors Enable/Shutdown logic input Undervoltage lockout Soft start GENERAL DESCRIPTION The ADP2105/ADP2106/ADP2107 are low quiescent current, synchronous, step-down dc-to-dc converters in a compact 4 mm × 4 mm LFCSP_VQ package. At medium-to-high load currents, these devices use a current-mode, constant-frequency pulse width modulation (PWM) control scheme for excellent stability and transient response. To ensure the longest battery life in portable applications, the ADP2105/ADP2106/ADP2107 use a pulse frequency modulation (PFM) control scheme under light load conditions that reduces switching frequency to save power. The ADP2105/ADP2106/ADP2107 run from input voltages of 2.7 V to 5.5 V, allowing single Li+/Li− polymer cell, multiple alkaline/NiMH cells, PCMCIA, and other standard power sources. The output voltage of ADP2105/ADP2106/ADP2107-ADJ is adjustable from 0.8 V to the input voltage, while the ADP2105/ ADP2106/ADP2107-xx are available in preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V. Each of these variations is available in three maximum current levels, 1 A (ADP2105), 1.5 A (ADP2106), and 2 A (ADP2107). The power switch and synchronous rectifier are integrated for minimal external part count and high efficiency. During logic-controlled shutdown, the input is disconnected from the output, and it draws less than 0.1 μA from the input source. Other key features include undervoltage lockout to prevent deep-battery discharge and programmable soft start to limit inrush current at startup. APPLICATIONS Mobile handsets PDAs and palmtop computers Telecommunication/Networking equipment Set top boxes Audio/Video consumer electronics TYPICAL PERFORMANCE CHARACTERISTICS 100 VIN = 3.3V 95 ON OFF 1 TYPICAL OPERATING CIRCUIT 0.1μF 10Ω VIN INPUT VOLTAGE = 2.7V TO 5.5V 10μF FB 16 15 14 13 VIN = 3.6V VOUT = 2.5V FB EN GND GND GND COMP 5 GND IN PWIN1 LX2 12 PGND 11 OUTPUT VOLTAGE = 2.5V 2μH EFFICIENCY (%) 90 2 VIN = 5V 85 3 ADP2107-ADJ LX1 10 PWIN2 9 SS 6 85kΩ FB VIN 40kΩ 10μF 10μF 4.7μF 4 80 06079-001 AGND NC 7 8 LOAD 0A TO 2A 70kΩ 120pF 1nF 0 200 400 600 800 1000 1200 1400 1600 1800 2000 NC = NO CONNECT LOAD CURRENT (mA) Figure 1. Efficiency vs. Load Current for the ADP2107 with VOUT = 2.5 V Figure 2. Circuit Configuration of ADP2107 with VOUT = 2.5 V Rev. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2006 Analog Devices, Inc. All rights reserved. 06079-002 75 ADP2105/ADP2106/ADP2107 TABLE OF CONTENTS Features .............................................................................................. 1 Applications....................................................................................... 1 General Description ......................................................................... 1 Typical Performance Characteristics ............................................. 1 Typical Operating Circuit................................................................ 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 5 Thermal Resistance ...................................................................... 5 Boundary Condition.................................................................... 5 ESD Caution.................................................................................. 5 Pin Configuration and Function Descriptions............................. 6 Typical Performance Characteristics ............................................. 7 Theory of Operation ...................................................................... 12 Control Scheme .......................................................................... 12 PWM Mode Operation.............................................................. 12 PFM Mode Operation................................................................ 12 Pulse-Skipping Threshold ......................................................... 12 100% Duty Cycle Operation (LDO Mode) ............................. 12 Slope Compensation .................................................................. 13 Features ........................................................................................ 13 Applications Information .............................................................. 15 External Component Selection................................................. 15 Setting the Output Voltage........................................................ 15 Inductor Selection ...................................................................... 16 Output Capacitor Selection....................................................... 17 Input Capacitor Selection.......................................................... 17 Input Filter................................................................................... 18 Soft Start ...................................................................................... 18 Loop Compensation .................................................................. 18 Bode Plots.................................................................................... 19 Load Transient Response .......................................................... 20 Efficiency Considerations ......................................................... 21 Thermal Considerations............................................................ 21 Design Example.......................................................................... 22 External Component Recommendations.................................... 24 Circuit Board Layout Recommendations ................................... 26 Evaluation Board ............................................................................ 27 Evaluation Board Schematic (ADP2107-1.8V)...................... 27 Recommended PCB Board Layout (Evaluation Board Layout)........................................................ 27 Application Circuits ....................................................................... 29 Outline Dimensions ....................................................................... 31 Ordering Guide .......................................................................... 31 REVISION HISTORY 7/06—Revision 0: Initial Version Rev. 0 | Page 2 of 32 ADP2105/ADP2106/ADP2107 SPECIFICATIONS VIN = 3.6 V @ TA = 25°C, unless otherwise noted. 1 Bold values indicate −40°C ≤ TJ ≤ +125°C. Table 1. Parameter INPUT CHARACTERISTICS Input Voltage Range Undervoltage Lockout Threshold Undervoltage Lockout Hysteresis 2 OUTPUT CHARACTERISTICS Output Regulation Voltage Conditions Min 2.7 2.2 2.0 Typ Max 5.5 2.6 2.5 Unit V V V mV V V V V V V V V %/A %/A %/A %/V V μA μA μA μA V μA μA μA μA mΩ mΩ μA A A A ns V V μA MHz μs VIN rising VIN falling 2.4 2.2 200 3.3 3.3 1.8 1.8 1.5 1.5 1.2 1.2 0.4 0.5 0.6 0.1 Load Regulation Line Regulation 3 Output Voltage Range FEEDBACK CHARACTERISTICS OUT_SENSE Bias Current ADP210x-3.3, load = 10 mA ADP210x-3.3, VIN = 3.5 V to 5.5 V, no load to full load ADP210x-1.8, load = 10 mA ADP210x-1.8, VIN = 2.7 V to 5.5 V, no load to full load ADP210x-1.5, load = 10 mA ADP210x-1.5, VIN = 2.7 V to 5.5 V, no load to full load ADP210x-1.2, load = 10 mA ADP210x-1.2, VIN = 2.7 V to 5.5 V, no load to full load ADP2105 ADP2106 ADP2107 Measured in servo loop ADP210x-ADJ ADP210x-1.2 ADP210x-1.5 ADP210x-1.8 ADP210x-3.3 ADP210x-ADJ ADP210x-ADJ ADP210x-ADJ, VFB = 0.9 V ADP210x-xx, output voltage 10% above regulation voltage VEN = 0 V P-channel switch N-channel synchronous rectifier VIN = 5.5 V, VLX = 0 V, 5.5 V P-channel switch, ADP2107 P-channel switch, ADP2106 P-channel switch, ADP2105 In PWM mode of operation, VIN = 5.5 V VIN = 2.7 V to 5.5 V VIN = 2.7 V to 5.5 V VIN = 5.5 V, VEN = 0 V, 5.5 V VIN = 2.7 V to 5.5 V CSS = 1 nF 3.267 3.201 1.782 1.746 1.485 1.455 1.188 1.164 3.333 3.399 1.818 1.854 1.515 1.545 1.212 1.236 0.8 3 4 5 10 0.8 0.3 VIN 6 8 10 20 0.816 +0.1 30 30 15 165 140 15 3.3 2.6 1.8 100 FB Regulation Voltage FB Bias Current INPUT CURRENT CHARACTERISTICS IN Operating Current IN Shutdown Current LX (SWITCH NODE) CHARACTERISTICS LX On Resistance 4 LX Leakage Current4 LX Peak Current Limit4 0.784 −0.1 20 20 0.1 100 90 0.1 2.9 2.25 1.5 2.6 2.0 1.3 LX Minimum On-Time4 ENABLE CHARACTERISTICS EN Input High Voltage EN Input Low Voltage EN Input Leakage Current OSCILLATOR FREQUENCY SOFT START PERIOD 2 −1 1 750 −0.1 1.2 1000 0.4 +1 1.4 1200 Rev. 0 | Page 3 of 32 ADP2105/ADP2106/ADP2107 Parameter THERMAL CHARACTERISTICS Thermal Shutdown Threshold Thermal Shutdown Hysteresis COMPENSATOR TRANSCONDUCTANCE (Gm) CURRENT SENSE AMPLIFIER GAIN (GCS)2 Conditions Min Typ 140 40 ADP2105 ADP2106 ADP2107 50 1.875 2.8125 3.625 Max Unit °C °C μA/V A/V A/V A/V 1 2 3 All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C. Guaranteed by design. The ADP2015/ADP2106/ADP2107 line regulation was measured in a servo loop on the ATE that adjusts the feedback voltage to achieve a specific comp voltage. 4 All LX (switch node) characteristics are guaranteed only when the LX1 and LX2 pins are tied together. 5 These specifications are guaranteed from −40°C to +85°C. Rev. 0 | Page 4 of 32 ADP2105/ADP2106/ADP2107 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter IN, EN, SS, COMP, OUT_SENSE/FB to AGND LX1, LX2 to PGND PWIN1, PWIN2 to PGND PGND to AGND GND to AGND PWIN1, PWIN2 to IN Operating Junction Temperature Range Storage Temperature Range Soldering Conditions Rating −0.3 V to +6 V −0.3 V to (VIN + 0.3 V) −0.3 V to +6 V −0.3 V to +0.3 V −0.3 V to +0.3 V −0.3 V to +0.3 V −40°C to +125°C −65°C to +150°C JEDEC J-STD-020 THERMAL RESISTANCE θJA is specified for the worst-case conditions, that is, a device soldered in a circuit board for surface-mount packages. Table 3. Thermal Resistance Package Type 16-Lead LFCSP_VQ/QFN Maximum Power Dissipation 1 θJA 1 40 1 Unit °C/W W θJA is specified for the worst-case conditions; that is, θJA is specified for device soldered in circuit board for surface mount packages. Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. BOUNDARY CONDITION Natural convection, 4-layer board, exposed pad soldered to the PCB. ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. 0 | Page 5 of 32 ADP2105/ADP2106/ADP2107 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS 16 OUT_SENSE/FB PIN 1 INDICATOR 13 PWIN1 15 GND 14 IN EN 1 GND 2 GND 3 GND 4 12 LX2 ADP2105/ ADP2106/ ADP2107 TOP VIEW (Not to Scale) 11 PGND 10 LX1 9 PWIN2 COMP 5 SS 6 AGND 7 NC 8 NC = NO CONNECT Figure 3. Pin Configuration Table 4. Pin Function Descriptions Pin No. 1 2, 3, 4, 15 5 Mnemonic ADP210x-xx ADP210x-ADJ EN EN GND GND Description Enable Input. Drive EN high to turn on the ADP2105/ADP2106/ADP2107. Drive EN low to turn it off and reduce the input current to 0.1 μA. Test Pins. These pins are used by Analog Devices, Inc. for internal testing and are not ground return pins. Tie these pins to the AGND plane as close to the ADP2105/ADP2106/ADP2107 as possible. Feedback Loop Compensation Node. COMP is the output of the internal transconductance error amplifier. Place a series RC network from COMP to AGND to compensate the converter. See the Loop Compensation section. Soft Start Input. Place a capacitor from SS to AGND to set the soft start period. A 1 nF capacitor sets a 1 ms soft start period. Analog Ground. Connect the ground of the compensation components, soft start capacitor, and the voltage divider on the FB pin to the AGND pin as close as possible to the ADP2105/ ADP2106/ADP2107. Also connect AGND to the exposed pad of ADP2105/ADP2106/ADP2107. No Connect. Not internally connected. Can be connected to other pins or left unconnected. Power Source Inputs. The source of the PFET high-side switch. Bypass each PWIN pin to the nearest PGND plane with a 4.7 μF or greater capacitor as close as possible to the ADP2105/ADP2106/ ADP2107. See the Input Capacitor Selection section. Switch Outputs. The drain of the P-channel power switch and N-channel synchronous rectifier. Tie the two LX pins together and connect the output LC filter between LX and the output voltage. Power Ground. Connect the ground return of all input and output capacitors to PGND pin, using a power ground plane as close as possible to the ADP2105/ADP2106/ADP2107. Also connect PGND to the exposed pad of the ADP2105/ADP2106/ADP2107. ADP2105/ADP2106/ADP2107 Power Input. The power source for the ADP2105/ADP2106/ ADP2107 internal circuitry. Connect IN and PWIN1 with a 10 Ω resistor as close as possible to the ADP2105/ADP2106/ADP2107. Bypass IN to AGND with a 0.1 μF or greater capacitor. See the Input Filter section. Output Voltage Sense or Feedback Input. For fixed output versions, connect OUT_SENSE to the output voltage. For adjustable versions, FB is the input to the error amplifier. Drive FB through a resistive voltage divider to set the output voltage. The FB regulation voltage is 0.8 V. COMP COMP 6 7 SS AGND SS AGND 8 9, 13 NC PWIN2, PWIN1 LX1, LX2 NC PWIN2, PWIN1 10, 12 LX1, LX2 11 PGND PGND 14 IN IN 16 OUT_SENSE FB Rev. 0 | Page 6 of 32 06079-003 ADP2105/ADP2106/ADP2107 TYPICAL PERFORMANCE CHARACTERISTICS 100 95 90 85 VIN = 3.6V VIN = 5.5V VIN = 4.2V VIN = 2.7V 100 95 90 85 VIN = 3.6V VIN = 2.7V EFFICIENCY (%) 80 75 70 65 60 06079-004 EFFICIENCY (%) 80 75 70 65 60 VIN = 5.5V VIN = 4.2V 55 50 1 10 55 50 1 10 100 1000 100 1000 LOAD CURRENT (mA) LOAD CURRENT (mA) Figure 4. Efficiency—ADP2105 (1.2 V Output) 100 95 90 85 VIN = 3.6V 100 95 90 85 Figure 7. Efficiency—ADP2105 (1.8 V Output) VIN = 3.6V VIN = 2.7V EFFICIENCY (%) EFFICIENCY (%) 80 75 70 65 60 VIN = 4.2V VIN = 5.5V 80 75 70 65 60 VIN = 4.2V VIN = 5.5V 06079-052 55 50 1 10 55 50 1 10 100 100 1000 1000 10000 LOAD CURRENT (mA) LOAD CURRENT (mA) Figure 5. Efficiency—ADP2105 (3.3 V Output) 100 95 90 85 VIN = 2.7V VIN = 4.2V VIN = 5.5V VIN = 3.6V 100 95 90 85 Figure 8. Efficiency—ADP2106 (1.2 V Output) EFFICIENCY (%) 80 75 70 65 60 EFFICIENCY (%) VIN = 5.5V VIN = 4.2V 80 75 70 65 60 06079-062 55 50 1 10 100 55 50 1 10 100 1000 10000 1000 10000 LOAD CURRENT (mA) LOAD CURRENT (mA) Figure 6. Efficiency—ADP2106 (1.8 V Output) Figure 9. Efficiency—ADP2106 (3.3 V Output) Rev. 0 | Page 7 of 32 06079-053 INDUCTOR: D62LCB, 2µH DCR: 28mΩ TA = 25°C VIN = 3.6V INDUCTOR: D62LCB, 3.3µH DCR: 47mΩ TA = 25°C 06079-008 INDUCTOR: CDRH5D18, 4.1μH DCR: 43mΩ TA = 25°C INDUCTOR: D62LCB, 2µH DCR: 28mΩ TA = 25°C 06079-061 INDUCTOR: SD14, 2.5µH DCR: 60mΩ TA = 25°C INDUCTOR: SD3814, 3.3µH DCR: 93mΩ TA = 25°C ADP2105/ADP2106/ADP2107 100 95 90 85 VIN = 3.6V EFFICIENCY (%) 100 95 VIN = 3.6V VIN = 2.7V VIN = 2.7V 90 85 80 75 70 65 60 06079-010 EFFICIENCY (%) 80 75 70 65 60 55 50 1 10 100 LOAD CURRENT (mA) INDUCTOR: SD12, 1.2µH DCR: 37mΩ TA = 25°C 1000 VIN = 5.5V VIN = 4.2V VIN = 4.2V VIN = 5.5V 55 50 1 10 100 10000 1000 10000 LOAD CURRENT (mA) Figure 10. Efficiency—ADP2107 (1.2 V) 100 95 90 85 OUTPUT VOLTAGE (V) 1.22 1.23 Figure 13. Efficiency—ADP2107 (1.8 V) 2.7V, –40°C 3.6V, –40°C 5.5V, –40°C 2.7V, +25°C 3.6V, +25°C 5.5V, +25°C 2.7V, +125°C 3.6V, +125°C 5.5V, +125°C EFFICIENCY (%) VIN = 5.5V VIN = 4.2V 1.21 80 75 70 65 60 55 50 1 1.20 1.19 VIN = 3.6V 06079-054 06079-082 INDUCTOR: CDRH5D28, 2.5µH DCR: 13mΩ TA = 25°C 10 100 LOAD CURRENT (mA) 1000 1.18 10000 1.17 0.01 0.1 1 10 100 1000 10000 LOAD CURRENT (mA) Figure 11. Efficiency—ADP2107 (3.3 V) 1.85 Figure 14. Output Voltage Accuracy—ADP2107 (1.2 V) 3.38 3.36 3.6V, –40°C 5.5V, –40°C 3.6V, +25°C 5.5V, +25°C 3.6V, +125°C 5.5V, +125°C 1.83 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 3.34 3.32 3.30 3.28 3.26 1.81 1.79 1.77 06079-064 06079-081 2.7V, –40°C 3.6V, –40°C 5.5V, –40°C 1.75 0.1 1 2.7V, +25°C 3.6V, +25°C 5.5V, +25°C 10 2.7V, +125°C 3.6V, +125°C 5.5V, +125°C 100 1000 3.24 3.22 0.01 10000 0.1 1 10 100 1000 10000 LOAD CURRENT (mA) LOAD CURRENT (mA) Figure 12. Output Voltage Accuracy—ADP2107 (1.8 V) Figure 15. Output Voltage Accuracy—ADP2107 (3.3 V) Rev. 0 | Page 8 of 32 06079-063 INDUCTOR: D62LCB, 1.5µH DCR: 21mΩ TA = 25°C ADP2105/ADP2106/ADP2107 10000 120 PMOS POWER SWITCH 100 SW ON RESISTANCE (mΩ) 1000 INPUT CURRENT (µA) 80 +25°C 100 –40°C 60 NMOS SYNCHRONOUS RECTIFIER 40 10 06079-016 1 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 4.8 5.2 0 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Figure 16. Quiescent Current vs. Input Voltage 0.802 0.801 0.800 0.799 0.798 0.797 0.796 0.795 –40 SWITCHING FREQUENCY (kHz) Figure 19. Switch On Resistance vs. Input Voltage 1260 1250 1240 1230 +125°C 1220 1210 1200 1190 2.7 –40°C +25°C FEEDBACK VOLTAGE (V) 06079-017 –20 0 20 40 60 80 100 120 125 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 TEMPERATURE (°C) INPUT VOLTAGE (V) Figure 17. Feedback Voltage vs. Temperature 1.75 1.70 1.65 PEAK CURRENT LIMIT (A) PEAK CURRENT LIMIT (A) Figure 20. Switching Frequency vs. Input Voltage 2.35 2.30 2.25 2.20 2.15 2.10 2.05 2.00 1.95 06079-073 1.60 1.55 1.50 1.45 1.40 1.35 1.30 1.25 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 5.7 INPUT VOLTAGE (V) ADP2105 (1A) ADP2106 (1.5A) 1.90 1.85 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 5.7 INPUT VOLTAGE (V) Figure 18. Peak Current Limit of ADP2105 Figure 21. Peak Current Limit of ADP2106 Rev. 0 | Page 9 of 32 06079-072 06079-021 06079-018 +125°C 20 ADP2105/ADP2106/ADP2107 3.00 2.95 2.90 135 PULSE SKIPPING THRESHOLD CURRENT (mA) 120 105 90 VOUT = 1.2V 75 60 45 30 06079-066 06079-083 PEAK CURRENT LIMIT (A) 2.85 ADP2107 (2A) 2.80 2.75 2.70 2.65 2.60 2.55 2.50 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 5.7 INPUT VOLTAGE (V) 06079-071 VOUT = 1.8V VOUT = 2.5V 15 0 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 5.7 INPUT VOLTAGE (V) Figure 22. Peak Current Limit of ADP2107 150 Figure 25. Pulse Skipping Threshold vs. Input Voltage for ADP2105 195 PULSE SKIPPING THRESHOLD CURRENT (mA) PULSE SKIPPING THRESHOLD CURRENT (mA) 135 120 105 90 75 60 45 30 06079-067 180 165 150 135 120 105 90 75 60 45 06079-068 VOUT = 1.2V VOUT = 1.2V VOUT = 1.8V VOUT = 1.8V VOUT = 2.5V VOUT = 2.5V 30 15 0 2.7 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 TA = 25°C 5.4 5.7 INPUT VOLTAGE (V) 15 0 2.7 TA = 25°C 3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4 5.7 INPUT VOLTAGE (V) Figure 23. Pulse Skipping Threshold vs. Input Voltage for ADP2106 LX NODE (SWITCH NODE) Figure 26. Pulse Skipping Threshold vs. Input Voltage for ADP2107 140 120 3 SWITCH ON RESISTANCE (mΩ) 100 80 PMOS POWER SWITCH INDUCTOR CURRENT Δ: 260mV @: 3.26V NMOS SYNCHRONOUS RECTIFIER 60 40 20 0 –40 1 OUTPUT VOLTAGE 06079-074 4 CH1 1V CH3 5V CH4 1AΩ M 10µs T 45.8% A CH1 1.78V –20 0 20 40 60 80 100 120 JUNCTION TEMPERATURE (°C) Figure 24. Short Circuit Response at Output Figure 27. Switch On Resistance vs. Temperature Rev. 0 | Page 10 of 32 ADP2105/ADP2106/ADP2107 LX NODE (SWITCH NODE) LX NODE (SWITCH NODE) 3 3 1 1 OUTPUT VOLTAGE (AC-COUPLED) OUTPUT VOLTAGE (AC-COUPLED) 06079-030 INDUCTOR CURRENT CH1 50mV CH3 2V CH4 200mAΩ M 2µs T 6% A CH3 3.88V 4 CH1 20mV CH3 2V CH4 1AΩ M 1µs T 17.4% A CH3 3.88V Figure 28. PFM Mode of Operation at Very Light Load (10 mA) Figure 31. PWM Mode of Operation at Medium/Heavy Load (1.5 A) LX NODE (SWITCH NODE) 3 3 CHANNEL 3 FREQUENCY = 336.6kHz LX NODE (SWITCH NODE) Δ: 2.86A @: 2.86A 1 OUTPUT VOLTAGE (AC-COUPLED) 1 INDUCTOR CURRENT OUTPUT VOLTAGE 06079-033 INDUCTOR CURRENT CH1 50mV CH3 2V M 400ns T 17.4% A CH3 3.88V CH4 200mAΩ CH1 1V CH3 5V CH4 1AΩ M 4µs T 45% A CH3 1.8V Figure 29. DCM Mode of Operation at Light Load (100 mA) LX NODE (SWITCH NODE) Figure 32. Current Limit Behavior of ADP2107 (Frequency Foldback) ENABLE VOLTAGE 3 OUTPUT VOLTAGE 3 1 1 OUTPUT VOLTAGE (AC-COUPLED) INDUCTOR CURRENT 06079-034 INDUCTOR CURRENT 4 CH1 20mV CH3 2V CH4 1AΩ M 2µs T 13.4% A CH3 1.84V CH1 1V CH3 5V CH4 500mAΩ M 400µs T 20.2% A CH1 1.84V Figure 30. Minimum Off Time Control at Dropout Figure 33. Startup and Shutdown Waveform (CSS = 1 nF → SS Time = 1 ms) Rev. 0 | Page 11 of 32 06079-035 4 06079-032 4 4 06079-031 4 INDUCTOR CURRENT ADP2105/ADP2106/ADP2107 THEORY OF OPERATION The ADP2105/ADP2106/ADP2107 are step-down, dc-to-dc converters that use a fixed frequency, peak current-mode architecture with an integrated high-side switch and low-side synchronous rectifier. The high 1.2 MHz switching frequency and tiny 16-lead, 4 mm × 4 mm LFCSP_VQ package allow for a small step-down dc-to-dc converter solution. The integrated high-side switch (P-channel MOSFET) and synchronous rectifier (N-channel MOSFET) yield high efficiency at medium-toheavy loads. Light load efficiency is improved by smoothly transitioning to variable frequency PFM mode. The ADP2105/ADP2106/ADP2107-ADJ operate with an input voltage from 2.7 V to 5.5 V and regulate an output voltage down to 0.8 V. The ADP2105/ADP2106/ADP2107 are also available with preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V. PFM MODE OPERATION The ADP2105/ADP2106/ADP2107 smoothly transition to the variable frequency PFM mode of operation when the load current decreases below the pulse-skipping threshold current, switching only as necessary to maintain the output voltage within regulation. When the output voltage dips below regulation, the ADP2105/ ADP2106/ADP2107 enter PWM mode for a few oscillator cycles to increase the output voltage back to regulation. During the wait time between bursts, both power switches are off, and the output capacitor supplies all the load current. Because the output voltage dips and recovers occasionally, the output voltage ripple in this mode is larger than the ripple in the PWM mode of operation. PULSE-SKIPPING THRESHOLD The output current at which the ADP2105/ADP2106/ADP2107 transition from variable frequency PFM control to fixed frequency PWM control is called the pulse-skipping threshold. The pulseskipping threshold has been optimized for excellent efficiency over all load currents. The variation of pulse-skipping threshold with input voltage and output voltage is shown in Figure 23, Figure 25, and Figure 26. CONTROL SCHEME The ADP2105/ADP2106/ADP2107 operate with a fixed frequency, peak current-mode PWM control architecture at medium-to-high loads for high efficiency, but shift to a variable frequency PFM control scheme at light loads for lower quiescent current. When operating in fixed frequency PWM mode, the duty cycle of the integrated switches is adjusted to regulate the output voltage, but when operating in PFM mode at light loads, the switching frequency is adjusted to regulate the output voltage. The ADP2105/ADP2106/ADP2107 operate in the PWM mode only when the load current is greater than the pulse-skipping threshold current. At load currents below this value, the converter smoothly transitions to the PFM mode of operation. 100% DUTY CYCLE OPERATION (LDO MODE) As the input voltage drops, approaching the output voltage, the ADP2105/ADP2106/ADP2107 smoothly transition to 100% duty cycle, maintaining the P-channel MOSFET switch on continuously. This allows the ADP2105/ADP2106/ADP2107 to regulate the output voltage until the drop in input voltage forces the P-channel MOSFET switch to enter dropout, as shown in the following equation: VIN(MIN) = IOUT × (RDS(ON) − P + DCRIND) + VOUT(NOM) The ADP2105/ADP2106/ADP2107 achieve 100% duty cycle operation by stretching the P-channel MOSFET switch on-time if the inductor current does not reach the peak inductor current level by the end of the clock cycle. Once this happens, the oscillator remains off until the inductor current reaches the peak inductor current level, at which time the switch is turned off and the synchronous rectifier is turned on for a fixed off-time. At the end of the fixed off-time, another cycle is initiated. As the ADP2105/ADP2106/ADP2107 approach dropout, the switching frequency decreases gradually to smoothly transition to 100% duty cycle operation. PWM MODE OPERATION In PWM mode, the ADP2105/ADP2106/ADP2107 operate at a fixed frequency of 1.2 MHz set by an internal oscillator. At the start of each oscillator cycle, the P-channel MOSFET switch is turned on, putting a positive voltage across the inductor. Current in the inductor increases until the current sense signal crosses the peak inductor current level that turns off the P-channel MOSFET switch and turns on the N-channel MOSFET synchronous rectifier. This puts a negative voltage across the inductor, causing the inductor current to decrease. The synchronous rectifier stays on for the rest of the cycle, unless the inductor current reaches zero, which causes the zero-crossing comparator to turn off the N-channel MOSFET, as well. The peak inductor current is set by the voltage on the COMP pin. The COMP pin is the output of a transconductance error amplifier that compares the feedback voltage with an internal 0.8 V reference. Rev. 0 | Page 12 of 32 ADP2105/ADP2106/ADP2107 SLOPE COMPENSATION Slope compensation stabilizes the internal current control loop of the ADP2105/ADP2106/ADP2107 when operating beyond 50% duty cycle to prevent sub-harmonic oscillations. It is implemented by summing a fixed scaled voltage ramp to the current sense signal during the on-time of the P-channel MOSFET switch. The slope compensation ramp value determines the minimum inductor that can be used to prevent sub-harmonic oscillations at a given output voltage. The slope compensation ramp values for ADP2105/ADP2106/ADP2107 follow. For more information, see the Inductor Selection section. For the ADP2105: Slope Compensation Ramp Value = 0.72 A/μs For the ADP2106: Slope Compensation Ramp Value = 1.07 A/μs For the ADP2107: Slope Compensation Ramp Value = 1.38 A/μs Short Circuit Protection The ADP2105/ADP2106/ADP2107 include frequency foldback to prevent output current run-away on a hard short. When the voltage at the feedback pin falls below 0.3 V, indicating the possibility of a hard short at the output, the switching frequency is reduced to 1/4 of the internal oscillator frequency. The reduction in the switching frequency gives more time for the inductor to discharge, preventing a runaway of output current. Undervoltage Lockout (UVLO) To protect against deep battery discharge, undervoltage lockout circuitry is integrated on the ADP2105/ADP2106/ADP2107. If the input voltage drops below the 2.2 V UVLO threshold, the ADP2105/ADP2106/ADP2107 shut down, and both the power switch and synchronous rectifier turn off. Once the voltage rises again above the UVLO threshold, the soft start period is initiated, and the part is enabled. Thermal Protection In the event that the ADP2105/ADP2106/ADP2107 junction temperatures rise above 140°C, the thermal shutdown circuit turns off the converter. Extreme junction temperatures can be the result of high current operation, poor circuit board design, and/or high ambient temperature. A 40°C hysteresis is included so that when thermal shutdown occurs, the ADP2105/ADP2106/ ADP2107 do not return to operation until the on-chip temperature drops below 100°C. When coming out of thermal shutdown, soft start is initiated. FEATURES Enable/Shutdown Drive EN high to turn on the ADP2105/ADP2106/ADP2107. Drive EN low to turn off the ADP2105/ADP2106/ADP2107, reducing input current below 0.1 μA. To force the ADP2105/ ADP2106/ADP2107 to automatically start when input power is applied, connect EN to IN. When shut down, the ADP2105/ ADP2106/ADP2107 discharge the soft start capacitor, causing a new soft start cycle every time they are re-enabled. Soft Start The ADP2105/ADP2106/ADP2107 include soft start circuitry to limit the output voltage rise time to reduce inrush current at startup. To set the soft start period, connect the soft start capacitor (CSS) from SS to AGND. When the ADP2105/ADP2106/ ADP2107 are disabled, or if the input voltage is below the undervoltage lockout threshold, CSS is internally discharged. When the ADP2105/ADP2106/ADP2107 are enabled, CSS is charged through an internal 0.8 μA current source, causing the voltage at SS to rise linearly. The output voltage rises linearly with the voltage at SS. Synchronous Rectification In addition to the P-channel MOSFET switch, the ADP2105/ ADP2106/ADP2107 include an integrated N-channel MOSFET synchronous rectifier. The synchronous rectifier improves efficiency, especially at low output voltage, and reduces cost and board space by eliminating the need for an external rectifier. Current Limit The ADP2105/ADP2106/ADP2107 have protection circuitry to limit the direction and amount of current flowing through the power switch and synchronous rectifier. The positive current limit on the power switch limits the amount of current that can flow from the input to the output, while the negative current limit on the synchronous rectifier prevents the inductor current from reversing direction and flowing out of the load. Rev. 0 | Page 13 of 32 ADP2105/ADP2106/ADP2107 COMP 5 SOFT START REFERENCE 0.8V 14 IN 9 PWIN2 SS 6 CURRENT SENSE AMPLIFIER 13 PWIN1 FB1 16 OUT_SENSE1 16 GM ERROR AMP PWM/ PFM CONTROL FOR PRESET VOLTAGES OPTIONS ONLY CURRENT LIMIT AGND 7 GND 2 GND 3 GND 4 NC 8 GND 15 DRIVER AND ANTISHOOT THROUGH 10 LX1 12 LX2 SLOPE COMPENSATION OSCILLATOR ZERO CROSS COMPARATOR EN 1 THERMAL SHUTDOWN 11 PGND 1FB FOR ADP210x-ADJ (ADJUSTABLE VERSION) AND OUT_SENSE FOR ADP210x-xx (FIXED VERSION). Figure 34. Block Diagram of the ADP2105/ADP2106/ADP2107 Rev. 0 | Page 14 of 32 06079-037 ADP2105/ADP2106/ADP2107 APPLICATIONS INFORMATION EXTERNAL COMPONENT SELECTION The external component selection for the ADP2105/ADP2106/ ADP2107 application circuits shown in Figure 35 and Figure 36 depend on input voltage, output voltage, and load current requirements. Additionally, tradeoffs between performance parameters like efficiency and transient response can be made by varying the choice of external components. into account when calculating resistor values. The FB bias current can be ignored for a higher divider string current, but this degrades efficiency at very light loads. To limit output voltage accuracy degradation due to FB bias current to less than 0.05% (0.5% maximum), ensure that the divider string current is greater than 20 μA. To calculate the desired resistor values, first determine the value of the bottom divider string resistor, RBOT, by RBOT = VFB I STRING SETTING THE OUTPUT VOLTAGE The output voltage of ADP2105/ADP2106/ADP2107-ADJ is externally set by a resistive voltage divider from the output voltage to FB. The ratio of the resistive voltage divider sets the output voltage, while the absolute value of those resistors sets the divider string current. For lower divider string currents, the small 10 nA (0.1 μA maximum) FB bias current should be taken where: VFB = 0.8 V, the internal reference. ISTRING is the resistor divider string current. 0.1μF 10Ω VIN INPUT VOLTAGE = 2.7V TO 5.5V CIN1 VOUT 16 15 14 13 ON OFF OUT_SENSE GND 1 EN 2 GND 3 GND 4 GND IN PWIN1 LX2 12 L OUTPUT VOLTAGE = 1.2V, 1.5V, 1.8V, 3.3V VOUT ADP2105/ ADP2106/ ADP2107 PGND 11 LX1 10 PWIN2 9 VIN COUT LOAD COMP 5 SS 6 AGND NC 7 8 CIN2 RCOMP CCOMP CSS NC = NO CONNECT Figure 35. Typical Applications Circuit for Fixed Output Voltage Options (ADP2105/ADP2106/ADP2107-xx) 0.1μF 10Ω VIN INPUT VOLTAGE = 2.7V TO 5.5V CIN1 FB 16 15 14 13 ON OFF 1 FB EN GND GND GND COMP 5 GND IN PWIN1 LX2 12 L OUTPUT VOLTAGE = 0.8V TO VIN RTOP VIN CIN2 FB RBOT COUT LOAD 2 3 ADP2105/ ADP2106/ ADP2107 PGND 11 LX1 10 PWIN2 9 4 SS 6 AGND NC 7 8 RCOMP CCOMP CSS NC = NO CONNECT Figure 36. Typical Applications Circuit for Adjustable Output Voltage Option (ADP2105/ADP2106/ADP2107-ADJ) Rev. 0 | Page 15 of 32 06079-038 06079-065 ADP2105/ADP2106/ADP2107 Once RBOT is determined, calculate the value of the top resistor, RTOP, by Ensure that the maximum rms current of the inductor is greater than the maximum load current, and the saturation current of the inductor is greater than the peak current limit of the converter used in the application. Table 5. Minimum Inductor Value for Common Output Voltage Options for the ADP2105 (1 A) VOUT 1.2 V 1.5 V 1.8 V 2.5 V 3.3 V 2.7 V 1.67 μH 1.68 μH 2.02 μH 2.80 μH 3.70 μH 3.6 V 2.00 μH 2.19 μH 2.25 μH 2.80 μH 3.70 μH VIN 4.2 V 2.14 μH 2.41 μH 2.57 μH 2.80 μH 3.70 μH 5.5 V 2.35 μH 2.73 μH 3.03 μH 3.41 μH 3.70 μH ⎡V − VFB ⎤ RTOP = RBOT ⎢ OUT ⎥ ⎦ ⎣ VFB The ADP2105/ADP2106/ADP2107-xx (where xx represents the fixed output voltage) include the resistive voltage divider internally, reducing the external circuitry required. Connect the OUT_SENSE to the output voltage as close as possible to the load for improved load regulation. INDUCTOR SELECTION The high switching frequency of ADP2105/ADP2106/ADP2107 allows for minimal output voltage ripple even with small inductors. The sizing of the inductor is a trade-off between efficiency and transient response. A small inductor leads to larger inductor current ripple that provides excellent transient response but degrades efficiency. Due to the high switching frequency of ADP2105/ADP2106/ADP2107, shielded ferrite core inductors are recommended for their low core losses and low EMI. As a guideline, the inductor peak-to-peak current ripple, ΔIL, is typically set to 1/3 of the maximum load current for optimal transient response and efficiency. ΔI L = VOUT × (V IN − VOUT ) I LOAD ( MAX ) ≈ 3 V IN × f SW × L Table 6. Minimum Inductor Value for Common Output Voltage Options for the ADP2106 (1.5 A) VOUT 1.2 V 1.5 V 1.8 V 2.5 V 3.3 V 2.7 V 1.11 μH 1.25 μH 1.49 μH 2.08 μH 2.74 μH 3.6 V 2.33 μH 1.46 μH 1.50 μH 2.08 μH 2.74 μH VIN 4.2 V 2.43 μH 1.61 μH 1.71 μH 2.08 μH 2.74 μH 5.5 V 1.56 μH 1.82 μH 2.02 μH 2.27 μH 2.74 μH ⇒ LIDEAL = 2.5 × VOUT × (VIN − VOUT ) μH VIN × I LOAD ( MAX ) Table 7. Minimum Inductor Value for Common Output Voltage Options for the ADP2107 (2 A) VOUT 1.2 V 1.5 V 1.8 V 2.5 V 3.3 V 2.7 V 0.83 μH 0.99 μH 1.19 μH 1.65 μH 2.18 μH 3.6 V 1.00 μH 1.09 μH 1.19 μH 1.65 μH 2.18 μH VIN 4.2 V 1.07 μH 1.21 μH 1.29 μH 1.65 μH 2.18 μH 5.5 V 1.17 μH 1.36 μH 1.51 μH 1.70 μH 2.18 μH where fSW is the switching frequency (1.2 MHz). The ADP2105/ADP2106/ADP2107 use slope compensation in the current control loop to prevent subharmonic oscillations when operating beyond 50% duty cycle. The fixed slope compensation limits the minimum inductor value as a function of output voltage. For the ADP2105: L > (1.12 μH/V) × VOUT For the ADP2106: L > (0.83 μH/V) × VOUT For the ADP2107: L > (0.66 μH/V) × VOUT Also, 4.7 μH or larger inductors are not recommended because they may cause instability in discontinuous conduction mode under light load conditions. Finally, it is important that the inductor be capable of handling the maximum peak inductor current, IPK, determined by the following equation: ⎛ ΔI ⎞ I PK = I LOAD ( MAX ) + ⎜ L ⎟ ⎝2⎠ Table 8. Inductor Recommendations for the ADP2105/ ADP2106/ADP2107 Vendor Sumida Toko Small-Sized Inductors ( < 5 mm × 5 mm) CDRH2D14, 3D16, 3D28 1069AS-DB3018, 1098AS-DE2812, 1070AS-DB3020 LPS3015, LPS4012, DO3314 SD3110, SD3112, SD3114, SD3118, SD3812, SD3814 Large-Sized Inductors ( > 5 mm × 5 mm) CDRH4D18, 4D22, 4D28, 5D18, 6D12 D52LC, D518LC, D62LCB DO1605T SD10, SD12, SD14, SD52 Coilcraft Cooper Bussmann Rev. 0 | Page 16 of 32 ADP2105/ADP2106/ADP2107 OUTPUT CAPACITOR SELECTION The output capacitor selection affects both the output voltage ripple and the loop dynamics of the converter. For a given loop crossover frequency (the frequency at which the loop gain drops to 0 dB), the maximum voltage transient excursion (overshoot) is inversely proportional to the value of the output capacitor. Therefore, larger output capacitors result in improved load transient response. To minimize the effects of the dc-to-dc converter switching, the crossover frequency of the compensation loop should be less than 1/10 of the switching frequency. Higher crossover frequency leads to faster settling time for a load transient response, but it can also cause ringing due to poor phase margin. Lower crossover frequency helps to provide stable operation but needs large output capacitors to achieve competitive overshoot specifications. Therefore, the optimal crossover frequency for the control loop of ADP2105/ADP2106/ADP2107 is 80 kHz, 1/15 of the switching frequency. For a crossover frequency of 80 kHz, Figure 37 shows the maximum output voltage excursion during a 1A load transient, as the product of the output voltage and the output capacitor is varied. Choose the output capacitor based on the desired load transient response and target output voltage. 18 17 16 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 15 It is also important, while choosing output capacitors, to account for the loss of capacitance due to output voltage dc bias. Figure 38 shows the loss of capacitance due to output voltage dc bias for a few X5R MLCC capacitors from Murata. 20 0 CAPACITANCE CHANGE (%) –20 1 –40 3 –60 2 –80 –100 0 2 VOLTAGE (VDC) 4 6 Figure 38. % Drop-In Capacitance vs. DC Bias for Ceramic Capacitors (Information Provided by Murata Corporation) % OVERSHOOT OF OUTPUT VOLTAGE For example, to get 20 μF output capacitance at an output voltage of 2.5 V, based on Figure 38, as well as giving some margin for temperature variance, it is suggested that a 22 μF and a 10 μF capacitor be used in parallel to ensure that the output capacitance is sufficient under all conditions for stable behavior. Table 9. Recommended Input and Output Capacitor Selection for the ADP2105/ADP2106/ADP2107 Capacitor 4.7 μF 10 V X5R 0805 10 μF 10 V X5R 0805 22 μF 6.3 V X5R 0805 Vendor Murata Taiyo Yuden GRM21BR61A475K LMK212BJ475KG GRM21BR61A106K GRM21BR60J226M LMK212BJ106KG JMK212BJ226MG 20 25 30 35 40 45 50 55 60 65 70 OUTPUT CAPACITOR × OUTPUT VOLTAGE (μC) 06079-070 Figure 37. % Overshoot for a 1 A Load Transient Response vs. Output Capacitor × Output Voltage INPUT CAPACITOR SELECTION The input capacitor reduces input voltage ripple caused by the switch currents on the PWIN pins. Place the input capacitors as close as possible to the PWIN pins. Select an input capacitor capable of withstanding the rms input current for the maximum load current in your application. For the ADP2105, it is recommended that each PWIN pin be bypassed with a 4.7 μF or larger input capacitor. For the ADP2106, bypass the PWIN pins with a 10 μF and a 4.7 μF capacitor, and for the ADP2107, bypass each PWIN pin with a 10 μF capacitor. As with the output capacitor, a low ESR ceramic capacitor is recommended to minimize input voltage ripple. X5R or X7R dialectrics are recommended, with a voltage rating of 6.3 V or 10 V. Y5V and Z5U dialectrics are not recommended, due to their poor temperature and dc bias characteristics. Refer to Table 9 for input capacitor recommendations. For example, if the desired 1A load transient response (overshoot) is 5% for an output voltage of 2.5 V, then from Figure 37 Output Capacitor × Output Voltage = 50 μC ⇒ Output Capacitor = 50 μ C 2 .5 ≈ 20 μ F The ADP2105/ADP2106/ADP2107 have been designed for operation with small ceramic output capacitors that have low ESR and ESL, thus comfortably able to meet tight output voltage ripple specifications. X5R or X7R dialectrics are recommended with a voltage rating of 6.3 V or 10 V. Y5V and Z5U dialectrics are not recommended, due to their poor temperature and dc bias characteristics. Table 9 shows a list of recommended MLCC capacitors from Murata and Taiyo Yuden. Rev. 0 | Page 17 of 32 06079-060 14.7µF 0805 X5R MURATA GRM21BR61A475K 210µF 0805 X5R MURATA GRM21BR61A106K 322µF 0805 X5R MURATA GRM21BR60J226M ADP2105/ADP2106/ADP2107 INPUT FILTER The IN pin is the power source for the ADP2105/ADP2106/ ADP2107 internal circuitry, including the voltage reference and current sense amplifier that are sensitive to power supply noise. To prevent high frequency switching noise on the PWIN pins from corrupting the internal circuitry of the ADP2105/ADP2106/ ADP2107, a low-pass RC filter should be placed between the IN pin and the PWIN1 pin. The suggested input filter consists of a small 0.1 μF ceramic capacitor placed between IN and AGND and a 10 Ω resistor placed between IN and PWIN1. This forms a 150 kHz low-pass filter between PWIN1 and IN that prevents any high frequency noise on PWIN1 from coupling into the IN pin. The transconductance error amplifier drives the compensation network that consists of a resistor (RCOMP) and capacitor (CCOMP) connected in series to form a pole and a zero, as shown in the following equation: ⎛ 1 ZCOMP (s) = ⎜ RCOMP + ⎜ sCCOMP ⎝ ⎞ ⎛ 1 + sRCOMP CCOMP ⎟=⎜ ⎟⎜ sCCOMP ⎠⎝ ⎞ ⎟ ⎟ ⎠ At the crossover frequency, the gain of the open loop transfer function is unity. This yields the following equation for the compensation network impedance at the crossover frequency: ⎛ (2π )FCROSS ⎞⎛ COUTVOUT ⎟⎜ ZCOMP (FCROSS ) = ⎜ ⎜ GG ⎟⎜ V m CS REF ⎝ ⎠⎝ where: ⎞ ⎟ ⎟ ⎠ SOFT START The ADP2105/ADP2106/ADP2107 include soft start circuitry to limit the output voltage rise time to reduce inrush current at startup. To set the soft start period, connect a soft start capacitor (CSS) from SS to AGND. The soft start period varies linearly with the size of the soft start capacitor, as shown in the following equation: TSS = CSS × 109 ms To get a soft start period of 1 ms, a 1 nF capacitor must be connected between SS and AGND. FCROSS = 80 kHz, the crossover frequency of the loop. COUTVOUT is determined from the Output Capacitor Selection section. To ensure that there is sufficient phase margin at the crossover frequency, place the Compensator Zero at 1/4 of the crossover frequency, as shown in the following equation: F (2 π)⎛ CROSS ⎜ ⎝4 ⎞R ⎟ COMP CCOMP = 1 ⎠ LOOP COMPENSATION The ADP2105/ADP2106/ADP2107 utilize a transconductance error amplifier to compensate the external voltage loop. The open loop transfer function at angular frequency, s, is given by Solving the above two simultaneous equations yields the value for the compensation resistor and compensation capacitor, as shown in the following equation: ⎛ (2 π)FCROSS RCOMP = 0.8 ⎜ ⎜ GG m CS ⎝ ⎞⎛ COUT VOUT ⎟⎜ ⎟⎜ V REF ⎠⎝ ⎞ ⎟ ⎟ ⎠ ⎛Z (s) ⎞⎛ V H (s) = GmGCS ⎜ COMP ⎟⎜ REF ⎜ sC ⎟⎜ OUT ⎠⎝ VOUT ⎝ where: ⎞ ⎟ ⎟ ⎠ CCOMP = 2 πFCROSS RCOMP VREF is the internal reference voltage (0.8 V). VOUT is the nominal output voltage. ZCOMP(s) is the impedance of the compensation network at the angular frequency, s. COUT is the output capacitor. Gm is the transconductance of the error amplifier (50 μA/V nominal). GCS is the effective transconductance of the current loop. GCS = 1.875 A/V for the ADP2105. GCS = 2.8125 A/V for the ADP2106. GCS = 3.625 A/V for the ADP2107. Rev. 0 | Page 18 of 32 ADP2105/ADP2106/ADP2107 BODE PLOTS 60 50 40 30 LOOP GAIN 0 ADP2106 60 50 40 LOOP GAIN ADP2105 0 45 PHASE MARGIN = 49° LOOP PHASE 90 135 180 LOOP PHASE (Degrees) 45 PHASE MARGIN = 48° LOOP PHASE 90 135 180 30 20 10 20 10 0 0 CROSSOVER OUTPUT VOLTAGE = 1.8V FREQUENCY = 87kHz –10 INPUT VOLTAGE = 5.5V LOAD CURRENT = 1A –20 INDUCTOR = 2.2µH (LPS4012) OUTPUT CAPACITOR = 22µF + 22µF –30 COMPENSATION RESISTOR = 180kΩ COMPENSATION CAPACITOR = 56pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. 300 06079-055 CROSSOVER OUTPUT VOLTAGE = 1.2V FREQUENCY = 79kHz INPUT VOLTAGE = 5.5V LOAD CURRENT = 1A –20 INDUCTOR = 3.3µH (SD3814) OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF –30 COMPENSATION RESISTOR = 267kΩ COMPENSATION CAPACITOR = 39pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. –10 300 06079-058 Figure 39. ADP2106 Bode Plot at VIN = 5.5 V, VOUT = 1.8 V and Load = 1 A 60 50 40 30 LOOP GAIN 0 Figure 42. ADP2105 Bode Plot at VIN = 5.5 V, VOUT = 1.2 V and Load = 1 A 60 50 40 LOOP GAIN PHASE MARGIN = 65° LOOP PHASE 0 45 90 135 180 ADP2106 ADP2107 LOOP PHASE (Degrees) 45 PHASE MARGIN = 52° LOOP PHASE 90 135 180 30 20 10 0 20 10 0 CROSSOVER OUTPUT VOLTAGE = 1.8V –10 INPUT VOLTAGE = 3.6V FREQUENCY = 83kHz LOAD CURRENT = 1A –20 INDUCTOR = 2.2µH (LPS4012) OUTPUT CAPACITOR = 22µF + 22µF –30 COMPENSATION RESISTOR = 180kΩ COMPENSATION CAPACITOR = 56pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. 300 06079-056 CROSSOVER OUTPUT VOLTAGE = 2.5V –10 INPUT VOLTAGE = 5V FREQUENCY = 76kHz LOAD CURRENT = 1A –20 INDUCTOR = 2µH (D62LCB) OUTPUT CAPACITOR = 10µF + 4.7µF –30 COMPENSATION RESISTOR = 70kΩ COMPENSATION CAPACITOR = 120pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 10% OVERSHOOT FOR A 1A LOAD TRANSIENT. 300 06079-059 Figure 40. ADP2106 Bode Plot at VIN = 3.6 V, VOUT = 1.8 V, and Load = 1 A 60 50 LOOP GAIN 40 30 0 Figure 43. ADP2107 Bode Plot at VIN = 5 V, VOUT = 2.5 V and Load = 1 A 60 50 40 LOOP GAIN PHASE MARGIN = 70° LOOP PHASE 0 45 90 135 180 ADP2105 ADP2107 LOOP PHASE (Degrees) 45 PHASE MARGIN = 51° LOOP PHASE 90 135 180 30 20 10 0 20 10 0 CROSSOVER OUTPUT VOLTAGE = 1.2V FREQUENCY = 71kHz –10 INPUT VOLTAGE = 3.6V LOAD CURRENT = 1A –20 INDUCTOR = 3.3µH (SD3814) OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF –30 COMPENSATION RESISTOR = 267kΩ COMPENSATION CAPACITOR = 39pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. 300 06079-057 CROSSOVER OUTPUT VOLTAGE = 3.3V –10 INPUT VOLTAGE = 5V FREQUENCY = 67kHz LOAD CURRENT = 1A –20 INDUCTOR = 2.5µH (CDRH5D28) OUTPUT CAPACITOR = 10µF + 4.7µF –30 COMPENSATION RESISTOR = 70kΩ COMPENSATION CAPACITOR = 120pF –40 1 10 100 (kHz) NOTES 1. EXTERNAL COMPONENTS WERE CHOSEN FOR A 10% OVERSHOOT FOR A 1A LOAD TRANSIENT. 300 06079-069 Figure 41. ADP2105 Bode Plot at VIN = 3.6 V, VOUT = 1.2 V, and Load = 1 A Figure 44. ADP2107 Bode Plot at VIN = 5 V, VOUT = 3.3 V, and Load = 1 A Rev. 0 | Page 19 of 32 LOOP PHASE (Degrees) LOOP GAIN (dB) LOOP GAIN (dB) LOOP PHASE (Degrees) LOOP GAIN (dB) LOOP GAIN (dB) LOOP PHASE (Degrees) LOOP GAIN (dB) LOOP GAIN (dB) ADP2105/ADP2106/ADP2107 LOAD TRANSIENT RESPONSE OUTPUT CURRENT 3 OUTPUT CURRENT 3 CH2 LOW –51mV OUTPUT VOLTAGE (AC-COUPLED) 2 OUTPUT VOLTAGE (AC-COUPLED) 2 CH2 LOW –93mV 06079-075 LX NODE (SWITCH NODE) CH1 2V CH2 50mV~ M 10µs CH3 1A A CH3 0.5A LX NODE (SWITCH NODE) CH1 2V CH2 50mV~ M 10µs CH3 1A A CH3 0.5A OUTPUT CAPACITOR: 22µF + 22µF + 4.7µF INDUCTOR: SD14, 2.5µH COMPENSATION RESISTOR: 270kΩ COMPENSATION CAPACITOR: 39pF OUTPUT CAPACITOR: 22µF + 4.7µF INDUCTOR: SD14, 2.5µH COMPENSATION RESISTOR: 135kΩ COMPENSATION CAPACITOR: 82pF Figure 45. 1 A Load Transient Response for ADP2105-1.2 with External Components Chosen for 5% Overshoot OUTPUT CURRENT 3 3 Figure 48. 1 A Load Transient Response for ADP2105-1.2 with External Components Chosen for 10% Overshoot OUTPUT CURRENT CH2 LOW –112mV 2 2 CH2 LOW –164mV OUTPUT VOLTAGE (AC-COUPLED) OUTPUT VOLTAGE (AC-COUPLED) 06079-077 LX NODE (SWITCH NODE) CH1 2V CH2 100mV~ M 10µs CH3 1A A CH3 0.5A LX NODE (SWITCH NODE) CH1 2V CH2 100mV~ M 10µs CH3 1A A CH3 0.5A OUTPUT CAPACITOR: 22µF + 22µF INDUCTOR: SD3814, 3.3µH COMPENSATION RESISTOR: 270kΩ COMPENSATION CAPACITOR: 39pF OUTPUT CAPACITOR: 10µF + 10µF INDUCTOR: SD3814, 3.3µH COMPENSATION RESISTOR: 135kΩ COMPENSATION CAPACITOR: 82pF Figure 46. 1 A Load Transient Response for ADP2105-1.8 with External Components Chosen for 5% Overshoot OUTPUT CURRENT 3 Figure 49. 1 A Load Transient Response for ADP2105-1.8 with External Components Chosen for 10% Overshoot OUTPUT CURRENT 3 OUTPUT VOLTAGE (AC-COUPLED) 2 CH2 LOW –178mV 2 CH2 LOW OUTPUT VOLTAGE (AC-COUPLED) –308mV LX NODE (SWITCH NODE) CH1 2V CH2 100mV~ M 10µs CH3 1A A CH3 0.5A LX NODE (SWITCH NODE) CH1 2V CH2 200mV~ M 10µs CH3 1A A CH3 0.5A OUTPUT CAPACITOR: 22µF + 4.7µF INDUCTOR: CDRH5D18, 4.1µH COMPENSATION RESISTOR: 270kΩ COMPENSATION CAPACITOR: 39pF OUTPUT CAPACITOR: 10µF + 4.7µF INDUCTOR: CDRH5D18, 4.1µH COMPENSATION RESISTOR: 135kΩ COMPENSATION CAPACITOR: 82pF Figure 47. 1 A Load Transient Response for ADP2105-3.3 with External Components Chosen for 5% Overshoot Rev. 0 | Page 20 of 32 Figure 50. 1 A Load Transient Response for ADP2105-3.3 with External Components Chosen for 10% Overshoot 06079-080 06079-079 1 1 06079-078 1 1 06079-076 1 1 ADP2105/ADP2106/ADP2107 EFFICIENCY CONSIDERATIONS Efficiency is defined as the ratio of output power to input power. The high efficiency of the ADP2105/ADP2106/ADP2107 has two distinct advantages. First, only a small amount of power is lost in the dc-to-dc converter package that reduces thermal constraints. In addition, high efficiency delivers the maximum output power for the given input power, extending battery life in portable applications. There are four major sources of power loss in dc-to-dc converters like the ADP2105/ADP2106/ADP2107. • • • • Power switch conduction losses Inductor losses Switching losses Transition losses The amount of power loss can by calculated by PSW = (CGATE − P + CGATE − N) × VIN2 × fSW where: (CGATE − P + CGATE − N) ~ 600 pF. fSW = 1.2 MHz, the switching frequency. Transition Losses Transition losses occur because the P-channel MOSFET power switch cannot turn on or turn off instantaneously. At the middle of a LX node transition, the power switch is providing all the inductor current, while the source to drain voltage of the power switch is half the input voltage, resulting in power loss. Transition losses increase with load current and input voltage and occur twice for each switching cycle. The amount of power loss can be calculated by PTRAN = VIN × I OUT × (tON + tOFF ) × f SW 2 Power Switch Conduction Losses Power switch conduction losses are caused by the flow of output current through the P-channel power switch and the N-channel synchronous rectifier, which have internal resistances (RDS(ON)) associated with them. The amount of power loss can be approximated by PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2 where D = VOUT/VIN. The internal resistance of the power switches increases with temperature but decreases with higher input voltage. Figure 19 in the Typical Performance Characteristics section shows the change in RDS(ON) vs. input voltage, while Figure 27 in the Typical Performance Characteristics section shows the change in RDS(ON) vs. temperature for both power devices. where tON and tOFF are the rise time and fall time of the LX node, which are approximately 3 ns. THERMAL CONSIDERATIONS In most applications, the ADP2105/ADP2106/ADP2107 do not dissipate a lot of heat due to their high efficiency. However, in applications with high ambient temperature, low supply voltage, and high duty cycle, the heat dissipated in the package is large enough that it can cause the junction temperature of the die to exceed the maximum junction temperature of 125°C. Once the junction temperature exceeds 140°C, the converter goes into thermal shutdown. It recovers only after the junction temperature has decreased below 100°C to prevent any permanent damage. Therefore, thermal analysis for the chosen application solution is very important to guarantee reliable performance over all conditions. The junction temperature of the die is the sum of the ambient temperature of the environment and the temperature rise of the package due to the power dissipation, as shown in the following equation: TJ = TA + TR where: TJ is the junction temperature. TA is the ambient temperature. TR is the rise in temperature of the package due to power dissipation in it. Inductor Losses Inductor conduction losses are caused by the flow of current through the inductor, which has an internal resistance (DCR) associated with it. Larger sized inductors have smaller DCR, which can improve inductor conduction losses. Inductor core losses are related to the magnetic permeability of the core material. Because the ADP2105/ADP2106/ADP2107 are high switching frequency dc-to-dc converters, shielded ferrite core material is recommended for its low core losses and low EMI. The total amount of inductor power loss can be calculated by PL = DCR × IOUT2 + Core Losses Switching Losses Switching losses are associated with the current drawn by the driver to turn on and turn off the power devices at the switching frequency. Each time a power device gate is turned on and turned off, the driver transfers a charge ΔQ from the input supply to the gate and then from the gate to ground. Rev. 0 | Page 21 of 32 ADP2105/ADP2106/ADP2107 The rise in temperature of the package is directly proportional to the power dissipation in the package. The proportionality constant for this relationship is defined as the thermal resistance from the junction of the die to the ambient temperature, as shown in the following equation: TR = θJA × PD where: TR is the rise in temperature of the package. PD is the power dissipation in the package. θJA is the thermal resistance from the junction of the die to the ambient temperature of the package. For example, consider an application where the ADP2107-1.8 is used with an input voltage of 3.6 V and a load current of 2 A. Also, assume that the maximum ambient temperature is 85°C. At a load current of 2 A, the most significant contributor of power dissipation in the dc-to-dc converter package is the conduction loss of the power switches. Using the graph of switch resistance vs. temperature (see Figure 27), as well as the equation of power loss given in the Power Switch Conduction Losses section, the power dissipation in the package can be calculated by PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2 = [109 mΩ × 0.5 + 90 mΩ × 0.5] × (2 A)2 ~ 400 mW The θJA for the LFCSP_VQ package is 40°C/W, as shown in Table 3. Thus, the rise in temperature of the package due to power dissipation is TR = θJA × PD = 40°C/W × 0.40 W = 16°C The junction temperature of the converter is TJ = TA + TR = 85°C + 16°C = 101°C which is below the maximum junction temperature of 125°C. Thus, this application operates reliably from a thermal point of view. RBOT = 2. See whether the output voltage desired is available as a fixed output voltage option. Because 2 V is not one of the fixed output voltage options available, choose the adjustable version of ADP2106. The first step in external component selection for an adjustable version converter is to calculate the resistance of the resistive voltage divider that sets the output voltage. 0.8 V VFB = = 40 kΩ I STRING 20 μA 3. ⎡ 2 V − 0.8 V ⎤ ⎡V − VFB ⎤ RTOP = RBOT ⎢ OUT ⎥ = 60 kΩ ⎥ = 40 kΩ × ⎢ ⎢ 0.8 V ⎥ ⎣ VFB ⎦ ⎣ ⎦ 4. Calculate the minimum inductor value as follows: For the ADP2106: L > (0.83 μH/V) × VOUT L > 0.83 μH/V × 2 V L > 1.66 μH Next, calculate the ideal inductor value that sets the inductor peak-to-peak current ripple, ΔIL, to1/3 of the maximum load current at the maximum input voltage. LIDEAL = 2.5 × VOUT × (VIN − VOUT ) μH = VIN × I LOAD ( MAX ) 2.5 × 2 × (4.2 − 2) μH = 2.18 μH 4. 2 × 1 .2 The closest standard inductor value is 2.2 μH. The maximum rms current of the inductor should be greater than 1.2 A, and the saturation current of the inductor should be greater than 2 A. One inductor that meets these criteria is the LPS4012-2.2 μH from Coilcraft. 5. Choose the output capacitor based on the transient response requirements. The worst-case load transient is 1.2 A, for which the overshoot must be less than 100 mV, which is 5% of the output voltage. Therefore, for a 1 A load transient, the overshoot must be less than 4% of the output voltage. For these conditions, Figure 37 gives Output Capacitor × Output Voltage = 60 μC DESIGN EXAMPLE Consider an application with the following specifications: Input Voltage = 3.6 V to 4.2 V. Output Voltage = 2 V. Typical Output Current = 600 mA. Maximum Output Current = 1.2 A. Soft Start Time = 2 ms. Overshoot ≤ 100 mV under all load transient conditions. 1. Choose the dc-to-dc converter that satisfies the maximum output current requirement. Because the maximum output current for this application is 1.2 A, the ADP2106 with a maximum output current of 1.5 A is ideal for this application. ⇒ Output Capacitor = 60 μC 2 .0 V ≈ 30 μF Next, taking into account the loss of capacitance due to dc bias, as shown in Figure 38, two 22 μF X5R MLCC capacitors from Murata (GRM21BR60J226M) are sufficient for this application. Rev. 0 | Page 22 of 32 ADP2105/ADP2106/ADP2107 6. Because the ADP2106 is being used in this application, the input capacitors are 10 μF and 4.7 μF X5R Murata capacitors (GRM21BR61A106K and GRM21BR61A475K). The input filter consists of a small 0.1 μF ceramic capacitor placed between IN and AGND and a 10 Ω resistor placed between IN and PWIN1. Choose a soft start capacitor of 2 nF to achieve a soft start time of 2 ms. 9. Finally, the compensation resistor and capacitor can be calculated as 7. ⎛ (2 π)FCROSS RCOMP = 0.8 ⎜ ⎜ GG m CS ⎝ ⎞⎛ COUT VOUT ⎟⎜ ⎟⎜ V REF ⎠⎝ ⎞ ⎟ ⎟ ⎠ 8. ⎛ ⎞⎛ 30 μF × 2 V ⎞ (2 π) × 80 kHz ⎟⎜ ⎟ = 215 kΩ = 0 .8 ⎜ ⎜ 50 μA / V × 2.8125 A / V ⎟⎜ 0.8 V ⎟ ⎝ ⎠⎝ ⎠ CCOMP = 2 2 = = 39 pF πFCROSS RCOMP π × 80 kHz × 215 kΩ Rev. 0 | Page 23 of 32 ADP2105/ADP2106/ADP2107 EXTERNAL COMPONENT RECOMMENDATIONS Table 10. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with 10% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36) Part ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2105-1.2 ADP2105-1.5 ADP2105-1.8 ADP2105-3.3 ADP2106-1.2 ADP2106-1.5 ADP2106-1.8 ADP2106-3.3 ADP2107-1.2 ADP2107-1.5 ADP2107-1.8 ADP2107-3.3 1 VOUT (V) 0.9 1.2 1.5 1.8 2.5 3.3 0.9 1.2 1.5 1.8 2.5 3.3 0.9 1.2 1.5 1.8 2.5 3.3 1.2 1.5 1.8 3.3 1.2 1.5 1.8 3.3 1.2 1.5 1.8 3.3 CIN1 1 (μF) 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 10 10 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 CIN2 2 (μF) 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 10 10 10 10 10 10 10 10 4.7 4.7 4.7 4.7 10 10 10 10 10 10 10 10 COUT 3 (μF) 22 + 10 22 + 4.7 10 + 10 10 + 10 10 + 4.7 10 + 4.7 22 + 10 22 + 4.7 10 + 10 10 + 10 10 + 4.7 10 + 4.7 22 + 10 22 + 4.7 10 + 10 10 + 10 10 + 4.7 10 + 4.7 22 + 4.7 10 + 10 10 + 10 10 + 4.7 22 + 4.7 10 + 10 10 + 10 10 + 4.7 22 + 4.7 10 + 10 10 + 10 10 + 4.7 L (μH) 2.0 2.5 3.0 3.3 3.6 4.1 1.5 1.8 2.0 2.2 2.5 3.0 1.2 1.5 1.5 1.8 1.8 2.5 2.5 3.0 3.3 4.1 1.8 2.0 2.2 3.0 1.5 1.5 1.8 2.5 RCOMP (kΩ) 135 135 135 135 135 135 90 90 90 90 90 90 70 70 70 70 70 70 135 135 135 135 90 90 90 90 70 70 70 70 CCOMP (pF) 82 82 82 82 82 82 100 100 100 100 100 100 120 120 120 120 120 120 82 82 82 82 100 100 100 100 120 120 120 120 RTOP 4 (kΩ) 5 20 35 50 85 125 5 20 35 50 85 125 5 20 35 50 85 125 - RBOT 5 (kΩ) 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 - 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L. 2 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L. 3 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L. 22 μF 0805 X5R 6.3 V Murata–GRM21BR60J226ME39L. 4 0.5% accuracy resistor. 5 0.5% accuracy resistor. Rev. 0 | Page 24 of 32 ADP2105/ADP2106/ADP2107 Table 11. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with 5% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36) Part ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2105-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2106-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2107-ADJ ADP2105-1.2 ADP2105-1.5 ADP2105-1.8 ADP2105-3.3 ADP2106-1.2 ADP2106-1.5 ADP2106-1.8 ADP2106-3.3 ADP2107-1.2 ADP2107-1.5 ADP2107-1.8 ADP2107-3.3 1 VOUT (V) 0.9 1.2 1.5 1.8 2.5 3.3 0.9 1.2 1.5 1.8 2.5 3.3 0.9 1.2 1.5 1.8 2.5 3.3 1.2 1.5 1.8 3.3 1.2 1.5 1.8 3.3 1.2 1.5 1.8 3.3 CIN1 1 (μF) 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 10 10 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 CIN2 2 (μF) 4.7 4.7 4.7 4.7 4.7 4.7 10 10 10 10 10 10 10 10 10 10 10 10 4.7 4.7 4.7 4.7 10 10 10 10 10 10 10 10 COUT 3 (μF) 22 + 22 + 22 22 + 22 + 4.7 22 + 22 22 + 22 22 + 10 22 + 4.7 22 + 22 + 22 22 + 22 + 4.7 22 + 22 22 + 22 22 + 10 22 + 4.7 22 + 22 + 22 22 + 22 + 4.7 22 + 22 22 + 22 22 + 10 22 + 4.7 22 + 22 + 4.7 22 + 22 22 + 22 22 + 4.7 22 + 22 + 4.7 22 + 22 22 + 22 22 + 4.7 22 + 22 + 4.7 22 + 22 22 + 22 22 + 4.7 L (μH) 2.0 2.5 3.0 3.3 3.6 4.1 1.5 1.8 2.0 2.2 2.5 3.0 1.2 1.5 1.5 1.8 1.8 2.5 2.5 3.0 3.3 4.1 1.8 2.0 2.2 3.0 1.5 1.5 1.8 2.5 RCOMP (kΩ) 270 270 270 270 270 270 180 180 180 180 180 180 140 140 140 140 140 140 270 270 270 270 180 180 180 180 140 140 140 140 CCOMP (pF) 39 39 39 39 39 39 56 56 56 56 56 56 68 68 68 68 68 68 39 39 39 39 56 56 56 56 68 68 68 68 RTOP 4 (kΩ) 5 20 35 50 85 125 5 20 35 50 85 125 5 20 35 50 85 125 - RBOT 5 (kΩ) 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 40 - 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L 10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L 2 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L 10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L 3 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L 10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L 22μF 0805 X5R 6.3V Murata – GRM21BR60J226ME39L 4 0.5% Accuracy Resistor 5 0.5% Accuracy Resistor Rev. 0 | Page 25 of 32 ADP2105/ADP2106/ADP2107 CIRCUIT BOARD LAYOUT RECOMMENDATIONS Good circuit board layout is essential in obtaining the best performance from the ADP2105/ADP2106/ADP2107. Poor circuit layout degrades the output ripple, as well as the electromagnetic interference (EMI) and electromagnetic compatibility (EMC) performance. Figure 52 and Figure 53 show the ideal circuit board layout for the ADP2105/ADP2106/ADP2107. Use this layout to achieve the highest performance. Refer to the following guidelines if adjustments to the suggested layout are needed. • Use separate analog and power ground planes. Connect the ground reference of sensitive analog circuitry (such as compensation and output voltage divider components) to analog ground; connect the ground reference of power components (such as input and output capacitors) to power ground. In addition, connect both the ground planes to the exposed pad of the ADP2105/ADP2106/ADP2107. For each PWIN pin, place an input capacitor as close to the PWIN pin as possible and connect the other end to the closest power ground plane. Place the 0.1 μF, 10 Ω low-pass input filter between the IN pin and the PWIN1 pin, as close to the IN pin as possible. Ensure that the high current loops are as short and as wide as possible. Make the high current path from CIN through L, COUT, and the PGND plane back to CIN as short as possible. To accomplish this, ensure that the input and output capacitors share a common PGND plane. • • Also, make the high current path from PGND pin of the ADP2105/ADP2106/ADP2107 through L and COUT back to the PGND plane as short as possible. To do this, ensure that the PGND pin of the ADP2105/ADP2106/ADP2107 is tied to the PGND plane as close as possible to the input and output capacitors. Place the feedback resistor divider network as close as possible to the FB pin to prevent noise pickup. Try to minimize the length of trace connecting the top of the feedback resistor divider to the output while keeping away from the high current traces and the switch node (LX) that can lead to noise pickup. To reduce noise pickup, place an analog ground plane on either side of the FB trace. For the low fixed voltage options (1.2 V and 1.5 V), poor routing of the OUT_SENSE trace can lead to noise pickup, adversely affecting load regulation. This can be fixed by placing a 1 nF bypass capacitor close to the OUT_SENSE pin. The placement and routing of the compensation components are critical for proper behavior of the ADP2105/ADP2106/ ADP2107. The compensation components should be placed as close to the COMP pin as possible. It is advisable to use 0402-sized compensation components for closer placement, leading to smaller parasitics. Surround the compensation components with analog ground plane to prevent noise pickup. Also, ensure that the metal layer under the compensation components is the analog ground plane. • • • Rev. 0 | Page 26 of 32 ADP2105/ADP2106/ADP2107 EVALUATION BOARD EVALUATION BOARD SCHEMATIC (ADP2107-1.8) C7 0.1µF VCC OUT J1 U1 16 15 14 13 R3 10Ω VCC INPUT VOLTAGE = 2.7V TO 5.5V VIN C1 10µF1 GND OUT_SENSE 1 GND IN PWIN1 LX2 12 PGND 11 EN GND GND GND 5 6 7 EN R2 100kΩ 2 ADP2107-1.8 3 LX1 10 VCC PWIN2 9 17 8 1 L12 2µH 2 OUTPUT VOLTAGE = 1.8V, 2A VOUT R4 0Ω C3 22µF1 C4 22µF1 GND R5 NS 4 OUT C2 10µF1 COMP SS AGND PADDLE NC R1 140kΩ C6 68pF C5 1nF NC = NO CONNECT 1 MURATA X5R 0805 10μF: GRM21BR61A106KE19L 22μF: GRM21BR60J226ME39L 2 2μH INDUCTOR D62LCB TOKO Figure 51. Evaluation Board Schematic of the ADP2107-1.8 (Bold Traces Are High Current Paths) RECOMMENDED PCB BOARD LAYOUT (EVALUATION BOARD LAYOUT) JUMPER TO ENABLE ENABLE VIN 100kΩ PULL-DOWN INPUT GROUND GROUND INPUT CAPACITOR POWER GROUND PLANE PLACE THE FEEDBACK RESISTORS AS CLOSE TO THE FB PIN AS POSSIBLE. RTOP RBOT CONNECT THE GROUND RETURN OF ALL POWER COMPONENTS SUCH AS INPUT AND OUTPUT CAPACITORS TO THE POWER GROUND PLANE. OUTPUT CAPACITOR CIN LX COUT OUTPUT ADP2105/ADP2106/ADP2107 RCOMP CCOMP PLACE THE COMPENSATION COMPONENTS AS CLOSE TO THE COMP PIN AS POSSIBLE. CSS PGND LX CIN INDUCTOR (L) VOUT COUT OUTPUT CAPACITOR ANALOG GROUND PLANE CONNECT THE GROUND RETURN OF ALL SENSITIVE ANALOG CIRCUITRY SUCH AS COMPENSATION AND OUTPUT VOLTAGE DIVIDER TO THE ANALOG GROUND PLANE. POWER GROUND INPUT CAPACITOR 06079-045 Figure 52. Recommended Layout of Top Layer of ADP2105/ADP2106/ADP2107 Rev. 0 | Page 27 of 32 06079-044 ADP2105/ADP2106/ADP2107 ENABLE VIN GND GND ANALOG GROUND PLANE POWER GROUND PLANE INPUT VOLTAGE PLANE CONNECTING THE TWO PWIN PINS AS CLOSE AS POSSIBLE. CONNECT THE EXPOSED PAD OF THE ADP2105/ADP2106/ADP2107 TO A LARGE GROUND PLANE TO AID POWER DISSIPATION. VIN CONNECT THE PGND PIN TO THE POWER GROUND PLANE AS CLOSE TO THE ADP2105/ADP2106/ADP2107 AS POSSIBLE. VOUT Figure 53. Recommended Layout of Bottom Layer of ADP2105/ADP2106/ADP2107 Rev. 0 | Page 28 of 32 06079-046 FEEDBACK TRACE: THIS TRACE CONNECTS THE TOP OF THE RESISTIVE VOLTAGE DIVIDER ON THE FB PIN TO THE OUTPUT. PLACE THIS TRACE AS FAR AWAY FROM THE LX NODE AND HIGH CURRENT TRACES AS POSSIBLE TO PREVENT NOISE PICKUP. ADP2105/ADP2106/ADP2107 APPLICATION CIRCUITS 0.1μF 10Ω VIN INPUT VOLTAGE = 5V 10μF1 VOUT 16 15 14 13 ON OFF 1 OUT_SENSE GND IN PWIN1 LX2 12 2.5μH2 PGND 11 VOUT OUTPUT VOLTAGE = 3.3V LOAD 0A TO 2A EN GND GND GND COMP 5 2 ADP2107-3.3 3 10μF1 LX1 10 4.7μF1 4 PWIN2 9 SS 6 VIN 10μF1 1 MURATA AGND NC 7 8 70kΩ 120pF 1nF X5R 0805 10μF: GRM21BR61A106KE19L 4.7μF: GRM21BR61A475KA73L 2 SUMIDA CDRH5D28: 2.5μH NOTES 1. NC = NO CONNECT. 2. EXTERNAL COMPONENTS WERE CHOSEN FOR A 10% OVERSHOOT FOR A 1A LOAD TRANSIENT. Figure 54. Application Circuit—VIN = 5 V, VOUT = 3.3 V, LOAD = 0 A to 2 A 0.1μF 10Ω VIN INPUT VOLTAGE = 3.6V 10μF1 VOUT 16 15 14 13 ON OFF 1 OUT_SENSE GND IN PWIN1 LX2 12 1.5μH2 PGND 11 VOUT OUTPUT VOLTAGE = 1.5V LOAD 0A TO 2A EN GND GND GND COMP 5 2 ADP2107-1.5 3 22μF1 LX1 10 22μF1 4 PWIN2 9 SS 6 VIN 10μF1 1 MURATA AGND NC 7 8 140kΩ 68pF 1nF X5R 0805 10μF: GRM21BR61A106KE19L 22μF: GRM21BR60J226ME39L 2 TOKO D62LCB OR COILCRAFT LPS4012 NOTES 1. NC = NO CONNECT. 2. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. Figure 55. Application Circuit—VIN = 3.6 V, VOUT = 1.5 V, LOAD = 0 A to 2 A 0.1μF 10Ω VIN INPUT VOLTAGE = 2.7V TO 4.2V 4.7μF1 VOUT 16 15 14 13 ON OFF 1 OUT_SENSE GND IN PWIN1 LX2 12 2.7μH2 PGND 11 VOUT OUTPUT VOLTAGE = 1.8V LOAD 0A TO 1A EN GND GND GND COMP 5 2 ADP2105-1.8 3 22μF1 LX1 10 22μF1 4 PWIN2 9 SS 6 VIN 4.7μF1 1 MURATA AGND NC 7 8 270kΩ 39pF 1nF X5R 0805 4.7μF: GRM21BR61A475KA73L 22μF: GRM21BR60J226ME39L 2 TOKO 1098AS-DE2812: 2.7μH NOTES 1. NC = NO CONNECT. 2. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. Figure 56. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.8 V, LOAD = 0 A to 1 A Rev. 0 | Page 29 of 32 06079-049 06079-048 06079-047 ADP2105/ADP2106/ADP2107 0.1μF 10Ω VIN INPUT VOLTAGE = 2.7V TO 4.2V 4.7μF1 VOUT 16 15 14 13 ON OFF 1 OUT_SENSE GND IN PWIN1 LX2 12 2.4μH2 PGND 11 VOUT OUTPUT VOLTAGE = 1.2V LOAD 0A TO 1A EN GND GND GND COMP 5 2 ADP2105-1.2 3 22μF1 LX1 10 4.7μF1 4 PWIN2 9 SS 6 VIN 4.7μF1 1 MURATA AGND NC 7 8 135kΩ 82pF 1nF X5R 0805 4.7μF: GRM21BR61A475KA73L 22μF: GRM21BR60J226ME39L 2 TOKO 1069AS-DB3018HCT OR TOKO 1070AS-DB3020HCT NOTES 1. NC = NO CONNECT. 2. EXTERNAL COMPONENTS WERE CHOSEN FOR A 10% OVERSHOOT FOR A 1A LOAD TRANSIENT. Figure 57. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.2 V, LOAD = 0 A to 1 A 0.1μF 10Ω VIN INPUT VOLTAGE = 5V 10μF1 FB 16 15 14 13 ON OFF 1 FB EN GND GND GND GND IN PWIN1 LX2 12 2.5μH2 PGND 11 OUTPUT VOLTAGE = 2.5V 85kΩ LX1 10 PWIN2 9 VIN 4.7μF1 FB 40kΩ 10μF1 22μF1 LOAD 0A TO 1.5A 2 ADP2106-ADJ 3 4 COMP SS 5 6 AGND NC 7 8 180kΩ 56pF 1nF 1 MURATA X5R 0805 4.7μF: GRM21BR61A475KA73L 10μF: GRM21BR61A106KE19L 22μF: GRM21BR60J226ME39L 2 COILTRONICS SD14: 2.5μH NOTES 1. NC = NO CONNECT. 2. EXTERNAL COMPONENTS WERE CHOSEN FOR A 5% OVERSHOOT FOR A 1A LOAD TRANSIENT. 06079-050 Figure 58. Application Circuit—VIN = 5 V, VOUT = 2.5 V, LOAD = 0 A to 1.5 A Rev. 0 | Page 30 of 32 06079-051 ADP2105/ADP2106/ADP2107 OUTLINE DIMENSIONS 4.00 BSC SQ 0.60 MAX 0.60 MAX 0.65 BSC 3.75 BSC SQ 0.75 0.60 0.50 (BOTTOM VIEW) PIN 1 INDICATOR 13 12 16 PIN 1 INDICATOR 1 TOP VIEW EXPOSED PAD 9 8 5 4 2.25 2.10 SQ 1.95 0.25 MIN 1.95 BSC 12° MAX 1.00 0.85 0.80 0.80 MAX 0.65 TYP 0.05 MAX 0.02 NOM COMPLIANT TO JEDEC STANDARDS MO-220-VGGC Figure 59. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 4 mm × 4 mm Body, Very Thin Quad (CP-16-4) Dimensions shown in millimeters ORDERING GUIDE Model ADP2105ACPZ-1.2-R7 1 ADP2105ACPZ-1.5-R71 ADP2105ACPZ-1.8-R71 ADP2105ACPZ-3.3-R71 ADP2105ACPZ-R71 ADP2106ACPZ-1.2-R71 ADP2106ACPZ-1.5-R71 ADP2106ACPZ-1.8-R71 ADP2106ACPZ-3.3-R71 ADP2106ACPZ-R71 ADP2107ACPZ-1.2-R71 ADP2107ACPZ-1.5-R71 ADP2107ACPZ-1.8-R71 ADP2107ACPZ-3.3-R71 ADP2107ACPZ-R71 ADP2105-1.8-EVAL ADP2105-EVAL ADP2106-1.8-EVAL ADP2106-EVAL ADP2107-1.8-EVAL ADP2107-EVAL 1 Output Current 1A 1A 1A 1A 1A 1.5 A 1.5 A 1.5 A 1.5 A 1.5 A 2A 2A 2A 2A 2A Junction Temperature Range −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C Output Voltage 1.2 V 1.5 V 1.8 V 3.3 V ADJ 1.2 V 1.5 V 1.8 V 3.3 V ADJ 1.2 V 1.5 V 1.8 V 3.3 V ADJ 1.8 V Adjustable, but set to 2.5 V 1.8 V Adjustable, but set to 2.5 V 1.8 V Adjustable, but set to 2.5 V Package Description 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ 16-Lead LFCSP_VQ Evaluation Board Evaluation Board Evaluation Board Evaluation Board Evaluation Board Evaluation Board 010606-0 SEATING PLANE 0.35 0.30 0.25 0.20 REF COPLANARITY 0.08 Package Option CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 CP-16-4 Z = Pb-free part. Rev. 0 | Page 31 of 32 ADP2105/ADP2106/ADP2107 NOTES ©2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D06079-0-7/06(0) Rev. 0 | Page 32 of 32