36 V,1 A, Synchronous,
Step-Down DC-to-DC Regulator
ADP2441
Data Sheet
FEATURES
TYPICAL CIRCUIT CONFIGURATION
APPLICATIONS
CBST
C4
RBOTTOM
COMP
EN
FREQ
VIN
ADP2441
SS/TRK
FB
RCOMP
VCC
AGND
RTOP
BST
VOUT
VIN
VIN
VOUT
SW
CIN
COUT
PGND
10581-001
CCOMP
RFREQ
Figure 1.
100
90
VOUT = 12V
VOUT = 5V
80
GENERAL DESCRIPTION
EFFICIENCY (%)
70
Point of load applications
Distributed power systems
Industrial control supplies
Standard rail conversion to 24 V/12 V/5 V/3.3 V
VOUT = 3.3V
60
50
40
30
VIN = 24V
fSW = 300kHz
20
The output voltage can be adjusted from 0.6 V to 0.9 V × VIN.
High efficiency is obtained with integrated low resistance Nchannel MOSFETs for both high-side and low-side devices.
The switching frequency is adjustable from 300 kHz to 1 MHz with
an external resistor. The ADP2441 also has an accurate power-good
(PGOOD) open-drain output signal.
At light load conditions, the regulator operates in pulse skip
mode by skipping pulses and reducing switching losses to improve
energy efficiency. In addition, at medium to heavy load conditions,
the regulator operates in fixed frequency pulse-width modulation
(PWM) mode to reduce electromagnetic interference (EMI).
10
0
0.02
0.2
LOAD (A)
1
10581-002
The ADP2441 is a constant frequency, current mode control,
synchronous, step-down dc-to-dc regulator that is capable of
driving loads up to 1 A with excellent line and load regulation
characteristics. The ADP2441 operates with a wide input voltage
range of 4.5 V to 36 V, which makes it ideal for regulating power
from a wide variety of sources. In addition, the ADP2441 has
very low minimum on time (50 ns) and is, therefore, suitable for
applications requiring a very high step-down ratio.
Rev. C
C3
PGOOD
Wide input voltage range of 4.5 V to 36 V
Low minimum on time of 50 ns
Maximum load current of 1 A
High efficiency of up to 94%
Adjustable output down to 0.6 V
±1% output voltage accuracy
Adjustable switching frequency of 300 kHz to 1 MHz
Pulse skip mode at light load for power saving
Precision enable input pin
Open-drain power good
External soft start with tracking
Overcurrent-limit protection
Shutdown current of less than 15 µA
UVLO and thermal shutdown
12-lead, 3 mm × 3 mm LFCSP package
Reference similar product ADP2442 with synchronization
capability
Figure 2. Efficiency vs. Load Current, VIN = 24 V
The ADP2441 uses hiccup mode to protect the IC from short
circuits or from overcurrent conditions on the output. The external
programmable soft start limits inrush current during startup for
a wide variety of load capacitances. Other key features include
tracking, input undervoltage lockout (UVLO), thermal shutdown
(TSD), and precision enable (EN), which can also be used as a
logic level shutdown input.
The ADP2441 is available in a 3 mm × 3 mm, 12-lead LFCSP
package and is rated for a junction temperature range of −40°C
to +125°C.
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ADP2441
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications Information .............................................................. 18
Applications ....................................................................................... 1
Selecting the Output Voltage .................................................... 18
General Description ......................................................................... 1
Setting the Switching Frequency .............................................. 18
Typical Circuit Configuration......................................................... 1
Soft Start ...................................................................................... 19
Revision History ............................................................................... 2
External Components Selection ............................................... 19
Specifications..................................................................................... 3
Boost Capacitor .......................................................................... 21
Absolute Maximum Ratings ............................................................ 5
VCC Capacitor............................................................................ 21
Thermal Resistance ...................................................................... 5
Loop Compensation .................................................................. 21
ESD Caution .................................................................................. 5
Large Signal Analysis of the Loop Compensation ................. 21
Pin Configuration and Function Descriptions ............................. 6
Design Example .............................................................................. 23
Typical Performance Characteristics ............................................. 7
Configuration and Components Selection ............................. 23
Internal Block Diagram ................................................................. 14
System Configuration ................................................................ 24
Theory of Operation ...................................................................... 15
Typical Application Circuits ......................................................... 25
Control Architecure ................................................................... 15
Design Example .......................................................................... 25
Adjustable Frequency................................................................. 16
Other Typical Circuit Configurations ..................................... 26
Power Good ................................................................................. 16
Power Dissipation and Thermal Considerations ....................... 29
Soft Start ...................................................................................... 16
Power Dissipation....................................................................... 29
Tracking ....................................................................................... 16
Thermal Considerations............................................................ 29
Undervoltage Lockout (UVLO) ............................................... 17
Evaluation Board Thermal Performance .................................... 30
Precision Enable/Shutdown ...................................................... 17
Circuit Board Layout Recommendations ................................... 31
Current-Limit and Short-Circuit Protection .......................... 17
Outline Dimensions ....................................................................... 32
Thermal Shutdown ..................................................................... 17
Ordering Guide .......................................................................... 32
REVISION HISTORY
8/15—Rev. B to Rev. C
Changes to Figure 45 ...................................................................... 14
6/14—Rev. A to Rev. B
Change to Features Section ............................................................. 1
11/12—Rev. 0 to Rev. A
Changes to Figure 1 .......................................................................... 1
Changed IVIN Maximum Parameter from 2.2 mA to2 mA.......... 3
Changes to Pin 3 and Pin 5 Descriptions ...................................... 6
Changes to Boost Capacitor Section ............................................ 21
Changes to Figure 66 ...................................................................... 31
Changes to Ordering Guide .......................................................... 32
6/12—Revision 0: Initial Version
Rev. C | Page 2 of 32
Data Sheet
ADP2441
SPECIFICATIONS
VIN = 4.5 V to 36 V, TJ = −40°C to +125°C, unless otherwise noted.
Table 1.
Parameter
POWER SUPPLY
Input Voltage Range
Supply Current
Shutdown Current
UVLO
Threshold
Hysteresis
INTERNAL REGULATOR
Regulator Output Voltage
OUTPUT
Output Voltage Range
Maximum Output Current
Feedback Regulation Voltage
Line Regulation
Load Regulation
ERROR AMPLIFIER
Feedback Bias Current
Transconductance
Open-Loop Voltage Gain 1
MOSFETS
High-Side Switch On Resistance 2
Low-Side Switch On Resistance2
Leakage Current
Minimum On Time 3
Minimum Off Time 4
CURRENT SENSE
Current Sense Amplifier Gain
Hiccup Time
Number Of Cumulative Current-Limit Cycles
to Go into Hiccup Mode
Peak Current Limit
FREQUENCY
Switching Frequency Range
Frequency Set Accuracy
SOFT START
Soft Start Current
PRECISION ENABLE
Input Threshold
Hysteresis
Leakage Current
Thermal Shutdown
Rising
Hysteresis
Symbol
Test Conditions/Comments
VIN
IVIN
ISHDN
VEN = 1.5 V not switching
VEN = AGND
VUVLO
VIN falling
VCC
VIN = 5 V to 36 V
VOUT
IOUT
VFB
Min
3.8
TJ = −40°C to +85°C
TJ = −40°C to +125°C
IFB_BIAS
gm
AVOL
VFB = 0.6 V
ICOMP = ±20 µA
200
RDS_H(ON)
RDS_L(ON)
ILKG
tON_MIN
tOFF_MIN
BST − SW = 5 V
VCC = 5 V
VEN = AGND
All switching frequencies
Unit
1.7
10
36
2
15
V
mA
µA
4
200
4.2
V
mV
5
5.5
V
0.9 × VIN
V
A
V
V
%/V
%/A
0.6
0.6
0.005
0.05
0.606
0.609
50
250
65
200
300
nA
µA/V
dB
170
120
1
50
165
270
180
25
65
175
mΩ
mΩ
μA
ns
ns
1.6
2
6
8
2.4
A/V
ms
Events
1.4
1.6
1.8
A
FREQ pin = 308 kΩ
FREQ pin = 92.5 kΩ
300
270
900
300
1000
1000
330
1100
kHz
kHz
kHz
VSS = 0 V
0.9
1
1.2
µA
1.15
1.20
100
0.1
1.25
V
mV
µA
GCS
fSW = 300 kHz to1 MHz
ICL
fSW
VEN(RISING)
VEN(HYST)
IIEN_LEAK
Max
4.5
0.6
1
0.594
0.591
ISS
Typ
VIN = VEN
TSD
TSD(HYST)
150
25
Rev. C | Page 3 of 32
1
°C
°C
ADP2441
Parameter
POWER GOOD
PGOOD High, FB Rising Threshold 5
PGOOD Low, FB Rising Threshold5
PGOOD High, FB Falling Threshold5
PGOOD Low, FB Falling Threshold5
PGOOD
Delay
High Leakage Current
Pull-Down Resistor
TRK
TRK Input Voltage Range
TRK to FB Offset Voltage
Data Sheet
Symbol
tPGOOD
IPGOOD(SRC)
IPGOOD(SNK)
Test Conditions/Comments
Min
Typ
Max
Unit
89
111
106
83
92
115
109
86
95
118
112
89
%
%
%
%
50
1
0.5
10
0.7
µs
µA
kΩ
VPGOOD = VCC
FB = 0 V
0
TRK = 0 mV to 500 mV
600
10
mV
mV
Guaranteed by design.
Measured between VIN and SW pins—includes bond wires and pin resistance.
3
Based on bench characterization. Measured with VIN = 12 V, VOUT = 1.2 V, load = 1 A, fSW = 1 MHz, and the output in regulation. Measurement does not include dead time.
4
Based on bench characterization. Measured with VIN = 15 V, VOUT = 12 V, load = 1 A, fSW = 600 kHz, and the output in regulation. Measurement does not include dead time.
5
This threshold is expressed as a percentage of the nominal output voltage.
1
2
Rev. C | Page 4 of 32
Data Sheet
ADP2441
ABSOLUTE MAXIMUM RATINGS
THERMAL RESISTANCE
Table 2.
Parameter
VIN to PGND
EN to AGND
SW to PGND
BST to PGND
VCC to AGND
BST to SW
FREQ, PGOOD, SS/TRK, COMP, FB to AGND
PGND to AGND
Operating Junction Temperature Range
Storage Temperature Range
Lead Temperature (Soldering, 10 sec)
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages, and is
based on a 4-layer standard JEDEC board.
Rating
−0.3 V to +40 V
−0.3 V to +40 V
−0.3 V to +40 V
−0.3 V to +45 V
−0.3 V to +6 V
−0.3 V to +6 V
−0.3 V to +6 V
±0.3 V
−40°C to +125°C
−65°C to +150°C
260°C
Table 3. Thermal Resistance
Package Type
12-Lead LFCSP
ESD CAUTION
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
Rev. C | Page 5 of 32
θJA
40
θJC
2.4
Unit
°C/W
ADP2441
Data Sheet
10 BST
11 VCC
12 AGND
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
FB 1
9 VIN
ADP2441
COMP 2
8 SW
TOP
VIEW
NOTES
1. THE EXPOSED PAD SHOULD BE CONNECTED
TO THE SYSTEM AGND PLANE AND PGND PLANE.
10581-003
SS/TRK 6
FREQ 5
7 PGND
PGOOD 4
EN 3
Figure 3. Pin Configuration, Top View
Table 4. Pin Function Descriptions
Pin No.
1
2
3
Mnemonic
FB
COMP
EN
4
5
PGOOD
FREQ
6
SS/TRK
7
PGND
8
SW
9
VIN
10
BST
11
VCC
12
AGND
EP
Description
Feedback Regulation Voltage is 0.6 V. Connect this pin to a resistor divider from the output of the dc-to-dc regulator.
Error Amplifier Compensation. Connect a resistor and capacitor in series to ground.
Precision Enable. This features offers ±5% accuracy when using a 1.25 V reference voltage. Pull this pin high to
enable the regulator and low to disable the regulator. Do not leave the EN pin floating.
Active High Power-Good Output. This pin is pulled low when the output is out of regulation.
Switching Frequency. A resistor to AGND sets the switching frequency (see the Setting the Switching Frequency
section). Do not leave the FREQ pin floating.
Soft Start/Tracking Input. A capacitor to ground is required to program the soft start time, which gradually ramps
up the output. A resistive divider to an external reference is required on this pin to track an external voltage.
Power Ground. Connect a decoupling ceramic capacitor as close as possible between the VIN pin and this pin.
Connect this pin directly to the exposed pad.
Switch. The midpoint for the drain of the low-side N-channel power MOSFET switch and the source for the high-side
N-channel power MOSFET switch.
Power Supply Input. Connect this pin to the input power source, and connect a bypass ceramic capacitor directly
from this pin to PGND, as close as possible to the IC. The operation voltage is 4.5 V to 36 V.
Boost. Connect a 10 nF ceramic capacitor between the BST and SW pins as close to the IC as possible to form a
floating supply for the high-side N-Channel power MOSFET driver. This capacitor is needed to drive the gate of the
N-channel power MOSFET above the supply voltage.
Output of the Internal Low Dropout Regulator. This pin supplies power for the internal controller and driver circuitry.
Connect a 1 µF ceramic capacitor between VCC and AGND and a 1 µF ceramic capacitor between VCC and PGND.
The VCC output is active when the EN pin voltage is more than 0.7 V.
Analog Ground. This pin is the internal ground for the control functions. Connect this pin directly to the exposed pad.
Exposed Thermal Pad. The exposed pad should be connected to AGND and PGND.
Rev. C | Page 6 of 32
Data Sheet
ADP2441
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
90
80
80
VIN = 12V
70
EFFICIENCY (%)
60
VIN = 24V
50
40
40
30
20
10
1
10581-004
0.1
LOAD (A)
0
0.01
90
VIN = 24V
VIN = 36V
50
40
VOUT = 5V
fSW = 300kHz
COILCRAFT MSS1038
50
40
20
1
10581-006
0.1
LOAD (A)
VOUT = 5V
fSW = 700kHz
COILCRAFT MSS1038
0
0.01
0.1
LOAD (A)
Figure 5. Efficiency vs. Load Current,
VOUT = 5 V, fSW = 300 kHz
Figure 8. Efficiency vs. Load Current,
VOUT = 5 V, fSW = 700 kHz
100
100
VIN = 24V
90
80
VIN = 24V
80
VIN = 36V
EFFICIENCY (%)
70
60
50
40
VOUT = 12V
fSW = 300kHz
COILCRAFT MSS1038
VIN = 36V
60
50
40
30
VOUT = 12V
fSW = 600kHz
20
10
COILCRAFT MSS1038
10
0.1
LOAD (A)
1
10581-008
EFFICIENCY (%)
VIN = 36V
10
0
0.01
0
0.01
VIN = 24V
60
30
10
20
1
70
EFFICIENCY (%)
EFFICIENCY (%)
60
30
VIN = 12V
80
70
70
1
100
VIN = 12V
80
90
1
Figure 7. Efficiency vs. Load Current,
VOUT = 3.3 V, fSW = 700 kHz
100
20
0.1
LOAD (A)
Figure 4. Efficiency vs. Load Current,
VOUT = 3.3 V, fSW = 300 kHz
30
VOUT = 3.3V
fSW = 700kHz
COILCRAFT MSS1038
10
0
0.01
90
VIN = 24V
50
10581-005
20
VOUT = 3.3V
fSW = 300kHz
COILCRAFT MSS1038
VIN = 12V
60
10581-007
EFFICIENCY (%)
70
30
VIN = 5V
10581-009
90
VIN = 5V
0
0.01
0.1
LOAD (A)
Figure 6. Efficiency vs. Load Current,
VOUT = 12 V, fSW = 300 kHz
Figure 9. Efficiency vs. Load Current,
VOUT = 12 V, fSW = 600 kHz
Rev. C | Page 7 of 32
ADP2441
Data Sheet
0.5
400
0.3
0.1
0
–0.1
VOUT = 5V
fSW = 700kHz
–0.2
–0.3
–0.4
0
0.2
0.4
0.6
0.8
1.0
LOAD (A)
250
fSW = 700kHz
200
150
100
VOUT = 3.3V
50
0
10581-010
–0.5
300
5
20
25
30
35
40
Figure 13. Pulse Skip Threshold, VOUT = 3.3 V
1.0
0.6
0.4
TA = +25°C
TA = –40°C
0.2
0
–0.2
–0.4
TA = +125°C
–0.8
–1.0
0
0.2
0.4
0.6
0.8
1.0
LOAD (A)
200
fSW = 700kHz
150
100
50
VOUT = 5V
0
10581-011
VIN = 24V
VOUT = 5V
fSW = 700kHz
–0.6
fSW = 300kHz
250
10
15
20
25
30
35
40
VIN (V)
10581-014
PSKIP THRESHOLD LOAD CURRENT (mA)
300
0.8
VOUT ERROR (%)
15
VIN (V)
Figure 10. Load Regulation for Different Supplies
Figure 14. Pulse Skip Threshold, VOUT = 5 V
Figure 11. Load Regulation for Different Temperatures
300
PSKIP THRESHOLD LOAD CURRENT (mA)
0.5
0.4
0.3
0.2
LOAD = 500mA
0.1
0
LOAD = 1A
–0.1
VIN = 24V
VOUT = 5V
fSW = 700kHz
–0.2
–0.3
–0.4
–0.5
7
12
17
22
27
32
VIN (V)
37
10581-012
VOUT ERROR (%)
10
250
fSW = 300kHz
200
150
fSW = 600kHz
100
50
VOUT = 12V
0
15
20
25
30
35
VIN (V)
Figure 15. Pulse Skip Threshold, VOUT = 12 V
Figure 12. Line Regulation, VOUT = 5 V for Different Loads
Rev. C | Page 8 of 32
40
10581-015
VOUT ERROR (%)
0.2
fSW = 300kHz
350
10581-013
0.4
PSKIP THRESHOLD LOAD CURRENT (mA)
VIN = 12V
VIN = 24V
VIN = 36V
Data Sheet
ADP2441
2.25
12
2.05
VIN = 36V
1.85
SUPPLY CURRENT (mA)
SHUTDOWN CURENT (µA)
10
8
6
4
1.65
1.45
VIN = 4.5V
VIN = 12V
1.25
1.05
VIN = 24V
0.85
VIN = 36V
0.65
VIN = 4.5V
2
0.45
0
–50
0.05
–50
50
100
150
TEMPERATURE (°C)
–30
–10
10
30
50
70
90
110
130
150
TEMPERATURE (°C)
Figure 16. Shutdown Current vs. Temperature
10581-016
0
10581-017
0.25
Figure 19. Supply Current vs. Temperature
4.5
1.24
1.22
ENABLE RISING THRESHOLD
1.20
ENABLE VOLTAGE (V)
UVLO THRESHOLD (V)
4.4
UVLO, RISING VIN
4.3
4.2
4.1
UVLO, FALLING VIN
1.18
1.16
1.14
1.12
ENABLE FALLING THRESHOLD
1.10
1.08
4.0
0
25
50
75
100
1.04
–50
125
–30
–10
10
TEMPERATURE (°C)
130
0.6
120
PGOOD THRESHOLD (%)
0.7
0.4
0.3
0.2
0.1
0.2
0.3
0.4
0.5
TRACK (V)
70
90
110
130
150
130
150
110
100
90
80
PGOOD FALL, FB INCREASING
PGOOD RISE, FB DECREASING
PGOOD RISE, FB INCREASING
PGOOD FALL, FB DECREASING
70
0.1
0.6
0.7
0.8
60
–50
10581-118
FB (V)
0.5
0
50
Figure 20. Enable Threshold vs. Temperature
Figure 17. UVLO Threshold vs. Temperature
0
30
TEMPERATURE (°C)
–30
–10
10
30
50
70
90
110
TEMPERATURE (°C)
Figure 21. PGOOD Threshold vs. Temperature
Figure 18. Tracking Range
Rev. C | Page 9 of 32
10581-021
–25
10581-019
10581-018
3.9
–50
1.06
ADP2441
Data Sheet
1200
1200
1100
fSW = 1MHz
1000
800
FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
fSW = 1MHz
1000
fSW = 700kHz
600
400
900
800
fSW = 700kHz
700
600
500
fSW = 300kHz
400
200
fSW = 300kHz
0
5
10
15
20
25
30
35
40
VIN (V)
200
–50
10581-022
0
–30
–10
10
30
50
70
90
110
130
150
TEMPERATURE (°C)
10581-023
300
Figure 25. Switching Frequency vs. Temperature
Figure 22. Switching Frequency vs. Supply
1.80
200
1.78
1.76
MINIMUM OFF
1.74
150
1.72
1.70
CURRENT (A)
ON TIME AND OFF TIME (ns)
175
125
100
75
1.68
VIN = 36V
1.66
1.64
1.62
VIN = 4.5V
1.60
1.58
MINIMUM ON
50
1.56
1.54
25
10
–10
30
50
70
90
130
110
150
TEMPERATURE (°C)
1.50
–50
10581-024
–30
150
Figure 26. Current Limit vs. Temperature
260
180
240
160
140
LOW-SIDE RDS(ON) (mΩ)
220
200
180
160
140
120
100
80
60
40
120
–25
0
25
50
75
100
125
TEMPERATURE (°C)
150
0
–50
–30
–10
10
30
50
70
90
TEMPERATURE (°C)
110
Figure 27. Low-Side RDS(ON) vs. Temperature
Figure 24. High-Side RDS(ON) vs. Temperature
Rev. C | Page 10 of 32
130
150
10581-026
20
10581-027
HIGH-SIDE RDS(ON) (mΩ)
100
TEMPERATURE (°C)
Figure 23. Minimum On Time and Minimum Off Time vs. Temperature
100
–50
50
0
10581-126
1.52
0
–50
Data Sheet
1
ADP2441
VOUT
VOUT
1
VIN = 24V
VOUT = 3.3V
fSW = 500kHz
LOAD = 100mA
INDUCTOR CURRENT
INDUCTOR CURRENT
4
4
VIN = 24V
VOUT = 3.3V
FSW = 500kHz
2
CH1 20.0mV
B
W CH2 10.0V
CH4 200mA Ω
M4.00µs A CH4
T 41.40%
10581-029
10581-028
SW
SW
2
CH1 20.0mV
120mA
Figure 28. Pulse Skip Mode,
VIN = 24 V, VOUT = 3.3 V, fSW = 500 kHz, No Load
B
W CH2 10.0V
M4.00µs A CH4
CH4 500mA Ω T 41.40%
120mA
Figure 31. Pulse Skip Mode,
VIN = 24 V, VOUT = 3.3 V, fSW = 500 kHz, Load = 100 mA
VOUT
VOUT
1
1
INDUCTOR CURRENT
INDUCTOR CURRENT
VIN = 24V
VOUT = 3.3V
fSW = 500kHz
LOAD = 1A
4
4
10581-030
CH1 20.0mV
B
W CH2 10.0V
CH4 500mA Ω
M1.00µs
A CH2
T 41.40%
10581-031
SW
SW
2
2
CH1 200mV
9.80V
Figure 29. PWM Mode,
VIN = 24 V, VOUT = 3.3 V, fSW = 500 kHz, Load = 1 A
B
W
CH2 10.0V
CH4 1.00A Ω
M2.00ms A CH1
T 49.40%
60.0mV
Figure 32. Hiccup Mode,
VIN = 24 V, VOUT = 3.3 V, fSW = 500 kHz, Output Short to PGND
VIN = 24V fSW = 700kHz
VOUT = 5V LOAD STEP = 300mA
VOUT
1
VOUT
VIN = 24V
VOUT = 5V
fSW = 700kHz
LOAD STEP = 500mA
1
LOAD
LOAD
SW
SW
4
CH1 100mV
B
W
M200µs A CH4
CH2 10V
CH4 500mA Ω T 79.80%
10581-033
4
10581-032
2
2
CH1 50.0mV
690mA
B
W
CH2 10.0V
M200µs
CH4 200mA Ω
A CH4
604mA
Figure 33. Load Transient Response,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz, Load Step = 300 mA
Figure 30. Load Transient Response,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz, Load Step = 500 mA
Rev. C | Page 11 of 32
ADP2441
Data Sheet
VOUT
1
VOUT
VIN = 12V
VOUT = 5V
fSW = 300kHz
LOAD STEP = 500mA
1
LOAD
SW
4
10581-034
4
2
B
CH1 100mV
W
M200µs
CH2 5.00V
CH4 500mA Ω
A CH4
VIN = 24V
VOUT = 12V
fSW = 600kHz
LOAD STEP = 500mA
SW
10581-037
LOAD
2
690mA
B
CH1 200mV
Figure 34. Load Transient Response,
VIN = 12 V, VOUT = 5 V, fSW = 300 kHz, Load Step = 500 mA
W
M200µs
CH2 10.0V
CH4 500mA Ω
A CH4
600mA
Figure 37. Load Transient Response,
VIN = 24 V, VOUT = 12 V, fSW = 600 kHz, Load Step = 500 mA
ENABLE
VOUT
1
3
VOUT
PGOOD
LOAD
4
VIN = 24V
VOUT = 12V
fSW = 300kHz
LOAD STEP = 500mA
SW
1
2
B
CH1 200mV
W
CH2 10.0V
M200µs
CH4 500mA Ω
A CH4
10581-038
VIN = 24V
VOUT = 5V
fSW = 700kHz
10581-035
4
CH1 2.00V BW
CH3 5.00V
550mA
Figure 35. Load Transient Response,
VIN = 24 V, VOUT = 12 V, fSW = 300 kHz, Load Step = 500 mA
M 200µs
A CH3
1.60V
CH4 2.00V
Figure 38. Power-Good Shutdown,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz
VIN
ENABLE
VIN = 24V
VOUT = 5V
fSW = 700kHz
VOUT
3
VIN = 36V
VOUT = 5V
fSW = 700kHz
3
VOUT
1
1
SW
PGOOD
CH1 2.00V BW
CH3 2.00V
M1.00ms
A CH3
10581-039
10581-036
4
2
CH1 2.00V BW
CH3 10.0V
1.64V
CH4 2.00V
Figure 36. Power Good Startup,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz
CH2 10.0V BW M4.00ms
A CH3
Figure 39. Startup with VIN,
VIN = 36 V, VOUT = 5 V, fSW = 700 kHz, No Load
Rev. C | Page 12 of 32
5.00V
Data Sheet
ADP2441
ENABLE
VIN
VIN = 36V
VOUT = 5V
fSW = 700kHz
LOAD = 5Ω
3
VIN = 24V
VOUT = 5V
fSW = 700kHz
VOUT
VOUT
3
SS
1
SW
1
4
CH2 10.0V BW M4.00ms
CH1 2.00V BW
CH3 10.0V
A CH3
10581-143
SW
10581-040
2
2
CH1 2.00V BW
CH3 5.00V
9.00V
Figure 40. Startup with VIN,
VIN = 36 V, VOUT = 5 V, fSW = 700 kHz, Load = 5 Ω
CH2 10.0V
CH4 2.00V
M1.00ms
A CH3
1.40V
Figure 43. Soft Start Startup with Precision Enable,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz, No Load, Internal SS
110
200
90
160
70
120
50
80
30
40
10
0
VOUT
SS
MAGNITUDE (dB)
VIN = 24V
VOUT = 5V
fSW = 700kHz
LOAD = 5Ω
SW
1
–40
–10
CROSSOVER = 58kHz: 1/12 fSW
PHASE MARGIN = 55°
VIN = 24V
VOUT = 5V
fSW = 700kHz
LOAD = 1A
–30
4
10581-041
–50
2
CH1 2.00V
CH3 5.00V
B
CH2 10.0V
CH4 2.00V
W
M200µs
A CH3
–70
–80
–120
–160
–200
–90
2.20V
1
10
100
FREQUENCY (kHz)
Figure 41. Shutdown with Precision Enable,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz, Load = 5 Ω
Figure 44. Magnitude and Phase vs. Frequency
ENABLE
3
VOUT
1
VIN = 24V
VOUT = 5V
fSW = 700kHz
LOAD = 5Ω
SS = 10nF
SS
4
10581-042
SW
2
CH1 2.00V BW
CH3 5.00V
CH2 10.0V
CH4 500mV
M2.00ms
A CH3
2.20V
Figure 42. Startup with Precision Enable,
VIN = 24 V, VOUT = 5 V, fSW = 700 kHz, Load = 5 Ω, SS = 10 nF
Rev. C | Page 13 of 32
10581-144
3
PHASE (Degrees)
ENABLE
ADP2441
Data Sheet
INTERNAL BLOCK DIAGRAM
VIN
AGND
VCC
INTERNAL LDO
BST
EN
UVLO
+
1.25V
BAND GAP
REFERENCE
OSC
FREQ
ENABLE
COMP
POWER STAGE
ISS
CLOCK
+
FB
–
SS/TRK
+
PULSE SKIP
ENABLE
1V
NMOS
STATE MACHINE GATE
CONTROL LOGIC
COMP
THRESHOLD
SW
+
VCC
+
PWM
VREF = 0.6V
NMOS
–
PWM
COMPARATOR
PGND
HICCUP
HICCUP
TIMER
–
SLOPE
COMPENSATION/
RAMP
GENERATOR
+
CURRENT SENSE
AMPLIFIER
PGOOD
115% OF
FEEDBACK
REFERENCE
CURRENT
VFB
86%OF
FEEDBACK
Figure 45. Block Diagram
Rev. C | Page 14 of 32
10581-043
CURRENT-LIMIT
COMPARATOR
Data Sheet
ADP2441
THEORY OF OPERATION
CONTROL ARCHITECURE
The ADP2441 is based on the emulated peak current mode
control architecture.
Fixed Frequency Mode
A basic block diagram of the control architecture is shown in
Figure 46. With medium to heavy loads, the ADP2441 operates
in the fixed switching frequency PWM mode. The output voltage,
VOUT, is sensed on the feedback pin, FB. An error amplifier
integrates the error between the feedback voltage and the reference
voltage (VREF = 0.6 V) to generate an error voltage at the COMP
pin. A current sense amplifier senses the valley inductor current
(IL) during the off period when the low-side power MOSFET is
on and the high-side power MOSFET is off. An internal oscillator
initiates a PWM pulse to turn off the low-side power MOSFET
and turn on the high-side power MOSFET at a fixed switching
frequency. When the high-side N-channel power MOSFET is
enabled, the valley inductor current information is added to an
emulated ramp signal, and then the PWM comparator compares
this value to the error voltage on the COMP pin. The output of
the PWM comparator modulates the duty cycle by adjusting the
trailing edge of the PWM pulse that turns off the high-side
power MOSFET and turns on the low-side power MOSFET.
Slope compensation is programmed internally into the emulated
ramp signal and is automatically selected, depending on the
input voltage, output voltage, and switching frequency. This
prevents subharmonic oscillations for near or greater than 50%
duty cycle operation. The one restriction of this feature is that
the inductor ripple current must be set between 0.2 A and 0.5 A
to provide sufficient current information to the loop.
CLOCK
COMP
VC
S
Q
VOUT
PWM DRIVER
R QB
IL
COMPARATOR
RSWL × IL
REF
gm V
FB
RAMP
EMULATION
SENSE_
BLOCK
OUT
GCS
VRAMP
10581-044
The ADP2441 includes programmable features, such as soft
start, output voltage, switching frequency, and power good.
These features are programmed externally via tiny resistors and
capacitors. The ADP2441 also includes protection features, such
as UVLO with hysteresis, output short-circuit protection, and
thermal shutdown.
VIN
Figure 46. Control Architecture Block Diagram
Pulse Skip Mode
The ADP2441 has built-in pulse skip circuitry that turns on during
light loads, switching only as necessary so that the output voltage
remains within regulation. This allows the regulator to maintain
high efficiency during operation with light loads by reducing
switching losses. The pulse skip circuitry includes a comparator,
which compares the COMP voltage to a fixed pulse skip threshold.
COMP
PULSE SKIP
THRESHOLD
DC
CONTROL
LOGIC
ADP2441
1V
10581-045
The ADP2441 is a fixed frequency, current mode control, stepdown, synchronous switching regulator that is capable of driving
1 A loads. The device operates with a wide input voltage range
from 4.5 V to 36 V, and its output is adjustable from 0.6 V to 0.9 V
× VIN. The integrated high-side N-channel power MOSFET and
the low-side N-channel power MOSFET yield high efficiency
with medium to heavy loads. Pulse skip mode is available to
improve efficiency at light loads.
Figure 47. Pulse Skip Comparator
With light loads, the output voltage discharges at a very slow rate
(load dependent). When the output voltage is within regulation,
the device enters sleep mode and draws a very small quiescent
current. As the output voltage drops below the regulation voltage,
the COMP voltage rises above the pulse skip threshold. The device
wakes up and starts switching until the output voltage is within
regulation.
As the load increases, the settling value of the COMP voltage
increases. At a particular load, COMP settles above the pulse skip
threshold, and the device enters the fixed frequency PWM mode.
Therefore, the load current at which COMP exceeds the pulse
skip threshold is defined as the pulse skip current threshold; the
value varies with the duty cycle and the inductor ripple current.
The measured value of pulse skip threshold over VIN is given in
Figure 13, Figure 14, and Figure 15.
Rev. C | Page 15 of 32
ADP2441
Data Sheet
ADJUSTABLE FREQUENCY
The ADP2441 features a programmable oscillator frequency with
a resistor connected between the FREQ and AGND pins.
At power-up, the FREQ pin is forced to 1.2 V and current flows
from the FREQ pin to AGND; the current value is based on the
resistor value on the FREQ pin. Then, the same current replicates
in the oscillator to set the switching frequency. Note that the
resistor connected to the FREQ pin should be placed as close as
possible to the FREQ pin (see the Applications Information
section for more information).
The soft start period ends when the soft start ramp voltage
exceeds the internal reference of 0.6 V. The ADP2441 also
features an internal default soft start time of 2 ms. For more
information, see the Applications Information section.
ENABLE
3
POWER GOOD
SS
Likewise, the PGOOD pin is pulled low to AGND when the
input voltage is below the internal UVLO threshold, when the
EN pin is low, or when a thermal shutdown event has occurred.
VOUT FALLING
116
110
100
90
84
POWER
GOOD
10581-149
A CH1
2.52V
TRACKING
The ADP2441 has a tracking feature that allows the output voltage
to track an external voltage. This feature is especially useful in a
system where power supply sequencing and tracking is required.
UNDERVOLTAGE
MASTER
VOLTAGE
PGOOD
COMP
SW
SS/TRK
FB
RTOP
RTRK_TOP
Figure 48. PGOOD Threshold
In a typical application, a pull-up resistor connected between the
PGOOD pin and an external supply is used to generate a logic
signal. This pull-up resistor should range in value from 30 kΩ
to 100 kΩ, and the external supply should be less than 5.5 V.
SOFT START
RTRK_BOT
VOUT
ADP2441
RBOTTOM
10581-048
OVERVOLAGE
M10.0ms
Figure 49. External Soft Start
10581-047
POWER
GOOD
CH2 1.00V
CH1 2.00V BW
CH3 5.00V
REF
100
2
The ADP2441 SS/TRK pin is connected to the internal error
amplifier. The internal error amplifier includes three inputs: the
internal reference voltage, the SS/TRK voltage, and the feedback
voltage. The error amplifier regulates the feedback voltage to the
lower of the other two voltages. To track a master voltage, tie
the SS/TRK pin to a resistor divider from the master voltage as
shown in Figure 50.
% OF VOUT SET
VOUT RISING
% OF VOUT SET
VIN = 24V
VOUT = 5V
fSW = 700kHz
LOAD = 1A
EXTERNAL SS = 10nF
1
The PGOOD pin is an open-drain output that indicates the
status of the output voltage. When the voltage of the FB pin is
between 92% and 109% of the internal reference voltage, the
PGOOD output is pulled high, provided there is a pull-up resistor
connected to the pin. When the voltage of the FB pin is not within
this range, the PGOOD output is pulled low to AGND. The
PGOOD threshold is shown in Figure 48.
UNDERVOLTAGE
VOUT
Figure 50. Tracking Feature Block Diagram
The ratio of the slave output voltage to the master voltage is a
function of the two dividers as follows:
The ADP2441 soft start feature allows the output voltage to ramp
up in a controlled manner, limiting the inrush current during
startup. An external capacitor connected between the SS/TRK
and AGND pins is required to program the soft start time.
The programmable soft start feature is useful when a load
requires a controlled voltage slew rate at startup. When the
regulator powers up and soft start is enabled, the internal 1 μA
current source charges the external soft start capacitor, establishing
a voltage ramp slope at the SS pin, as shown in Figure 49.
Rev. C | Page 16 of 32
VOUT
VMASTER
RTOP
1 +
R
BOTTOM
=
R
1 + TRK _ TOP
R
TRK _ BOT
(1)
Data Sheet
ADP2441
Coincident Tracking
PRECISION ENABLE/SHUTDOWN
The most common mode of tracking is coincident tracking. In this
method, the slope of the slave voltage matches that of the master
voltage, as shown in Figure 51. As the master voltage rises, the
slave voltage rises identically. Eventually, the slave voltage reaches
its regulation voltage, at which point the internal reference takes
over the regulation while the SS/TRK input continues to increase,
thus preventing itself from influencing the output voltage.
The ADP2441 features a precision enable pin (EN) that can be used
to enable or shut down the device. The ±5% accuracy lends
itself to using a resistor divider from the VIN pin (or another
external supply) to program a desired UVLO threshold that is
higher than the fixed internal UVLO of 4.2 V. The hysteresis is
100 mV.
VOLTAGE (V)
MASTER VOLTAGE
SLAVE VOLTAGE
If a resistor divider is not used, a logic signal can be applied. A logic
high enables the device, and a logic low forces the device into
shutdown mode.
VIN
BST
VIN
SW
10581-049
TIME
VOUT
ADP2441
R1
EN
FB
Figure 51. Coincident Tracking
R2
FREQ AGND COMP
Ratiometric Tracking
10581-051
For coincident tracking, select resistor values such that RTRK_TOP
= RTOP and RTRK_BOT = RBOTTOM in Equation 1.
Figure 53. Precision Enable Used as a Programmable UVLO
In the ratiometric tracking scheme, the master and the slave
voltages rise with different slopes.
CURRENT-LIMIT AND SHORT-CIRCUIT
PROTECTION
The ADP2441 has a current-limit comparator that compares the
current sensed across the low-side power MOSFET to the
internally set reference current. If the sensed current exceeds
the reference current, the high-side power MOSFET is not
turned on in the next cycle and the low-side power MOSFET
stays on until the inductor current ramps down below the
current-limit level.
VOLTAGE (V)
MASTER VOLTAGE
TIME
10581-050
SLAVE VOLTAGE
Figure 52. Ratiometric Tracking
For ratiometric tracking in which the master voltage rises faster
than the slave voltage (as shown in Figure 52), select RTRK_TOP ≥
RTOP and RTRK_BOT = RBOTTOM in Equation 1.
UNDERVOLTAGE LOCKOUT (UVLO)
The UVLO function prevents the IC from turning on while the
input voltage is below the specified operating range to avoid an
undesired operating mode. If the input voltage drops below the
specified range, the UVLO function shuts off the device. The
rising input voltage threshold for the UVLO function is 4.2 V
with 200 mV hysteresis. The 200 mV of hysteresis prevents the
regulator from turning on and off repeatedly with slow voltage
ramp on the VIN pin.
If the output is overloaded and the peak inductor current exceeds
the preset current limit for more than eight consecutive clock
cycles, the hiccup mode current-limit condition occurs. The
output goes to sleep for 6 ms, during which time the output is
discharged, the average power dissipation is reduced, and the
device wakes up with a soft start period. If the current-limit
condition is triggered again, the output goes to sleep and wakes up
after 6 ms. Figure 32 shows the current-limit hiccup mode when
the output is shorted to PGND.
THERMAL SHUTDOWN
If the ADP2441 junction temperature rises above 150°C, the
thermal shutdown circuit turns off the switching regulator. Extreme
junction temperatures can be the result of high current operation,
poor circuit board design, or high ambient temperature. A 25°C
hysteresis is included so that when a thermal shutdown occurs,
the ADP2441 does not return to normal operation until the
junction temperature drops below 125°C. Soft start is active upon
each restart cycle.
Rev. C | Page 17 of 32
ADP2441
Data Sheet
APPLICATIONS INFORMATION
The ratio of the resistive voltage divider sets the output voltage,
and the absolute value of the resistors sets the divider string
current. For lower divider string currents, the small 50 nA
(0.1 μA maximum) FB bias current should be taken into account
when calculating the resistor values. The FB bias current can be
ignored for a higher divider string current; however, using small
feedback resistors degrades efficiency at very light loads.
To limit degradation of the output voltage accuracy due to FB
bias current to less than 0.005% (0.5% maximum), ensure that
the divider string current is greater than 20 μA. To calculate the
desired resistor values, first determine the value of the bottom
resistor, RBOTTOM, as follows:
(2)
I STRING
70
60
50
40
30
20
0
0
200
400
600
800
1000
1200
FREQUENCY (kHz)
Figure 55. Duty Cycle vs. Switching Frequency
Calculate the value of the frequency resistor using the following
equation:
92,500
f SW
(4)
Table 6 and Figure 56 provide examples of frequency resistor
values, which are based on the switching frequency.
(3)
Table 6. Frequency Resistor Selection
RFREQ (kΩ)
308
132
92.5
VOUT
ADP2441
FB
Frequency
300 kHz
700 kHz
1 MHz
1200
RBOTTOM
SS/TRK
RFREQ
1100
1000
CSS
EXTERNAL
SUPPLY
FREQUENCY (kHz)
FREQ
10581-052
PGOOD
Figure 54. Voltage Divider
Table 5. Output Voltage Selection
Voltage (V)
12
5
3.3
1.2
DMIN
10
where RFREQ is in kΩ, and fSW is in kHz.
Then calculate the value of the top resistor, RTOP, as follows:
RTOP
DMAX
80
RFREQ =
where:
VREF is the internal reference and equals 0.6 V.
ISTRING is the resistor divider string current.
V
− VREF
RTOP = R BOTTOM × OUT
VREF
90
RTOP (kΩ)
190
73
45
10
RBOTTOM (kΩ)
10
10
10
10
900
800
700
600
500
400
10581-053
R BOTTOM =
VREF
100
10581-155
The output voltage is set using a resistor divider connected between
the output voltage and the FB pin (see Figure 54). The resistor
divider divides down the output voltage to the 0.6 V FB regulation
voltage. The output voltage can be set to as low as 0.6 V and as
high as 90% of the power input voltage.
However, the choice is also influenced by whether there is a
need for small external components. For example, for small, area
limited power solutions, higher switching frequencies are required.
DUTY CYCLE (%)
SELECTING THE OUTPUT VOLTAGE
300
200
50
100
150
200
250
RESISTANCE (kΩ)
SETTING THE SWITCHING FREQUENCY
The choice of the switching frequency depends on the required
dc-to-dc conversion ratio and is limited by the minimum and
maximum controllable duty cycle, as shown in Figure 55. This is
due to the requirement of minimum on time and minimum off
time for current sensing and robust operation.
Rev. C | Page 18 of 32
Figure 56. Frequency vs. Resistance
300
350
Data Sheet
ADP2441
SOFT START
The soft start function limits the input inrush current and prevents
output overshoot at startup. The soft start time is programmed
by connecting a small ceramic capacitor between the SS/TRK
and AGND pins, with the value of this capacitor defining the
soft start time, tSS, as follows:
I
VREF
= SS
t SS
C SS
(5)
where:
VREF is the internal reference voltage and equals 0.6 V.
ISS is the soft start current and equals 1 μA.
CSS is the soft start capacitor value.
Table 7. Soft Start Time Selection
Soft Start Capacitor (nF)
5
10
20
Soft Start Time (ms)
3
6
12
Alternatively, the user can float the SS/TRK pin and use the
internal soft start time of 2 ms.
EXTERNAL COMPONENTS SELECTION
Input Capacitor Selection
The input current to a buck regulator is pulsating in nature. The
current is zero when the high-side switch is off and is
approximately equal to the load current when the switch is on.
Because switching occurs at reasonably high frequencies (300 kHz
to 1 MHz), the input bypass capacitor usually supplies most of
the high frequency current (ripple current), allowing the input
power source to supply only the average (dc) current. The input
capacitor needs a sufficient ripple current rating to handle the
input ripple and needs an ESR that is low enough to mitigate the
input voltage ripple. In many cases, different types of capacitors
are placed in parallel to minimize the effective ESR and ESL.
Table 10 shows a list of recommended MLCC capacitors from
Murata and Taiyo Yuden.
For large step load transients, add more bulk capacitance by, for
example, using electrolytic or polymer capacitors. Make sure that
the ripple current rating of the bulk capacitor exceeds the
minimum input ripple current of a particular design.
Inductor Selection
The high switching frequency of the ADP2441 allows for minimal
output voltage ripple even when small inductors are used. Selecting
the size of the inductor involves considering the trade-off between
efficiency and transient response. A smaller inductor results in
larger inductor current ripple, which provides excellent transient
response but degrades efficiency. Due to the high switching
frequency of the ADP2441, using shielded ferrite core inductors
is recommended because of their low core losses and low EMI.
The inductor ripple current also affects the stability of the loop
because the ADP2441 uses the emulated peak current mode
architecture. In the traditional approach of slope compensation,
the user sets the inductor ripple current and then sets the slope
compensation using an external ramp resistor. In most cases, the
inductor ripple current is typically set to be 1/3 of the maximum
load current for optimal transient response and efficiency. The
ADP2441 has internal slope compensation, which assumes that
the inductor ripple current is set to 0.3 A (30% of the maximum
load of 1 A), eliminating the need for an external ramp resistor.
For the ADP2441, choose an inductor such that the peak-to-peak
ripple current of the inductor is between 0.2 A and 0.5 A for
stable operation.
Therefore, calculate the inductor value as follows:
∆I L =
I OUT × D × (1 − D)
(VPP − I OUT × D × RESR ) f SW
2 × VOUT × (VIN − VOUT )
5 × VOUT × (VIN − VOUT )
≤L≤
VIN × f SW
VIN × f SW
(6)
where:
VPP is the desired input ripple voltage.
RESR is the equivalent series resistance of the capacitor.
IOUT is the maximum load current.
C IN _ MIN =
I OUT × D × (1 − D)
VPP × f SW
LIDEAL =
3.3 × VOUT × (VIN − VOUT )
VIN × f SW
where:
VIN is the input voltage.
VOUT is the desired output voltage.
fSW is the regulator switching frequency.
It is recommended to use a ceramic bypass capacitor because
the ESR associated with this type of capacitor is near zero,
simplifying the equation to
For applications with a wide input (VIN) range, choose the
inductor based on the geometric mean of the input voltage
extremes.
(7)
In addition, it is recommended to use a ceramic capacitor with a
voltage rating that is 1.5 times the input voltage with X5R and X7R
dielectrics. Using Y5V and Z5U dielectrics is not recommended
due to their poor temperature and dc bias characteristics.
(8)
0.2 A ≤ ΔIL ≤ 0.5 A
The minimum input capacitance required for a particular load is
C IN _ MIN =
VOUT × (VIN − VOUT )
VIN × f SW × L
VIN (GEOMETRIC ) = VIN _ MAX × VIN _ MIN
where:
VIN_MAX is the maximum input voltage.
VIN_MIN is the minimum input voltage.
Rev. C | Page 19 of 32
(9)
ADP2441
Data Sheet
The inductor value is based on VIN(GEOMETRIC) as follows:
LIDEAL =
Output Capacitor Selection
3.3 × VOUT × (VIN (GEOMETRIC ) − VOUT )
VIN (GEOMETRIC ) × f SW
Table 8. Inductor Values for Various VIN, VOUT, and fSW
Combinations
fSW (kHz)
300
300
300
300
300
300
300
300
600
600
600
600
600
600
600
1000
1000
1000
1000
VIN (V)
12
12
24
24
24
36
36
36
12
12
24
24
24
36
36
12
24
24
36
VOUT (V)
3.3
5
3.3
5
12
3.3
5
12
3.3
5
3.3
5
12
3.3
5
5
5
12
5
Inductor Value
Min (µH)
Max (µH)
22
27
27
33
27
33
39
47
56
68
27
33
39
47
68
82
12
15
15
18
15
18
18
22
27
33
15
18
22
27
6.8
10
10
12
18
22
12
15
To avoid inductor saturation and ensure proper operation, choose
the inductor value so that neither the saturation current nor the
maximum temperature rated current ratings are exceeded.
Inductor manufacturers specify both of these ratings in data
sheets or the rating can be calculated as follows:
I L _ PEAK
∆I
= I LOAD( MAX ) + L
2
(10)
where:
ILOAD(MAX) is the maximum dc load current.
ΔIL is the peak-to-peak inductor ripple current.
Table 9. Recommended Inductors
Value (µH)
10
18
33
15
Small Size Inductors
(10 mm × 10 mm)
MSS1260
MSS1260
MSS1260
MSS1260
The output capacitor selection affects both the output voltage
ripple and the loop dynamics of the regulator. The ADP2441 is
designed to operate with small ceramic output capacitors that
have low ESR and ESL; therefore, the device can easily meet tight
output voltage ripple specifications. For best performance, use
X5R or X7R dielectric capacitors with a voltage rating that is
1.5 times the output voltage and avoid using Y5V and Z5U
dielectric capacitors, which have poor temperature and dc bias
characteristics. Table 10 lists some recommended capacitor from
Murata and Taiyo Yuden.
For acceptable maximum output voltage ripple, determine the
minimum output capacitance, COUT(MIN), as follows:
1
∆VRIPPLE ≅ ∆I L × ESR +
8
×
f
×
COUT ( MIN )
SW
(11)
Therefore,
COUT ( MIN ) ≅
∆I L
8 × f SW × (∆VRIPPLE − ∆I L × ESR)
(12)
where:
ΔVRIPPLE is the allowable peak-to-peak output voltage ripple.
ΔIL is the inductor ripple current.
ESR is the equivalent series resistance of the capacitor.
fSW is the switching frequency of the regulator.
If there is a step load requirement, choose the output capacitor
value based on the value of the step load. For the maximum
acceptable output voltage droop/overshoot caused by the step load,
3
COUT ( MIN ) ≅ ∆I OUT (STEP ) ×
f SW × ∆VDROOP
(13)
where:
ΔIOUT(STEP) is the load step.
fSW is the switching frequency of the regulator.
ΔVDROOP is the maximum allowable output voltage droop/overshoot.
Select the largest output capacitance given by Equation 12 and
Equation 13. When choosing the type of ceramic capacitor for the
output filter of the regulator, select one with a nominal capacitance
that is 20% to 30% larger than the calculated value because the
effective capacitance degrades with dc voltage and temperature.
Figure 57 shows the capacitance loss due to the output voltage
dc bias for three X7R MLCC capacitors from Murata.
Rev. C | Page 20 of 32
Data Sheet
ADP2441
LOOP COMPENSATION
30.0
The ADP2441 uses peak current mode control architecture for
excellent load and line transient response. This control architecture
has two loops: an external voltage loop and an inner current loop.
The inner current loop senses the current in the low-side switch
and controls the duty cycle to maintain the average inductor
current. Slope compensation is added to the inner current loop
to ensure stable operation when the duty cycle is above 50%.
19.2
22µF/25V
13.8
10µF/25V
3.00
0
5
10
15
DC BIAS VOLTAGE (V)
20
25
10581-157
8.40
The external voltage loop senses the output voltage and adjusts
the duty cycle to regulate the output voltage to the desired value. A
transconductance amplifier with an external series RC network
connected to the COMP pin compensates the external voltage loop.
ADP2441
Figure 57. Capacitance vs. DC Bias Voltage
COMP
For example, to attain 20 μF of output capacitance with an output
voltage of 5 V while providing some margin for temperature
variation, use a 22 μF capacitor with a voltage rating of 25 V
and a 10 μF capacitor with a voltage rating of 25 V in parallel.
This configuration ensures that the output capacitance is sufficient
under all conditions and, therefore, that the device exhibits stable
behavior.
gm
0.6V
BOOST CAPACITOR
The boost pin (BST) is used to power up the internal driver for the
high-side power MOSFET. In the ADP2441, the high-side power
MOSFET is an N-channel device to achieve high efficiency in
mid and high duty cycle applications. To power up the high-side
driver, a capacitor is required between the BST and SW pins.
The size of this boost capacitor is critical because it affects the
light load functionality and efficiency of the device. Therefore,
choose a boost ceramic capacitor with a value between 10 nF to
22 nF with a voltage rating of 50 V and place the capacitor as
close as possible to the IC. It is recommended to use a boost
capacitor within this range because a capacitor beyond 22 nF
can cause the LDO to reach the current-limit threshold.
Figure 58. RC Compensation Network
LARGE SIGNAL ANALYSIS OF THE LOOP
COMPENSATION
The control loop can be broken down into the following three
sections:
•
•
•
VOUT to VCOMP
VCOMP to IL
IL to VOUT
VIN
INDUCTOR
CURRENT
SENSE
PULSE-WIDTH
MODULATOR
VOUT
COUT
VCOMP
VCC CAPACITOR
IL
RLOAD
Gmgm
VREF = 0.6V
The ADP2441 has an internal regulator to power up the internal
controller and the low-side driver. The VCC pin is the output of
the internal regulator. The internal regulator provides the pulse
current when the low-side driver turns on. Therefore, it is
recommended that a 1 µF ceramic capacitor be placed between
the VCC and PGND pins as close as possible to the IC and that
a 1 µF ceramic capacitor be placed between the VCC and
AGND pins.
Rev. C | Page 21 of 32
ADP2441
RCOMP
CCOMP
10581-055
Capacitor
10 µF/25 V
22 µF/25 V
47 µF/6.3 V
4.7 µF/50 V
CCOMP
AGND
Table 10. Recommended Output Capacitors for ADP2441
Vendor
Murata
Taiyo Yuden
GRM32DR71E106KA12L
TMK325B7106KN-TR
GRM32ER71E226KE15L
TMK325B7226MM-TR
GCM32ER70J476KE19L
JMK325B7476MM-TR
GRM31CR71H475KA12L
UMK325B7475MMT
RCOMP
VFB
10581-054
CAPACITANCE (µF)
24.6
Figure 59. Large Signal Model
ADP2441
Data Sheet
Correspondingly, there are three transfer functions:
VCOMP (s) VREF
=
× g m × Z COMP (s)
VOUT (s) VOUT
(14)
I L (s )
= GCS
VCOMP (s)
(15)
VOUT (s)
= Z FILT (s)
I L (s )
(16)
where:
gm is the transconductance of the error amplifier and equals
250 µA/V.
GCS is the current sense gain and equals 2 A/V.
VOUT is the output voltage of the regulator.
VREF is the internal reference voltage and equals 0.6 V.
ZCOMP(s) is the impedance of the RC compensation network that
forms a pole at the origin and a zero as expressed in Equation 17.
1 + s × RCOMP × CCOMP
ZCOMP (s) =
s × CCOMP
(17)
ZFILT(s) is the impedance of the output filter and is expressed as
Z FILT (s) =
RLOAD
1 + s × RLOAD × COUT
At the crossover frequency, the gain of the open-loop transfer
function is unity.
H(fCROSSOVER) = 1
This yields Equation 21 for the RC compensation network
impedance at the crossover frequency.
ZCOMP ( fCROSSOVER ) =
ZCOMP ( fCROSSOVER ) =
f ZERO =
1 + 2 × π × fCROSSOVER × RCOMP × CCOMP
2 × π × fCROSSOVER × CCOMP
(22)
1
2 × π × RCOMP × CCOMP
≈
fCROSSOVER
8
(23)
Solving Equation 21, Equation 22, and Equation 23 yields the
value for the resistor and capacitor in the RC compensation
network, as shown in Equation 24 and Equation 25.
(18)
(19)
(21)
To ensure that there is sufficient phase margin at the crossover
frequency, place the compensator zero at 1/8 of the crossover
frequency, as shown in the following equation:
RCOMP = 0.9 ×
CCOMP =
The overall loop gain, H(s), is obtained by multiplying the three
transfer functions previously mentioned as follows:
VREF
× Z COMP (s) × Z FILT (s)
VOUT
2 × π × fCROSSOVER × COUT VOUT
×
VREF
g m × GCS
Placing s = fCROSSOVER in Equation 17,
where s is the angular frequency, which can be written as s = 2πf.
H (s) = g m × GCS ×
(20)
2 × π × f CROSSOVER COUT × VOUT
×
g m × GCS
VREF
1
2 × π × f ZERO × RCOMP
(24)
(25)
Using these equations allows calculating the compensations for
the voltage loop.
When the switching frequency (fSW), output voltage (VOUT),
output inductor (L), and output capacitor (COUT) values are
selected, the unity crossover frequency can be set to 1/12 of the
switching frequency.
Rev. C | Page 22 of 32
Data Sheet
ADP2441
DESIGN EXAMPLE
Consider an application with the following specifications:
Soft Start Capacitor
•
•
•
•
•
•
•
For a given soft start time, the soft start capacitor can be calculated
using Equation 5,
VIN =24 V ± 10%
VOUT = 5 V ± 1%
Switching frequency = 700 kHz
Load = 800 mA typical
Maximum load current = 1 A
Soft start time = 6 ms
Overshoot ≤ 2% under all load transient conditions
I
VREF
= SS
t SS
C SS
C SS =
CONFIGURATION AND COMPONENTS SELECTION
CSS =
Resistor Divider
L IDEAL =
VREF
0. 6
=
= 10 kΩ
I STRING 60 μA
Choosing the switching frequency involves considering the
trade-off between efficiency and component size. Low frequency
improves the efficiency by reducing the gate losses but requires
a large inductor. The choice of high frequency is limited by the
minimum and maximum duty cycle.
= 18.66 μH ≈ 18.3 μH
The input filter consists of a small 0.1 µF ceramic capacitor
placed as close as possible to the IC.
The minimum input capacitance required for a particular load is
CIN _ MIN =
Duty Cycle
DNOMINAL = 20.8%
DMIN = 19%
DMAX = 23%
Based on the estimated duty cycle range, choose the switching
frequency according to the minimum and maximum duty cycle
limitations, as shown in Figure 55. For example, a 700 kHz,
frequency is well within the maximum and minimum duty
cycle limitations.
RFREQ
24 V × 700 kHz
Input Capacitor Selection
Table 11. Duty Cycle
92,500
=
f SW
3.3 × 5 V × (24 − 5) V
In Equation 9, VIN = 24 V, VOUT = 5 V, ILOAD(MAX) = 1 A, and fSW =
700 kHz, which results in L = 18.66 µH. When L = 18 μH (the
closest standard value) in Equation 8, ΔIL = 0.314 A. Although
the maximum output current required is 1 A, the maximum
peak current is 1.6 A. Therefore, the inductor should be rated
for higher than 1.6 A current.
Switching Frequency
Using Equation 4,
= 10 nF
3.3 × VOUT × (VIN − VOUT )
VIN × f SW
LIDEAL =
5 V − 0. 6 V
= 73.3 kΩ
RTOP = 10 kΩ ×
0. 6 V
VIN
24 V (Nominal)
26 V (10% Above Nominal)
22 V (10% Less than Nominal)
0. 6 V
Select the inductor by using Equation 9.
Using Equation 2 and Equation 3,
V
− VREF
RTOP = R BOTTOM × OUT
VREF
1 μA × 6 ms
Inductor Selection
The first step in selecting the external components is to
calculate the resistance of the resistor divider that sets the
output voltage.
RBOTTOM =
I SS × t SS
VREF
IOUT × D × (1 − D)
VPP × f SW
where:
VPP = 50 mV.
IOUT = 1 A.
D = 0.23.
fSW = 700 kHz.
Therefore,
C IN _ MIN =
1 A × 0.22 × (1 − 0.22)
0.05 V × 700 kHz
≈ 4.9 μF
Choosing an input capacitor of 10 µF with a voltage rating of
50 V ensures sufficient capacitance over voltage and temperature.
RFREQ = 132 kΩ
Rev. C | Page 23 of 32
ADP2441
Data Sheet
Output Capacitor Selection
Select the output capacitor by using Equation 12 and Equation 13:
COUT ( MIN ) ≅
∆I L
Table 13. Calculated Parameter Value
8 × f SW × (∆VRIPPLE − ∆I L × ESR)
Parameter
fCROSSOVER
fZERO
VREF
gm
Equation 12 is based on the output voltage ripple (ΔVRIPPLE),
which is 1% of the output voltage.
3
COUT ( MIN ) ≅ ∆I OUT (STEP )
f SW × ∆VDROOP
Selecting the crossover frequency to be 1/12 of the switching
frequency and placing the zero frequency at 1/8 of the crossover
frequency ensures that there is enough phase margin in the system.
Equation 13 calculates the capacitor selection based on the
transient load performance requirement of 2%. Perform these
calculations, and then use the equation that yields the larger
capacitor size to select a capacitor.
In this example, the values listed in Table 12 are substituted for
the variables in Equation 12 and Equation 13.
GCS
COUT
VOUT
Test Conditions/Comments
Fixed at 0.3 A for the ADP2441
1% of VOUT
2% of VOUT
Value
0.3 A
50 mV
100 mV
5 mΩ
700 kHz
The calculation based on the output voltage ripple (see
Equation 12) dictates that the minimum output capacitance is
COUT ( MIN ) ≅
0. 3 A
8 × 700 kHz × (50 mV − 0.3 A × 5 mΩ)
= 1.1 μF
whereas the calculation based on the transient load (see
Equation 13) dictates that the minimum output capacitance is
COUT ( MIN ) ≅ 0.5 ×
3
≈ 22 µF
700 kHz × 0.1 V
RCOMP = 0.9 ×
COUT = 1.5 × 22 µF = 32 µF
Compensation Selection
CCOMP =
CCOMP =
2 × π × 58.3 22 × 5
×
≈ 121 kΩ
250 × 2
0. 6
1
= 185 pF ≈ 180 pF
2 × π × 7.3 × 118
SYSTEM CONFIGURATION
Configure the system as follows:
1.
2.
3.
5.
Connect a capacitor of 1 µF between the VCC and PGND
pins and another capacitor of 1 µF between the VCC and
AGND pins. For best performance, use ceramic X5R or
X7R capacitors with a 25 V voltage rating.
Connect a ceramic capacitor of 10 nF with a 50 V voltage
rating between the BST and SW pins.
Connect a resistor between the FREQ and AGND pins as
close as possible to the IC.
If using the power-good feature, connect a pull-up resistor
of 50 kΩ to an external supply of 5 V.
Connect a capacitor of 10 nF between the SS and AGND pins.
If the tracking feature is needed, connect a resistor divider
between the TRK pin and another supply, as shown in
Figure 50.
See Figure 60 for a schematic of this design example and Table 14
for the calculated component values.
Calculate the compensation component values for the feedback
loop by using the following equations:
RCOMP = 0.9 ×
2 A/V
22 µF
5V
The closest standard resistor value is 118 kΩ. Therefore,
4.
To meet both requirements, use the value determined by the
latter equation. As shown in Figure 57, capacitance degrades
with dc bias; therefore, choose a capacitor that is 1.5 times the
calculated value.
Value
58.3 kHz
7.3 kHz
0.6 V
250 µA/V
Based on the values listed in Table 13, calculate the compensation value:
Table 12. Requirements
Parameter
Ripple Current
Voltage Ripple
Voltage Droop Due
to Load Transient
ESR
fSW
Test Conditions/Comments
1/12 of fSW
1/8 of fCROSSOVER
Fixed reference
Transconductance of error
amplifier
Current sense gain
Output capacitor
Output voltage
2 × π × f CROSSOVER COUT × VOUT
×
g m × GCS
VREF
1
2 × π × f ZERO × RCOMP
Rev. C | Page 24 of 32
Data Sheet
ADP2441
TYPICAL APPLICATION CIRCUITS
DESIGN EXAMPLE
VIN = 24 V ± 10%, VOUT = 5 V, fSW = 700 kHz.
C3
1µF/25V
BST
VCC
VIN
SS/TRK
FREQ
C1
4.7µF/
50V
L1
18µH
VOUT
5V, 1A
SW
C6
0.1µF
C7
22µF
C8
10µF
PGND
EXT
R7
50kΩ
PGOOD
R9
132kΩ
C11
10nF
TRK
10581-057
R2
73.3kΩ
COMP
EN
C2
4.7µF/
50V
ADP2441
PGOOD
C10
180pF
AGND
FB
R5
118kΩ
R3
10kΩ
VIN
24V
C5
10nF/50V
C4
1µF/25V
Figure 60. ADP2441 Typical Application Circuit, VIN = 24 V ± 10%, VOUT = 5 V, fSW = 700 kHz
Table 14. Calculated Component Values for Figure 60
Qty.
2
2
2
1
1
1
1
1
1
1
1
1
1
Reference Designator
C1, C2
C3, C4
C5, C11
C7
C8
L1
C6
C10
R9
R5
R2
R3
R7
Value
4.7 µF
1 µF
10 nF
22 µF
10 µF
18.3 µH
0.1 µF
185 pF
132 kΩ
118 kΩ
74 kΩ
10 kΩ
50 kΩ
Description
Capacitor ceramic, X7R, 50 V
Capacitor ceramic, 1 µF, 25 V, X7R, 10%, 0603
Capacitor ceramic, 10,000 pF, 50 V, 10%, X7R, 603
Capacitor ceramic, 22 µF, 25 V, X7R, 1210
Capacitor ceramic, 10 µF, 25 V, X7R, 1210
Inductor, 18.3 µH
Capacitor ceramic, 0.1 µF, 50 V, X7R, 0805
Capacitor ceramic, 50 V
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Rev. C | Page 25 of 32
Part Number
GRM31CR71H475KA12L
GRM188R71E105KA12D
ECJ-1VB1H103K
GRM32ER71E226K
GRM32DR71E106KA12L
CoilCraft MSS1260T-183NLB
ECJ-2FB1H104K
Vishay, Panasonic
ADP2441
Data Sheet
OTHER TYPICAL CIRCUIT CONFIGURATIONS
VIN = 24 V ± 10%, VOUT = 12 V, fSW = 600 kHz.
C3
1µF/25V
C4
1µF/25V
R2
191kΩ
VIN
24V
C2
4.7µF/
50V
C1
4.7µF/
50V
BST
VCC
SS/TRK
FREQ
COMP
EN
VIN
ADP2441
PGOOD
C10
220pF
AGND
FB
R5
121kΩ
R3
10kΩ
C5
10nF/50V
L1
33.3µH
VOUT
12V, 1A
SW
C6
0.1µF
PGND
C7
22µF/
25V
EXT
R7
50kΩ
FSW
600kHz
PGOOD
C11
10nF
TRK
10581-058
R9
154kΩ
Figure 61. ADP2441 Typical Application Circuit, VIN = 24 V ± 10%, VOUT = 12 V, fSW = 600 kHz
Table 15. Calculated Component Values for Figure 61
Qty.
2
2
2
1
1
1
1
1
1
1
1
1
Reference Designator
C1, C2
C3, C4
C5, C11
C7
L1
C6
C10
R9
R5
R2
R3
R7
Value
4.7 µF
1 µF
10 nF
22 µF
33.3 µH
0.1 µF
220 pF
154 kΩ
121 kΩ
191 kΩ
10 kΩ
50 kΩ
Description
Capacitor ceramic, X7R, 50 V
Capacitor ceramic, 1 µF, 25 V, X7R, 10%, 0603
Capacitor ceramic, 10,000 pF, 50 V, 10%, X7R, 0603
Capacitor ceramic, 22 µF, 25 V, X7R, 1210
Inductor, 33.3 µH
Capacitor ceramic, 0.1 µF, 50 V, X7R, 0805
Capacitor ceramic, 50 V
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Rev. C | Page 26 of 32
Part Number
GRM31CR71H475KA12L
GRM188R71E105KA12D
ECJ-1VB1H103K
GRM32ER71E226K
CoilCraft MSS1038-333ML
ECJ-2FB1H104K
Vishay, Panasonic
Data Sheet
ADP2441
VIN = 12 V ± 10%, VOUT = 5 V, fSW = 500 kHz.
C3
1µF/25V
C4
1µF/25V
R2
73.3kΩ
C2
4.7µF/
50V
C1
4.7µF/
50V
L1
18µH
VOUT
5V, 1A
SW
C6
0.1µF
PGND
EXT
R7
50kΩ
VIN
12V
BST
VCC
FREQ
SS/TRK
COMP
EN
VIN
ADP2441
PGOOD
C10
270pF
AGND
FB
R5
118kΩ
R3
10kΩ
C5
10nF/50V
C7
22µF
C8
22µF
FSW
500kHz
PGOOD
TRK
10581-059
R9
185kΩ
C11
10nF
Figure 62. ADP2441 Typical Application Circuit, VIN = 12 V ± 10%, VOUT = 5 V, fSW = 500 kHz
Table 16. Calculated Component Values for Figure 62
Qty.
2
2
2
1
1
1
1
1
1
1
1
1
1
Reference Designator
C1, C2
C3, C4
C5, C11
C7
C8
L1
C6
C10
R9
R5
R2
R3
R7
Value
4.7 µF
1 µF
10 nF
22 µF
22 µF
18.3 µH
0.1 µF
270 pF
185 kΩ
118 kΩ
74 kΩ
10 kΩ
50 kΩ
Description
Capacitor ceramic, X7R, 50 V
Capacitor ceramic, 1 µF, 25 V, X7R, 10%, 0603
Capacitor ceramic, 10,000 pF, 50 V, 10%, X7R, 0603
Capacitor ceramic, 22 µF, 25 V, X7R, 1210
Capacitor ceramic, 22 µF, 25 V, X7R, 1210
Inductor, 18.3 µH
Capacitor ceramic, 0.1 µF, 50 V, X7R, 0805
Capacitor ceramic, 50 V
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Rev. C | Page 27 of 32
Part Number
GRM31CR71H475KA12L
GRM188R71E105KA12D
ECJ-1VB1H103K
GRM32ER71E226K
CoilCraft MSS1038-183ML
ECJ-2FB1H104K
Vishay, Panasonic
ADP2441
Data Sheet
VIN = 36 V ± 10%, VOUT = 3.3 V, fSW = 300 kHz.
C3
1µF/25V
C4
1µF/25V
R2
45kΩ
EXT
R7
50kΩ
VIN
36V
C2
4.7µF/
50V
C1
4.7µF/
50V
BST
VCC
SS/TRK
FREQ
COMP
EN
VIN
ADP2441
PGOOD
C10
560pF
AGND
FB
R5
91kΩ
R3
10kΩ
C5
10nF/50V
L1
33.3µH
VOUT
3.3V, 1A
SW
C6
0.1µF
PGND
C7
47µF
C8
47µF
FSW
300kHz
PGOOD
C11
10nF
TRK
10581-060
R9
300kΩ
Figure 63. ADP2441 Typical Application Circuit, VIN = 36 V ± 10%, VOUT = 3.3 V, fSW = 300 kHz
Table 17. Calculated Component Values for Figure 63
Qty.
2
2
2
1
1
1
1
1
1
1
1
1
1
Reference Designator
C1, C2
C3, C4
C5, C11
C7
C8
L1
C6
C10
R9
R5
R2
R3
R7
Value
4.7 µF
1 µF
10 nF
47 µF
47 µF
33.3 µH
0.1 µF
560 pF
300 kΩ
91 kΩ
45 kΩ
10 kΩ
50 kΩ
Description
Capacitor ceramic, X7R, 50 V
Capacitor ceramic, 1 µF, 25 V, X7R, 10%, 0603
Capacitor ceramic, 10,000 pF, 50 V, 10%, X7R, 0603
Capacitor ceramic, 47 µF, 6.3 V, X7R, 1210
Capacitor ceramic, 47 µF, 6.3 V, X7R, 1210
Inductor, 33.3 µH
Capacitor ceramic, 0.1 µF, 50 V, X7R, 0805
Capacitor ceramic, 50 V
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Resistor, 1/10 W, 1%, 0603, SMD
Rev. C | Page 28 of 32
Part Number
GRM31CR71H475KA12L
GRM188R71E105KA12D
ECJ-1VB1H103K
GRM32ER70J476KE20L
GRM32ER70J476KE20L
CoilCraft MSS1038T-333ML
ECJ-2FB1H104K
Vishay, Panasonic
Data Sheet
ADP2441
POWER DISSIPATION AND THERMAL CONSIDERATIONS
POWER DISSIPATION
Transition Losses
The efficiency of a dc-to-dc regulator is
where:
PIN is the input power.
POUT is the output power.
Transition losses occur because the N-channel MOSFET power
switch cannot turn on or off instantaneously. During a switch
node transition, the power switch provides all of the inductor
current, and the source-to-drain voltage of the power switch is
half the input, resulting in power loss. Transition losses increase
as the load current and input voltage increase, and these losses
occur twice for each switching cycle.
The power loss of a dc-to-dc regulator is
The transition losses can be calculated as follows:
Efficiency
POUT
100%
PIN
(26)
PLOSS = PIN − POUT
PTRANS
There are four main sources of power loss in a dc-to-dc regulator:
THERMAL CONSIDERATIONS
The power dissipated by the regulator increases the die junction
temperature, TJ, above the ambient temperature, TA, as follows:
Inductor Losses
Inductor conduction losses are caused by the flow of current
through the inductor DCR (internal resistance).
The inductor power loss (excluding core loss) is
(27)
Power Switch Conduction Losses
Power switch conductive losses are due to the output current, IOUT,
flowing through the N-channel MOSFET power switches that
have internal resistance, RDS(ON). The amount of power loss can
be approximated as follows:
PCOND = [RDS(ON) – High Side × D + RDS(ON) − Low Side × (1 − D)] × IOUT2 (28)
Switching Losses
Switching losses are associated with the current drawn by the
driver to turn the power devices on and off at the switching
frequency. Each time a power device gate is turned on and off,
the driver transfers a charge (ΔQ) from the input supply to the
gate and then from the gate to ground.
The amount of switching loss can by calculated as follows:
PSW = QG_TOTAL × VIN × fSW
(30)
where tON and tOFF are the rise time and fall time of the switch
node and are each approximately 10 ns for a 24 V input.
Inductor losses
Power switch conduction losses
Switching losses
Transition losses
PL = IOUT2 × DCRL
VIN
I OUT (t ON t OFF ) f SW
2
(29)
TJ = TA + TR
(31)
where the temperature rise, TR, is proportional to the power
dissipation, PD, in the package.
The proportionality coefficient is defined as the thermal
resistance from the junction temperature of the die to the
ambient temperature as follows:
TR = θJA + PD
(32)
where θJA is the junction-to-ambient thermal resistance and
equals 40°C/W for the JEDEC board (see Table 3).
When designing an application for a particular ambient
temperature range, calculate the expected ADP2441 power
dissipation (PD) due to the conduction, switching, and transition
losses using Equation 28, Equation 29, and Equation 30, and
then estimate the temperature rise using Equation 31 and
Equation 32. Improved thermal performance can be achieved
by implementing good board layout. For example, on the ADP2441
evaluation board (ADP2441-EVALZ), the measured θJA is
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