ADXL150JQC

a 5 g to 50 g, Low Noise, Low Power, Single/Dual Axis iMEMS® Accelerometers ADXL150/ADXL250 FUNCTIONAL BLOCK DIAGRAMS TP (DO NOT CONNECT) +VS 0.1 F SENSOR +VS 2 FEATURES Complete Acceleration Measurement System on a Single Monolithic IC 80 dB Dynamic Range Pin Programmable 50 g or 25 g Full Scale Low Noise: 1 m g /√ Hz Typical Low Power: +2.5V, R2 CONNECTS TO GND. (e) FOR VPIN 10 < +2.5V, R2 CONNECTS TO +VS. DESIRED FS OUTPUT RANGE SCALE FACTOR 76mV/g 100mV/g 200mV/g 400mV/g 25g 20g 10g 5g EXT AMP GAIN 2.0 2.6 5.3 10.5 R1 VALUE 49.9k 38.3k 18.7k 9.53k Figure 17. “Quick Zero g Calibration” Connection Adjusting the Zero g Bias Level When a true dc (gravity) response is needed, the output from the accelerometer must be dc coupled to the external amplifier’s input. For high gain applications, a zero g offset trim will also be needed. The external offset trim permits the user to set the zero g offset voltage to exactly +2.5 volts (allowing the maximum output swing from the external amplifier without clipping with a +5 supply). With a dc coupled connection, any difference between the zero g output and +2.5 V will be amplified along with the signal. To obtain the exact zero g output desired or to allow the maximum output voltage swing from the external amplifier, the zero g offset will need to be externally trimmed using the circuit of Figure 20. The external amplifier’s maximum output swing should be limited to ± 2 volts, which provides a safety margin of ± 0.25 volts before clipping. With a +2.5 volt zero g level, the maximum gain will equal: 2 Volts 38 mV /g Times the Max Applied Acceleration in g The device scale factor and zero g offset levels can be calibrated using the earth’s gravity, as explained in the section “calibrating the ADXL150/ADXL250.” Using the Zero g “Quick-Cal” Method In Figure 18 (accelerometer alone, no external op amp), a trim potentiometer connects directly to the accelerometer’s zero g null pin. The “quick offset calibration” scheme shown in Figure 17 is preferred over using a potentiometer, which could change its setting over time due to vibration. The “quick offset calibration” method requires measuring only the output voltage of the ADXL150/ADXL250 while it is oriented normal to the earth’s gravity. Then, by using the simple equations shown in the figures, the correct resistance value for R2 can be calculated. In Figure 17, an external op amp is used to amplify the signal. A resistor, R2, is connected to the op amp’s summing junction. The other side of R2 connects to either ground or +VS depending on which direction the offset needs to be shifted. TP (DO NOT CONNECT) 5 +VS C1 0.1 F 14 ADXL150 GAIN AMP SENSOR +VS 2 5k 10 DEMODULATOR 25k CLOCK BUFFER AMP 8 VOUT 9 SELF-TEST COM 7 C2 0.1 F OFFSET NULL RIN AT PIN 8 30k 10k 200k +VS Figure 18. Offset Nulling the ADXL150/ADXL250 Using a Trim Potentiometer REV. 0 –9– ADXL150/ADXL250 DEVICE BANDWIDTH VS. MEASUREMENT RESOLUTION Although an accelerometer is usually specified according to its full-scale g level, the limiting resolution of the device, i.e., its minimum discernible input level, is extremely important when measuring low g accelerations. 100mg 660mg approximately 1.6 times the 3 dB bandwidth. For example, the typical rms noise of the ADXL150 using a 100 Hz one pole post filter is: Noise rms = 1 mg/ Hz × 100 1.6 = 12.25 mg () () NOISE LEVEL – Peak to Peak NOISE LEVEL – rms Because the ADXL150/ADXL250’s noise is, for all practical purposes, Gaussian in amplitude distribution, the highest noise amplitudes have the smallest (yet nonzero) probability. Peakto-peak noise is therefore difficult to measure and can only be estimated due to its statistical nature. Table I is useful for estimating the probabilities of exceeding various peak values, given the rms value. Table I. 10mg 66mg Nominal Peak-toPeak Value 2.0 × rms 4.0 × rms 6.0 × rms 6.6 × rms 8.0 × rms % of Time that Noise Will Exceed Nominal Peak-to-Peak Value 32% 4.6% 0.27% 0.1% 0.006% 1mg 10 100 3dB BANDWIDTH – Hz 6.6mg 1k Figure 19. ADXL150/ADXL250 Noise Level vs. 3 dB Bandwidth (Using a “Brickwall” Filter) The limiting resolution is predominantly set by the measurement noise “floor,” which includes the ambient background noise and the noise of the ADXL150/ADXL250 itself. The level of the noise floor varies directly with the bandwidth of the measurement. As the measurement bandwidth is reduced, the noise floor drops, improving the signal-to-noise ratio of the measurement and increasing its resolution. The bandwidth of the accelerometer can be easily reduced by adding low-pass or bandpass filtering. Figure 19 shows the typical noise vs. bandwidth characteristic of the ADXL150/ ADXL250. The output noise of the ADXL150/ADXL250 scales with the square root of the measurement bandwidth. With a single pole roll-off, the equivalent rms noise bandwidth is π divided by 2 or RMS and peak-to-peak noise (for 0.1% uncertainty) for various bandwidths are estimated in Figure 19. As shown by the figure, device noise drops dramatically as the operating bandwidth is reduced. For example, when operated in a 1 kHz bandwidth, the ADXL150/ADXL250 typically have an rms noise level of 32 m g. When the device bandwidth is rolled off to 100 Hz, the noise level is reduced to approximately 10 mg. Alternatively, the signal-to-noise ratio may be improved considerably by using a microprocessor to perform multiple measurements and then to compute the average signal level. Low-Pass Filtering The bandwidth of the accelerometer can easily be reduced by using post filtering. Figure 20 shows how the buffer amplifier can be connected to provide 1-pole post filtering, zero g offset trimming, and output scaling. The table provides practical component values +VS R2 1M Cf 5k 10 TP (DO NOT CONNECT) 5 RT 200k 0g TRIM +VS 2 +VS C1 0.1 F 14 ADXL150 GAIN AMP SENSOR R1a 75k R1b 50k R3 100k +VS 0.1 F 2 7 DEMODULATOR 25k 7 8 BUFFER AMP CLOCK 9 SCALE FACTOR TRIM (OPTIONAL) SELF-TEST COM OFFSET NULL 0.1 F +VS 2 3 OP196 4 6 VOUT DESIRED F.S. OUTPUT RANGE SCALE FACTOR 76mV/g 100mV/g 200mV/g 400mV/g 25g 20g 10g 5g EXT AMP GAIN 2.0 2.6 5.3 10.5 R3 Cf ( F) Cf ( F) Cf ( F) VALUE 100Hz 30Hz 10Hz 200k 261k 536k 1M 0.0082 0.027 0.0056 0.022 0.0033 0.010 0.082 0.056 0.033 0.0015 0.0056 0.015 Figure 20. One-Pole Post Filter Circuit with SF and Zero g Offset Trims – 10– REV. 0 ADXL150/ADXL250 for various full-scale g levels and approximate circuit bandwidths. For bandwidths other than those listed, use the formula: Cf = ( 2 π R 3 Desired 3 dB Bandwidth in Hz ) 1 or simply scale the value of capacitor Cf accordingly; i.e., for an application with a 50 Hz bandwidth, the value of Cf will need to be twice as large as its 100 Hz value. If further noise reduction is needed while maintaining the maximum possible bandwidth, a 2- or 3-pole post filter is recommended. These provide a much steeper roll-off of noise above the pole frequency. Figure 21 shows a circuit that provides 2-pole post filtering. Component values for the 2-pole filter were selected to operate the first op amp at unity gain. Capacitors C3 and C4 were chosen to provide 3 dB bandwidths of 10 Hz, 30 Hz, 100 Hz and 300 Hz. The second op amp offsets and scales the output to provide a +2.5 V ± 2 V output over a wide range of full-scale g levels. APPLICATION HINTS ADXL250 Power Supply Pins Figure 22 shows how both the zero g offset and output sensitivity of the ADXL150/ADXL250 vary with changes in supply voltage. If they are to be used with nonratiometric devices, such as an ADC with a built-in 5 V reference, then both components should be referenced to the same source, in this case the ADC reference. Alternatively, the circuit can be powered from an external +5 volt reference. 2.65 40.25 2.60 39.50 2.55 0g OFFSET 38.75 SENSITIVITY 2.50 38.00 2.45 37.25 2.40 36.50 2.35 5.25 5.20 5.15 5.10 5.05 5.00 4.95 4.90 4.85 4.80 4.75 POWER SUPPLY VOLTAGE 35.75 When wiring the ADXL250, be sure to connect BOTH power supply terminals, Pins 14 and 13. Ratiometric Operation Figure 22. Typical Ratiometric Operation Ratiometric operation means that the circuit uses the power supply as its voltage reference. If the supply voltage varies, the accelerometer and the other circuit components (such as an ADC, etc.) track each other and compensate for the change. Since any voltage variation is transferred to the accelerometer’s output, it is important to reduce any power supply noise. Simply following good engineering practice of bypassing the power supply right at Pin 14 of the ADXL150/ADXL250 with a 0.1 µF capacitor should be sufficient. TP (DO NOT CONNECT) 5 R3 82.5k +VS 2 C4 5k R1 82.5k 10 +VS C1 0.1 F 14 ADXL150 GAIN AMP SENSOR +VS 0.1 F R2 42.2k 2 8 DEMODULATOR 25k CLOCK BUFFER AMP 8 C3 3 TYPICAL FILTER VALUES BW 300Hz 100Hz 30Hz 10Hz 1/2 OP296 1 C3 C4 9 0.027 F 0.0033 F 0.082 F 0.01 F 0.27 F 0.82 F 0.033 F 0.1 F EXT AMP GAIN 2.0 2.6 5.3 10.5 7 2-POLE FILTER +VS 2 +VS 2 SELF-TEST COM OFFSET NULL C2 0.1 F SCALING AMPLIFIER 7 5 DESIRED F.S. OUTPUT RANGE SCALE FACTOR 76mV/g 100mV/g 200mV/g 400mV/g ±25g ±20g ±10g ±5g R5 VALUE 200k 261k 536k 1M OUTPUT 1/2 OP296 4 6 R4 100k R6 1M +VS 200k 0g TRIM R5 Figure 21. Two-Pole Post Filter Circuit REV. 0 –11– ADXL150/ADXL250 Additional Noise Reduction Techniques CALIBRATING THE ADXL150/ADXL250 Shielded wire should be used for connecting the accelerometer to any circuitry that is more than a few inches away—to avoid 60 Hz pickup from ac line voltage. Ground the cable’s shield at only one end and connect a separate common lead between the circuits; this will help to prevent ground loops. Also, if the accelerometer is inside a metal enclosure, this should be grounded as well. Mounting Fixture Resonances A common source of error in acceleration sensing is resonance of the mounting fixture. For example, the circuit board that the ADXL150/ADXL250 mounts to may have resonant frequencies in the same range as the signals of interest. This could cause the signals measured to be larger than they really are. A common solution to this problem is to damp these resonances by mounting the ADXL150/ADXL250 near a mounting post or by adding extra screws to hold the board more securely in place. When testing the accelerometer in your end application, it is recommended that you test the application at a variety of frequencies to ensure that no major resonance problems exist. REDUCING POWER CONSUMPTION If a calibrated shaker is not available, both the zero g level and scale factor of the ADXL150/ADXL250 may be easily set to fair accuracy by using a self-calibration technique based on the 1 g acceleration of the earth’s gravity. Figure 24 shows how gravity and package orientation affect the ADXL150/ADXL250’s output. With its axis of sensitivity in the vertical plane, the ADXL150/ADXL250 should register a 1 g acceleration, either positive or negative, depending on orientation. With the axis of sensitivity in the horizontal plane, no acceleration (the zero g bias level) should be indicated. The use of an external buffer amplifier may invert the polarity of the signal. 8 7 1 14 8 7 1 14 8 14 1 7 14 1 7 8 0g 0g (a) (b) +1g (c) –1g (d) The use of a simple power cycling circuit provides a dramatic reduction in the accelerometer’s average current consumption. In low bandwidth applications such as shipping recorders, a simple, low cost circuit can provide substantial power reduction. If a microprocessor is available, it can supply a TTL clock pulse to toggle the accelerometer’s power on and off. A 10% duty cycle, 1 ms on, 9 ms off, reduces the average current consumption of the accelerometer from 1.8 mA to 180 µA, providing a power reduction of 90%. Figure 23 shows the typical power-on settling time of the ADXL150/ADXL250. VS 5.0 4.5 4.0 VOLTAGE – Volts 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 0 0.04 0.08 0.12 0.16 0.20 TIME – ms 0.24 0.28 0.32 0.36 VOUT + 50g 0.5V VOUT = 0g 0.5V VOUT – 50g Figure 24. Using the Earth’s Gravity to SelfCalibrate the ADXL150/ADXL250 Figure 24 shows how to self-calibrate the ADXL150/ADXL250. Place the accelerometer on its side with its axis of sensitivity oriented as shown in “a.” (For the ADXL250 this would be the “X” axis—its “Y” axis is calibrated in the same manner, but the part is rotated 90° clockwise.) The zero g offset potentiometer RT is then roughly adjusted for midscale: +2.5 V at the external amp output (see Figure 20). Next, the package axis should be oriented as in “c” (pointing down) and the output reading noted. The package axis should then be rotated 180° to position “d” and the scale factor potentiometer, R1b, adjusted so that the output voltage indicates a change of 2 gs in acceleration. For example, if the circuit scale factor at the external buffer’s output is 100 mV per g, the scale factor trim should be adjusted so that an output change of 200 mV is indicated. Self-Test Function Figure 23. Typical Power-On Settling with Full-Scale Input. Time Constant of Post Filter Dominates the Response When a Signal Is Present. A Logic “1” applied to the self-test (ST) input will cause an electrostatic force to be applied to the sensor that will cause it to deflect. If the accelerometer is experiencing an acceleration when the self-test is initiated, the output will equal the algebraic sum of the two inputs. The output will stay at the self-test level as long as the ST input remains high, and will return to the actual acceleration level when the ST voltage is removed. Using an external amplifier to increase output scale factor may cause the self-test output to overdrive the buffer into saturation. The self-test may still be used in this case, but the change in the output must then be monitored at the accelerometer’s output instead of the external amplifier’s output. Note that the value of the self-test delta is not an exact indication of the sensitivity (mV/g) and therefore may not be used to calibrate the device for sensitivity error. – 12– REV. 0 ADXL150/ADXL250 MINIMIZING EMI/RFI The architecture of the ADXL150/ADXL250, and its use of synchronous demodulation, makes the device immune to most electromagnetic (EMI) and radio frequency (RFI) interference. The use of synchronous demodulation allows the circuit to reject all signals except those at the frequency of the oscillator driving the sensor element. However, the ADXL150/ADXL250 have a sensitivity to noise on the supply lines that is near its internal clock frequency (approximately 100 kHz) or its odd harmonics and can exhibit baseband errors at the output. These error signals are the beat frequency signals between the clock and the supply noise. Such noise can be generated by digital switching elsewhere in the system and must be attenuated by proper bypassing. By inserting a small value resistor between the accelerometer and its power supply, an RC filter is created. This consists of the resistor and the accelerometer’s normal 0.1 µF bypass capacitor. For example if R = 20 Ω and C = 0.1 µF, a filter with a pole at 80 kHz is created, which is adequate to attenuate noise on the supply from most digital circuits, with proper ground and supply layout. Power supply decoupling, short component leads, physically small (surface mount, etc.) components and attention to good grounding practices all help to prevent RFI and EMI problems. Good grounding practices include having separate analog and digital grounds (as well as separate power supplies or very good decoupling) on the printed circuit boards. INTERFACING THE ADXL150/ADXL250 SERIES iMEMS ACCELEROMETERS WITH POPULAR ANALOG-TODIGITAL CONVERTERS. Basic Issues In selecting an appropriate ADC to use with our accelerometer we need to find a device that has a resolution better than the measurement resolution but, for economy’s sake, not a great deal better. For most applications, an 8- or 10-bit converter is appropriate. The decision to use a 10-bit converter alone, or to use a gain stage together with an 8-bit converter, depends on which is more important: component cost or parts count and ease of assembly. Table II shows some of the tradeoffs involved. Table II. 8-Bit Converter and 10-Bit (or 12-Bit) Op Amp Preamp Converter Advantages: Low Cost Converter Disadvantages: Needs Op Amp Needs Zero g Trim Higher Cost Converter No Zero g Trim Required Adding amplification between the accelerometer and the ADC will reduce the circuit’s full-scale input range but will greatly reduce the resolution requirements (and therefore the cost) of the ADC. For example, using an op amp with a gain of 5.3 following the accelerometer will increase the input drive to the ADC from 38 mV/g to 200 mV/g. Since the signal has been gained up, but the maximum full-scale (clipping) level is still the same, the dynamic range of the measurement has also been reduced by 5.3. Table III. Typical System Resolution Using Some Popular ADCs Being Driven with and without an Op Amp Preamp Converter SF mV/Bit Preamp in (5 V/2n) Gain mV/g 19.5 mV 19.5 mV 19.5 mV 19.5 mV None 2 2.63 5.26 None 2 2.63 5.26 None 2 2.63 5.26 38 76 100 200 38 76 100 200 38 76 100 200 FS Range in g’s ± 50 ± 25 ± 20 ± 10 ± 50 ± 25 ± 20 ± 10 ± 50 ± 25 ± 20 ± 10 System Resolution in g’s (p-p) 0.51 0.26 0.20 0.10 0.13 0.06 0.05 0.02 0.03 0.02 0.01 0.006 The ADXL150/ADXL250 Series accelerometers were designed to drive popular analog-to-digital converters (ADCs) directly. In applications where both a ± 50 g full-scale measurement range and a 1 kHz bandwidth are needed, the VOUT terminal of the accelerometer is simply connected to the VIN terminal of the ADC as shown in Figure 25a. The accelerometer provides its (nominal) factory preset scale factor of +2.5 V ± 38 mV/g which drives the ADC input with +2.5 V ± 1.9 V when measuring a 50 g full-scale signal (38 mV/g × 50 g = 1.9 V). As stated earlier, the use of post filtering will dramatically improve the accelerometer’s low g resolution. Figure 25b shows a simple post filter connected between the accelerometer and the ADC. This connection, although easy to implement, will require fairly large values of Cf, and the accelerometer’s signal will be loaded down (causing a scale factor error) unless the ADC’s input impedance is much greater than the value of Rf. ADC input impedance’s range from less than 1.5 kΩ up to greater than 15 kΩ with 5 kΩ values being typical. Figure 25c is the preferred connection for implementing low-pass filtering with the added advantage of providing an increase in scale factor, if desired. Calculating ADC Requirements Converter Type 2n 8 Bit 256 256 256 256 10 Bit 1,024 4.9 mV 1,024 4.9 mV 1,024 4.9 mV 1,024 4.9 mV 12 Bit 4,096 1.2 mV 4,096 1.2 mV 4,096 1.2 mV 4,096 1.2 mV The resolution of commercial ADCs is specified in bits. In an ADC, the available resolution equals 2n, where n is the number of bits. For example, an 8-bit converter provides a resolution of 28 which equals 256. So the full-scale input range of the converter divided by 256 will equal the smallest signal it can resolve. Table III is a chart showing the required ADC resolution vs. the scale factor of the accelerometer with or without a gain amplifier. Note that the system resolution specified in the table refers REV. 0 –13– ADXL150/ADXL250 to that provided by the converter and preamp (if used). It is necessary to use sufficient post filtering with the accelerometer to reduce its noise floor to allow full use of the converter’s resolution (see post filtering section). The use of a gain stage following the accelerometer will normally require the user to adjust the zero g offset level (either by trimming or by resistor selection—see previous sections). For many applications, a modern “economy priced” 10-bit converter, such as the AD7810 allows you to have high resolution without using a preamp or adding much to the overall circuit cost. In addition to simplicity and cost, it also meets two other necessary requirements: it operates from a single +5 V supply and is very low power. +VS +VS XL VOUT ADC a. Direct Connection, No Signal Amplification or Post Filtering +VS +VS RF ADC Cf INPUT RESISTANCE XL VOUT b. Single-Pole Post Filtering, No Signal Amplification +VS 0g OFFSET ADJUST R1 RF VOUT VOS NULL PIN Cf +VS ADC XL c. Single-Pole Post Filtering and Signal Amplification Figure 25. Interfacing the ADXL150/ADXL250 Series Accelerometers to an ADC – 14– REV. 0 ADXL150/ADXL250 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 14-Lead Cerpac (QC-14) 0.390 (9.906) MAX 14 8 0.291 (7.391) 0.285 (7.239) 1 7 0.419 (10.643) 0.394 (10.008) PIN 1 0.300 (7.62) 0.195 (4.953) 0.115 (2.921) 0.345 (8.763) 0.290 (7.366) 0.020 (0.508) 0.004 (0.102) 0.215 (5.461) 0.119 (3.023) 0.050 (1.27) BSC 0.020 (0.508) 0.013 (0.330) 0.0125 (0.318) 0.009 (0.229) 0.050 (1.270) 0.016 (0.406) 8 0 SEATING PLANE REV. 0 –15– PRINTED IN U.S.A. C2949–8–4/98