LT1074/LT1076
Step-Down Switching
Regulator
ponents, are included on the chip. The topology is a classic
positive “buck” configuration but several design innovations allow this device to be used as a positive-to-negative
converter, a negative boost converter, and as a flyback
converter. The switch output is specified to swing 40V
below ground, allowing the LT1074 to drive a tappedinductor in the buck mode with output currents up to 10A.
FEATURES
■
■
■
■
■
■
■
■
5A Onboard Switch (LT1074)
Operates Up to 60V Input
100kHz Switching Frequency
Greatly Improved Dynamic Behavior
Available in Low Cost 5 and 7-Lead Packages
Only 8.5mA Quiescent Current
Programmable Current Limit
Micropower Shutdown Mode
U
APPLICATIO S
■
■
■
■
■
Buck Converter with Output Voltage Range of 2.5V
to 50V
Tapped-Inductor Buck Converter with 10A Output
at 5V
Positive-to-Negative Converter
Negative Boost Converter
Multiple Output Buck Converter
U
DESCRIPTIO
The LT®1074 is a 5A (LT1076 is rated at 2A) monolithic
bipolar switching regulator which requires only a few
external parts for normal operation. The power switch, all
oscillator and control circuitry, and all current limit com-
The LT1074 uses a true analog multiplier in the feedback
loop. This makes the device respond nearly instantaneously to input voltage fluctuations and makes loop gain
independent of input voltage. As a result, dynamic behavior of the regulator is significantly improved over previous
designs.
On-chip pulse by pulse current limiting makes the LT1074
nearly bust-proof for output overloads or shorts. The input
voltage range as a buck converter is 8V to 60V, but a selfboot feature allows input voltages as low as 5V in the
inverting and boost configurations.
The LT1074 is available in low cost TO-220 or DD packages
with frequency pre-set at 100kHz and current limit at 6.5A
(LT1076 = 2.6A). A 7-pin TO-220 package is also available
which allows current limit to be adjusted down to zero. In
addition, full micropower shutdown can be programmed.
See Application Note 44 for design details.
A fixed 5V output, 2A version is also available. See LT1076-5.
, LTC and LT are registered trademarks of Linear Technology Corporation.
U
TYPICAL APPLICATIO
Buck Converter Efficiency
Basic Positive Buck Converter
VIN
VSW
LT1074
GND
+
VC
MBR745*
FB
R3
2.7k
C3†
200µF
5V
5A
C2
0.01µF
R1
2.8k
1%
R2
2.21k
1% +
C1
500µF
25V
* USE MBR340 FOR LT1076
** COILTRONICS #50-2-52 (LT1074)
#100-1-52 (LT1076)
PULSE ENGINEERING, INC.
#PE-92114 (LT1074)
#PE-92102 (LT1076)
HURRICANE #HL-AK147QQ (LT1074)
#HL-AG210LL (LT1076)
†
RIPPLE CURRENT RATING ≥ IOUT/2
EFFICIENCY (%)
L1**
50µH (LT1074)
100µH (LT1076)
10V TO 40V
LT1074
100
VOUT = 12V, V IN = 20V
90
80
VOUT = 5V, V IN = 15V
70
L = 50µH TYPE 52 CORE
DIODE = MBR735
60
50
0
1
2
3
5
4
6
OUTPUT LOAD CURRENT (A)
LT1074•TA01
LT1074•TPC27
sn1074 1074fds
1
LT1074/LT1076
U
W W
W
ABSOLUTE
AXI U RATI GS
(Note 1)
Input Voltage
LT1074/ LT1076 .................................................. 45V
LT1074HV/LT1076HV ......................................... 64V
Switch Voltage with Respect to Input Voltage
LT1074/ LT1076 .................................................. 64V
LT1074HV/LT1076HV ......................................... 75V
Switch Voltage with Respect to Ground Pin (VSW Negative)
LT1074/LT1076 (Note 7) ..................................... 35V
LT1074HV/LT1076HV (Note 7) ........................... 45V
Feedback Pin Voltage ..................................... –2V, +10V
Shutdown Pin Voltage (Not to Exceed VIN) .............. 40V
ILIM Pin Voltage (Forced) ............................................ 5.5V
Maximum Operating Ambient Temperature Range
Commercial ................................................. 0°C to 70°C
Industrial ................................................ –40°C to 85°C
Military (OBSOLETE) ..................... –55°C to 125°C
Maximum Operating Junction Temperature Range
Commercial ............................................... 0°C to 125°C
Industrial .............................................. –40°C to 125°C
Military (OBSOLETE) .................... – 55°C to 150°C
Maximum Storage Temperature ............... –65°C to 150°C
Lead Temperature (Soldering, 10 sec) ...................... 300°C
U
W
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
FRONT VIEW
TAB IS
GND
5
VIN
4
VSW
3
GND
2
VC
1
FB/SENSE
4
VSW
LT1074: θJC = 2.5°C, θJA = 35°C/W
LT1076: θJC = 4°C, θJA = 35°C/W
LT1076CR
LT1076IR
LT1076HVCR
LT1076HVIR
SHDN
VC
FB/SENSE
GND
ILIM
VSW
VIN
R PACKAGE
7-LEAD PLASTIC DD
OBSOLETE PACKAGE
FRONT VIEW
SHDN
VC
FB
GND
ILIM
VSW
VIN
LT1074CT7
LT1074HVCT7
LT1074IT7
LT1074HVIT7
LT1076CT7
LT1076HVCT7
LT1074CK
LT1074HVCK
LT1074MK
LT1074HVMK
LT1076CK
LT1076HVCK
LT1076MK
LT1076HVMK
Consider the T5 Package for Alternate Source
FRONT VIEW
TAB IS
GND
LT1076: θJC = 4°C, θJA = 30°C/W
7
6
5
4
3
2
1
CASE
IS GND
3
K PACKAGE
4-LEAD TO-3 METAL CAN
FRONT VIEW
TAB IS
GND
2
FB
LT1076: θJC = 4°C, θJA = 30°C/W
TAB IS
GND
VIN
1
LT1076CQ
LT1076IQ
Q PACKAGE
5-LEAD PLASTIC DD
7
6
5
4
3
2
1
ORDER PART
NUMBER
BOTTOM VIEW
VC
5
VIN
4
VSW
3
GND
2
VC
1
FB
T PACKAGE
5-LEAD PLASTIC TO-220
LEADS ARE FORMED STANDARD FOR
STRAIGHT LEADS, ORDER FLOW 06
LT1074CT
LT1074HVCT
LT1074IT
LT1074HVIT
LT1076CT
LT1076HVCT
LT1076IT
LT1076HVIT
LT1074: θJC = 2.5°C, θJA = 50°C/W
LT1076: θJC = 4°C, θJA = 50°C/W
T7 PACKAGE
7-LEAD PLASTIC TO-220
LT1074: θJC = 2.5°C, θJA = 50°C/W
LT1076: θJC = 4°C, θJA = 50°C/W
*Assumes package is soldered to 0.5 IN2 of 1 oz. copper over internal ground plane or over back side plane.
Consult LTC Marketing for parts specified with wider operating temperature ranges.
sn1074 1074fds
2
LT1074/LT1076
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Tj = 25°C, VIN = 25V, unless otherwise noted.
PARAMETER
CONDITIONS
Switch “On” Voltage (Note 2)
LT1074
ISW = 1A, Tj ≥ 0°C
ISW = 1A, Tj < 0°C
ISW = 5A, Tj ≥ 0°C
ISW = 5A, Tj < 0°C
LT1076
ISW = 0.5A
ISW = 2A
LT1074
VIN ≤ 25V, VSW = 0
VIN = VMAX, VSW = 0 (Note 8)
LT1076
VIN = 25V, VSW = 0
VIN = VMAX, VSW = 0 (Note 8)
Switch “Off” Leakage
MIN
TYP
●
●
5
10
MAX
UNITS
1.85
2.1
2.3
2.5
V
V
V
V
1.2
1.7
V
V
300
500
µA
µA
150
250
µA
µA
Supply Current (Note 3)
VFB = 2.5V, VIN ≤ 40V
40V < VIN < 60V
VSHUT = 0.1V (Device Shutdown) (Note 9)
●
●
●
8.5
9
140
11
12
300
mA
mA
µA
Minimum Supply Voltage
Normal Mode
Startup Mode (Note 4)
●
●
7.3
3.5
8
4.8
V
V
Switch Current Limit (Note 5)
LT1074
ILIM Open
RLIM = 10k (Note 6)
RLIM = 7k (Note 6)
●
5.5
6.5
4.5
3
8.5
A
A
A
LT1076
ILIM Open
RLIM = 10k (Note 6)
RLIM = 7k (Note 6)
●
2
2.6
1.8
1.2
3.2
A
A
A
●
85
90
100
Tj ≤ 125°C
Tj > 125°C
VFB = 0V through 2kΩ (Note 5)
●
●
90
85
85
110
120
125
kHz
kHz
kHz
kHz
Switching Frequency Line Regulation
8V ≤ VIN ≤ VMAX (Note 8)
●
0.1
%/V
Error Amplifier Voltage Gain (Note 7)
1V ≤ VC ≤ 4V
Maximum Duty Cycle
Switching Frequency
20
0.03
2000
Error Amplifier Transconductance
Error Amplifier Source and Sink Current
%
Source (VFB = 2V)
Sink (VFB = 2.5V)
V/V
3700
5000
8000
µmho
100
0.7
140
1
225
1.6
µA
mA
0.5
2
µA
2.155
2.21
2.265
V
Feedback Pin Bias Current
VFB = VREF
●
Reference Voltage
VC = 2V
●
Reference Voltage Tolerance
VREF (Nominal) = 2.21V
All Conditions of Input Voltage, Output
Voltage, Temperature and Load Current
●
±0.5
±1
±1.5
±2.5
%
%
8V ≤ VIN ≤ VMAX (Note 8)
●
0.005
0.02
%/V
●
1.5
–4
V
mV/°C
24
V
Reference Voltage Line Regulation
VC Voltage at 0% Duty Cycle
Over Temperature
Multiplier Reference Voltage
Shutdown Pin Current
VSH = 5V
VSH ≤ VTHRESHOLD (≅2.5V)
●
●
5
10
20
50
µA
µA
Shutdown Thresholds
Switch Duty Cycle = 0
Fully Shut Down
●
●
2.2
0.1
2.45
0.3
2.7
0.6
V
V
Thermal Resistance Junction to Case
LT1074
LT1076
2.5
4.0
°C/W
°C/W
sn1074 1074fds
3
LT1074/LT1076
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: To calculate maximum switch “on” voltage at currents between
low and high conditions, a linear interpolation may be used.
Note 3: A feedback pin voltage (VFB) of 2.5V forces the VC pin to its low
clamp level and the switch duty cycle to zero. This approximates the zero
load condition where duty cycle approaches zero.
Note 4: Total voltage from VIN pin to ground pin must be ≥ 8V after startup for proper regulation.
Note 5: Switch frequency is internally scaled down when the feedback pin
voltage is less than 1.3V to avoid extremely short switch on times. During
testing, VFB is adjusted to give a minimum switch on time of 1µs.
R – 1k
R – 1k
(LT1074), ILIM ≈ LIM
(LT1076).
Note 6: ILIM ≈ LIM
5.5k
2k
Note 7: Switch to input voltage limitation must also be observed.
Note 8: VMAX = 40V for the LT1074/76 and 60V for the LT1074HV/76HV.
Note 9: Does not include switch leakage.
W
BLOCK DIAGRA
INPUT SUPPLY
LT1074
320 µ A
10 µ A
0.3V
+
µ-POWER
SHUTDOWN
–
6V
REGULATOR
AND BIAS
500 Ω
6V TO ALL
CIRCUITRY
CURRENT
LIMIT
COMP
CURRENT
LIMIT
SHUTDOWN
2.35V
+
0.04
+
C2
250 Ω
–
–
I LIM*
SHUTDOWN*
4.5V
10k
FREQ SHIFT
100kHz
OSCILLATOR
S
SYNC
R
R/S
Q
LATCH
R
G1
3V(p-p)
VIN
+
+
2.21V
–
FB
Z
ANALOG
X MULTIPLIER
XY
Z
Y
A1
ERROR
AMP
VC
400 Ω
15 Ω
C1
–
PULSE WIDTH
COMPARATOR
SWITCH
OUTPUT
(VSW )
24V (EQUIVALENT)
LT1076
0.1 Ω
*AVAILABLE ON PACKAGES WITH PIN
COUNTS GREATER THAN 5.
100 Ω
SWITCH
OUTPUT (VSW )
LT1074 • BD01
sn1074 1074fds
4
LT1074/LT1076
U
W
BLOCK DIAGRA
DESCRIPTIO
A switch cycle in the LT1074 is initiated by the oscillator
setting the R/S latch. The pulse that sets the latch also
locks out the switch via gate G1. The effective width of this
pulse is approximately 700ns, which sets the maximum
switch duty cycle to approximately 93% at 100kHz switching frequency. The switch is turned off by comparator C1,
which resets the latch. C1 has a sawtooth waveform as one
input and the output of an analog multiplier as the other
input. The multiplier output is the product of an internal
reference voltage, and the output of the error amplifier, A1,
divided by the regulator input voltage. In standard buck
regulators, this means that the output voltage of A1
required to keep a constant regulated output is independent of regulator input voltage. This greatly improves line
transient response, and makes loop gain independent of
input voltage. The error amplifier is a transconductance
type with a GM at null of approximately 5000µmho. Slew
current going positive is 140µA, while negative slew
current is about 1.1mA. This asymmetry helps prevent
overshoot on start-up. Overall loop frequency compensation is accomplished with a series RC network from VC to
ground.
voltages by feeding the FB signal into the oscillator and
creating a linear frequency downshift when the FB signal
drops below 1.3V. Current trip level is set by the voltage on
the ILIM pin which is driven by an internal 320µA current
source. When this pin is left open, it self-clamps at about
4.5V and sets current limit at 6.5A for the LT1074 and 2.6A
for the LT1076. In the 7-pin package an external resistor
can be connected from the ILIM pin to ground to set a lower
current limit. A capacitor in parallel with this resistor will
soft-start the current limit. A slight offset in C2 guarantees
that when the ILIM pin is pulled to within 200mV of ground,
C2 output will stay high and force switch duty cycle to zero.
Switch current is continuously monitored by C2, which
resets the R/S latch to turn the switch off if an overcurrent
condition occurs. The time required for detection and
switch turn off is approximately 600ns. So minimum
switch “on” time in current limit is 600ns. Under dead
shorted output conditions, switch duty cycle may have to
be as low as 2% to maintain control of output current. This
would require switch on time of 200ns at 100kHz switching frequency, so frequency is reduced at very low output
The switch used in the LT1074 is a Darlington NPN (single
NPN for LT1076) driven by a saturated PNP. Special
patented circuitry is used to drive the PNP on and off very
quickly even from the saturation state. This particular
switch arrangement has no “isolation tubs” connected to
the switch output, which can therefore swing to 40V below
ground.
The “Shutdown” pin is used to force switch duty cycle to
zero by pulling the ILIM pin low, or to completely shut down
the regulator. Threshold for the former is approximately
2.35V, and for complete shutdown, approximately 0.3V.
Total supply current in shutdown is about 150µA. A 10µA
pull-up current forces the shutdown pin high when left
open. A capacitor can be used to generate delayed startup. A resistor divider will program “undervoltage lockout”
if the divider voltage is set at 2.35V when the input is at the
desired trip point.
sn1074 1074fds
5
LT1074/LT1076
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TYPICAL PERFOR A CE CHARACTERISTICS
VC Pin Characteristics
Feedback Pin Characteristics
2.0
500
150
1.5
400
VFB ADJUSTED FOR
IC = 0 AT VC = 2V
0
SLOPE ≈ 400kΩ
–100
VFB ≥ 2.5V
200
CURRENT (µA)
50
–50
300
1.0
CURRENT (mA)
100
CURRENT (mA)
VC Pin Characteristics
200
0.5
0
–0.5
–400
–2.0
–200
1
2
3
5
4
6
7
8
0
9
1
2
3
5
4
7
6
30
–5
20
–10
–20
40
5
6
7
8
9
10
LT1074•TPC03
Tj = 25°C
CURRENT FLOWS OUT
OF SHUTDOWN PIN
50
0
–15
SHUTDOWN
THRESHOLD
–20
–25
Tj = 25°C
–100
–150
–200
–250
–300
–350
–40
30
4
ILIM Pin Characteristics
–35
–40
20
3
100
–30
DETAILS OF THIS
AREA SHOWN IN
OTHER GRAPH
10
2
–50
–10
0
1
VOLTAGE (V)
CURRENT (µA)
0
THIS POINT MOVES
WITH VIN
0
Shutdown Pin Characteristics
0
CURRENT (µA)
CURRENT (µA)
Shutdown Pin Characteristics
40
10
–500
9
LT1074•TPC02
LT1074•TPC01
VIN = 50V
8
VOLTAGE (V)
VOLTAGE (V)
–30
0
–100
–300
–1.5
–150
0
100
–200
–1.0
VFB ≤ 2V
START OF
FREQUENCY SHIFTING
50
60
70
0
80
0.5
1.0
VOLTAGE (V)
1.5
2.0
2.5
3.0
3.5
4.0
VOLTAGE (V)
–400
–2 –1
0
1
2
3
4
5
6
7
8
VOLTAGE (V)
LT1074•TPC04
LT1074•PC05
LT1074•TPC06
Supply Current
20
18
INPUT CURRENT (mA)
16
14
DEVICE NOT SWITCHING
12
VC = 1V
10
8
6
4
2
0
0
10
20
30
40
50
60
INPUT VOLTAGE (V)
LT1074•TPC11
sn1074 1074fds
6
LT1074/LT1076
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage vs
Temperature
Supply Current (Shutdown)
3.0
Tj = 25°C
2.24
250
2.5
150
100
“ON” VOLTAGE (V)
2.23
200
VOLTAGE (V)
2.22
2.21
2.20
LT1074
1.5
LT1076
1.0
2.18
2.17
–50 –25
0
0
10
20
40
30
50
60
INPUT VOLTAGE (V)
0
25
50
75
TRI WAVE
–20
SQUARE
WAVE
–40
–50
–60
–70
200
120
7k
150
115
100
110
θ
6k
5k
50
4k
0
GM
2k
–100
90
1k
–150
85
0
PEAK-TO-PEAK RIPPLE AT FB PIN (mV)
10k
100k
–200
10M
1M
80
–50 –25
0
25
50
75
100 125 150
JUNCTION TEMPERATURE (°C)
LT1074•TPC18
LT1074•TPC17
Feedback Pin Frequency Shift
Current Limit vs Temperature*
160
8
140
7
OUTPUT CURRENT LIMIT (A)
SWITCHING FREQUENCY (kHz)
95
FREQUENCY (Hz)
LT1074•TPC16
6
100
–50
1k
5
105
3k
60 80 100 120 140 160 180 200
120
100
80
150°C
–55°C
40
4
Switching Frequency vs
Temperature
8k
–80
60
3
LT1074•TPC28
PHASE (°)
TRANSCONDUCTANCE (µmho)
0
20 40
2
SWITCH CURRENT (A)
Error Amplifier Phase and GM
10
0
1
LT1074•TPC14
20
–30
0
JUNCTION TEMPERATURE (°C)
Reference Shift with Ripple
Voltage
–10
0.5
100 125 150
LT1074•TPC13
CHANGE IN REFERENCE VOLTAGE (mV)
2.0
2.19
50
FREQUENCY (kHz)
INPUT CURRENT (µA)
Switch “On” Voltage
2.25
300
25°C
20
I LIM PIN OPEN
6
5
R LIM = 10kΩ
4
3
2
R LIM= 5kΩ
1
0
0
0.5
1.0
1.5
2.0
2.5
3.0
FEEDBACK PIN VOLTAGE (V)
*MULTIPLY CURRENTS BY 0.4 FOR LT1076
0
–50 –25 0
25 50 75 100 125 150
JUNCTION TEMPERATURE (°C)
LT1074•TPC19
LT1074•TPC22
sn1074 1074fds
7
LT1074/LT1076
U
U
PI DESCRIPTIO S
VIN PIN
The VIN pin is both the supply voltage for internal control
circuitry and one end of the high current switch. It is
important, especially at low input voltages, that this pin be
bypassed with a low ESR, and low inductance capacitor to
prevent transient steps or spikes from causing erratic
operation. At full switch current of 5A, the switching
transients at the regulator input can get very large as
shown in Figure 1. Place the input capacitor very close to
the regulator and connect it with wide traces to avoid extra
inductance. Use radial lead capacitors.
( )(
dl
dt
LP
∆VOUT =
(∆VGND )(VOUT )
2.21
To ensure good load regulation, the ground pin must be
connected directly to the proper output node, so that no
high currents flow in this path. The output divider resistor
should also be connected to this low current connection
line as shown in Figure 2.
LT1074
GND
)
FB
R2
STEP =
( ISW ) ( ESR )
RAMP =
( ISW ) ( TON )
HIGH CURRENT
RETURN PATH
C
NEGATIVE OUTPUT NODE
WHERE LOAD REGULATION
WILL BE MEASURED
LT1074•PD02
LT1074•PD01
Figure 1. Input Capacitor Ripple
LP = Total inductance in input bypass connections
and capacitor.
“Spike” height (dI/dt • LP) is approximately 2V per
inch of lead length for LT1074 and 0.8V per inch for
LT1076.
“Step” for ESR = 0.05Ω and ISW = 5A is 0.25V.
“Ramp” for C = 200µF, TON = 5µs, and ISW = 5A,
is 0.12V.
Input current on the VIN Pin in shutdown mode is the sum
of actual supply current (≈140µA, with a maximum of
300µA), and switch leakage current. Consult factory for
special testing if shutdown mode input current is critical.
GROUND PIN
It might seem unusual to describe a ground pin, but in the
case of regulators, the ground pin must be connected
properly to ensure good load regulation. The internal
reference voltage is referenced to the ground pin; so any
error in ground pin voltage will be multiplied at the output;
Figure 2. Proper Ground Pin Connection
FEEDBACK PIN
The feedback pin is the inverting input of an error amplifier
which controls the regulator output by adjusting duty
cycle. The noninverting input is internally connected to a
trimmed 2.21V reference. Input bias current is typically
0.5µA when the error amplifier is balanced (IOUT = 0). The
error amplifier has asymmetrical GM for large input signals to reduce startup overshoot. This makes the amplifier
more sensitive to large ripple voltages at the feedback pin.
100mVp-p ripple at the feedback pin will create a 14mV
offset in the amplifier, equivalent to a 0.7% output voltage
shift. To avoid output errors, output ripple (P-P) should be
less than 4% of DC output voltage at the point where the
output divider is connected.
See the “Error Amplifier” section for more details.
Frequency Shifting at the Feedback Pin
The error amplifier feedback pin (FB) is used to downshift
the oscillator frequency when the regulator output voltage
is low. This is done to guarantee that output short-circuit
sn1074 1074fds
8
LT1074/LT1076
U
U
PI DESCRIPTIO S
current is well controlled even when switch duty cycle
must be extremely low. Theoretical switch “on” time for a
buck converter in continuous mode is:
tON =
VOUT + VD
VIN • f
VD = Catch diode forward voltage ( ≈ 0.5V)
f = Switching frequency
At f = 100kHz, tON must drop to 0.2µs when VIN = 25V
and the output is shorted (VOUT = 0V). In current limit,
the LT1074 can reduce tON to a minimum value of
≈0.6µs, much too long to control current correctly for
VOUT = 0. To correct this problem, switching frequency
is lowered from 100kHz to 20kHz as the FB pin drops
from 1.3V to 0.5V. This is accomplished by the circuitry
TO
OSCILLATOR
VOUT
+2V
VC
+
ERROR
AMPLIFIER
–
2.21V
Q1
R1
R3
3k
EXTERNAL
DIVIDER
FB
R2
2.21k
SHUTDOWN PIN
The shutdown pin is used for undervoltage lockout, micropower shutdown, soft-start, delayed start, or as a general
purpose on/off control of the regulator output. It controls
switching action by pulling the ILIM pin low, which forces
the switch to a continuous “off” state. Full micropower
shutdown is initiated when the shutdown pin drops below
0.3V.
The V/I characteristics of the shutdown pin are shown in
Figure 4. For voltages between 2.5V and ≈VIN, a current of
10µA flows out of the shutdown pin. This current increases to ≈25µA as the shutdown pin moves through the
2.35V threshold. The current increases further to ≈30µA at
the 0.3V threshold, then drops to ≈15µA as the shutdown
voltage fall below 0.3V. The 10µA current source is included to pull the shutdown pin to its high or default state
when left open. It also provides a convenient pull-up for
delayed start applications with a capacitor on the shutdown pin.
When activated, the typical collector current of Q1 in
Figure 5, is ≈2mA. A soft-start capacitor on the ILIM pin will
delay regulator shutdown in response to C1, by
≈(5V)(CLIM)/2mA. Soft-start after full micropower shutdown is ensured by coupling C2 to Q1.
0
LT1074•PD03
Tj = 25°C
CURRENT FLOWS OUT
OF SHUTDOWN PIN
–5
Figure 3. Frequency Shifting
shown in Figure 3.
Q1 is off when the output is regulating (VFB = 2.21V). As
the output is pulled down by an overload, VFB will eventually reach 1.3V, turning on Q1. As the output continues to
drop, Q1 current increases proportionately and lowers the
frequency of the oscillator. Frequency shifting starts when
the output is ≈ 60% of normal value, and is down to its
minimum value of ≅ 20kHz when the output is ≅ 20% of
normal value. The rate at which frequency is shifted is
determined by both the internal 3k resistor R3 and the
external divider resistors. For this reason, R2 should not
be increased to more than 4kΩ, if the LT1074 will be
subjected to the simultaneous conditions of high input
voltage and output short-circuit.
CURRENT (µA)
–10
–15
SHUTDOWN
THRESHOLD
–20
–25
–30
–35
–40
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
VOLTAGE (V)
LT1074•PC05
Figure 4. Shutdown Pin Characteristics
sn1074 1074fds
9
LT1074/LT1076
U
U
PI DESCRIPTIO S
Hysteresis in undervoltage lockout may be accomplished
by connecting a resistor (R3) from the ILIM pin to the
shutdown pin as shown in Figure 7. D1 prevents the
shutdown divider from altering current limit.
V IN
300 µ A
10 µ A
SHUTDOWN
PIN
–
ILIM
PIN
C1
2.3V
VIN
R1
+
SHUT
Q1
6V
–
EXTERNAL
CLIM
R3
I LIM
R2
C2
0.3V
LT1074
D1*
OPTIONAL CURRENT
LIMIT RESISTOR
+
LT1074•PD09
*1N4148
TO TOTAL
REGULATOR
SHUTDOWN
Figure 7. Adding Hysteresis
LT1074•PD07
Figure 5. Shutdown Circuitry
Undervoltage Lockout
Undervoltage lockout point is set by R1 and R2 in Figure 6.
To avoid errors due to the 10µA shutdown pin current, R2
is usually set at 5k, and R1 is found from:
R1 = R2
(VTP − VSH)
VSH
⎛ R1⎞
Trip Po int = VTP = 2.35V ⎜ 1 + ⎟
⎝ R2⎠
If R3 is added, the lower trip point (VIN descending) will be
the same. The upper trip point (VUTP) will be:
⎛ R1 R1⎞
⎛ R1⎞
VUTP = VSH ⎜ 1 + ⎟ − 0.8V⎜ ⎟
⎝ R2 R3 ⎠
⎝ R3 ⎠
If R1 and R2 are chosen, R3 is given by:
VTP = Desired undervoltage lockout voltage
VSH = Threshold for lockout on the
shutdown pin = 2.45V
R3 =
If quiescent supply current is critical, R2 may be increased
up to 15kΩ, but the denominator in the formula for R2
should replace VSH with VSH – (10µA)(R2).
R1
VIN
(VSH − 0.8V)(R1)
⎛ R1⎞
VUTP − VSH ⎜ 1 + ⎟
⎝ R2⎠
Example: An undervoltage lockout is required such that
the output will not start until VIN = 20V, but will continue
to operate until VIN drops to 15V. Let R2 = 2.32k.
SHUT
(
LT1074
R2
5k
R1 = 2.34k
GND
LT1074•PD08
Figure 6. Undervoltage Lockout
R3 =
(
)(
15V − 2.35V
2.35V
2.35 − 0.8 12.5
)( )
⎛ 12.5 ⎞
20 − 2.35⎜ 1 +
⎟
⎝ 2.32⎠
) = 12.5k
= 3.9k
sn1074 1074fds
10
LT1074/LT1076
U
U
PI DESCRIPTIO S
ILIM PIN
The ILIM pin is used to reduce current limit below the
preset value of 6.5A. The equivalent circuit for this pin is
shown in Figure 8.
TO LIMIT
CIRCUIT
from forcing current back into the ILIM pin. To calculate a
value for RFB, first calculate RLIM, the RFB:
RFB =
VIN
320 µ A
*Change 0.44 to 0.16, and 0.5 to 0.18 for LT1076.
D2
Example: ILIM = 4A, ISC = 1.5A, RLIM = (4)(2k) + 1k = 9k
Q1
D1
R1
8K
(ISC − 0.44 *)(RL ) (RL in kΩ)
0.5 *(RL − 1kΩ) − ISC
4.3V
RFB =
D3
6V
(1.5 − 0.44)(9kΩ) (3.8kΩ)
0.5(9k − 1k) − 1.5
I LIM
VOUT
LT1047•PD12
Figure 8. ILIM Pin Circuit
When ILIM is left open, the voltage at Q1 base clamps at 5V
through D2. Internal current limit is determined by the
current through Q1. If an external resistor is connected
between ILIM and ground, the voltage at Q1 base can be
reduced for lower current limit. The resistor will have a
voltage across it equal to (320µA)(R), limited to ≈5V when
clamped by D2. Resistance required for a given current
limit is:
RLIM = ILIM(2kΩ) + 1kΩ (LT1074)
RLIM = ILIM(5.5kΩ) + 1kΩ (LT1076)
As an example, a 3A current limit would require
3A(2k) + 1k = 7kΩ for the LT1074. The accuracy of these
formulas is ±25% for 2A ≤ ILIM ≤ 5A (LT1074) and
7A ≤ ILIM ≤ 1.8A (LT1076), so ILIM should be set at least
25% above the peak switch current required.
Foldback current limiting can be easily implemented by
adding a resistor from the output to the ILIM pin as shown
in Figure 9. This allows full desired current limit (with or
without RLIM) when the output is regulating, but reduces
current limit under short-circuit conditions. A typical value
for RFB is 5kΩ, but this may be adjusted up or down to set
the amount of foldback. D2 prevents the output voltage
LT1074
FB
I LIM
R FB
R LIM
D2
1N4148
LT1074•PD13
Figure 9. Foldback Current Limit
Error Amplifier
The error amplifier in Figure 10 is a single stage design
with added inverters to allow the output to swing above
and below the common mode input voltage. One side of
the amplifier is tied to a trimmed internal reference voltage
of 2.21V. The other input is brought out as the FB (feedback) pin. This amplifier has a GM (voltage “in” to current
“out”) transfer function of ≈5000µmho. Voltage gain is
determined by multiplying GM times the total equivalent
output loading, consisting of the output resistance of Q4
and Q6 in parallel with the series RC external frequency
compensation network. At DC, the external RC is ignored,
and with a parallel output impedance for Q4 and Q6 of
400kΩ, voltage gain is ≈2000. At frequencies above a few
hertz, voltage gain is determined by the external compensation, RC and CC.
sn1074 1074fds
11
LT1074/LT1076
U
U
PI DESCRIPTIO S
5.8V
Q4
90 µ A
90 µ A
Q3
50 µ A
Q2
Q1
X1.8
VC
D1
FB
50 µ A
90 µ A
D2
Q6
2.21V
EXTERNAL
FREQUENCY
COMPENSATION
RC
140 µ A
CC
300 Ω
ALL CURRENTS SHOWN ARE AT NULL CONDITION
LT1074 • PD11
Figure 10. Error Amplifier
Gm
AV =
at mid frequencies
2π • f • C C
A V = G m • RC at high frequencies
Phase shift from the FB pin to the VC pin is 90° at mid
frequencies where the external CC is controlling gain, then
drops back to 0° (actually 180° since FB is an inverting
input) when the reactance of CC is small compared to RC.
The low frequency “pole” where the reactance of CC is
equal to the output impedance of Q4 and Q6 (rO), is:
fPOLE =
1
rO ≈ 400kΩ
2π • rO • C
Although fPOLE varies as much as 3:1 due to rO variations,
mid-frequency gain is dependent only on Gm, which is
specified much tighter on the data sheet. The higher
frequency “zero” is determined solely by RC and CC.
fZERO =
The error amplifier has asymmetrical peak output current.
Q3 and Q4 current mirrors are unity-gain, but the Q6
mirror has a gain of 1.8 at output null and a gain of 8 when
the FB pin is high (Q1 current = 0). This results in a
maximum positive output current of 140µA and a maximum negative (sink) output current of ≅1.1mA. The asymmetry is deliberate—it results in much less regulator
output overshoot during rapid start-up or following the
release of an output overload. Amplifier offset is kept low
by area scaling Q1 and Q2 at 1.8:1.
Amplifier swing is limited by the internal 5.8V supply for
positive outputs and by D1 and D2 when the output goes
low. Low clamp voltage is approximately one diode drop
(≈0.7V – 2mV/°C).
Note that both the FB pin and the VC pin have other internal
connections. Refer to the frequency shifting and synchronizing discussions.
1
2π • RC • C C
sn1074 1074fds
12
LT1074/LT1076
U
TYPICAL APPLICATIO S
Tapped-Inductor Buck Converter
L2
5µH
L1*
VIN
20V† TO 35V
VIN
VSW
3
D2
35V
5W
LT1074HV
GND
+
VC
C3
200µF
50V
R1
2.8k
D1**
+
FB
R3
1k
C2
0.2µF
VOUT
5V, 10A†
1
D3
1N5819
R2
2.21k
C1
4400µF
(2 EA
2200µF,
16V)
+
C4
390µF
16V
0.01µF
* PULSE ENGINEERING #PE±65282
** MOTOROLA MBR2030CTL
†
IF INPUT VOLTAGE IS BELOW 20V,
MAXIMUM OUTPUT CURRENT WILL BE REDUCED. SEE AN44
LT1074 •TA02
Positive-to-Negative Converter with 5V Output
+
+
VIN
4.5V to
40V
C1
220µF
50V
L1
25µH
5A††
VIN
VSW
+
LT1074
GND
VC
R3*
2.74k
R1**
5.1k
R2**
10k
OPTIONAL FILTER
VFB
D1†
MBR745
C3
0.1µF
C2
1000µF
10V
5µH
C4**
0.01µF
R4
1.82k*
– 200µF
+ 10V
–5V,1A***
* = 1% FILM RESISTORS
D1 = MOTOROLA-MBR745
C1 = NICHICON-UPL1C221MRH6
C2 = NICHICON-UPL1A102MRH6
L1 = COILTRONICS-CTX25-5-52
†
††
LOWER REVERSE VOLTAGE RATING MAY BE USED FOR LOWER INPUT VOLTAGES.
LOWER CURRENT RATING IS ALLOWED FOR LOWER OUTPUT CURRENT. SEE AN44.
LOWER CURRENT RATING MAY BE USED FOR LOWER OUTPUT CURRENT. SEE AN44.
** R1, R2, AND C4 ARE USED FOR LOOP FREQUENCY COMPENSATION WITH LOW INPUT VOLTAGE,
BUT R1 AND R2 MUST BE INCLUDED IN THE CALCULATION FOR OUTPUT VOLTAGE DIVIDER VALUES.
FOR HIGHER OUTPUT VOLTAGES, INCREASE R1, R2, AND R3 PROPORTIONATELY.
FOR INPUT VOLTAGE > 10V, R1, R2, AND C4 CAN BE ELIMINATED, AND COMPENSATION IS
DONE TOTALLY ON THE V C PIN.
R3 = VOUT –2.37 (KΩ)
R1 = (R3) (1.86)
R2 = (R3) (3.65)
** MAXIMUM OUTPUT CURRENT OF 1A IS DETERMINED BY MINIMUM INPUT
VOLTAGE OF 4.5V. HIGHER MINIMUM INPUT VOLTAGE WILL ALLOW MUCH HIGHER
OUTPUT CURRENTS. SEE AN44.
LT1074 • TA03
sn1074 1074fds
13
LT1074/LT1076
U
PACKAGE DESCRIPTIO
K Package
4-Lead TO-3 Metal Can
(Reference LTC DWG # 05-08-1311)
0.760 – 0.775
(19.30 – 19.69)
0.320 – 0.350
(8.13 – 8.89)
0.060 – 0.135
(1.524 – 3.429)
0.420 – 0.480
(10.67 – 12.19)
0.038 – 0.043
(0.965 – 1.09)
1.177 – 1.197
(29.90 – 30.40)
0.655 – 0.675
(16.64 – 19.05)
0.470 TP
P.C.D.
0.151 – 0.161
(3.84 – 4.09)
DIA 2 PLC
0.167 – 0.177
(4.24 – 4.49)
R
0.490 – 0.510
(12.45 – 12.95)
R
72°
18°
K4(TO-3) 1098
OBSOLETE PACKAGE
Q Package
5-Lead Plastic DD Pak
(Reference LTC DWG # 05-08-1461)
0.256
(6.502)
0.060
(1.524)
0.060
(1.524)
TYP
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
15° TYP
0.060
(1.524)
0.183
(4.648)
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAT SINK
(
+0.008
0.004 –0.004
+0.203
0.102 –0.102
)
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
0.045 – 0.055
(1.143 – 1.397)
(
+0.012
0.143 –0.020
+0.305
3.632 –0.508
)
0.067
(1.70)
0.028 – 0.038 BSC
(0.711 – 0.965)
0.013 – 0.023
(0.330 – 0.584)
0.050 ± 0.012
(1.270 ± 0.305)
Q(DD5) 1098
sn1074 1074fds
14
LT1074/LT1076
U
PACKAGE DESCRIPTIO
R Package
7-Lead Plastic DD Pak
(Reference LTC DWG # 05-08-1462)
0.256
(6.502)
0.060
(1.524)
TYP
0.060
(1.524)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.045 – 0.055
(1.143 – 1.397)
15° TYP
0.060
(1.524)
0.183
(4.648)
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
(
+0.203
0.102 –0.102
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAT SINK
)
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
+0.008
0.004 –0.004
(
+0.012
0.143 –0.020
+0.305
3.632 –0.508
)
0.050
(1.27)
0.026 – 0.036 BSC
(0.660 – 0.914)
0.050 ± 0.012
(1.270 ± 0.305)
0.013 – 0.023
(0.330 – 0.584)
R (DD7) 1098
T Package
5-Lead Plastic TO-220 (Standard)
(Reference LTC DWG # 05-08-1421)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.460 – 0.500
(11.684 – 12.700)
0.570 – 0.620
(14.478 – 15.748)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.78 – 18.491)
SEATING PLANE
0.152 – 0.202
0.260 – 0.320 (3.861 – 5.131)
(6.60 – 8.13)
0.095 – 0.115
(2.413 – 2.921)
0.155 – 0.195*
(3.937 – 4.953)
0.013 – 0.023
(0.330 – 0.584)
BSC
0.067
(1.70)
0.028 – 0.038
(0.711 – 0.965)
0.135 – 0.165
(3.429 – 4.191)
* MEASURED AT THE SEATING PLANE
T5 (TO-220) 0399
sn1074 1074fds
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1074/LT1076
U
TYPICAL APPLICATIO
Negative Boost Converter
R1
12.7k
100pF
VIN
VFB
R2
2.21k
LT1074
C3
200µF
15V
VC
GND
+
VSW
+
L1
25µH
C2
1nF
R3
750Ω
0.01µF
D1*
C1
1000µF
25V
VOUT
–15V**
VIN
–5V TO –15V
* MBR735
** IOUT (MAX) = 1A TO 3A DEPENDING
ON INPUT VOLTAGE. SEE AN44
+
100µF
5µH
OPTIONAL OUTPUT FILTER
LT1074 • TA04
U
PACKAGE DESCRIPTIO
T7 Package
7-Lead Plastic TO-220 (Standard)
(Reference LTC DWG # 05-08-1422)
0.165 – 0.180
(4.191 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.390 – 0.415
(9.906 – 10.541)
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.570 – 0.620
(14.478 – 15.748)
0.460 – 0.500
(11.684 – 12.700)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.780 – 18.491)
SEATING PLANE
0.152 – 0.202
0.260 – 0.320 (3.860 – 5.130)
(6.604 – 8.128)
BSC
0.026 – 0.036
(0.660 – 0.914)
0.050
(1.27)
0.135 – 0.165
(3.429 – 4.191)
0.095 – 0.115
(2.413 – 2.921)
0.155 – 0.195*
(3.937 – 4.953)
0.013 – 0.023
(0.330 – 0.584)
*MEASURED AT THE SEATING PLANE
T7 (TO-220) 0399
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
VIN Up to 25V, IOUT Up to 1.25A, SO-8
LT1374/LT1374HV
4.5A, 500kHz Step-Down Switching Regulators
VIN Up to 25V (32V for HV), IOUT Up to 4.25A, SO-8/DD
LT1370
6A, 500kHz High Efficiency Switching Regulator
6A/42V Internal Switch, 7-Lead DD/TO-220
LT1676
Wide Input Range, High Efficiency Step-Down Regulator
VIN from 7.4V to 60V, IOUT Up to 0.5A, SO-8
LT1339
High Power Synchronous DC/DC Controller
VIN Up to 60V, IOUT Up to 50A, Current Mode
LT1765
3A, 1.25MHz, Step-Down Regulator
VIN = 3V to 25V, VµF =1.2V, TSSOP-16E, SO8 Package
sn1074 1074fds
16
Linear Technology Corporation
LT/CPI 0202 1.5K REV D • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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