LT1248
Power Factor Controller
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FEATURES
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DESCRIPTIO
The LT®1248 provides active power factor correction for
universal off-line power systems. By using fixed high
frequency PWM current averaging, without the need for
slope compensation, the LT1248 achieves far lower line
current distortion with a smaller magnetic element than
systems that use either peak-current detection or zero
current switching approaches in both continuous and
discontinuous modes of operation.
High Power Factor Over Wide Load Range
with Line Current Averaging
International Operation Without Switches
Instantaneous Overvoltage Protection
Minimal Line Current Dead Zone
Typical 250µA Start-Up Supply Current
Rejects Line Switching Noise
Synchronization Capability
Low Quiescent Current: 9mA
Fast 1.5A Peak Current Gate Driver
The LT1248 uses a multiplier containing a square gain
function from the voltage amplifier to reduce the AC gain
at light output load and thus maintains low line current
distortion and high system stability. The LT1248 also
provides filtering capability to reject line switching noise
which can cause instability when fed into the multiplier.
Line current dead zone is minimized with low bias voltage
at the current input to the multiplier.
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APPLICATIO S
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Universal Power Factor Corrected Power Supplies
Preregulators Up To 1500W
The LT1248 provides many protection features including
peak current limiting and overvoltage protection, and can
be operated at frequencies as high as 300kHz.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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BLOCK DIAGRA
VCC
+
16V TO 10V
–
2.6V/
2.2V
–
EN/SYNC
10
VAOUT
VREF
7
9
4
3
2
GND
VCC
1
15
–
+
RUN
2.2V
+
7µA
+
M1
–
11
–
IAC
+
IA
EA
32k
7.5V
6
7.9V
I 2I
I = A B
IB M 200µA2
+
–
–
SS
13
5
7.5V
VREF
VSENSE
OVP
8
CAOUT PKLIM
MOUT ISENSE
12µA
5V
+
IM
–
CA
+
R
R
–
+
+
–
0.7V
RUN
SYNC
ONE SHOT
200ns
Q
S
16
GTDR
OSC
1 6V
14
CSET
12
RSET
1248 BD
1
LT1248
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RATI GS
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ABSOLUTE
PACKAGE/ORDER I FOR ATIO
(Note 1)
Supply Voltage ....................................................... 27V
GTDR Current Continuous ..................................... 0.5A
GTDR Output Energy(Per Cycle) .............................. 5µJ
IAC, RSET, PKLIM Input Current ............................. 20mA
VSENSE, EN/SYNC, OVP Input Voltage ................... VMAX
ISENSE, MOUT Input Current .................................. ± 5mA
Operating Junction Temperature Range
LT1248C ................................................ 0°C to 100°C
LT1248I ........................................... – 40°C to 125°C
Thermal Resistance (Junction-to-Ambient)
N Package .................................................. 100°C/W
S Package ................................................... 120°C/W
Storage Temperature Range ..................–65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART
NUMBER
TOP VIEW
GND
1
16 GTDR
PKLIM
2
15 VCC
CAOUT
3
14 CSET
ISENSE
4
13 SS
MOUT
5
12 RSET
IAC
6
11 VSENSE
VAOUT
7
10 EN/SYNC
OVP
8
9
LT1248CN
LT1248IN
LT1248CS
LT1248IS
VREF
N PACKAGE
16-LEAD PDIP
S PACKAGE
16-LEAD NARROW PLASTIC SO
TJMAX = 125°C, θJA = 100°C/W (N)
TJMAX = 125°C, θJA = 120°C/W (S)
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, RSET = 15k to GND,
CSET = 1nF to GND, IAC = 100µA, ISENSE = 0V, CAOUT = 3.5V, VAOUT = 5V, OVP = 7.5V, no load on any outputs, unless otherwise noted.
PARAMETER
Overall
Supply Current (VCC in Undervoltage Lockout)
Supply Current (Inactive)
Supply Current, On
VCC Turn-On Threshold (Undervoltage Lockout)
VCC Turn-Off Threshold
EN/SYNC Threshold, Rising
EN/SYNC Threshold Hysteresis
EN/SYNC Input Current
Voltage Amplifier
Voltage Amp Offset Voltage
Input Bias Current
Voltage Gain
Voltage Amp Unity-Gain Bandwidth
Voltage Amp Output High (Internally Clamped)
Voltage Amp Output Low
Voltage Amp Short-Circuit Current
SS Current
Current Amplifier
Current Amp Offset Voltage
ISENSE Bias Current
Current Amp Voltage Gain
Current Amp Unity-Gain Bandwidth
Current Amp Output High
Current Amp Output Low
2
CONDITIONS
VCC = Lockout Voltage – 0.2V
EN/SYNC = 0V, VCC ≤ VMAX
11.5V ≤ VCC ≤ VMAX, CAOUT = 1V
MIN
TYP
MAX
UNITS
15.5
9.5
2.2
0.25
0.5
8.5
16.5
10.5
2.6
0.40
–1
– 25
0.45
1.5
12.0
17.5
11.5
2.85
mA
mA
mA
V
V
V
V
µA
µA
●
●
●
●
●
●
EN/SYNC = 0V
3V ≤ EN/SYNC ≤ 7V
●
–5
– 50
VAOUT = 3.5V
VSENSE = 0V to 7V
●
–8
●
70
●
11.3
●
VAOUT = 0V
SS = 2.5V
●
●
5
5
●
●
80
●
●
7.2
– 25
100
3
13.3
1.1
14
12
±1
– 25
110
3
8.5
1.1
5
50
8
– 250
2
30
30
±4
– 250
2
mV
nA
dB
MHz
V
V
mA
µA
mV
nA
dB
MHz
V
V
LT1248
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, RSET = 15k to GND,
CSET = 1nF to GND, IAC = 100µA, ISENSE = 0V, CAOUT = 3.5V, VAOUT = 5V, OVP = 7.5V, no load on any outputs, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
●
5
14
30
mA
●
– 0.3
1
V
7.60
V
Current Amplifier
Current Amp Short-Circuit Current
CAOUT = 0V
Input Range, ISENSE, MOUT (Linear Operation)
Reference
Reference Output Voltage
IREF = 0mA, TA = 25°C
7.39
7.50
– 20
5
VREF Load Regulation
– 5mA < IREF < 0mA
VREF Line Regulation
11.5V < VCC < VMAX
●
5
mV
VREF Short-Circuit Current
VREF = 0V
●
12
VREF Worst Case
Load, Line, Temperature
●
7.32
●
– 15
15
mV
– 50
– 100
µA
20
mV
28
50
mA
7.5
7.68
V
Current Limit
PKLIM Offset Voltage
PKLIM Input Current
PKLIM = – 0.1V
PKLIM to GTDR Propagation Delay
PKLIM Falling from 50mV to – 50mV
400
ns
IAC = 100µA, RSET = 15k
35
µA
●
Multiplier
Multiplier Output Current
Multiplier Output Current Offset
RAC = 1M from IAC to GND
●
Multiplier Maximum Output Current
IAC = 450µA, RSET = 15k, VAOUT = 7V, MOUT = 0V
●
– 286
Multiplier Gain Constant (Note 2)
IAC Input Resistance
– 0.05
– 0.5
µA
– 260
– 235
µA
V –2
0.035
IAC from 50µA to 1mA
15
32
50
kΩ
85
58
100
68
115
78
kHz
kHz
4.35
4.7
5.0
V
1.25
1.4
1.55
V
4.5
5.6
6.5
V
1.6
f NOM
Oscillator
Oscillator Frequency
RSET = 15k, CSET = 1000pF
RSET = 15k, CSET = 1500pF
●
●
CSET Ramp Peak-to-Peak Amplitude
CSET Ramp Valley Voltage
Synchronization Pulse Threshold on EN/SYNC Pin
Synchronization Frequency Range
Pulse Low = 3.5V, High = 7V, Width > 200ns
RSET = 15k, CSET = 1000pF
●
1.2
●
1.04
Overvoltage Comparator
Comparator Trip Voltage Ratio (VTRIP / VREF)
Hysteresis
OVP Bias Current
1.05
1.06
0.35
OVP = 7.5V
– 50
●
OVP Propagation Delay
V
– 250
100
nA
ns
Gate Driver
Max GTDR Output Voltage
0mA Load, 18V < VCC
●
12
15
17.5
V
GTDR Output High
– 200mA Load, 11.5V ≤ VCC ≤ 15V
●
VCC – 3.0
GTDR Output Low (Device Unpowered)
VCC = 0V, 50mA Load (Sinking)
●
0.9
1.5
V
GTDR Output Low (Device Active)
200mA Load (Sinking)
10mA Load
●
●
0.5
0.2
1
0.4
V
V
V
Peak GTDR Current
10nF from GTDR to GND
2
A
GTDR Rise and Fall Time
1nF from GTDR to GND
25
ns
96
%
GTDR Max Duty Cycle
90
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired
IM
Note 2: Multiplier Gain Constant: K =
IAC (VAOUT – 2)2
3
LT1248
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TYPICAL PERFOR A CE CHARACTERISTICS
Current Amplifier Open-Loop
Gain and Phase
Voltage Amplifier Open-Loop
Gain and Phase
100
0
80
100
–20
0
80
–20
GAIN
40
–60
20
10
100
1k
10k 100k
FREQUENCY (Hz)
1M
60
–40
40
–60
–80
20
–100
0
–120
10M
–20
PHASE
0
–20
GAIN (dB)
–40
–80
PHASE
–100
10
1148 G02
Reference Voltage vs
Temperature
Multiplier Current
7.536
300
7.524
VAOUT = 5.5V
VAOUT = 7V
7.512
VAOUT = 6.5V
7.500
VAOUT = 6V
IM (µA)
REFERENCE VOLTAGE (V)
–120
10M
1M
1k
10k 100k
FREQUENCY (Hz)
100
1148 G01
7.488
7.476
VAOUT = 5V
150
VAOUT = 4.5V
7.464
VAOUT = 4V
7.452
VAOUT = 3.5V
7.440
7.428
–75 –50 –25 0 25 50 75 100 125 150
JUNCTION TEMPERATURE (°C)
0
0
VAOUT = 3V
VAOUT = 2.5V
500
250
IAC (µA)
1248 G04
1248 G03
Supply Current vs Supply Voltage
8
7
TJ = 125°C
GTDR VOLTAGE (V)
SUPPLY CURRENT (mA)
VCC = 18V
TJ = 25°C
6
5
4
0.9
17.0
0.8
16.5
16.0
15.5
14.5
2
14.0
1
13.5
13.0
21
SUPPLY VOLTAGE (V)
32
TJ = 125°C
15.0
3
10
1.0
17.5
GTDR VOLTAGE (V)
TJ = –55°C
1248 G05
4
1.1
18.0
10
0
GTDR Sink Current
GTDR Source Current
18.5
11
9
PHASE (DEG)
60
PHASE (DEG)
GAIN (dB)
GAIN
TJ = 25°C
0.7
0.6
0.5
TA = –55°C
0.4
0.3
TJ = –55°C
0.2
TA = 25°C
0.1
0
–120
–180
–240
– 60
SOURCE CURRENT (mA)
–300
1248 G06
0
TA = 125°C
0
60
120
180
240
SINK CURRENT (mA)
300
1248 G07
LT1248
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TYPICAL PERFOR A CE CHARACTERISTICS
Start-Up Supply Current vs
Supply Voltage
GTDR Rise and Fall Time
SUPPLY CURRENT (µA)
TIME (ns)
300
FALL TIME
200
RISE TIME
100
NOTE: GTDR SLEWS
BETWEEN 1V AND 16V
0
0
10
20
30
40
LOAD CAPACITANCE (nF)
550
500
500
450
450
400
400
350
300
–55°C
25°C
250
200
125°C
150
250
200
150
50
50
4
2
0
200
6 8 10 12 14 16 18 20
SUPPLY VOLTAGE (V)
1248 G08
600
1000
1800
1400
CSET CAPACITANCE (pF)
Shutdown Mode Supply Current
and Reference Voltage
1.1
1.1
0.99
1.0
0.98
0.9
0.9
0.8
0.8
0.7
0.7
0.97
0.96
0.95
0.94
RSET = 10k
RSET = 15k
RSET = 20k
RSET = 30k
0.92
0.91
0.90
200
600
1000
1800
1400
CSET CAPACITANCE (pF)
EN/SYNC ≤ 1.8V
1.0
0.6
0.6
SUPPLY CURRENT
–55°C ≤ TJ ≤ 25°C
TJ = 125°C
0.5
0.4
0.3
0.2
0.2
REFERENCE VOLTAGE
TJ ≤ 125°C
0.1
0
2200
0
16
SUPPLY VOLTAGE (V)
SHUTDOWN
THRESHOLD
–32
SS CURRENT (µA)
EN/SYNC CURRENT (µA)
–36
SYNCHRONIZATION
THRESHOLD
–28
–24
TJ = – 55°C
–20
TJ = 25°C
–16
TJ = 125°C
–12
1.5
–20
1.0
–18
0.5
–16
TJ = –55°C
–14
TJ = 25°C
–12
TJ = 125°C
–10
–8
–6
0
–1.0
–1.5
–2.0
–2.5
–4
–3.0
–4
–2
–3.5
0
0
1
2
3 4 5
6 7 8
EN/SYNC VOLTAGE (V)
9
10
1248 G13
0
4
SS VOLTAGE (V)
8
1248 G14
TJ = 125°C
TJ = 25°C
TJ = –55°C
–0.5
–8
0
0
32
MOUT Pin Characteristics
–22
MOUT CURRENT (mA)
–40
0.1
1248 G12
SS Pin Characteristics
–44
0.4
0.3
1248 G11
Synchronization and Shutdown
Thresholds at EN/SYNC Pin
0.5
REFERENCE VOLTAGE (V)
SUPPLY CURRENT (mA)
1.00
0.93
2200
1248 G10
1248 G09
GTDR Maximum Duty Cycle vs
RSET and CSET
MAXIMUM DUTY CYCLE
300
100
0
RSET = 10k
RSET = 15k
RSET = 20k
RSET = 30k
350
100
0
50
Frequency vs RSET and CSET
FREQUENCY (kHz)
400
–4.0
–2.4
1.2
–1.2
0
MOUT VOLTAGE (V)
2.4
1248 G15
5
LT1248
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TYPICAL PERFOR A CE CHARACTERISTICS
RSET Voltage vs Current
PKLIM Pin Characteristics
120
–360
TJ = 125°C
TJ = 25°C
TJ = –55°C
100
–240
60
PKLIM CURRENT (µA)
VRSET – VREF (mV)
80
40
20
0
–20
–40
–180
–120
–60
0
60
120
–60
180
–80
240
–100
0
–0.2
–0.8
–0.4
–0.6
RSET CURRENT (mA)
TJ = 125°C
TJ = 25°C
TJ = –55°C
–300
–1.0
1248 G16
300
–0.8
0.4
–0.4
0
PKLIM VOLTAGE (V)
0.8
1248 G17
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Pin 1 (GND).
Pin 2 (PKLIM): The threshold of the peak current limit
comparator is GND. To set current limit, a resistor divider
can be connected from VREF to current sense resistor.
Pin 3 (CAOUT): This is the output of the current amplifier
that senses and forces the line current to follow the
reference signal that comes from the multiplier by commanding the pulse width modulator. When CAOUT is low,
the modulator has zero duty cycle.
Pin 4 (ISENSE): This is the inverting input of the current
amplifier. This pin is clamped at – 0.6V by an ESD protection diode.
Pin 5 (MOUT): This is the multiplier high impedance
current output and the noninverting input of the current
amplifier. This pin is clamped at – 0.6V and 2V.
Pin 6 (IAC): This is the AC line voltage sensing input to the
multiplier. It is a current input that is biased at 2V to
minimize the crossover dead zone caused by low line
voltage. At the pin, a 32k resistor is in series with the
current input, so that a lowpass RC can be used to filter out
the switching noise from the high impedance lines.
6
Pin 7 (VAOUT): This is the output of the voltage error
amplifier. The output is clamped at 13.5V. When the
output goes below 2.5V, the multiplier output current is
zero.
Pin 8 (OVP): This is the input to the overvoltage comparator. The threshold is 1.05 times the reference voltage.
When the comparator trips, the multiplier is quickly inhibited and outputs no current. Figure 4 in the Applications
Information section shows how to set overvoltage threshold with only one additional resistor.
Pin 9 (VREF): This is the 7.5V reference. When either VCC
or EN/SYNC goes low, VREF will stay at 0V. VREF biases
most of the internal circuity and can source up to 5mA
externally.
Pin 10 (EN/SYNC): This pin has two functions. When it
goes below 2.6V, the chip goes into shutdown mode and
draws little current. Pulses at this pin that go below the 5V
threshold will synchronize the chip. The synchronizing
pulses should have an on-time of at least 200ns for the
LT1248 resetting circuit to work.
Pin 11 (VSENSE): This is the inverting input to the voltage
amplifier.
LT1248
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Pin 12 (RSET): A resistor from RSET to GND sets the
oscillator charging current and the maximum multiplier
output current which is used to limit the maximum line
current.
IM(MAX) = 3.75V/RSET
Pin 13 (SS): Soft-Start. When either VCC or EN/SYNC goes
low, the SS pin will stay at 0V. With a capacitor from the
pin to GND, the 12µA charging current slowly brings up the
SS to 8V; below 7.5V SS is the reference input to the
voltage amplifier. At supply dropout or EN/SYNC low, the
soft start capacitor will be quickly discharged.
Pin 15 (VCC): This is the supply for the chip. The LT1248
has a very fast gate driver required to fast charge high
power MOSFET gate capacitance. High current spikes
occur during charging. For good supply bypass, a 0.1µF
ceramic capacitor in parallel with a low ESR electrolytic
capacitor, 56µF or higher is required in close proximity to
IC GND.
Pin 16 (GTDR): The MOSFET gate driver is a 1.5A fast
totem pole output. It is clamped at 15V, but capacitive
loads like MOSFET gates may cause overshoot. A gate
series resistor of at least 5Ω will prevent the overshoot.
Pin 14 (CSET): The capacitor from this pin to GND, and
RSET, determine oscillator frequency. The oscillator ramp
is 5V, and the frequency = 1.5/(RSET • CSET).
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APPLICATI
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Error Amplifier
Multiplier
The error amplifier has a 100dB DC gain and 3MHz unitygain frequency. The output is internally clamped at 13.5V.
The noninverting input is tied to the 7.5V VREF through a
diode and can be pulled down from the SS (soft-start) pin.
The multiplier is a current multiplier with high noise
immunity in a high power switching environment. The
current gain is: IM = (IAC • IEA2)/(200µA)2, with IEA = (VAOUT
– 2V)/25k. With a square function, because of the lower
gain at light power load, system stability is maintained and
line current distortion caused by the line frequency AC
The current amplifier has a 110dB DC gain, 3MHz unitygain frequency, and a 2V/µs slew rate. It is internally
clamped at 8.5V. Note that in the current averaging operation, high gain at twice the line frequency is necessary to
minimize line current distortion. Because CAOUT may need
to swing 5V over one line cycle at high line condition,
14mV AC will be needed at the inputs of the current
amplifier for a gain of 350 at 120Hz. Especially at light load
when the current loop reference signal is small, lower gain
will distort the reference signal and line current. If signal
gain at switching frequency is too high, the system behaves more like a current mode system and can cause
subharmonic oscillation. Therefore, the current amplifier
should be compensated to have a gain of less than 15 at
the switching frequency, but more than 250 at twice the
line frequency.
300
VAOUT = 5.5V
VAOUT = 7V
VAOUT = 6.5V
VAOUT = 6V
IM (µA)
Current Amplifier
VAOUT = 5V
150
VAOUT = 4.5V
VAOUT = 4V
VAOUT = 3.5V
0
0
250
IAC (µA)
VAOUT = 3V
VAOUT = 2.5V
500
1248 G04
Figure 1. Multiplier Current IM vs IAC and VAOUT
7
LT1248
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APPLICATI
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ripple fed back to the error amplifier is minimized. Note
that switching ripple on the high impedance lines could get
into the multiplier from the IAC pin and cause instability.
The LT1248 provides an internal 25k resistor in series with
the low impedance multiplier current input so that only a
capacitor from the IAC pin to GND is needed to filter out the
noise. The maximum multiplier output current, which
limits the system line current, is set by the RSET according
to the formula: IM(MAX) = 3.75V/RSET.
Oscillator Frequency and Maximum Line
Current Settling
Oscillator frequency is set by RSET and CSET. Ramp amplitude is 5V and CSET charging current is set by VREF/RSET.
Typical discharging time for CSET = 1nF is 250ns. RSET
should always be determined first to set the maximum
multiplier output current for system line current limit. For
a 300W preregulator, with RSET = 15k, IM(MAX) = 3.75V/15k
= 250µA. With a 4k resistor RREF from MOUT to the 0.2Ω
line current sense resistor RS, the line current limit is: (IM
• 4k)/RS. As a general rule, RS is chosen according to:
RS = IM(MAX) • RREF • VLINE(MIN)
K(1.414)POUT(MAX)
R1
10k
7.5V
VREF
+
–
RS
0.2Ω
IPKLIM
ILINE
C1 IS TO REJECT NOISE, CURRENT
LIMIT DELAY IS ABOUT 2µs.
–
+
1248 F02
Figure 2
8
PKLIM
C1
1nF
Always use RSET to set the primary line current limit. The
PKLIM comparator is only for secondary protection. The
secondary limit should be higher than the primary limit;
6.5A is good (5A for primary limit) for a 300W regulator.
When line current reaches the primary limit, VOUT drops to
keep the line current constant, and system stability is still
maintained by the current loop which is controlled by the
current amplifier. When line current reaches the secondary limit, the comparator controls the system and loop
hysteresis may occur and can cause audible noise.
Synchronization
The LT1248 can be synchronized to a frequency that is up
to 1.6 times the natural frequency. With a 200ns one-shot
timer on-chip, the LT1248 provides flexibility on the
synchronizing pulse width. Because the EN/SYNC pin also
serves the chip shutdown function, the pulses at the pin
should not go below 3V and must go below 5V with widths
greater than 200ns. The Figure 3 circuit will synchronize
the LT1248.
VREF
where POUT(MAX) is the maximum power output and K is
usually between 1.1 and 1.3 depending on efficiency and
resistor tolerance. With RSET selected, CSET can then be
determined by: CSET = 1.5/(Frequency • RSET). For 100kHz,
CSET = 1.5/(100kHz • 15k) = 1nF. For optional double
protection, the LT1248 provides a current limit comparator. When the comparator trips at 0V, the GTDR pin quickly
goes low to shut off the MOS switch. A resistor divider
from VREF to RS (Figure 2) senses the voltage across the
line current sense resistor and the current limit is set by:
ILINE = [(7.5V/R1) + 50µA](R2/RS), where 50µA is IPKLIM.
R2
1.6k
With ILINE and RS chosen, let R1 = 10k, then R2 =
(ILINE • RS )/0.8mA.
30k
1N4148
200k
VCC
EN/SYNC
1N4685
3.6V
SYNC PULSE
AT LEAST 200ns
VN2222
1248 F03
Figure 3
Overvoltage Protection
Because of the slow loop response necessary for power
factor correction, output overshoot can occur with sudden
load removal or reduction. To protect the power components and output load, the LT1248 provides an overvoltage comparator which senses the output voltage and
quickly shuts off the current switch. In Figure 4, because
there is no DC current going through R3, R1 and R2 set the
regulator output DC level: VOUT = VREF[(R1 + R2)/R2], with
R1 = 1M, R2 = 20k, VOUT is 382V.
LT1248
W
U
U
UO
APPLICATI
S I FOR ATIO
Note that VSENSE is the summing node and it stays at 7.5V.
When overshoot occurs on VOUT, the overcurrent from R1
will go through R2 as well as R3. Amplifier feedback will
keep VSENSE locked at 7.5V. The equivalent AC resistance,
seen by the comparator input pin OVP, is R2 in parallel
with R3, which is 10k. Therefore, with the comparator trip
level of 1.05VREF and R3 of 20k, the comparator trips when
VOUT overshoot exceeds 10%. Overvoltage trip level:
R2 + R3
%VOUT = 5%
R3
MOUT is a high impedance current output. In the current
loop, offset line current is determined by multiplier offset
current and input offset voltage of the current amplifier.
A – 4mV current amplifier VOS translates into 20mA line
current and 5W input power for 250V line if 0.2Ω sense
resistor is used. Under no load or when the load power is
less than this offset input power, VOUT would slowly
charge up to an overvoltage state because the overvoltage
comparator can only reduce multiplier output current to
zero. This does not guarantee zero output current if the
current amplifier has offset. To regulate VOUT under this
condition, the amplifier M1 (see Block Diagram), becomes
active in the current loop when VAOUT goes down to 2.2V.
The M1 can put out up to 7µA to the resistor at the ISENSE
pin to cancel any current amplifier negative VOS and keep
VOUT error to within 2V.
Undervoltage Lockout
The LT1248 turns on when VCC is higher than 16V and
remains on until VCC falls below 10V, whereupon the chip
enters the lockout state. In the lockout state, the LT1248
only draws 250µA, the oscillator is off, and the VREF and
the GTDR pins remain low to keep the power MOSFET off.
Start-Up and Supply Voltage
The LT1248 draws only 250µA before the chip starts at
16V on VCC. To trickle start, a 90k resistor from the power
line to VCC supplies the trickle current and C4 holds the VCC
up while switching starts. Then the auxiliary winding takes
over and supplies the operating current. Note that D3 and
the large value C3, in both Figures 5 and 6, are only
necessary for systems that have sudden large load variation down to minimum load and/or very light load conditions. Under these conditions, the loop may exhibit a start/
restart mode because switching remains off long enough
for C4 to discharge below 10V. The C3 will hold VCC up
until switching resumes. For less severe load variations,
D3 is replaced with a short and C3 is omitted. The turns
ratio between the primary winding and the auxiliary winding determines VCC according to:
LINE
MAIN INDUCTOR
NP
NS
R1
90k, 1W
D1
D3
D2
C1
0.47µF
REGULATOR OUTPUT
VOUT = 382V
R1
1M
R3
20k
C1
2µF
+
+
C2
2µF
330k
C3
390µF
+
C4
56µF
1248 F05
VSENSE
–
+
Figure 5
VAOUT
C2
1000pF
ERROR AMP
VREF = 7.5V
R2
20k
VCC
+
0.047µF
MAIN INDUCTOR
LINE
LT1248
OVP
R1
90k
1W
–
D2
+
1.05VREF
OVERVOLTAGE
COMPARATOR
D3
+
D1
C3
390µF
18V
+
VCC
C4
56µF
1248 F04
1248 F06
Figure 4
Figure 6
9
LT1248
U
W
U
UO
APPLICATI
S I FOR ATIO
VOUT/(VCC – 2V) = NP/NS.
For 382V VOUT and 18V VCC, Np/Ns ≈ 19.
In Figure 6, a new technique for supply voltage eliminates
the need for an extra inductor winding. It uses capacitor
charge transfer to generate a constant current source
which feeds a Zener diode. Current to the Zener is equal
to (VOUT – VZ)(C)(f), where VZ is Zener voltage and f is
switching frequency. For VOUT = 382V, VZ = 18V, C =
1000pF, and f = 100kHz, Zener current will be 36mA. This
is enough to operate the LT1248, including the FET gate
drive. Normally soft-start is not needed because the
LT1248 has overcurrent limit and overvoltage protection.
If soft-start is used with a 0.01µF capacitor on SS pin,
VOUT ramps up slower during start-up. Then C4 has to
hold VCC longer, and the circuit may not start. Increasing
C4 to 100µF ensures start-up, but start-up time will be
extended if the same 90k trickle charge resistor is used.
Output Capacitor
The peak-to-peak 120Hz output ripple is determined by:
VP-P = (2) (ILOAD(DC))(Z)
where ILOAD(DC): DC load current.
Z: capacitor impedance at 120Hz.
For 180µF at 300W load, ILOAD(DC) = 300W/385V = 0.78A,
VP-P = 2 • 0.78A • 7.4Ω = 11.5V. If less ripple is desired,
higher capacitance should be used. The selection of the
output capacitor should also be based on the operating
ripple current through the capacitor. The ripple current
can be divided into three major components. The first is at
120Hz; it’s RMS value is related to the DC load current as
follows:
I1RMS ≈ 0.71 • ILOAD(DC)
The second component contains the PF switching frequency ripple current and its harmonics. Analysis of the
ripple is complicated because it is modulated with a 120Hz
signal. However computer numerical integration and Fourier analysis approximate the RMS value reasonably close
to the bench measurements. The RMS value is about 0.82A
at a typical condition of 120VAC, 200W load. This ripple is
line-voltage dependent, and the worst case is at low line.
I2RMS = 0.82A at 120VAC, 200W
10
The third component is the switching ripple from the load,
if the load is a switching regulator.
I3RMS ≈ ILOAD(DC)
For the United Chemicon KMH 400V capacitor series,
ripple current multiplier for currents at 100kHz is 1.43. The
equivalent 120Hz ripple current can be then found:
IRMS = √(I1RMS)2 + (I2RMS/1.43)2 + (I3RMS/1.43)2
For a typical system that runs at an average load of 200W
and 385V output:
ILOAD(DC) = 0.52A
I1RMS ≈ 0.71 • 0.52A = 0.37A
I2RMS ≈ 0.82A at 120VAC
I3RMS ≈ ILOAD(DC) = 0.52A
IRMS = √(0.37A)2 +(0.82A/1.43)2 +(0.52A/1.43)2 = 0.77A
The 120Hz ripple current rating at 105°C ambient is 0.95A
for the 180µF KMH 400V capacitor. The expected life of the
output capacitor may be calculated from the thermal
stress analysis:
(105°C+∆TK) – (TA+∆TO)
L = LO • 2
10
where:
L: expected life time
LO: hours of load life at rated ripple current and rated
ambient temperature.
∆TK: Capacitor internal temperature rise at rated condition. ∆TK = (I2R)/(KA). Where I is the rated current,
R is capacitor ESR, and KA is a volume constant.
TA: Operating ambient temperature.
∆TO: Capacitor internal temperature rise at operating
condition.
In our example LO = 2000 hours and ∆TK = 10°C at rated
0.95A. ∆TO can then be calculated from:
∆TK = (IRMS/0.95A)2 • ∆TK = (0.77A/0.95A)2 • 10°C = 6.6°C
Assuming the operating ambient temperature is 60°C, the
approximate life time is:
LO ≈ 2000 • 2
(105°C +10°C) – (60°+ 6.6°C)
10
≈ 57,000 hours
For longer life, a capacitor with a higher ripple current
rating or parallel capacitors should be used.
LT1248
UO
TYPICAL APPLICATI
300W, 382V Preregulator
90V
TO
270V
MURH860
750µH*
T
+
VOUT
EMI
FILTER
6A
–
0.47µF
20k
RREF
4k
330k
4k
100pF
VCC = 18V**
1nF
0.1µF
VAOUT
VCC
+
16V TO 10V
–
10
11
1M
2.2V
7.5V
OVP
+
13
GND
15
VCC
–
7µA
I 2I
I = A B
IB M 200µA2
–
+
CA
+
R
R
–
32k
–
5V
IM
+
Q
GTDR
16
10Ω
S
RUN
+
–
12µA
SS
1
+
0.7V–
4.7nF
50k
PKLIM
2
3
CAOUT
+
IA
–
+
7.9V
8
ISENSE
M1
EA
6
4
–
VSENSE
IAC
5
56µF
35V
RUN
+
EN/SYNC
MOUT
+
20k
7.5V
VREF
–
2.6V/2.2V
VREF
9
7
180µF
20k
1%
RS
0.2Ω
0.047µF
+
1M
1%
IRF840
†
OSC
ONE SHOT
200ns
16V
SYNC
1N5819
0.01µF
CSET
* 1. COILTRONICS CTX02-12236-1 (TYPE 52 CORE)
AIR MOVEMENT NEEDED AT POWER LEVEL GREATER THAN 250W.
2. COILTRONICS CTX02-12295 (MAGNETICS Kool Mµ® 77930 CORE)
** SEE START-UP AND SUPPLY VOLTAGE SECTION FOR VCC GENERATOR.
† THIS SCHOTTKY DIODE IS TO CLAMP GTDR WHEN MOS SWITCH
TURNS OFF. PARASITIC INDUCTANCE AND GATE CAPACITANCE MAY
TURN ON CHIP SUBSTRATE DIODE AND CAUSE ERRATIC OPERATIONS
IF GTDR IS NOT CLAMPED.
1000pF
14
12
RSET
15k
1248 TA01
Kool Mµ is a registered trademark of Magnetics, Inc.
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LT1248
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
N Package
16-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.130 ± 0.005
(3.302 ± 0.127)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
+0.889
8.255
–0.381
0.045 – 0.065
(1.143 – 1.651)
0.020
(0.508)
MIN
+0.035
0.325 –0.015
)
0.770*
(19.558)
MAX
0.065
(1.651)
TYP
0.125
(3.175)
MIN
15
14
13
12
11
10
1
2
3
4
5
6
7
9
0.255 ± 0.015*
(6.477 ± 0.381)
0.018 ± 0.003
(0.457 ± 0.076)
0.100
(2.54)
BSC
16
8
N16 1098
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
16
15
14
13
12
11
10
9
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.050
(1.270)
BSC
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
S16 1098
1
2
3
4
5
6
7
8
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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Universal Off-Line Inputs with Outputs to 100W
LT1249
PFC in SO-8
Simplified PFC Design with Minimal Part Count
LT1508
Power Factor and PWM Controller
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LT1509
Power Factor and PWM Controller
Complete Solution for Universal Off-Line Switching Power Supplies
12
Linear Technology Corporation
1248fd LT/GP 0799 2K REV D • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1993
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