LT1302/LT1302-5
Micropower
High Output Current
Step-Up Adjustable and
Fixed 5V DC/DC Converters
U
DESCRIPTIO
FEATURES
■
■
■
■
■
■
■
■
5V at 600mA or 12V at 120mA from 2-Cell Supply
200µA Quiescent Current
Logic Controlled Shutdown to 15µA
Low VCESAT Switch: 310mV at 2A Typical
Burst ModeTM Operation at Light Load
Current Mode Operation for Excellent
Line and Load Transient Response
Available in 8-Lead SO or PDIP
Operates with Supply Voltage as Low as 2V
UO
APPLICATI
■
■
■
■
The internal low loss NPN power switch can handle
current in excess of 2A and switch at frequencies up to
400kHz. Quiescent current is just 200µA and can be
further reduced to 15µA in shutdown.
Available in 8-pin PDIP or 8-pin SO packaging, the LT1302/
LT1302-5 have the highest switch current rating of any
similarly packaged switching regulators presently on the
market.
Notebook and Palmtop Computers
Portable Instruments
Personal Digital Assistants
Cellular Telephones
Flash Memory
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
UO
■
S
The LT®1302/LT1302-5 are micropower step-up DC/DC
converters that maintain high efficiency over a wide
range of output current. They operate from a supply
voltage as low as 2V and feature automatic shifting
between Burst Mode operation at light load, and current
mode operation at heavy load.
TYPICAL APPLICATI
6
7
2 CELLS
+ C1
100µF
C3
0.1µF
D1
VIN
SW
SHDN
PGND
GND
1
+ C2
100µF
IT
LT1302-5
8
2-Cell to 5V Converter Efficiency
NC
5
SENSE
VC
90
3
88
SHUTDOWN
2
RC
20k
CC
0.01µF
VIN = 3V
86
4
EFFICIENCY (%)
L1
10µH
84
VIN = 2.5V
82
VIN = 2V
80
78
76
74
72
OUTPUT
5V
600mA
LT1302 • F01
C1 = C2 = SANYO OS-CON
L1 = COILTRONICS CTX10-3
COILCRAFT DO3316-103
D1 = MOTOROLA MBRS130LT3
70
1
10
100
LOAD CURRENT (mA)
1000
LT1302 • TA02
Figure 1. 2-Cell to 5V/600mA DC/DC Converter
1
LT1302/LT1302-5
U
U
RATI GS
W
W W
W
AXI U
U
ABSOLUTE
PACKAGE/ORDER I FOR ATIO
VIN Voltage ............................................................. 10V
SW Voltage ............................................................. 25V
FB Voltage .............................................................. 10V
SHDN Voltage ......................................................... 10V
VC Voltage ................................................................ 4V
IT Voltage .................................................................. 4V
Maximum Power Dissipation ............................ 700mW
Operating Temperature Range .................... 0°C to 70°C
Storage Temperature Range ............... – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
TOP VIEW
GND 1
8
PGND
VC 2
7
SW
SHDN 3
6
VIN
(SENSE*)FB 4
5
IT
N8 PACKAGE
S8 PACKAGE
8-LEAD PDIP
8-LEAD PLASTIC SO
*FIXED VERSION
PINS 1 AND 8 ARE INTERNALLY
CONNECTED IN SOIC PACKAGE
ORDER PART
NUMBER
LT1302CN8
LT1302CS8
LT1302CN8-5
LT1302CS8-5
S8 PART MARKING
1302
13025
TJMAX = 125°C, θJA = 100°C/W (N8)
TJMAX = 125°C, θJA = 80°C/W (S8)
Consult factory for Industrial and Military grade parts.
DC ELECTRICAL CHARACTERISTICS
SYMBOL
IQ
PARAMETER
Quiescent Current
VIN
Input Voltage Range
TA = 25°C, VIN = 2.5V, unless otherwise noted.
CONDITIONS
VSHDN = 0.5V, VFB = 1.3V
VSHDN = 1.8V
MIN
●
●
●
2.0
2.2
1.22
●
4.85
●
175
160
75
●
VFB
VOS
DC
tON
tOFF
VCESAT
Feedback Voltage (LT1302)
Feedback Pin Bias Current (LT1302)
Output Sense Voltage (LT1302-5)
Output Ripple Voltage (LT1302-5)
Sense Pin Resistance to Ground (LT1302-5)
Offset Voltage
Comparator Hysteresis
Oscillator Frequency
Maximum Duty Cycle
Switch On Time
Switch Off Time
Output Line Regulation
Switch Saturation Voltage
VC = 0.4V
VFB = 1V
VC = 0.4V
VC = 0.4V
See Block Diagram
(Note 1)
Current Limit Not Asserted (Note 2)
Current Limit Not Asserted
2 < VIN < 8V
ISW = 2A
●
TYP
200
15
1.24
100
5.05
50
420
15
5
220
86
3.9
0.7
0.06
310
●
Switch Leakage Current
Switch Current Limit
VSHDNH
VSHDNL
ISHDN
Error Amplifier Voltage Gain
Shutdown Pin High
Shutdown Pin Low
Shutdown Pin Bias Current
VSW = 5V, Switch Off
VC = 0.4V (Burst Mode Operation)
VC = 1.25V (Full Power) (Note 3)
0.9V ≤ VC ≤ 1.2V, ∆VC/∆VFB
●
●
2.0
50
1.8
0.1
1
2.8
75
●
VSHDN = 5V
VSHDN = 2V
VSHDN = 0V
I T Pin Resistance to Ground
The ● denotes specifications which apply over the 0°C to 70°C
temperature range.
Note 1: Hysteresis is specified at DC. Output ripple depends on capacitor
size and ESR.
2
●
●
●
8
3
0.1
3.9
MAX
300
25
8
1.26
5.25
265
310
95
0.15
400
475
10
3.9
0.5
20
1
UNITS
µA
µA
V
V
V
nA
V
mV
kΩ
mV
mV
kHz
kHZ
%
µs
µs
%/V
mV
mV
µA
A
A
V/ V
V
V
µA
µA
µA
kΩ
Note 2: The LT1302 operates in a variable frequency mode. Switching
frequency depends on load inductance and operating conditions and may
be above specified limits.
Note 3: Minimum switch current 100% tested. Maximum switch current
guaranteed by design.
LT1302/LT1302-5
U W
TYPICAL PERFORMANCE CHARACTERISTICS
No-Load Quiescent Current
Circuit of Figure 1
Switch Saturation Voltage
400
TA = 25°C
TA = 25°C
450
ISW = 2A
400
350
SATURATION VOLTAGE (mV)
500
400
VCESAT (V)
QUIESCENT CURRENT (µA)
Switch Saturation Voltage
600
500
300
250
200
300
200
150
100
100
350
300
250
200
150
50
0
2.0
0
2.5
3.0
3.5
4.0
SUPPLY VOLTAGE (V)
4.5
5.0
0
1
2
3
SWITCH CURRENT (A)
100
–50
4
–25
0
25
50
TEMPERATURE (°C)
75
100
1302 G02
1302 G03
1302 G01
LT1302 Feedback Voltage
LT1302-5 Sense Pin Resistance
1.250
Quiescent Current
600
300
500
250
VIN = 2.5V
SWITCH OFF
SENSE RESISTANCE (kΩ)
FEEDBACK VOLTAGE (V)
1.240
1.235
1.230
1.225
1.220
1.215
1.210
QUIESCENT CURRENT (µA)
1.245
400
300
200
200
150
100
50
100
1.205
1.200
–50
–25
0
25
50
TEMPERATURE (°C)
75
0
–50
100
–25
0
25
50
TEMPERATURE (°C)
1302 G04
Error Amplifier Offset Voltage
LT1302-5 Output Voltage
10
5
0
25
50
TEMPERATURE (°C)
75
100
1302 G07
100
4.5
5.050
5.025
5.000
4.975
4.0
3.5
3.0
4.950
2.5
4.925
–25
75
Maximum On-Time
ON-TIME (µs)
OUTPUT VOLTAGE (V)
15
0
25
50
TEMPERATURE (°C)
5.0
5.075
20
–25
1302 G06
5.100
25
OFFSET VOLTAGE (mV)
0
–50
100
1302 G05
30
0
–50
75
4.900
–50
–25
0
25
50
TEMPERATURE (°C)
75
100
1302 G08
2.0
–50
–25
0
25
50
TEMPERATURE (°C)
75
100
1302 G09
3
LT1302/LT1302-5
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Pin Bias Current
Oscillator Frequency
Maximum Duty Cycle
100
300
20
TA = 25°C
18
FREQUENCY (kHz)
DUTY CYCLE (%)
90
80
70
60
SHUTDOWN CURRENT (µA)
275
250
225
200
175
16
14
12
10
8
6
4
2
50
–50
–25
0
25
50
TEMPERATURE (°C)
75
100
150
–50
–25
75
0
25
50
TEMPERATURE (°C)
1302 G10
0
100
0
1
6
4
2
3
5
SHUTDOWN VOLTAGE (V)
1302 G11
LT1302-5 Output Voltage vs
Load Current
7
8
1302 G12
Maximum Output Power*
Boost Mode
5.20
20
16
5.10
OUTPUT POWER (W)
OUTPUT VOLTAGE (V)
5.15
5.05
5.00
VIN = 4V
4.95
VIN = 2.2V
VIN = 3V
12
8
4.90
4
4.85
4.80
0
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
LOAD CURRENT (A)
1302 G13
0
2
6
4
INPUT VOLTAGE (V)
* APPROXIMATE
8
10
1302 G14
U
U
U
PI FU CTIO S
GND (Pin 1): Signal Ground. Feedback resistor and 0.1µF
ceramic bypass capacitor from VIN should be connected
directly to this pin.
VC (Pin 2): Frequency Compensation Pin. Connect series
RC to GND. Keep trace short.
SHDN (Pin 3): Shutdown. Pull high to effect shutdown; tie
to ground for normal operation.
FB/Sense (Pin 4): Feedback/Sense. On the LT1302 this
pin connects to CMP1 input. On the LT1302-5 this pin
connects to the output resistor string.
4
IT (Pin 5): Normally left floating. Addition of a 3.3k resistor
to GND forces the LT1302 into current mode at light loads.
Efficiency drops at light load but increases at medium
loads. See Applications Information section.
VIN (Pin 6): Supply Pin. Must be bypassed with: (1) a 0.1µF
ceramic to GND, and (2) a large value electrolytic to PGND.
When VIN is greater than 5V, a low value resistor (2Ω to
10Ω) is recommended to isolate the VIN pin from input
supply noise.
LT1302/LT1302-5
U
U
U
PI FU CTIO S
SW (Pin 7): Switch Pin. Connect inductor and diode here.
Keep layout short and direct.
and 8 are thermally connected to the die. One square inch
of PCB copper provides an adequate heat sink for the
device.
PGND (Pin 8): Power Ground. Pins 8 and 1 should be
connected under the package. In the SO package, pins 1
W
BLOCK DIAGRA SM
D1
L1
VIN
C1
+
6
C3
7
VIN
36mV
R5
730Ω
A2
–
1.24V
REFERENCE
+
VOS
15mV
R2
220kHz
OSCILLATOR
Q5
SHUTDOWN
Q4
160X
DRIVER
VIN
Q1
A1
SHDN
Q3
VIN
–
3
A3
HYSTERETIC
COMPARATOR 2µA
FB
4
C5
100pF
OFF
ENABLE
–
R1
SW
R4
1.75Ω
+
CMP1
VOUT
+
C2
0.1µF
Q2
BIAS
+
ERROR
AMPLIFIER
300Ω
1
GND
2
VC
R3
22k
C4
0.01µF
5
IT
3.6k
8
PGND
1302 F02
Figure 2. LT1302 Block Diagram
5
LT1302/LT1302-5
W
BLOCK DIAGRA SM
SENSE
VIN
SW
4
6
7
36mV
R4
1.75Ω
+
R5
730Ω
A2
R1
315k
–
1.24V
REFERENCE
CMP1
+
OFF
ENABLE
–
3
Q5
–
SHUTDOWN
Q3
Q4
160X
DRIVER
VIN
VIN
Q1
A1
SHDN
A3
HYSTERETIC
COMPARATOR 2µA
VOS
15mV
R2
105k
220kHz
OSCILLATOR
Q2
BIAS
+
ERROR
AMPLIFIER
300Ω
3.6k
1
2
5
8
GND
VC
IT
PGND
1302 F03
Figure 3. LT1302-5 Block Diagram
OPERATIO
U
The LT1302’s operation can best be understood by
examining the block diagram in Figure 2. The LT1302
operates in one of two modes, depending on load. With
light loads, comparator CMP1 controls the output; with
heavy loads, control is passed to error amplifier A1.
Burst Mode operation consists of monitoring the FB pin
voltage with hysteretic comparator CMP1. When the FB
voltage, related to the output voltage by external attenuator R1 and R2, falls below the 1.24V reference voltage,
the oscillator is enabled. Switch Q4 alternately turns on,
causing current buildup in inductor L1, then turns off,
allowing the built-up current to flow into output capacitor C3 via D1. As the output voltage increases, so does
the FB voltage; when it exceeds the reference plus
6
CMP1’s hysteresis (about 5mV) CMP1 turns the oscillator off. In this mode, peak switch current is limited to
approximately 1A by A2, Q2, and Q3. Q2’s current, set at
34µA, flows through R5, causing A2’s negative input to
be 25mV lower than VIN. This node must fall more than
36mV below VIN for A2 to trip and turn off the oscillator.
The remaining 11mV is generated by Q3’s current flowing through R4. Emitter-area scaling sets Q3’s collector
current to 0.625% of switch Q4’s current. When Q4’s
current is 1A, Q3’s current is 6.25mA, creating an 11mV
drop across R4 which, added to R5’s 25mV drop, is
enough to trip A2.
When the output load is increased to the point where the
1A peak current cannot support the output voltage,
LT1302/LT1302-5
OPERATIO
U
CMP1 stays on and the peak switch current is regulated
by the voltage on the VC pin (A1’s output). VC drives the
base of Q1. As the VC voltage rises, Q2 conducts less
current, resulting in less drop across R5. Q4’s peak
current must then increase in order for A2 to trip. This
current mode control results in good stability and immunity to input voltage variations. Because this is a linear,
closed-loop system, frequency compensation is required.
A series RC from VC to ground provides the necessary
pole-zero combination.
The LT1302-5 incorporates feedback resistors R1 and
R2 into the device. Output voltage is set at 5.05V in Burst
Mode, dropping to 4.97V in current mode.
U
W
U
U
APPLICATIONS INFORMATION
Inductor Selection
Inductors used with the LT1302 must fulfill two requirements. First, the inductor must be able to handle current
of 2.5A to 3A without runaway saturation. Rod or drum
core units usually saturate gradually and it is acceptable to
exceed manufacturers’ published saturation currents by
20% or so. Second, it should have low DCR, under 0.05Ω
so that copper loss is kept low. Inductance value is not
critical. Generally, for low voltage inputs down to 2V, a
10µH inductor is recommended (such as Coilcraft DO3316103). For inputs above 4V to 5V use a 22µH unit (such as
Coilcraft DO3316-223). Switching frequency can reach up
to 400kHz so the core material should be able to handle
high frequency without loss. Ferrite or molypermalloy
cores are a better choice than powdered iron. If EMI is a
concern a toroidal inductor is suggested, such as Coiltronics
CTX20-4.
For a boost converter, duty cycle can be calculated by the
following formula:
V
DC = 1– IN
VOUT
A special situation exists where the VOUT/VIN differential is
high, such as a 2V-to-12V converter. The required duty
cycle is higher than the LT1302 can provide, so the
converter must be designed for discontinuous operation.
This means that inductor current goes to zero during the
switch off-time. In the 2V-to-12V case, inductance must
be low enough so that current in the inductor can reach
2A in a single cycle. Inductor value can be defined by:
L≤
(V
IN − VSW
)× t
ON
2A
With the 2V input a value of 3.3µH is acceptable. Since the
inductance is so low, usually a smaller core size can be
used. Efficiency will not be as high as for the continuous
case since peak currents will necessarily be higher.
Table 1 lists inductor suppliers along with appropriate part
numbers.
Table 1. Recommended Inductors
VENDOR
Coilcraft
Coiltronics
Dale
Sumida
PART NO.
DO3316-103
DO3316-153
DO3316-223
CTX10-2
CTX20-4
LPT4545-100LA
LPT4545-200LA
CD105-100
CD105-150
CDR125-220
VALUE(µH)
10
15
22
10
20
10
20
10
15
22
PHONE NO.
(708) 639-6400
(407) 241-7876
(605) 665-9301
(708) 956-0666
Capacitor Selection
The output capacitor should have low ESR for proper
performance. A high ESR capacitor can result in “modehopping” between current mode and Burst Mode at high
load currents because the output voltage will increase by
ISW × ESR when the inductor current is flowing into the
diode. Figure 4 shows output voltage of an LT1302-5
boost converter with two 220µF AVX TPS capacitors at the
output. Ripple voltage at a 510mA load is about 30mVP-P
7
LT1302/LT1302-5
U
W
U
U
APPLICATIONS INFORMATION
and there is no low frequency component. The total ESR
is under 0.03Ω. If a single 100µF aluminum electrolytic
capacitor is used instead, the converter mode-hops between current mode and Burst Mode due to high ESR,
causing the voltage comparator to trip as shown in Figure
5. The ripple voltage is now over 500mVP-P and contains
a low frequency component. Maximum allowable output
capacitor ESR can be calculated by the following formula:
ESRMAX =
VOS × VOUT
VREF × 1A
where,
VOS = 15mV
VREF = 1.24V
Input Capacitor
The input supply should be decoupled with a good quality
electrolytic capacitor close to the LT1302 to provide a
stable input supply. Long leads or traces from power
source to the switcher can have considerable impedance
at the LT1302’s switching frequency. The input capacitor
provides a low impedance at high frequency. A 0.1µF
ceramic capacitor is required right at the VIN pin. When the
input voltage can be above 5V, a 10Ω/1µF decoupling
network for VIN is recommended as detailed in Figure 6.
This network is also recommended when driving a transformer.
VIN > 5V
10Ω
VIN
+
VOUT
50mV/DIV
AC COUPLED
47µF
TO
100µF
SW
+
1µF
LT1302
GND
• • •
PGND
1302 F06
ILOAD
510mA
10mA
Figure 6. A 10Ω/1µF Decoupling Network at VIN Is
Recommended When Input Voltage Is Above 5V
500µs/DIV
1302 F04
Figure 4. Low ESR Output Capacitor Results in Stable
Operation. Ripple Voltage is Under 30mVP-P
SERIES
TPS
OS-CON
595D
TYPE
Surface Mount
Through Hole
Surface Mount
PHONE NO.
(803) 448-9411
(619) 661-6835
(603) 224-1961
Diode Selection
510mA
10mA
500µs/DIV
1302 F05
Figure 5. Inexpensive Electrolytic Capacitor Has High
ESR, Resulting in Mode-Hop, Ripple Voltage Amplitude Is
Over 500mVP-P and Includes Low Frequency Component
8
Table 2. Recommended Capacitors
VENDOR
AVX
Sanyo
Sprague
VOUT
200mV/DIV
AC COUPLED
ILOAD
Table 2 lists capacitor vendors along with device types.
A 2A Schottky diode such as Motorola MBRS130LT3 has
been found to be the best available. Other choices include
1N5821 or MBRS130T3. Do not use “general purpose”
diodes such as 1N4001. They are much too slow for use
in switching regulator applications.
LT1302/LT1302-5
U
U
W
U
APPLICATIONS INFORMATION
Frequency Compensation
Obtaining proper RC values for the frequency compensation network is largely an empirical procedure, since
variations in input and output voltage, topology, capacitor
ESR and inductance make a simple formula elusive. As an
example, consider the case of a 2.5V to 5V boost converter
supplying 500mA. To determine optimum compensation,
the circuit is built and a transient load is applied to the
circuit. Figure 7 shows the setup.
In Figure 7a, the VC pin is simply left floating. Although
output voltage is maintained and transient response is
good, switch current rises instantaneously to the internal
current limit upon application of load. This is an undesirable situation as it places maximum stress on the switch
and the other power components. Additionally, efficiency
is well down from its optimal value. Next, a 0.1µF capacitor
is connected with no resistor. Figure 7b details response.
Although the circuit eventually stabilizes, the loop is quite
underdamped. Initial output “sag” exceeds 5%. Aberrant
VIN
2.5V
VIN
D1
0.1µF
C2
220µF
+
The VC pin is sensitive to high frequency noise. Some
layouts may inject enough noise to modulate peak switch
current at 1/2 the switching frequency. A small capacitor
connected from VC to ground will eliminate this. Do not
exceed 1/10 of the compensation capacitor value.
C3
220µF
IT
LT1302-5
PGND
GND
+
Finally, a 0.01µF/24k series network results in the response shown in Figure 7f. This has optimal damping,
undershoot less than 100mV and settles in less than 1ms.
SHDN
SW
C1
330µF
In Figure 7c, the 0.1µF capacitor has been replaced by a
0.01µF unit. Undershoot is less but the response is still
underdamped. Figure 7d shows the results of the 0.1µF
capacitor and a 10k resistor in series. Now some amount
of damping is observed, and behavior is more controlled.
Figure 7e details response with a 0.01µF/10k series network. Undershoot is down to around 100mV, or 2%. A
slight underdamping is still noticeable.
NC
L1
10µH
+
behavior in the 4th graticule is the result of the LT1302’s
Burst Mode comparator turning off all switching as output
voltage rises above its threshold.
SENSE
VC
500Ω
10Ω
2W
R
PULSE
GENERATOR
C
MTP3055EL
C1, C2, C3 = AVX TPS SERIES
D1 = MOTOROLA MBRS130LT3
L1 = COILCRAFT DO3316-103K
50Ω
1302 F07
Figure 7. Boost Converter with Simulated Load
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
510mA
510mA
ILOAD
ILOAD
10mA
2ms/DIV
1302 F07a
Figure 7a. VC Pin Left Unconnected. Output Shows
Low Frequency Components Under Load
10mA
2ms/DIV
1302 F07b
Figure 7b. 0.1µF from VC to Ground.
Better, but More Improvement Needed
9
LT1302/LT1302-5
U
U
W
U
APPLICATIONS INFORMATION
IT Pin
VOUT
100mV/DIV
AC COUPLED
ILOAD
The IT pin is used to disable Burst Mode, forcing the
LT1302 to operate in current mode even at light load. To
disable Burst Mode, 3.3k resistor R1 is connected from IT
to gound. More conservative frequency compensation
must be used when in this mode. A 0.1µF capacitor and
4.7k resistor from VC to ground has been found to be
adequate. Low frequency Burst Mode ripple can be
reduced or eliminated using this technique in many applications.
510mA
10mA
2ms/DIV
1302 F07c
Figure 7c. 0.01µF from VC to Ground.
Underdamped Response Requires Series R
To illustrate, the transient load response of Figure 8’s
circuit is pictured without and with R1. Figure 8a shows
output voltage and inductor current without the resistor.
Note the 6kHz burst rate when the converter is delivering
25mA. By adding the 3.3k resistor, the low frequency
bursting is eliminated, as shown in Figure 8b. This feature
is useful in systems that contain audio circuitry. At very
light or zero load, switching frequency drops and eventu-
VOUT
100mV/DIV
AC COUPLED
510mA
ILOAD
10mA
2ms/DIV
1302 F07d
Figure 7d. 0.1µF with 10k Series RC.
Classic Overdamped Response
VIN
2.5V
10µH
+
VOUT
100mV/DIV
AC COUPLED
VIN
C1
330µF
MBRS130LT3
+
220µF
10V
0.1µF
+
LT1302-5
PGND
GND
220µF
10V
510mA
ILOAD
SENSE
SW
VC
IT
4.7k
R1
3.3k
0.1µF
10mA
1302 F08
2ms/DIV
1302 F07e
Figure 7e. 0.01µF, 10k Series RC Shows Good
Transient Response. Slight Underdamping
Still Noticeable
VOUT
100mV/DIV
AC COUPLED
ILOAD
Figure 8. Addition of R1 Eliminates Low Frequency
Output Ripple in This 2.5V to 5V Boost Converter
VOUT
100mV/DIV
AC COUPLED
INDUCTOR
CURRENT
1A/DIV
510mA
10mA
ILOAD 525mA
25mA
2ms/DIV
1302 F07f
Figure 7f. 0.01µF, 24k Series RC
Results in Optimum Response
10
VOUT
5V
600mA
1ms/DIV
1302 F08a
Figure 8a. IT Pin Floating. Note 6kHz Burst Rate at
ILOAD = 25mA. 0.1µF/4.7k Compensation Network
Causes 220mV Undershoot
LT1302/LT1302-5
U
U
W
U
APPLICATIONS INFORMATION
ally reaches audio frequencies, but at a much lighter load
than without the IT feature. At some input voltage/load
current combinations, some residual bursting may occur
at frequencies out of the audio band.
Figure 8c details efficiency with and without the addition of
R1. Burst Mode operation keeps efficiency high at light
load with IT floating. Efficiency falls off at light load with
R1 added because the LT1302 cannot transition into Burst
Mode.
VOUT
100mV/DIV
AC COUPLED
INDUCTOR
CURRENT
1A/DIV
Layout
525mA
ILOAD 25mA
1ms/DIV
1302 F08b
Figure 8b. 3.3k Resistor from IT Pin to Ground Forces
LT1302 into Current Mode Regardless of Load. Audio
Frequency Component Eliminated
90
IT FLOATING
80
EFFICIENCY (%)
The IT pin cannot be used as a soft-start. Large capacitors
connected to the pin will cause erratic operation. If operating the device in Burst Mode, let the pin float. Keep high
dV/dt signals away from the pin.
70
3.3k IT TO GND
60
50
40
30
1
10
100
OUTPUT CURRENT (mA)
1000
1302 F08c
Figure 8c. 3.3k Resistor for IT to Ground Increases
Efficiency at Moderate Load, Decreases at Light Load
The high speed, high current switching associated with
the LT1302 mandates careful attention to layout. Follow
the suggested component placement in Figure 9 for proper
operation. High current functions are separated by the
package from sensitive control functions. Feedback resistors R1 and R2 should be close to the feedback pin (pin4).
Noise can easily be coupled into this pin if care is not taken.
A small capacitor (100pF to 200pF) from FB to ground
provides a high frequency bypass. If the LT1302 is operated off a three-cell or higher input, R3 (2Ω to 10Ω) in
series with VIN is recommended. This isolates the device
from noise spikes on the input supply. Do not put in R3 if
the device must operate from a 2V input, as input current
will cause the voltage at the LT1302’s VIN pin to go below
2V. The 0.1µF ceramic bypass capacitor C3 (use X7R, not
Z5U) should be mounted as close as possible to the
package. When R3 is used, C3 should be a 1µF tantalum
unit. Grounding should be segregated as illustrated. C3’s
ground trace should not carry switch current. Run a
VIN
R2
C3
R3
2Ω
L1
+
4
3
6
LT1302
C1
D1
5
7
2
8
1
R1
200pF
SHUTDOWN
RC
CC
+
C2
VOUT
GND (BATTERY AND LOAD RETURN)
1302 F09
Figure 9. Suggested Component Placement for LT1302
11
LT1302/LT1302-5
U
W
U
U
APPLICATIONS INFORMATION
separate ground trace up under the package as shown.
The battery and load return should go to the power side of
the ground copper.
Thermal Considerations
The LT1302 contains a thermal shutdown feature which
protects against excessive internal (junction) temperature. If the junction temperature of the device exceeds the
protection threshold, the device will begin cycling between normal operation and an off state. The cycling is not
harmful to the part. The thermal cycling occurs at a slow
rate, typically 10ms to several seconds, which depends on
the power dissipation and the thermal time constants of
the package and heat sinking. Raising the ambient temperature until the device begins thermal shutdown gives a
good indication of how much margin there is in the
thermal design.
For surface mount devices heat sinking is accomplished
by using the heat spreading capabilities of the PC board
and its copper traces. Experiments have shown that the
heat spreading copper layer does not need to be electrically connected to the tab of the device. The PCB material
can be very effective at transmitting heat between the pad
area attached to pins 1 and 8 of the device, and a ground
or power plane layer either inside or on the opposite side
of the board. Although the actual thermal resistance of the
PCB material is high, the length/area ratio of the thermal
resistance between the layer is small. Copper board stiffeners and plated through holes can also be used to spread
the heat generated by the device.
Table 3 lists thermal resistance for the SO package.
Measured values of thermal resistance for several different board sizes and copper areas are listed for each
surface mount package. All measurements were taken in
still air on 3/32" FR-4 board with 1oz copper. This data can
be used as a rough guideline in estimating thermal resistance. The thermal resistance for each application will be
affected by thermal interactions with other components as
well as board size and shape.
12
Table 3. S8 Package, 8-Lead Plastic SO
COPPER AREA
TOPSIDE*
BACKSIDE
BOARD AREA
THERMAL RESISTANCE
(JUNCTION-TO-AMBIENT)
2500 sq. mm 2500 sq. mm 2500 sq. mm
60°C/W
1000 sq. mm 2500 sq. mm 2500 sq. mm
62°C/W
225 sq. mm
2500 sq. mm 2500 sq. mm
65°C/W
100 sq. mm
2500 sq. mm 2500 sq. mm
69°C/W
100 sq. mm
1000 sq. mm 2500 sq. mm
73°C/W
100 sq. mm
225 sq. mm
2500 sq. mm
80°C/W
100 sq. mm
100 sq. mm
2500 sq. mm
83°C/W
* Pins 1 and 8 attached to topside copper
N8 Package, 8-Lead DIP:
Thermal Resistance (Junction-to-Ambient) = 100°C/W
Calculating Temperature Rise
Power dissipation internal to the LT1302 in a boost
regulator configuration is approximately equal to:
2
V
V
2
OUT + VD
OUT + VD
−
PD = IOUT
R
IOUT VOUT R
V − IOUT VOUT R
VIN −
IN
VIN
VIN
+
(
)
IOUT VOUT + VD − VIN
27
The first term in this equation is due to switch “onresistance.” The second term is from the switch driver. R
is switch resistance, typically 0.15Ω. VD is the diode
forward drop.
The temperature rise can be calculated from:
∆T = PD × θJA
where:
∆T = Temperature Rise
PD = Device Power Dissipation
θJA = Thermal Resistance (Junction-to-Ambient)
LT1302/LT1302-5
U
W
U
U
APPLICATIONS INFORMATION
As an example, consider a boost converter with the
following specifications:
VIN = 3V
VOUT = 6V
IOUT = 700mA
Total power loss in the LT1302, assuming R = 0.15Ω and
VD = 0.45V, is:
(
PD = 700mA
2
0.7 6 + 0.45 − 3
6 + 0.45
6 + 0.45
0.15Ω
−
+
×
×
.
.
.
.
0
7
6
0
15
0
7
6
0
15
27
×
×
3 −
3−
3
3
)(
2
)
( )(
)
= 223mW + 89mW = 312mW
Using the CS8 package with 100 sq. mm topside and
backside heat sinking:
∆T = (312mW)(84°C/W) = 25.9°C rise
With the N8 package:
∆T = 31.2°C
At a 70°C ambient, die temperature would be 101.2°C.
13
LT1302/LT1302-5
U
TYPICAL APPLICATIONS
Single Cell to 5V/150mA Converter
5V/150mA
OUTPUT
L1
3.3µH
D1
220Ω
10Ω
R1
301k
1%
2N3906
(169k FOR 3.3V)
100k
1.5V
CELL
100k
IL
SET
100k
VIN
SW
VIN
SW1
LT1073
FB
GND
SHDN
FB
LT1302
AO
IT
PGND
SW2
56.2k
1%
VC
GND
100pF
4.99k
1%
20k
+
C1
47µF
L1 = COILCRAFT DO3316-332
D1 = MOTOROLA MBRS130LT3
C2
220µF
+
0.1µF
0.01µF
C1 = AVX TPSD476M016R0150
C2 = AVX TPSE227M010R0100
COILCRAFT (708) 639-2361
36.5k
1%
1302 TA03
2V to 12V/120mA Converter
L1
3.3µH
6
7
2 CELLS
+ C1
100µF
C3
0.1µF
D1
VIN
SW
33µF
LT1302
8
PGND
GND
+ C2
FB
3
SHUTDOWN
4
VC
2
RC
20k
R1
100k
1%
100pF
33µF
CC
0.02µF
C1 = AVX TPSD107M010R0100
C2 = AVX TPSD336M025R0200
D1 = MOTOROLA MBRS130LT3
L1 = COILCRAFT DO3316-332
OUTPUT
12V
120mA
14
IT
SHDN
1
+ C2
NC
5
LT1302 • TA04
R2
866k
1%
LT1302/LT1302-5
U
TYPICAL APPLICATIONS
3 Cell to 3.3V Buck-Boost Converter with Auxiliary 12V Regulated Output
VIN
2.5V-8V
10Ω
SHUTDOWN
SHDN
FB
169k
1%
200pF
VC
GND
6
T1D
T1E
4
5
+ C3
47µF
16V
VIN
SW
D2
LT1302
100k
1%
7
13V
0.1µF
IT
PGND
2
+ C1
100µF
16V
D1
24k
4700pF
IN
1
T1C
T1A
12V
120mA
OUT
+ 22µF
9
25V
330k
1%
LT1121
SHDN
3
8
T1B
+
ADJ
GND
+ C2
330µF
6.3V
3.3µF
150k
1%
10
1302 TA05
3.3V OUTPUT
400mA
T1 =
D1, D2 =
C1 =
C2 =
C3 =
DALE LPE-6562-A069, 1:3:1:1:1 TURNS RATIO, 10µH PRIMARY. DALE (605) 665-9301
MOTOROLA MBRS130LT3
AVX TPSE107016R0100
AVX TPSE337006R0100
AVX TPSD476016R0150
2 Li-Ion Cell to 5.8V/600mA DC/DC Converter
C2
220µF
10V
L1
22µH
VIN
4V TO 9V
+
10Ω
+
C1
100µF
16V
SW
FB
VIN
IT
+
1µF
365k
1%
L2
22µH
MBRS130LT3
VOUT
5.8V
600mA
LT1302
SHDN
GND
SHUTDOWN
L1, L2=COILCRAFT DO3316-223
C1=AVX TPSE107016R0100
C2, C3=AVX TPSE227010R0100
VC
PGND
+
20k
100k
1%
C3
220µF
10V
10nF
DUTY CYCLE =
VOUT
VIN + VOUT
1302 TA07
PEAK SWITCH VOLTAGE = VIN + VOUT
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1302/LT1302-5
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead Plastic DIP
0.400*
(10.160)
MAX
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
(
+0.025
0.325 –0.015
8.255
+0.635
–0.381
0.130 ± 0.005
(3.302 ± 0.127)
0.045 – 0.065
(1.143 – 1.651)
0.125
(3.175)
MIN
0.045 ± 0.015
(1.143 ± 0.381)
)
0.018 ± 0.003
(0.457 ± 0.076)
0.100 ± 0.010
(2.540 ± 0.254)
0.015
(0.380)
MIN
N8 0694
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTURSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm).
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157*
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
SO8 0294
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
16
Linear Technology Corporation
LT/GP 0295 10K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
LINEAR TECHNOLOGY CORPORATION 1995
工商网监
湘ICP备2023018690号
