LT1578/LT1578-2.5
1.5A, 200kHz Step-Down
Switching Regulator
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FEATURES
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DESCRIPTIO
The LT ®1578 is a 200kHz monolithic buck mode switching
regulator. A 1.5A switch is included on the die along with
all the necessary oscillator, control and logic circuitry. The
topology is current mode for fast transient response and
good loop stability. The LT1578 is a modified version of the
LT1507 that has been optimized for noise sensitive applications. It will operate over a 4V to 15V input range.
1.5A Switch Current
High Efficiency—Low Loss 0.2Ω Switch
Constant 200kHz Switching Frequency
4V to 15V Input VoltageRange
Minimum Output: 1.21V
Low Supply Current: 1.9mA
Low Shutdown Current: 20µA
Easily Synchronizable Up to 400kHz
Cycle-by-Cycle Current Limit
Reduced EMI Generation
Low Thermal Resistance SO-8 Package
Uses Small Low Value Inductors
In addition, the reference voltage has been lowered to allow the device to produce output voltages down to 1.2V.
Quiescent current has been reduced by a factor of two.
Switch on resistance has been reduced by 30%. Switch transition times have been slowed to reduce EMI generation.
The oscillator frequency has been reduced to 200kHz to
maintain high efficiency over a wide output current range.
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APPLICATIO S
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The pinout has been changed to improve PC layout by allowing the high current, high frequency switching circuitry
to be easily isolated from low current, noise sensitive control circuitry. The new SO-8 package includes a fused
ground lead that significantly reduces the thermal resistance
of the device to extend the ambient operating temperature
range. Standard surface mount external parts can be used
including the inductor and capacitors.
Portable Computers
Battery-Powered Systems
Battery Chargers
Distributed Power Systems
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
Efficiency vs Load Current
3.3V Buck Converter
C3*
10µF TO
50µF
+
C2
0.33µF
VIN
BOOST
90
85
D2
1N914
L1**
15µH
OUTPUT**
3.3V, 1.25A
VSW
LT1578
OPEN = ON
SHDN
GND
* RIPPLE CURRENT RATING ≥ IOUT/2
** INCREASE L1 TO 30µH FOR LOAD
CURRENTS ABOVE 0.6A AND TO
60µH ABOVE 1A
SEE APPLICATIONS INFORMATION
FB
VC
R1
8.66k
CC
100pF
D1
1N5818
R2
4.99k
+
C1
100µF, 10V
SOLID
TANTALUM
1578 TA01
80
EFFICIENCY (%)
INPUT
5V TO 15V
75
70
65
60
VOUT = 3.3V
VIN = 5V
L = 25µH
55
50
0
0.25
0.50 0.75 1.00
LOAD CURRENT (A)
1.25
1.50
1578 TA02
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LT1578/LT1578-2.5
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PACKAGE/ORDER INFORMATION
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(Note 1)
Input Voltage .......................................................... 16V
BOOST Pin Above Input Voltage ............................. 10V
SHDN Pin Voltage ..................................................... 7V
SENSE Pin Voltage .................................................... 4V
FB Pin Voltage (Adjustable Part) ............................ 3.5V
FB Pin Current (Adjustable Part) ............................ 1mA
SYNC Pin Voltage ..................................................... 7V
Operating Junction Temperature Range
LT1578C ............................................... 0°C to 125° C
LT1578I ........................................... – 40°C to 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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ABSOLUTE MAXIMUM RATINGS
ORDER PART
NUMBER
TOP VIEW
VSW 1
8 SYNC
VIN 2
7 SHDN
BOOST 3
GND 4
LT1578CS8
LT1578IS8
LT1578CS8-2.5
LT1578IS8-2.5
6 FB/SENSE
5 VC
S8 PACKAGE
8-LEAD PLASTIC SO
S8 PART MARKING
θJA = 80°C/ W WITH FUSED GROUND PIN
CONNECTED TO GROUND PLANE OR
LARGE LANDS
1578
1578I
157825
578I25
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.
PARAMETER
Feedback Voltage
CONDITIONS
All Conditions
●
All Conditions
●
Sense Voltage (Fixed 2.5)
Sense Pin Resistance
Reference Voltage Line Regulation
Feedback Input Bias Current
Error Amplifier Voltage Gain (Notes 2, 10)
Error Amplifier Transconductance (Note 10)
4.3V ≤ VIN ≤ 15V
●
●
∆I (VC) = ±10µA
●
VC Pin to Switch Current Transconductance
Error Amplifier Source Current
Error Amplifier Sink Current
VC Pin Switching Threshold
VC Pin High Clamp
Switch Current Limit
Slope Compensation (Note 8)
Switch On Resistance (Note 7)
MIN
1.195
1.18
2.46
2.44
5.7
200
800
400
VFB = 1.1V
VFB = 1.4V
Duty Cycle = 0
●
●
40
50
VC Open, VFB = 1.1V, DC ≤ 50%
DC = 80%
ISW = 1.5A
●
1.5
TYP
1.21
2.5
9.5
0.01
0.5
400
1050
1.5
110
130
0.8
2.1
2
0.3
0.2
●
Maximum Switch Duty Cycle
Minimum Switch Duty Cycle (Note 9)
Switch Frequency
Switch Frequency Line Regulation
Frequency Shifting Threshold on FB Pin
Minimum Input Voltage (Note 3)
Minimum Boost Voltage (Note 4)
2
VFB = 1.1V
●
90
86
●
180
160
VC Set to Give 50% Duty Cycle
4.3V ≤ VIN ≤ 15V
∆f = 10kHz
●
●
●
ISW ≤ 1.5A
●
0.4
94
94
8
200
0
0.74
4.0
2.3
MAX
1.225
1.24
2.54
2.56
13.7
0.03
2
UNITS
V
V
V
V
kΩ
%/V
µA
1300
1700
µMho
µMho
A/ V
µA
µA
V
V
A
A
Ω
Ω
%
%
%
kHz
kHz
%/ V
V
V
V
190
200
3.5
0.35
0.45
220
240
0.15
1.0
4.3
3.0
LT1578/LT1578-2.5
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.
PARAMETER
Boost Current (Note 5)
CONDITIONS
ISW = 0.5A
ISW = 1.5A
VIN Supply Current (Note 6)
Shutdown Supply Current
MIN
TYP
9
27
1.9
20
2.34
0.13
0.25
2.42
0.37
0.45
1.5
●
●
●
VSHDN = 0V, VIN ≤ 15V, VSW = 0V, VC Open
●
Lockout Threshold
Shutdown Thresholds
VC Open
VC Open Device Shutting Down
Device Starting Up
●
●
●
Synchronization Threshold
Synchronizing Range
SYNC Pin Input Resistance
250
MAX
18
50
2.7
50
75
2.50
0.60
0.7
2.2
400
40
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: Gain is measured with a VC swing equal to 200mV above the
switching threshold level to 200mV below the upper clamp level.
Note 3: Minimum input voltage is not measured directly, but is guaranteed
by other tests. It is defined as the voltage where internal bias lines are still
regulated so that the reference voltage and oscillator frequency remain
constant. Actual minimum input voltage to maintain a regulated output will
depend on output voltage and load current. See Applications Information.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
Note 5: Boost current is the current flowing into the boost pin with the pin
held 5V above input voltage. It flows only during switch on time.
Note 6: Input supply current is the bias current drawn by the input pin
with switching disabled.
UNITS
mA
mA
mA
µA
µA
V
V
V
V
kHz
kΩ
Note 7: Switch on resistance is calculated by dividing VIN to VSW voltage
by the forced current (1.5A). See Typical Performance Characteristics for
the graph of switch voltage at other currents.
Note 8: Slope compensation is the current subtracted from the switch
current limit at 80% duty cycle. See Maximum Output Load Current in the
Applications Information section for further details.
Note 9: Minimum on-time is 400ns typical. For a 200kHz operating
frequency this means the minimum duty cycle is 8%. In frequency
foldback mode, the effective duty cycle will be less than 8%.
Note 10: Transconductance and voltage gain refer to the internal amplifier
exclusive of the voltage divider. To calculate gain and transconductance
referred to the sense pin on the fixed voltage parts, divide values shown by
the ratio 2.5/1.21.
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Voltage Drop
Switch Peak Current Limit
0.5
TYPICAL
25°C
0.3
–20°C
0.2
0.1
0
2.0
FEEDBACK VOLTAGE (V)
SWITCH PEAK CURRENT (A)
125°C
0.4
SWITCH VOLTAGE (V)
Feedback Pin Voltage
1.23
2.5
MINIMUM
1.5
1.0
0.5
0
0
0.25
0.50 0.75 1.00
SWITCH CURRENT (A)
1.25
1.50
1576 G01
0
20
60
40
DUTY CYCLE (%)
80
100
1576 G02
1.22
1.21
1.20
1.19
0
25
50
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
125
1576 G03
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LT1578/LT1578-2.5
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TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Pin Bias Current
(VSHDN = Lockout Threshold)
Shutdown Pin Bias Current
(VSHDN = Shutdown Threshold)
SHDN PIN CURRENT (µA)
3
2
1
0.8
160
0.7
140
120
100
80
60
CURRENT REQUIRED TO FORCE
SHUTDOWN (FLOWS OUT OF PIN).
AFTER SHUTDOWN, CURRENT
DROPS TO A FEW µA
40
20
0
0
25
50
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
0
0
25
50
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
125
1576 G04
Standby Thresholds
ON
2.43
STANDBY
2.42
2.41
25
VSHDN = 0V
15
10
5
100
ROUT
570k
COUT
2.4pF
50
ERROR AMPLIFIER EQUIVALENT CIRCUIT
0
0
0.1
0.2
0.3
SHUTDOWN VOLTAGE (V)
0.4
1576 G010
Frequency Foldback
250
SWITCHING FREQUENCY
1200
1000
800
600
400
200
150
100
50
200
RLOAD = 50Ω
–500
1k
10k
FREQUENCY (Hz)
TRANSCONDUCTANCE (µMho)
GAIN
PHASE (DEG)
GAIN (µMho)
0
1400
VC
100k
–50
1M
1576 G09
4
5
Error Amplifier Transconductance
150
100
10
15
1600
PHASE
1500
10
15
1576 G08
200
0
5
10
INPUT VOLTAGE (V)
VIN = 10V
20
0
Error Amplifier Transconductance
)
Shutdown Supply Current
20
0
125
1576 G06
30
125
2000
(
0.2
25
1576 G07
VFB 1 × 10–3
SHUTDOWN
0.3
0
0
50
25
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
125
INPUT SUPPLY CURRENT (µA)
INPUT SUPPLY CURRENT (µA)
SHUTDOWN PIN VOLTAGE (V)
2.44
2.40
50
100
–50 –25
25
75
0
JUNCTION TEMPERATURE (°C)
START-UP
0.4
Shutdown Supply Current
2.45
500
0.5
1576 G05
2.46
1000
0.6
0.1
SWITCHING FREQUENCY (kHz)
OR CURRENT (µA)
SHDN PIN CURRENT (µA)
AT 2.44V STANDBY THRESHOLD
(CURRENT FLOWS OUT OF PIN)
Shutdown Thresholds
180
SHUTDOWN PIN VOLTAGE (V)
4
0
25
75 100
0
50
–50 –25
JUNCTION TEMPERATURE (°C)
FEEDBACK PIN CURRENT
125
1576 G11
0
0
1.0
0.5
1.5
FEEDBACK VOLTAGE (V)
2.0
1576 G12
LT1578/LT1578-2.5
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TYPICAL PERFORMANCE CHARACTERISTICS
Switching Frequency
3.0
220
200
180
4.50
2.5
4.25
INPUT VOLTAGE (V)
SWITCH CURRENT LIMIT (A)
240
FREQUENCY (kHz)
Minimum Input Voltage to Start
with 3.3V Output
Switch Current Limit Foldback
2.0
1.5
1.0
4.00
3.75
0.5
160
0
25
50
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
0
125
1.0
0.4
0.6
0.8
0.2
FEEDBACK PIN VOLTAGE (V)
0
Maximum Output Current
at VOUT = 2.5V
1.6
L = 15µH
0.8
0.6
0.4
L = 30µH
L = 30µH
1.2
OUTPUT CURRENT (A)
1.0
1.4
1.4
OUTPUT CURRENT (A)
L = 30µH
L = 60µH
L = 60µH
L = 60µH
1.2
L = 15µH
1.0
0.8
0.6
0.4
9
0
12
15
8
10
12
14
0.8
0.6
0.4
4
6
8
10
12
14
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1578 G17
1578 G16
1578 G15
VC Pin Shutdown Threshold
BOOST Pin Current
30
1.0
THRESHOLD VOLTAGE (V)
25
BOOST PIN CURRENT (mA)
L = 15µH
1.0
0
6
4
INPUT VOLTAGE (V)
20
15
10
5
0
1.2
0.2
0.2
0.2
6
1000
1576 G14
1.6
1.6
1.4
10
100
LOAD CURRENT (mA)
1
Maximum Output Current
at VOUT = 3.3V
Maximum Output Current
at VOUT = 5V
OUTPUT CURRENT (A)
3.50
1578 G19
1576 G13
0
1.2
0
0.25
0.50 0.75 1.00
SWITCH CURRENT (A)
1.25
1.50
1576 G20
0.8
0.6
0.4
0.2
0
0
25
50
75 100
–50 –25
JUNCTION TEMPERATURE (°C)
125
1576 G21
Kool Mµ is a registered trademark of Magnetics, Inc.
Metglas is a registered trademark of AlliedSignal, Inc.
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LT1578/LT1578-2.5
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PIN FUNCTIONS
VSW (Pin 1): The switch pin is the emitter of the on-chip
power NPN switch. This pin is driven up to the input pin
voltage during switch on time. Inductor current drives the
switch pin negative during switch off time. Negative voltage is clamped with the external catch diode. Maximum
negative switch voltage allowed is – 0.8V.
VIN (Pin 2): This is the collector of the on-chip power NPN
switch. This pin powers the internal circuitry and internal
regulator. At NPN switch on and off, high dI/dt edges occur
through this pin. Keep the external bypass and catch diode
close to this pin. Trace inductance in this path will create
a voltage spike at switch off, adding to the VCE voltage
across the internal NPN.
BOOST (Pin 3): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch. Without this added voltage, the
typical switch voltage loss would be about 1.5V. The
additional boost voltage allows the switch to saturate with
its voltage drop approximating that of a 0.2Ω FET structure. Efficiency improves from 75% for conventional bipolar designs to > 88% for the LT1578.
GND (Pin 4): The GND pin connection needs consideration
for two reasons. First, it acts as the reference for the
regulated output, so load regulation will suffer if the
“ground” end of the load is not at the same voltage as the
GND pin of the IC. This condition will occur when load
current or other currents flow through metal paths between the GND pin and the load ground point. Keep the
ground path short between the GND pin and the load and
use a ground plane when possible. The second consideration is EMI caused by GND pin current spikes. Internal
capacitance between the VSW pin and the GND pin creates
very narrow (100MHz oscilloscope must be
used, and waveforms should be observed on the leads of
the package. This switch off spike will also cause the SW
node to go below ground. The LT1578 has special circuitry
inside which mitigates this problem, but negative voltages
over 1V lasting longer than 10ns should be avoided. Note
that 100MHz oscilloscopes are barely fast enough to see
the details of the falling edge overshoot in Figure 7.
A second, much lower frequency ringing is seen during
switch off time if load current is low enough to allow the
inductor current to fall to zero during part of the switch off
time (see Figure 8). Switch and diode capacitance resonate with the inductor to form damped ringing at 1MHz to
10 MHz. This ringing is not harmful to the regulator and it
has not been shown to contribute significantly to EMI. Any
attempt to damp it with an RC snubber will slightly degrade
efficiency.
INPUT BYPASSING AND VOLTAGE RANGE
RISE AND FALL
WAVEFORMS ARE
SUPERIMPOSED
(PULSE WIDTH IS
NOT 350ns)
5V/DIV
50ns/DIV
1578 F07
Figure 7. Switch Node Response
5V/DIV
SWITCH NODE
VOLTAGE
INDUCTOR
CURRENT
50mA/DIV
1µs/DIV
1578 F08
Figure 8. Discontinuous Mode Ringing
Input Bypass Capacitor
Step-down converters draw current from the input supply
in pulses. The average height of these pulses is equal to
load current, and the duty cycle is equal to VOUT/ VIN. Rise
and fall times of the current are very fast. A local bypass
capacitor across the input supply is necessary to ensure
proper operation of the regulator and minimize the ripple
current fed back into the input supply. The capacitor also
forces switching current to flow in a tight local loop,
minimizing EMI.
Do not cheat on the ripple current rating of the input
bypass capacitor, but also do not be overly concerned with
the value in microfarads. The input capacitor is intended
to absorb all the switching current ripple, which can have
an RMS value as high as one half of the load current. Ripple
current ratings on the capacitor must be observed to
ensure reliable operation. In many cases it is necessary to
parallel two capacitors to obtain the required ripple rating.
Both capacitors must be of the same value and manufacturer to guarantee power sharing. The actual value of the
capacitor in microfarads is not particularly important
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LT1578/LT1578-2.5
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APPLICATIONS INFORMATION
because at 200kHz, any value above 15µF is essentially
resistive. RMS ripple current rating is the critical parameter. Actual RMS current can be calculated from:
(
)
IRIPPLE RMS = IOUT VOUT VIN − VOUT / VIN
( )
2
The term inside the radical has a maximum value of 0.5
when input voltage is twice output, and stays near 0.5 for
a relatively wide range of input voltages. It is common
practice therefore to simply use the worst-case value and
assume that RMS ripple current is one half of load current.
At maximum output current of 1.5A for the LT1578, the
input bypass capacitor should be rated at 0.75A ripple
current. Note however, that there are many secondary
considerations in choosing the final ripple current rating.
These include ambient temperature, average versus peak
load current, equipment operating schedule, and required
product lifetime. For more details, see Application Notes
19 and 46, and Design Note 95.
series for instance, see Table 3), but even these units may
fail if the input voltage surge approaches the maximum
voltage rating of the capacitor. AVX recommends derating
capacitor voltage by 2:1 for high surge applications. The
highest voltage rating is 50V, so 25V may be a practical
input voltage upper limit when using solid tantalum capacitors for input bypassing.
Larger capacitors may be necessary when the input voltage is very close to the minimum specified on the data
sheet. Small voltage dips during switch on time are not
normally a problem, but at very low input voltage they may
cause erratic operation because the input voltage drops
below the minimum specification. Problems can also
occur if the input-to-output voltage differential is near
minimum. The amplitude of these dips is normally a
function of capacitor ESR and ESL because the capacitive
reactance is small compared to these terms. ESR tends to
be the dominate term and is inversely related to physical
capacitor size within a given capacitor type.
Input Capacitor Type
SYNCHRONIZING
Some caution must be used when selecting the type of
capacitor used at the input to regulators. Aluminum
electrolytics are lowest cost, but are physically large to
achieve adequate ripple current rating, and size constraints (especially height) may preclude their use.
Ceramic capacitors are now available in larger values, and
their high ripple current and voltage rating make them
ideal for input bypassing. Cost is fairly high and footprint
may also be somewhat large. Solid tantalum capacitors
would be a good choice, except that they have a history of
occasional spectacular failures when they are subjected to
large current surges during power-up. The capacitors can
short and then burn with a brilliant white light and lots of
nasty smoke. This phenomenon occurs in only a small
percentage of units, but it has led some OEMs to forbid
their use in high surge applications. The input bypass
capacitors of regulators can see these high surges when
a battery or high capacitance source is connected. Several
manufacturers have developed a line of solid tantalum
capacitors specially tested for surge capability (AVX TPS
The SYNC pin is used to synchronize the internal oscillator
to an external signal. The SYNC input must pass from a
logic level low, through the maximum synchronization
threshold with a duty cycle between 10% and 90%. The
input can be driven directly from a logic level output. The
synchronizing range is equal to initial operating frequency
up to 400kHz. This means that minimum practical sync
frequency is equal to the worst-case high self-oscillating
frequency (250kHz), not the typical operating frequency of
200kHz. Caution should be used when synchronizing
above 280kHz because at higher sync frequencies the
amplitude of the internal slope compensation used to
prevent subharmonic switching is reduced. This type of
subharmonic switching only occurs at input voltages less
than twice output voltage. Higher inductor values will tend
to eliminate this problem. See Frequency Compensation
section for a discussion of an entirely different cause of
subharmonic switching before assuming that the cause is
insufficient slope compensation. Application Note 19 has
more details on the theory of slope compensation.
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APPLICATIONS INFORMATION
At power-up, when VC is being clamped by the FB pin (see
Figure 2, Q2), the sync function is disabled. This allows the
frequency foldback to operate in the shorted output condition. During normal operation, switching frequency is
controlled by the internal oscillator until the FB pin reaches
0.7V, after which the SYNC pin becomes operational. If no
synchronization is required, this pin should be connected
to ground.
Power dissipation in the LT1578 chip comes from four
sources: switch DC loss, switch AC loss, boost circuit
current, and input quiescent current. The following formulas show how to calculate each of these losses. These
formulas assume continuous mode operation, so they
should not be used for calculating efficiency at light load
currents.
Switch loss:
PSW =
( ) (VOUT) + 60ns(IOUT)(VIN)(f)
2
VIN
Boost current loss:
2
PBOOST =
(
VOUT IOUT / 50
)
VIN
Quiescent current loss:
PQ = VIN 0.55 • 10−3 + VOUT 1.6 • 10−3
2
VOUT 0.004
+
VIN
(
2
10
= 0.1 + 012
. = 0.22W
)
RSW = Switch resistance (≈ 0.2Ω)
60ns = Equivalent switch current/voltage overlap time
f = Switch frequency
Example: with VIN = 10V, VOUT = 5V and IOUT = 1A:
(5) (1/ 50) = 0.05W
PBOOST =
2
10
(5) (0.004)
PQ = 10 0.55 • 10−3 + 5 1.6 • 10−3 +
2
THERMAL CALCULATIONS
RSW IOUT
(0.2)(1) (5) + 60 • 10−9 (1)(10)2 00 • 10 3
PSW =
10
= 0.02W
Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W.
Thermal resistance for LT1578 package is influenced by
the presence of internal or backside planes. With a full
plane under the SO package, thermal resistance will be
about 80°C/W. No plane will increase resistance to about
120°C/W. To calculate die temperature, add in worst-case
ambient temperature:
TJ = TA + θJA (PTOT)
With the SO-8 package (θJA = 80°C/W), at an ambient
temperature of 50°C,
TJ = 50 + 80 (0.29) = 73.2°C
Die temperature is highest at low input voltage, so use
lowest continuous input operating voltage for thermal
calculations.
FREQUENCY COMPENSATION
Loop frequency compensation of switching regulators
can be a rather complicated problem because the reactive
components used to achieve high efficiency also introduce multiple poles into the feedback loop. The inductor
and output capacitor on a conventional step-down converter actually form a resonant tank circuit that can exhibit
peaking and a rapid 180° phase shift at the resonant
frequency. By contrast, the LT1578 uses a “current mode”
architecture to help alleviate the phase shift created by the
inductor. The basic connections are shown in Figure 9.
Figure 10 shows a Bode plot of the phase and gain of the
power section of the LT1578, measured from the VC pin to
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LT1578/LT1578-2.5
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the output. Gain is set by the 1.5A/V transconductance of
the LT1578 power section and the effective complex
impedance from output to ground. Gain rolls off smoothly
above the 160Hz pole frequency set by the 100µF output
capacitor. Phase drop is limited to about 85°. Phase
recovers and gain levels off at the zero frequency (≈16kHz)
set by capacitor ESR (0.1Ω).
Error amplifier transconductance phase and gain are shown
in Figure 11. The error amplifier can be modeled as a
transconductance of 1000µMho, with an output impedance of 570kΩ in parallel with 2.4pF. In all practical
applications, the compensation network from the VC pin to
ground has a much lower impedance than the output
impedance of the amplifier at frequencies above 200Hz.
This means that the error amplifier characteristics themselves do not contribute excess phase shift to the loop, and
the phase/gain characteristics of the error amplifier section are completely controlled by the external compensation network.
In Figure 12, full loop phase/gain characteristics are
shown with a compensation capacitor of 100pF, giving the
error amplifier a pole at 2.8kHz, with phase rolling off to
90° and staying there. The overall loop has a gain of 66dB
at low frequency, rolling off to unity-gain at 58kHz. The
phase plot shows a two-pole characteristic until the ESR
of the output capacitor brings it back to single pole above
16kHz. Phase margin is about 77° at unity-gain.
2000
LT1578
VSW
1500
150
R1
ESR
–
1.21V
+
GAIN
1000
100
(
500
ROUT
570k
VC
COUT
2.4pF
50
+
C1
0
ERROR AMPLIFIER EQUIVALENT CIRCUIT
0
R2
RC
CF
)
VFB 1 × 10–3
PHASE (DEG)
FB
VC
GND
200
PHASE
OUTPUT
ERROR
AMPLIFIER
GAIN (µMho)
CURRENT MODE
POWER STAGE
gm = 1.5A/V
RLOAD = 50Ω
–500
CC
10
100
1k
10k
FREQUENCY (Hz)
–50
1M
100k
1578 F11
1578 F09
Figure 9. Model for Loop Response
Figure 11. Error Amplifier Gain and Phase
40
40
VIN = 10V
VOUT = 5V
IOUT = 500mA
180
60
135
–40
0
PHASE (DEG)
PHASE
–80
–20
–40
10
100
1k
10k
FREQUENCY (Hz)
–120
100k
LOOP GAIN (dB)
GAIN
PHASE
40
VIN = 10V
VOUT = 5V
IOUT = 500mA
COUT = 100µF
10V, AVX TPS
CC = 100pF
L = 30µH
20
0
45
GAIN
0
–20
10
100
1k
10k
FREQUENCY (Hz)
90
100k
–45
1M
1578 F12
1578 F07
Figure 10. Response from VC Pin to Output
20
Figure 12. Overall Loop Characteristics
LOOP PHASE (DEG)
0
20
GAIN (dB)
80
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Analog experts will note that around 7kHz, phase dips
close to the zero phase margin line. This is typical of
switching regulators, especially those that operate over a
wide range of loads. This region of low phase is not a
problem as long as it does not occur near unity-gain. In
practice, the variability of output capacitor ESR tends to
dominate all other effects with respect to loop response.
Variations in ESR will cause unity-gain to move around,
but at the same time phase moves with it so that adequate
phase margin is maintained over a very wide range of ESR
(≥ ±3:1).
What About a Resistor in the Compensation Network?
It is common practice in switching regulator design to add
a “zero” to the error amplifier compensation to increase
loop phase margin. This zero is created in the external
network in the form of a resistor (RC) in series with the
compensation capacitor. Increasing the size of this resistor generally creates better and better loop stability, but
there are two limitations on its value. First, the combination of output capacitor ESR and a large value for RC may
cause loop gain to stop rolling off altogether, creating a
gain margin problem. An approximate formula for RC
where gain margin falls to zero is:
(
) (GMP)(GMA)(ESR)(1.21)
R C Loop Gain = 1 =
VOUT
GMP = Transconductance of power stage = 1.5A/V
GMA = Error amplifier transconductance = 1(10–3)
ESR = Output capacitor ESR
1.21 = Reference voltage
With VOUT = 5V and ESR = 0.1Ω, a value of 27.5k for RC
would yield zero gain margin, so this represents an upper
limit. There is a second limitation however which has
nothing to do with theoretical small signal dynamics. This
resistor sets high frequency gain of the error amplifier,
including the gain at the switching frequency. If the
switching frequency gain is high enough, an excessive
amout of output ripple voltage will appear at the VC pin
resulting in improper operation of the regulator. In a
marginal case, subharmonic switching occurs, as
evidenced by alternating pulse widths seen at the switch
node. In more severe cases, the regulator squeals or
hisses audibly even though the output voltage is still
roughly correct. None of this will show on a Bode plot
since this is an amplitude insensitive measurement. Tests
have shown that if ripple voltage on the VC is held to less
than 100mVP-P, the LT1578 will generally be well behaved.
The formula below will give an estimate of VC ripple
voltage when RC is added to the loop, assuming that RC is
large compared to the reactance of CC at 200kHz.
(
(RC)(GMA)(VIN − VOUT)(ESR)(1.21)
)
(VIN)(L)(f)
VC RIPPLE =
GMA = Error amplifier transconductance (1000µMho)
If a series compensation resistor of 15k gave the best
overall loop response, with adequate gain margin, the
resulting VC pin ripple voltage with VIN = 10V, VOUT = 5V,
ESR = 0.1Ω, L = 30µH, would be:
15k )(1 • 10 −3 )(10 − 5)(0.1)(1.21)
(
VC(RIPPLE ) =
= 0.151V
−6 200 • 103
10
30
10
•
( )(
)(
)
This ripple voltage is high enough to possibly create
subharmonic switching. In most situations a compromise
value (< 10k in this case) for the resistor gives acceptable
phase margin and no subharmonic problems. In other
cases, the resistor may have to be larger to get acceptable
phase response, and some means must be used to control
ripple voltage at the VC pin. The suggested way to do this
is to add a capacitor (CF) in parallel with the RC /CC network
on the VC pin. The pole frequency for this capacitor is
typically set at one-fifth of the switching frequency so that
it provides significant attenuation of the switching ripple,
but does not add unacceptable phase shift at the loop
unity-gain frequency. With RC = 15k,
CF =
5
(2π)(f)(RC )
=
(
5
)( )
2π 200 • 103 15k
= 265pF
21
LT1578/LT1578-2.5
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How Do I Test Loop Stability?
The “standard” compensation for LT1578 is a 100pF
capacitor for CC, with RC = 0. While this compensation will
work for most applications, the “optimum” value for loop
compensation components depends, to various extents,
on parameters which are not well controlled. These include inductor value (±30% due to production tolerance,
load current and ripple current variations), output capacitance (±20% to ±50% due to production tolerance,
temperature, aging and changes at the load), output
capacitor ESR (±200% due to production tolerance,
temperature and aging), and finally, DC input voltage and
output load current . This makes it important for the
designer to check out the final design to ensure that it is
“robust” and tolerant of all these variations.
One way to check switching regulator loop stability is by
pulse loading the regulator output while observing the
transient response at the output, using the circuit shown
in Figure 13. The regulator loop is “hit” with a small
transient AC load current at a relatively low frequency,
50Hz to 1kHz. This causes the output to jump a few
millivolts, then settle back to the original value, as shown
in Figure 14. A well behaved loop will settle back cleanly,
whereas a loop with poor phase or gain margin will “ring”
as it settles. The number of rings indicates the degree of
stability, and the frequency of the ringing shows the
approximate unity-gain frequency of the loop. Amplitude
of the signal is not particularly important, as long as the
amplitude is not so high that the loop behaves nonlinearly.
RIPPLE FILTER
470Ω
SWITCHING
REGULATOR
ADJUSTABLE
INPUT SUPPLY
+
ADJUSTABLE
DC LOAD
100µF TO
1000µF
3300pF
TO X1
OSCILLOSCOPE
PROBE
4.7k
330pF
50Ω
TO
OSCILLOSCOPE
SYNC
100Hz TO 1kHz
100mV TO 1VP-P
1578 F13
Figure 13. Loop Stability Test Circuit
VOUT AT
IOUT = 500mA
BEFORE FILTER
VOUT AT
IOUT = 500mA
AFTER FILTER
VOUT AT
IOUT = 50mA
AFTER FILTER
LOAD PULSE
THROUGH 50Ω
f ≈ 780Hz
10mV/DIV
5A/DIV
0.2ms/DIV
Figure 14. Loop Stability Check
22
1578 F14
LT1578/LT1578-2.5
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The output of the regulator contains both the desired low
frequency transient information and a reasonable amount
of high frequency (200kHz) ripple. The ripple makes it
difficult to observe the small transient, so a two-pole,
100kHz filter has been added. This filter is not particularly
critical; even if it attenuated the transient signal slightly,
this wouldn’t matter because amplitude is not critical.
light loads is not particularly sensitive to component variation, so if it looks reasonable under a transient test, it will
probably not be a problem in production. Note that frequency of the light load ringing may vary with component
tolerance but phase margin generally hangs in there.
After verifying that the setup is working correctly, start
varying load current and input voltage to see if you can find
any combination that makes the transient response look
suspiciously “ringy.” This procedure may lead to an adjustment for best loop stability or faster loop transient
response. Nearly always you will find that loop response
looks better if you add in several kΩ for RC. Do this only
if necessary, because as explained before, RC above 1k
may require the addition of CF to control VC pin ripple.
If everything looks OK, use a heat gun and cold spray on
the circuit (especially the output capacitor) to bring out
any temperature-dependent characteristics.
The circuit in Figure 15 is a classic positive-to-negative
topology using a grounded inductor. It differs from the
standard approach in the way the IC chip derives its
feedback signal. Because the LT1578 accepts only positive feedback signals, the ground pin must be tied to the
regulated negative output. A resistor divider to ground or,
in this case, the sense pin, then provides the proper
feedback voltage for the chip.
Keep in mind that this procedure does not take initial
component tolerance into account. You should see fairly
clean response under all load and line conditions to ensure
that component variations will not cause problems. One
note here: according to Murphy, the component most
likely to be changed in production is the output capacitor,
because that is the component most likely to have manufacturer variations (in ESR) large enough to cause problems. It would be a wise move to lock down the sources of
the output capacitor in production. Also, try varying component values by a factor of 2 and see if the behavior is still
acceptable. Double and halve the values of RC and CC and
output capacitors. If the regulator still works correctly, it
will likely be good in production.
A possible exception to the “clean response” rule is at very
light loads, as evidenced in Figure 14 with ILOAD = 50mA.
Switching regulators tend to have dramatic shifts in loop
response at very light loads, mostly because the inductor
current becomes discontinuous. One common result is very
slow but stable characteristics. A second possibility is low
phase margin, as evidenced by ringing at the output with
transients. The good news is that the low phase margin at
POSITIVE-TO-NEGATIVE CONVERTER
D1
1N4148
INPUT
5.5V TO
15V
C3
10µF TO
50µF
BOOST
VIN
C2
L1*
0.33µF 15µH
VSW
R1
15.8k
LT1578
+
FB
GND
VC
CC
RC
R2
4.99k
D2
1N5818
+
C1
100µF
10V TANT
×2
* INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS.
SEE APPLICATIONS INFORMATION
** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE
AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION
OUTPUT**
– 5V, 0.5A
1578 F15
Figure 15. Positive-to-Negative Converter
Inverting regulators differ from buck regulators in the
basic switching network. Current is delivered to the output
as square waves with a peak-to-peak amplitude much
greater than load current. This means that maximum load
current will be significantly less than the LT1578’s 1.5A
maximum switch current, even with large inductor values.
The buck converter in comparison, delivers current to the
output as a triangular wave superimposed on a DC level
equal to load current, and load current can approach 1.5A
23
LT1578/LT1578-2.5
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Maximum load current:
( )(
) (
)( )( )
)(
VIN VOUT
VOUT VIN − 0.35
IP −
2 VOUT + VIN f L
IMAX =
VOUT + VIN − 0.35 VOUT + VF
(
(
)(
)
)
IP = Maximum rated switch current
VIN = Minimum input voltage
VOUT = Output voltage
VF = Catch diode forward voltage
0.35 = Switch voltage drop at 1.5A
Example: with VIN(MIN) = 5.5V, VOUT = 5V, L = 30µH,
VF = 0.5V, IP = 1.5A: IMAX = 0.6A. Note that this equation
does not take into account that maximum rated switch
current (IP) on the LT1578 is reduced slightly for duty
cycles above 50%. If duty cycle is expected to exceed 50%
(input voltage less than output voltage), use the actual IP
value from the Electrical Characteristics table.
Operating duty cycle:
DC =
VOUT + VF
VIN − 0.3 + VOUT + VF
(This formula uses an average value for switch loss, so it
may be several percent in error.)
This duty cycle is close enough to 50% that IP can be
assumed to be 1.5A.
OUTPUT DIVIDER
If the adjustable part is used, the resistor connected to
VOUT (R2) should be set to approximately 5k. R1 is
calculated from:
R1 =
(
)
R2 VOUT − 1.21
1.21
INDUCTOR VALUE
Unlike buck converters, positive-to-negative converters
cannot use large inductor values to reduce output ripple
voltage. At 200kHz, values larger than 75µH make almost
no change in output ripple. The graph in Figure 16 shows
peak-to-peak output ripple voltage for a 5V to – 5V converter versus inductor value. The criteria for choosing the
150
OUTPUT RIPPLE VOLTAGE (mVP-P)
with large inductors. Output ripple voltage for the positiveto-negative converter will be much higher than a buck
converter. Ripple current in the output capacitor will also
be much higher. The following equations can be used to
calculate operating conditions for the positive-to-negative
converter.
5V TO –5V CONVERTER
OUTPUT CAPACITOR’S
ESR = 0.1Ω
120
DISCONTINUOUS
ILOAD = 0.1A
90
DISCONTINUOUS
ILOAD = 0.25A
60
30
CONTINUOUS
ILOAD > 0.38A
0
0
15
45
60
30
INDUCTOR SIZE (µH)
75
1578 F16
With the conditions above:
DC =
24
5 + 0.5
= 51%
5.5 − 0.3 + 5 + 0.5
Figure 16. Ripple Voltage on Positive-to-Negative Converter
LT1578/LT1578-2.5
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inductor is therefore typically based on ensuring that peak
switch current rating is not exceeded. This gives the
lowest value of inductance that can be used, but in some
cases (lower output load currents) it may give a value that
creates unnecessarily high output ripple voltage. A compromise value is often chosen that reduces output ripple.
As you can see from the graph, large inductors will not
give arbitrarily low ripple, but small inductors can give
high ripple.
The difficulty in calculating the minimum inductor size
needed is that you must first know whether the switcher
will be in continuous or discontinuous mode at the critical
point where switch current is 1.5A. The first step is to use
the following formula to calculate the load current where
the switcher must use continuous mode. If your load
current is less than this, use the discontinuous mode
formula to calculate the minimum inductor value needed.
If the load current is higher, use the continuous mode
formula.
Output current where continuous mode is needed:
(V ) (I )
4(V + V )(V + V
2
ICONT =
IN
IN
OUT
2
P
IN
OUT + VF
)
Minimum inductor discontinuous mode:
L MIN =
( )( )
(f)(I )
2 VOUT IOUT
2
P
Minimum inductor continuous mode:
L MIN =
(V )(V )
IN
OUT
(
VOUT + VF
2 f VIN + VOUT IP − IOUT 1 +
VIN
( )(
)
For the example above, with maximum load current of
0.25A:
(5.5) (1.5)
= 0.38A
4(5.5 + 5)(5.5 + 5 + 0.5)
2
ICONT =
2
This says that discontinuous mode can be used and the
minimum inductor needed is found from:
L MIN =
( )( ) = 5.6µH
200 • 103 1.5 2
( )
2 5 0.25
In practice, the inductor should be increased by about 30%
over the calculated minimum to handle losses and variations in value. This suggests a minimum inductor of 7.3µH
for this application, but looking at the ripple voltage chart
shows that output ripple voltage could be reduced by a factor of two by using a 30µH inductor. There is no rule of thumb
here to make a final decision. If modest ripple is needed and
the larger inductor does the trick, this is probably the best
solution. If ripple is noncritical use the smaller inductor. If
ripple is extremely critical, a second stage filter may have
to be added in any case, and the lower value of inductance
can be used. Keep in mind that the output capacitor is the
other critical factor in determining output ripple voltage.
Ripple shown on the graph (Figure 16) is with a capacitor’s
ESR of 0.1Ω. This is reasonable for AVX type TPS “D” or
“E” size surface mount solid tantalum capacitors, but the
final capacitor chosen must be looked at carefully for ESR
characteristics.
)
25
LT1578/LT1578-2.5
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Ripple Current in the Input and Output Capacitors
Diode Current
Positive-to-negative converters have high ripple current in
both the input and output capacitors. For long capacitor
lifetime, the RMS value of this current must be less than
the high frequency ripple current rating of the capacitor.
The following formula will give an approximate value for
RMS ripple current. This formula assumes continuous
conduction mode and a large inductor value. Small inductors will give somewhat higher ripple current, especially in
discontinuous mode. The exact formulas are very complex and appear in Application Note 44, pages 30 and 31.
For our purposes here, a simple fudge factor (ff) is added.
The value for ff is about 1.2 for load currents above 0.38A
(in continuous conduction mode) and L ≥10µH. It increases to about 2.0 for smaller inductors at lower load
currents (in discontinuous conduction mode).
Average diode current is equal to load current. Peak diode
current will be considerably higher.
( )( )
Capacitor IRMS = ff IOUT
ff = Fudge factor (1.2 to 2.0)
26
VOUT
VIN
Peak diode current:
Continuous Mode =
IOUT
(VIN + VOUT ) + (VIN)(VOUT )
VIN
2(L)( f)( VIN + VOUT )
Discontinuous Mode =
( )( )
(L)(f)
2 IOUT VOUT
Keep in mind that during start-up and output overloads,
the average diode current may be much higher than with
normal loads. Care should be used if diodes rated less than
1A are used, especially if continuous overload conditions
must be tolerated.
LT1578/LT1578-2.5
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PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
SO8 1298
27
LT1578/LT1578-2.5
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TYPICAL APPLICATION
Dual Output SEPIC␣ Converter
coupling losses. C4 provides a low impedance path to
maintain an equal voltage swing in L1B, improving regulation. In a flyback converter, during switch on time, all the
converter’s energy is stored in L1A only, since no current
flows in L1B. At switch off, energy is transferred by
magnetic coupling into L1B, powering the – 5V rail. C4
pulls L1B positive during switch on time, causing current
to flow, and energy to build in L1B and C4. At switch off,
the energy stored in both L1B and C4 supply the – 5V rail.
This reduces the current in L1A and changes L1B current
waveform from square to triangular. For details on this
circuit see Design Note 100.
The circuit in Figure 17 generates both positive and
negative 5V outputs with a single piece of magnetics. The
inductor L1 is a 33µH surface mount inductor from
Coiltronics. It is manufactured with two identical windings
that can be connected in series or parallel. The topology for
the 5V output is a standard buck converter. The – 5V
topology would be a simple flyback winding coupled to the
buck converter if C4 were not present. C4 creates the
SEPIC (Single-Ended Primary Inductance Converter) topology which improves regulation and reduces ripple
current in L1. Without C4, the voltage swing on L1B
compared to L1A would vary due to relative loading and
INPUT
6V TO 15V
VIN
BOOST
C2
0.33µF
+
L1A*
33µH
R1
15.8k
FB
VC
C3
22µF
35V TANT
OUTPUT
5V
VSW
LT1578
SHDN
GND
D2
1N914
CC
100pF
D1
1N5818
+
C1**
100µF
10V TANT
+
C5**
100µF
10V TANT
R2
4.99k
GND
* L1 IS A SINGLE CORE WITH TWO WINDINGS
COILTRONICS CTX33-2
** AVX TSPD107M010
† IF LOAD CAN GO TO ZERO, AN OPTIONAL
PRELOAD OF 1k TO 5k MAY BE USED TO
IMPROVE LOAD REGULATION
C4**
100µF
+
L1B* D3
1N5818
OUTPUT
–5V†
1578 F17
Figure 17. Dual Output SEPIC Converter
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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40V Input, 100kHz, 5A and 2A
LTC1174
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LT1372/LT1377
500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology
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High Efficiency Step-Down Switching Regulator
15V, 1.5A, 500kHz Switch
LT1676/LT1776
High Efficiency Step-Down Switching Regulators
7.4V to 60V Input, 100kHz/200kHz
LTC1772
SOT-23 Low Voltage Step-Down DC/DC Controller
550kHz, Drives PFET, 6-Lead SOT-23 Package; up to 4.5A Output Current
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High Efficiency Step-Down Converter
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LT1777
Low Noise Step-Down Switching Regulator
48V Input, Internally Limited dV/dt, Programmable di/dt
Burst Mode is a trademark of Linear Technology Corporation.
28
Linear Technology Corporation
1578f LT/TP 0100 4K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1999