LT1940/LT1940L
Dual Monolithic 1.4A,
1.1MHz Step-Down
Switching Regulator
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FEATURES
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DESCRIPTIO
The LT®1940 is a dual current mode PWM step-down
DC/DC converter with internal 2A power switches. Both converters are synchronized to a single 1.1MHz oscillator and
run with opposite phases, reducing input ripple current. The
output voltages are set with external resistor dividers, and
each regulator has independent shutdown and soft-start
circuits. Each regulator generates a power-good signal
when its output is in regulation, easing power supply sequencing and interfacing with microcontrollers and DSPs.
Wide Input Voltage Range
LT1940: 3.6V to 25V
LT1940L: 3.6V to 7V
Two 1.4A Output Switching Regulators with
Internal Power Switches
Constant 1.1MHz Switching Frequency
Anti-Phase Switching Reduces Ripple
Independent Shutdown/Soft-Start Pins
Independent Power Good Indicators Ease
Supply Sequencing
Uses Small Inductors and Ceramic Capacitors
Small 16-Lead Thermally Enhanced
TSSOP Surface Mount Package
The LT1940’s 1.1MHz switching frequency allows the use
of tiny inductors and capacitors, resulting in a very small
dual 1.4A output solution. Constant frequency and ceramic
capacitors combine to produce low, predictable output
ripple voltage. With its wide input range of 3.6V to 25V, the
LT1940 regulates a wide variety of power sources, from
4-cell batteries and 5V logic rails to unregulated wall transformers, lead acid batteries and distributed-power supplies.
The LT1940L is intended to operate from regulated 5V
supplies. A current mode PWM architecture provides fast
transient response with simple compensation components
and cycle-by-cycle current limiting. Frequency foldback and
thermal shutdown provide additional protection.
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APPLICATIO S
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Disk Drives
DSP Power Supplies
Wall Transformer Regulation
Distributed Power Regulation
DSL Modems
Cable Modems
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
VIN
7V TO 25V
4.7µF
VIN
CMDSH-3
BOOST1
LT1940
0.1µF
Efficiency vs Load Current
OUT2
5V
1.4A
100
VIN = 12V
0.1µF
3.3µH
4.7µH
SW1
90
SW2
UPS140
EFFICIENCY (%)
OUT1
3.3V
1.4A
CMDSH-3
BOOST2
UPS140
16.5k
30.1k
FB1
FB2
VC1
VC2
RUN/SS1
PG1
RUN/SS2
PG2
VOUT = 5V
80
VOUT = 3.3V
70
10.0k
10µF
15k
330pF
1nF
GND
15k
330pF
10µF
10.0k
60
100k
1940 F01
POWER
GOOD
0
0.5
1.0
LOAD CURRENT (A)
1.5
1940 F01b
Figure 1. 3.3V and 5V Dual Output Step-Down Converter with Output Sequencing (LT1940)
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LT1940/LT1940L
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ABSOLUTE
AXI U RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
VIN Voltage
LT1940 .................................................... (–0.3), 25V
LT1940L ................................................... (–0.3), 7V
BOOST Pin Voltage
LT1940 ................................................................ 35V
LT1940L ............................................................. 16V
BOOST Pin Above SW Pin
LT1940 ................................................................ 25V
LT1940L ............................................................. 16V
PG Pin Voltage
LT1940 ................................................................ 25V
LT1940L ............................................................... 7V
SW Voltage ................................................................VIN
RUN/SS, FB Pins ...................................................... 6V
Maximum Junction Temperature .......................... 125°C
Operating Temperature Range (Note 2) ...–40°C to 85°C
Storage Temperature Range ..................–65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
BOOST1
1
16 FB1
SW1
2
15 VC1
VIN
3
14 PG1
VIN
4
VIN
5
12 RUN/SS2
VIN
6
11 PG2
SW2
7
10 VC2
BOOST2
8
9
17
LT1940EFE
LT1940LEFE
13 RUN/SS1
FE PART
MARKING
FB2
1940EFE
1940LEFE
FE PACKAGE
16-LEAD PLASTIC TSSOP
EXPOSED PAD (PIN 17) IS GND
MUST BE SOLDERED TO PCB
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VBOOST = 8V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
Minimum Operating Voltage
MIN
TYP
MAX
UNITS
3.4
3.6
V
Quiescent Current
Not Switching
3.8
4.8
mA
Shutdown Current
VRUNSS = 0V
30
45
µA
1.250
1.250
1.250
1.270
1.270
1.270
V
V
V
240
1200
nA
●
Feedback Voltage
0°C to 70°C
–40°C to 85°C
●
●
FB Pin Bias Current
VFB = 1.25V, VC = 0.4V
●
Reference Line Regulation
LT1940: VIN 5V to 25V
LT1940L: VIN 4V to 7V
1.230
1.225
1.215
0.005
0.005
Error Amp GM
330
Error Amp Voltage Gain
180
%/V
%/V
uMhos
VC Source Current
VFB = 1V
42
µA
VC Sink Current
VFB = 1.5V
60
µA
VC Pin to Switch Current Gain
2.4
A/V
VC Switching Threshold
0.75
V
VC Clamp Voltage
1.8
V
Switching Frequency
VFB = 1.1V
●
Switching Phase
Maximum Duty Cycle
Frequency Shift Threshold on FB
●
fSW = 1MHz
1
0.95
1.1
1.1
1.25
1.35
MHz
MHz
150
180
210
Deg
78
88
%
0.5
V
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LT1940/LT1940L
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VBOOST = 8V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
Foldback Frequency
VFB = 0V
Switch Current Limit
Note 3
Switch VCESAT
ISW = 1A
MIN
TYP
1.8
ISW = 1A
BOOST Pin Current
ISW = 1A
RUN/SS Current
RUN/SS Threshold
0.3
kHz
3.2
A
210
320
mV
10
µA
1.8
2.5
V
20
30
mA
2.3
µA
0.6
V
PG Threshold Offset
VFB Rising
125
160
mV
PG Voltage Output Low
VFB = 1V, IPG = 250µA
0.22
0.4
V
PG Pin Leakage
VPG = 2V
0.1
1
µA
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LT1940E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
90
UNITS
2.4
Switch Leakage Current
Minimum Boost Voltage Above Switch (Note 4)
MAX
150
Note 3: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at high duty cycle.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency, VOUT = 3.3V
Efficiency, VOUT = 1.8V
90
VOUT = 1.8V
L = 2.2µH (SUMIDA CR43-2R2)
TA = 25°C
VOUT = 5V
L = 4.7µH (SUMIDA CR43-4R7)
TA = 25°C
VOUT = 3.3V
L = 3.3µH (SUMIDA CR43-3R3)
TA = 25°C
90
80
90
VIN = 5V
70
EFFICIENCY (%)
VIN = 5V
EFFICIENCY (%)
EFFICIENCY (%)
Efficiency, VOUT = 5V
100
100
80
VIN = 12V
70
VIN = 18V
VIN = 8V
80
VIN = 12V
VIN = 18V
70
60
60
50
0
0.5
1.0
LOAD CURRENT (A)
1.5
1940 G01
50
60
0
0.5
1.0
LOAD CURRENT (A)
1.5
1940 G02
0
0.5
1.0
LOAD CURRENT (A)
1.5
1940 G03
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LT1940/LT1940L
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TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Load Current,
VOUT = 1.8V
Maximum Load Current,
VOUT = 3.3V
Switch VCESAT
400
1.8
1.8
TA = 25°C
1.6
LOAD CURRENT (A)
LOAD CURRENT (A)
1.6
L = 2.2µH
L = 1.5µH
1.4
L = 1µH
L = 4.7µH
L = 3.3µH
1.4
L = 2.2µH
1.2
1.2
1.0
SWITCH VOLTAGE (mV)
SLOPE COMPENSATION REQUIRES
L > 2.2µH FOR VIN < 7 WITH VOUT = 3.3V
1.0
0
2
4
8
10 12
6
INPUT VOLTAGE (V)*
14
16
300
200
100
0
0
5
15
20
10
INPUT VOLTAGE (V)*
25
0
1.0
0.5
1.5
1940 G06
1940 G05
1940 G04
VOUT vs Temperature
Current Limit vs Duty Cycle
Boost Pin Current
40
2.0
SW CURRENT (A)
3.40
3.0
CHANNEL 1, FIGURE 1, VIN = 12V
20
TYPICAL
3.35
2.0
VOUT (V)
CURRENT LIMIT (A)
BOOST CURRENT (mA)
2.5
30
MINIMUM
1.5
3.30
1.0
10
3.25
0.5
0
0
0
1.0
1.5
0.5
SWITCH CURRENT (A)
0
2.0
20
60
40
DUTY CYCLE (%)
80
Frequency vs Temperature
1.0
100
125
1940 G10
50
75
100
125
IRUN/SS vs Temperature
1.4
3.0
1.2
2.5
1.0
0.8
0.6
0.4
2.0
1.5
1.0
0.5
0.2
0
75
0
25
50
TEMPERATURE (°C)
25
1940 G09
RUN/SS CURRENT (µA)
SWITCHING FREQUENCY (MHz)
FREQUENCY (MHz)
1.1
0
TEMPERATURE (°C)
Frequency Foldback
1.3
1.2
–25
1940 G08
1940 G07
0.9
–50 –25
3.20
–50
100
0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
1.2
1940 G11
0
–50
–25
75
0
25
50
TEMPERATURE (°C)
100
125
1940 G12
*Maximum VIN is 7V for LT1940L.
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TYPICAL PERFOR A CE CHARACTERISTICS
RUN/SS Thresholds vs
Temperature
6.0
MINIMUM INPUT VOLTAGE (V)
RUNN/SS THRESHOLDS (V)
1.2
1.0
TO SWITCH
0.8
0.6
TO RUN
0.4
0.2
50
25
75
0
TEMPERATURE (°C)
100
125
5.5
VIN TO START
5.0
BOOST DIODE
TIED TO OUTPUT
4.5
BOOST DIODE
TIED TO INPUT
4.0
VIN TO RUN
3.5
3.0
7.5
TA = 25°C
DBOOST = BAT54
MINIMUM INPUT VOLTAGE (V)
1.4
0
–50 –25
Minimum Input Voltage,
VOUT = 5V
Minimum Input Voltage,
VOUT = 3.3V
7.0 VIN TO START
6.5
BOOST DIODE
TIED TO OUTPUT
6.0
BOOST DIODE
TIED TO INPUT
5.5
10
100
LOAD CURRENT (mA)
1940 G13
1000
1940 G14
VIN TO RUN
5.0
4.5
1
TA = 25°C
DBOOST = BAT54
1
10
100
LOAD CURRENT (mA)
1000
1940 G14
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PI FU CTIO S
BOOST1, BOOST2 (Pins 1, 8): The BOOST pins are used
to provide drive voltages, higher than the input voltage, to
the internal bipolar NPN power switches.Tie through a
diode from VOUT or from VIN.
SW1, SW2 (Pins 2, 7): The SW pins are the outputs of the
internal power switches. Connect these pins to the inductors, catch diodes and boost capacitors.
VIN (Pins 3, 4, 5, 6): The VIN pins supply current to the
LT1940’s internal regulator and to the internal power
switches. These pins must be tied to the same source, and
must be locally bypassed.
FB1, FB2 (Pins 9, 16): The LT1940 regulates each feedback pin to 1.25V. Connect the feedback resistor divider
taps to these pins.
VC1, VC2 (Pins 10, 15): The VC pins are the outputs of the
internal error amps. The voltages on these pins control the
peak switch currents. These pins are normally used to
compensate the control loops, but can also be used to
override the loops. Pull these pins to ground with an open
drain to shut down each switching regulator.
PG1, PG2 (Pins 11, 14): The Power Good pins are the
open collector outputs of an internal comparator. PG
remains low until the FB pin is within 10% of the final
regulation voltage. As well as indicating output regulation,
the PG pins can be used to sequence the two switching
regulators. These pins can be left unconnected. The PG
outputs are valid when VIN is greater than 2.4V and either
of the RUN/SS pins is high. The PG comparators are
disabled in shutdown.
RUN/SS1, RUN/SS2 (Pins 12, 13): The RUN/SS pins are
use to shut down the individual switching regulators and
the internal bias circuits. They also provide a soft-start
function. To shut down either regulator, pull the RUN/SS
pin to ground with an open drain or collector. Tie a
capacitor from these pins to ground to limit switch current
during start-up. If neither feature is used, leave these pins
unconnected.
GND (Pin 17): The Exposed Pad of the package provides
both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be
soldered to the circuit board for proper operation.
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LT1940/LT1940L
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BLOCK DIAGRA
VIN
2µA
RUN/SS2
INT REG
AND REF
MASTER
OSC
CLK1
CLK2
2µA
VIN
RUN/SS1
IN
CIN
0.75V
∑
SLOPE
R
S
C1
BOOST
D2
Q
SLAVE
OSC
CLK
C3
SW
L1
OUT
+
ERROR
AMP
0.5V
VC
FB
R1
–
–
C1
D1
R2
+
CF
–
RC
1.25V
CC RUN/SS
+
125mV
ILIMIT
CLAMP
PG
+
GND
1940 F02
–
Figure 2. Block Diagram of the LT1940 with Associated External Components (One of Two Switching Regulators Shown)
The LT1940 is a dual, constant frequency, current mode
buck regulator with internal 2A power switches. The two
regulators share common circuitry including input source,
voltage reference and oscillator, but are otherwise independent. This section describes the operation of the
LT1940; refer to the Block Diagram.
If the RUN/SS (run/soft-start) pins are both tied to ground,
the LT1940 is shut down and draws 30µA from the input
source tied to VIN. Internal 2µA current sources charge
external soft-start capacitors, generating voltage ramps at
these pins. If either RUN/SS pin exceeds 0.6V, the internal
bias circuits turn on, including the internal regulator,
1.25V reference and 1.1MHz master oscillator. In this
state, the LT1940 draws 3.5mA from VIN, whether one or
both RUN/SS pins are high. Neither switching regulator
will begin to operate until its RUN/SS pin reaches ~0.8V.
The master oscillator generates two clock signals of
opposite phase.
The two switchers are current mode step-down regulators. This means that instead of directly modulating the
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LT1940/LT1940L
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BLOCK DIAGRA
duty cycle of the power switch, the feedback loop controls
the peak current in the switch during each cycle. This
current mode control improves loop dynamics and provides cycle-by-cycle current limit.
The Block Diagram shows only one of the two switching
regulators. A pulse from the slave oscillator sets the RS
flip-flop and turns on the internal NPN bipolar power
switch. Current in the switch and the external inductor
begins to increase. When this current exceeds a level
determined by the voltage at VC, current comparator C1
resets the flip-flop, turning off the switch. The current in
the inductor flows through the external Schottky diode,
and begins to decrease. The cycle begins again at the next
pulse from the oscillator. In this way the voltage on the VC
pin controls the current through the inductor to the output.
The internal error amplifier regulates the output voltage by
continually adjusting the VC pin voltage.
The threshold for switching on the VC pin is 0.75V, and an
active clamp of 1.8V limits the output current. The VC pin
is also clamped to the RUN/SS pin voltage. As the internal
current source charges the external soft-start capacitor,
the current limit increases slowly.
Each switcher contains an independent oscillator. This
slave oscillator is normally synchronized to the master
oscillator. However, during start-up, short-circuit or overload conditions, the FB pin voltage will be near zero and an
internal comparator gates the master oscillator clock
signal. This allows the slave oscillator to run the regulator
at a lower frequency. This frequency foldback behavior
helps to limit switch current and power dissipation under
fault conditions.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
internal bipolar NPN power switch for efficient operation.
A power good comparator trips when the FB pin is at 90%
of its regulated value. The PG output is an open collector
transistor that is off when the output is in regulation,
allowing an external resistor to pull the PG pin high. Power
good is valid when the LT1940 is enabled (either RUN/SS
pin is high) and VIN is greater than ~2.4V.
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APPLICATIO S I FOR ATIO
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1%
resistors according to:
R1 = R2(VOUT/1.25 – 1)
R2 should be 10.0kΩ or less to avoid bias current errors.
Reference designators refer to the Block Diagram in
Figure␣ 2.
Input Voltage Range
The minimum input voltage is determined by either the
LT1940’s minimum operating voltage of ~3.5V, or by its
maximum duty cycle. The duty cycle is the fraction of time
that the internal switch is on and is determined by the input
and output voltages:
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.3V at maximum load). This leads to a minimum input
voltage of:
VINMIN = (VOUT + VD)/DCMAX - VD + VSW
with DCMAX = 0.78.
A more detailed analysis includes inductor loss and the
dependence of the diode and switch drop on operating
current. A common application where the maximum duty
cycle limits the input voltage range is the conversion of
5V to 3.3V. The maximum load current that the LT1940
can deliver at 3.3V depends on the accuracy of the 5V
input supply. With a low loss inductor (DCR less than
80mΩ), the LT1940 can deliver 1A for VIN > 4.7V and
1.4A for VIN > 4.85V.
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LT1940/LT1940L
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APPLICATIO S I FOR ATIO
The maximum input voltage is determined by the absolute
maximum ratings of the VIN and BOOST pins and by the
minimum duty cycle DCMIN = 0.15:
VINMAX = (VOUT + VD)/DCMIN – VD + VSW.
This limits the maximum input voltage to ~14V with
VOUT = 1.8V and ~19V with VOUT = 2.5V. Note that this is
a restriction on the operating input voltage; the circuit will
tolerate transient inputs up to the absolute maximum
rating. For the LT1940L, the maximum input voltage is 7V.
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L = (VOUT + VD)/1.2
where VD is the voltage drop of the catch diode (~0.4V) and
L is in µH. With this value the maximum load current will
be ~1.4A, independent of input voltage. The inductor’s
RMS current rating must be greater than your maximum
load current and its saturation current should be about
30% higher. To keep efficiency high, the series resistance
(DCR) should be less than 0.1Ω. Table 1 lists several
vendors and types that are suitable.
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A
larger value provides a slightly higher maximum load
current, and will reduce the output voltage ripple. If your
load is lower than 1.4A, then you can decrease the value of
the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one with
a lower DCR resulting in higher efficiency. Be aware that if
the inductance differs from the simple rule above, then the
maximum load current will depend on input voltage. There
are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum
load current as a function of input voltage and inductor
value for several popular output voltages. Also, low
inductance may result in discontinuous mode operation,
which is okay, but further reduces maximum load current.
For details of maximum output current and discontinuous
mode operation, see Linear Technology Application
Note 44. Finally, for duty cycles greater than 50%
(VOUT/VIN < 0.5), there is a minimum inductance required
to avoid subharmonic oscillations. See AN19. The discussion below assumes continuous inductor current.
The current in the inductor is a triangle wave with an
average value equal to the load current. The peak switch
current is equal to the output current plus half the peak-topeak inductor ripple current. The LT1940 limits its switch
current in order to protect itself and the system from
overload faults. Therefore, the maximum output current
that the LT1940 will deliver depends on the current limit,
the inductor value, and the input and output voltages. L is
chosen based on output current requirements, output
voltage ripple requirements, size restrictions and efficiency goals.
When the switch is off, the inductor sees the output
voltage plus the catch diode drop. This gives the peak-topeak ripple current in the inductor:
∆IL = (1 – DC)(VOUT + VD)/(L • f)
where f is the switching frequency of the LT1940 and L is
the value of the inductor. The peak inductor and switch
current is
ISWPK = ILPK = IOUT + ∆IL/2.
To maintain output regulation, this peak current must be
less than the LT1940’s switch current limit ILIM. ILIM is at
least 1.8A at low duty cycle and decreases linearly to 1.5A
at DC = 0.8. The maximum output current is a function of
the chosen inductor value:
IOUTMAX = ILIM – ∆IL/2 = 1.8A • (1 – 0.21 • DC) – ∆IL/2
If the inductor value is chosen so that the ripple current is
small, then the available output current will be near the
switch current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT1940 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continuous.
Discontinuous operation occurs when IOUT is less than
∆IL/2 as calculated above.
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APPLICATIO S I FOR ATIO
Table 1. Inductors.
Part Number
Value
(µH)
ISAT
(A) DC
DCR
(Ω)
Height
(mm)
CR43-1R4
1.4
2.52
0.056
3.5
CR43-2R2
2.2
1.75
0.071
3.5
CR43-3R3
3.3
1.44
0.086
3.5
CR43-4R7
4.7
1.15
0.109
3.5
CDRH3D16-1R5
1.5
1.55
0.040
1.8
CDRH3D16-2R2
2.2
1.20
0.050
1.8
CDRH3D16-3R3
3.3
1.10
0.063
1.8
CDRH4D28-3R3
3.3
1.57
0.049
3.0
CDRH4D28-4R7
4.7
1.32
0.072
3.0
CDRH5D28-5R3
5.3
1.9
0.028
3.0
CDRH5D18-4R1
4.1
1.95
0.042
2.0
DO1606T-152
1.5
2.10
0.060
2.0
DO1606T-222
2.2
1.70
0.070
2.0
DO1606T-332
3.3
1.30
0.100
2.0
DO1606T-472
4.7
1.10
0.120
2.0
DO1608C-152
1.5
2.60
0.050
2.9
DO1608C-222
2.2
2.30
0.070
2.9
DO1608C-332
3.3
2.00
0.080
2.9
DO1608C-472
4.7
1.50
0.090
2.9
1812PS-222M
2.2
1.7
0.070
3.81
1008PS-182M
1.8
2.1
0.090
2.74
LQH32CN1R0M11L
1.0
1.00
0.078
2.2
LQH32CN2R2M11L
2.2
0.79
0.126
2.2
LQH43CN1R5M01L
1.5
1.00
0.090
2.8
LQH43CN2R2M01L
2.2
0.90
0.110
2.8
LQH43CN3R3M01L
3.3
0.80
0.130
2.8
Sumida
Coilcraft
Murata
Input Capacitor Selection
Bypass the input of the LT1940 circuit with a 4.7µF or
higher ceramic capacitor of X7R or X5R type. A lower value
or a less expensive Y5V type can be used if there is
additional bypassing provided by bulk electrolytic or
tantalum capacitors. The following paragraphs describe
the input capacitor considerations in more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT1940 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
The input capacitor must have low impedance at the
switching frequency to do this effectively, and it must have
an adequate ripple current rating. With two switchers
operating at the same frequency but with different phases
and duty cycles, calculating the input capacitor RMS
current is not simple. However, a conservative value is the
RMS input current for the channel that is delivering most
power (VOUT • IOUT). This is given by:
CINRMS = IOUT √[VOUT • (VIN – VOUT)]/VIN < IOUT/2
and is largest when VIN = 2VOUT (50% duty cycle). As the
second, lower power channel draws input current, the
input capacitor’s RMS current actually decreases as the
out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum
load current from a single channel is ~1.4A, RMS ripple
current will always be less than 0.7A.
The high frequency of the LT1940 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 10µF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors make them the
preferred choice. The low ESR results in very low voltage
ripple and the capacitors can handle plenty of ripple
current. They are also comparatively robust and can be
used in this application at their rated voltage. X5R and X7R
types are stable over temperature and applied voltage, and
give dependable service. Other types (Y5V and Z5U) have
very large temperature and voltage coefficients of capacitance, so they may have only a small fraction of their
nominal capacitance in your application. While they will
still handle the RMS ripple current, the input voltage ripple
may become fairly large, and the ripple current may end up
flowing from your input supply or from other bypass
capacitors in your system, as opposed to being fully
sourced from the local input capacitor.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example a 1µF ceramic capacitor in parallel with a low ESR
tantalum capacitor. For the electrolytic capacitor, a value
larger than 10µF will be required to meet the ESR and
ripple current requirements. Because the input capacitor
1940fa
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is likely to see high surge currents when the input source
is applied, tantalum capacitors should be surge rated. The
manufacturer may also recommend operation below the
rated voltage of the capacitor. Be sure to place the 1µF
ceramic as close as possible to the VIN and GND pins on
the IC for optimal noise immunity.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT1940. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see AN88.
Output Capacitor Selection
For 5V and 3.3V outputs with greater than 1A output, a
10µF 6.3V ceramic capacitor (X5R or X7R) at the output
results in very low output voltage ripple and good transient
response. For lower voltages, 10µF is adequate but increasing COUT to 15µF or 22µF will improve transient
performance. Other types and values can be used; the
following discusses tradeoffs in output ripple and transient performance.
The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy
in order satisfy transient loads and to stabilize the LT1940’s
control loop. Because the LT1940 operates at a high
frequency, you don’t need much output capacitance. Also,
the current mode control loop doesn’t require the presence of output capacitor series resistance (ESR). For these
reasons, you are free to use ceramic capacitors to achieve
very low output ripple and small circuit size.
Estimate output ripple with the following equations:
VRIPPLE = ∆IL/(8f COUT) for ceramic capacitors, and
VRIPPLE = ∆IL ESR for electrolytic capacitors (tantalum
and aluminum);
where ∆IL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low, and the
RMS current rating of the output capacitor is usually not
of concern.
Another constraint on the output capacitor is that it must
have greater energy storage than the inductor; if the stored
energy in the inductor is transferred to the output, you
would like the resulting voltage step to be small compared
to the regulation voltage. For a 5% overshoot, this requirement becomes COUT > 10L(ILIM/VOUT)^2.
Finally, there must be enough capacitance for good transient performance. The last equation gives a good starting
point. Alternatively, you can start with one of the designs
in this data sheet and experiment to get the desired
performance. This topic is covered more thoroughly in the
section on loop compensation.
The high performance (low ESR), small size and robustness of ceramic capacitors make them the preferred type
for LT1940 applications. However, all ceramic capacitors
are not the same. As mentioned above, many of the higher
value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U
types lose a large fraction of their capacitance with applied
voltage and temperature extremes. Because the loop
stability and transient response depend on the value of
COUT, you may not be able to tolerate this loss. Use X7R
and X5R types.
You can also use electrolytic capacitors. The ESRs of most
aluminum electrolytics are too large to deliver low output
ripple. Tantalum and newer, lower ESR organic electrolytic capacitors intended for power supply use are suitable, and the manufacturers will specify the ESR. The
choice of capacitor value will be based on the ESR required
for low ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger
than a ceramic capacitor that would give you similar ripple
performance. One benefit is that the larger capacitance
may give better transient response for large changes in
load current. Table 2 lists several capacitor vendors.
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Table 2. Low-ESR Surface Mount Capacitors
Boost Pin Considerations
Vendor
Type
Series
Taiyo Yuden
Ceramic X5R, X7R
AVX
Ceramic X5R, X7R
Tantalum
TPS
Kemet
Tantalum
Ta Organic
Al Organic
T491,T494,T495
T520
A700
Sanyo
Ta or Al Organic
POSCAP
Panasonic
Al Organic
SP CAP
TDK
Ceramic X5R, X7R
Catch Diode
Use a 1A Schottky diode for the catch diode (D1 in
Figure 2). The diode must have a reverse voltage rating
greater than the maximum input voltage. The ON Semiconductor MBRM120LT3 (20V) and MBRM130LT3 (30V)
are good choices; they have a tiny package with good
thermal properties. Many vendors have surface mount
versions of the 1N5817 (20V) and 1N5818 (30V) 1A
Schottky diodes such as the Microsemi UPS120 that are
suitable.
The capacitor and diode tied to the BOOST pin generate a
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
CMDSH-3 or FMMD914) will work well. Figure 3 shows
three ways to arrange the boost circuit. The BOOST pin
must be more than 2.5V above the SW pin for full efficiency. For outputs of 3.3V and higher the standard circuit
(Figure 3a) is best. For outputs between 2.8V and 3.3V,
use a small Schottky diode (such as the BAT-54). For lower
output voltages the boost diode can be tied to the input
(Figure␣ 3b). The circuit in Figure 3a is more efficient
because the BOOST pin current comes from a lower
voltage source. Finally, as shown in Figure 3c, the anode
of the boost diode can be tied to another source that is at
least 3V. For example, if you are generating 3.3V and 1.8V
and the 3.3V is on whenever the 1.8V is on, the 1.8V boost
diode can be connected to the 3.3V output. In any case,
you must also be sure that the maximum voltage at the
BOOST pin is less than the maximum specified in the
Absolute Maximum Ratings section.
D2
D2
C3
BOOST
VIN
VIN
VOUT
SW
VIN
VIN
SW
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
(3b)
(3a)
D2
D2
VIN2
>VIN + 3V
VIN2 > 3V
BOOST
BOOST
C3
LT1940
VIN
VOUT
GND
GND
VIN
C3
BOOST
LT1940
LT1940
LT1940
SW
VOUT
VIN
VIN
GND
SW
VOUT
GND
1940 F03
VBOOST – VSW ≅ VIN2
MAX VBOOST ≅ VIN2 + VIN
MINIMUM VALUE FOR VIN2 = 3V
1940 F03
MAX VBOOST – VSW ≅ VIN2
MAX VBOOST ≅ VIN2
MINIMUM VALUE FOR VIN2 = VIN + 3V
(3c)
(3d)
Figure 3. Generating the Boost Voltage
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Frequency Compensation
The LT1940 uses current mode control to regulate the
output. This simplifies loop compensation. In particular,
the LT1940 does not require the ESR of the output
capacitor for stability so you are free to use ceramic
capacitors to achieve low output ripple and small circuit
size.
Frequency compensation is provided by the components
tied to the VC pin. Generally a capacitor and a resistor in
series to ground determine loop gain. In addition, there is
a lower value capacitor in parallel. This capacitor is not part
Loop compensation determines the stability and transient
performance. Designing the compensation network is a
bit complicated and the best values depend on the application and in particular the type of output capacitor. A
practical approach is to start with one of the circuits in this
data sheet that is similar to your application and tune the
compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough
discussion of loop compensation and describes how to
test the stability using a transient load.
Figure 4 shows an equivalent circuit for the LT1940
control loop. The error amp is a transconductance amplifier with finite output impedance. The power section,
consisting of the modulator, power switch and inductor, is
modeled as a transconductance amplifier generating an
output current proportional to the voltage at the VC pin.
Note that the output capacitor integrates this current, and
that the capacitor on the VC pin (CC) integrates the error
amplifier output current, resulting in two poles in the loop.
In most cases a zero is required and comes from either the
output capacitor ESR or from a resistor in series with CC.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. A phase lead
capacitor (CPL) across the feedback divider may improve
the transient response.
LT1940
CURRENT MODE
POWER STAGE
gm = 2.5mho
VSW
OUTPUT
ERROR
AMPLIFIER
R1
500k
GND
1.25V
VC
RC
CPL
FB
gm =
340µmho
+
The minimum input voltage of an LT1940 application is
limited by the minimum operating voltage (< 3.6V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by the
boost circuit. If the input voltage is ramped slowly, or the
LT1940 is turned on with its RUN/SS pin when the output
is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. The Typical Performance Characteristics section shows plots of the minimum load current to
start and to run as a function of input voltage for 3.3V and
5V outputs. In many cases the discharged output capacitor will present a load to the switcher which will allow it to
start. The plots show the worst-case situation where VIN
is ramping very slowly. Use a Schottky diode (such as the
BAT-54) for the lowest start-up voltage.
of the loop compensation but is used to filter noise at the
switching frequency.
–
The boost circuit can also run directly from a DC voltage
that is higher than the input voltage by more than 3V, as in
Figure 3d. The diode is used to prevent damage to the
LT1940 in case VIN2 is held low while VIN is present. The
circuit saves several components (both BOOST pins can
be tied to D2). However, efficiency may be lower and
dissipation in the LT1940 may be higher. Also, if VIN2 is
absent, the LT1940 will still attempt to regulate the output,
but will do so with very low efficiency and high dissipation
because the switch will not be able to saturate, dropping
1.5V to 2V in conduction.
ESR
+
C1
C1
R2
CF
POLYMER
OR
TANTALUM
CERAMIC
CC
1940 F05
Figure 4. Model for Loop Response
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Soft-Start and Shutdown
The RUN/SS (Run/Soft-Start) pins are used to place the
individual switching regulators and the internal bias circuits in shutdown mode. They also provide a soft-start
function. To shut down either regulator, pull the RUN/SS
pin to ground with an open-drain or collector. If both
RUN/SS pins are pulled to ground, the LT1940 enters its
shutdown mode with both regulators off and quiescent
current reduced to ~30µA. Internal 2µA current sources
pull up on each pin. If either pin reaches ~0.5V, the internal
bias circuits start and the quiescent current increases to
~3.5mA.
If a capacitor is tied from the RUN/SS pin to ground, then
the internal pull-up current will generate a voltage ramp on
this pin. This voltage clamps the VC pin, limiting the peak
switch current and therefore input current during start up.
A good value for the soft-start capacitor is COUT/10,000,
where COUT is the value of the output capacitor.
The RUN/SS pins can be left floating if the shutdown
feature is not used. They can also be tied together with a
single capacitor providing soft-start. The internal current
sources will charge these pins to ~2.5V.
The RUN/SS pins provide a soft-start function that limits
peak input current to the circuit during start-up. This helps
to avoid drawing more current than the input source can
supply or glitching the input supply when the LT1940 is
enabled. The RUN/SS pins do not provide an accurate
delay to start or an accurately controlled ramp at the
output voltage, both of which depend on the output
capacitance and the load current. However, the power
good indicators can be used to sequence the two outputs,
as described below.
Power Good Indicators
The PG pin is the open collector output of an internal
comparator. PG remains low until the FB pin is within 10%
of the final regulation voltage. Tie the PG pin to any supply
with a pull-up resistor that will supply less than 250µA.
Note that this pin will be open when the LT1940 is placed
in shutdown mode (both RUN/SS pins at ground) regardless of the voltage at the FB pin. Power good is valid when
the LT1940 is enabled (either RUN/SS pin is high) and VIN
is greater than ~2.4V.
Output Sequencing
The PG and RUN/SS pins can be used to sequence the two
outputs. Figure 5 shows several circuits to do this. In each
case channel 1 starts first. Note that these circuits sequence the outputs during start-up. When shut down the
two channels turn off simultaneously.
In Figure 5a, a larger capacitor on RUN/SS2 delays channel 2 with respect to channel 1. The soft-start capacitor on
RUN/SS2 should be at least twice the value of the capacitor
on RUN/SS1. A larger ratio may be required, depending on
the output capacitance and load on each channel. Make
sure to test the circuit in the system before deciding on
final values for these capacitors.
The circuit in Figure 5b requires the fewest components,
with both channels sharing a single soft-start capacitor.
The power good comparator of channel 1 disables channel
2 until output 1 is in regulation.
For independent control of channel 2, use the circuit in
Figure 5c. The capacitor on RUN/SS1 is smaller than the
capacitor on RUN/SS2. This allows the LT1940 to start up
and enable its power good comparator before RUN/SS2
gets high enough to allow channel 2 to start switching.
Channel 2 only operates when it is enabled with the
external control signals and output 1 is in regulation.
The circuit in Figure 5a leaves both power good indicates
free. However, the circuits in Figures 5b and 5c have
another advantage. As well as sequencing the two outputs
at start-up, they also disable channel 2 if output 1 falls out
of regulation (due to a short circuit or a collapsing input
voltage).
Finally, be aware that the circuit in Figure 5d does not
work, because the power good comparators are disabled
in shutdown. When the system is placed in shutdown
mode by pulling down on RUN/SS1, then output 1 will go
low, PG1 will pull down on RUN/SS2, and the LT1940 will
enter its low current shutdown state. This disables PG1,
and RUN/SS2 ramps up again to enable the LT1940. The
circuit will oscillate and pull extra current from the input.
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RUN/SS1
RUN/SS1
OFF ON
1nF
OFF ON
VC2
1nF
RUN/SS2
GND
RUN/SS2 PG1
GND
2.2nF
(5a) Channel 2 is Delayed
(5b) Fewest Components
RUN/SS1
OFF ON
RUN/SS1
OFF ON
1nF
OFF2 ON2
1nF
PG1
PG1
RUN/SS2
GND
RUN/SS2
GND
1.5nF
1.5nF
(5c) Independent Control of Channel 2
1940 F05
(5d) Doesn't Work !
Figure 5. Several Methods of Sequencing the Two Outputs. Channel 1 Starts First.
PARASITIC DIODE
D4
VIN
VIN
SW
VOUT
LT1940
1940 F06
Figure 6. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output.
Shorted Input Protection
PCB Layout
If the inductor is chosen so that it won’t saturate excessively, the LT1940 will tolerate a shorted output. There is
another situation to consider in systems where the output
will be held high when the input to the LT1940 is absent.
If the VIN and one of the RUN/SS pins are allowed to float,
then the LT1940’s internal circuitry will pull its quiescent
current through its SW pin. This is fine if your system can
tolerate a few mA of load in this state. With both RUN/SS
pins grounded, the LT1940 enters shutdown mode and
the SW pin current drops to ~30µA. However, if the VIN pin
is grounded while the output is held high, then parasitic
diodes inside the LT1940 can pull large currents from the
output through the SW pin and the VIN pin. A Schottky
diode in series with the input to the LT1940 will protect the
LT1940 and the system from a shorted or reversed input.
For proper operation and minimum EMI, care must be
taken during printed circuit board (PCB) layout. Figure 7
shows the high-di/dt paths in the buck regulator circuit.
Note that large, switched currents flow in the power
switch, the catch diode and the input capacitor. The loop
formed by these components should be as small as
possible. These components, along with the inductor and
output capacitor, should be placed on the same side of the
circuit board, and their connections should be made on
that layer. Place a local, unbroken ground plane below
these components, and tie this ground plane to system
ground at one location, ideally at the ground terminal of the
output capacitor C2. Additionally, the SW and BOOST
nodes should be kept as small as possible. Figure 8 shows
recommended component placement with trace and via
locations.
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VIN
VIN
SW
GND
SW
GND
(7a)
(7b)
VSW
VIN
IC1
C1
L1
SW
D1
GND
(7c)
C2
1940 F07
Figure 7. Subtracting the Current when the Switch is ON (a) From the Current when the Switch is OFF (b) Reveals the Path
of the High Frequency Switching Current (c) Keep This Loop Small. The Voltage on the SW and BOOST Nodes will also be
Switched; Keep these Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane.
Thermal Considerations
The PCB must also provide heat sinking to keep the
LT1940 cool. The exposed metal on the bottom of the
package must be soldered to a ground plane. This ground
should be tied to other copper layers below with thermal
vias; these layers will spread the heat dissipated by the
LT1940. Place additional vias near the catch diodes.
Adding more copper to the top and bottom layers and tying
this copper to the internal planes with vias can reduce
thermal resistance further. With these steps, the thermal
resistance from die (or junction) to ambient can be reduced to θJA = 45°C/W.
The power dissipation in the other power components—
catch diodes, boost diodes and inductors,␣ cause additional copper heating and can further increase what the IC
sees as ambient temperature. See the LT1767 data sheet’s
Thermal Considerations section.
Single, Low-Ripple 2.8A Output
VOUT1
VIA TO LOCAL GROUND PLANE
VIA TO VIN
GND
VOUT2
1940 F08
Figure 8. A Good PCB Layout Ensures Proper Low EMI Operation
The LT1940 can generate a single, low-ripple 2.8A output
if the outputs of the two switching regulators are tied
together and share a single output capacitor. By tying the
two FB pins together and the two VC pins together, the two
channels will share the load current. There are several
advantages to this two-phase buck regulator. Ripple currents at the input and output are reduced, reducing voltage
ripple and allowing the use of smaller, less expensive
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capacitors. Although two inductors are required, each will
be smaller than the inductor required for a single-phase
regulator. This may be important when there are tight
height restrictions on the circuit. The Typical Applications
section shows circuits with maximum heights of 1.4mm,
1.8mm and 2.1mm.
There is one special consideration regarding the twophase circuit. When the difference between the input
voltage and output voltage is less than 2.5V, then the boost
circuits may prevent the two channels from properly
sharing current. If, for example, channel 1 gets started
first, it can supply the load current, while channel 2 never
switches enough current to get its boost capacitor charged.
In this case, channel 1 will supply the load until it reaches
current limit, the output voltage drops, and channel 2 gets
started. The solution is to generate a boost supply generated from either SW pin that will service both BOOST pins.
The low profile, single output 5V to 3.3V converter shown
in the Typical Applications section shows how to do this.
Generating an Output Under 1.25V
The LT1940 regulates its feedback pins to 1.25V. Two
resistors can be used to program an output that is higher
than 1.25V. Generating an output voltage that is less than
the internal reference is generally more difficult, but the
LT1940 can easily generate an output voltage less than
1.25V if the other output is greater than 1.25V. Figure 9
shows how.
VOUT1, which must be greater than 1.25V, is used as a
reference voltage for the feedback divider from VOUT2 to
the FB2 pin (R3 and R4). Calculate the resistor values with
these equations:
R2/R1 = VOUT1/1.25V - 1
R4/R3 = (1.25V – VOUT2)/(VOUT1 – 1.25V)
R5 prevents the current through R3 and R4 from pulling
VOUT2 high when there is no load current.
R5 < (R3 + R4) VOUT2/(VOUT1 – VOUT2).
If VOUT1 is out of regulation (during start-up or if it is
overloaded or shorted) then VOUT2 will regulate to a higher
voltage than intended. To avoid this, the power good
output from the channel 1 (PG1) is tied to the compensation pin (VC2) of the channel 2. This disables channel 2
until VOUT1 is in regulation. Accuracy is good, especially
when R4/R3 is small.
For example, for VOUT1 = 3.3V and VOUT2 = 1.2V, choose
R1 = 10k, R2 = 16.5k, R3 = 10k, R4 = 243Ω and R5 = 4.7k.
Other Linear Technology Publications
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
note DN100 shows how to generate a dual (+ and –) output
supply using a buck regulator.
R3
VOUT1
SW1
LT1940
R2
FB1
R1
FB2
SW2
PG1 GND VC2
R4
VOUT2
R5
1940 F09
Figure 9. This circuit can be used when VOUT1 is greater than 1.25V and VOUT2 is less than 1.25V.
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3.3V and 1.8V Outputs with Sequencing
VIN
4.7V TO 14V
C3
4.7µF
D3
VIN
BOOST1
OUT1
1.8V
1.4A
L1
2.2µH
D4
LT1940
0.1µF
L2
3.3µH
0.1µF
Start-Up Waveforms
SW2
SW1
D1
VIN
2V/DIV
D2
10.0k
C1
22µF
OUT2
3.3V
1A
(1.4A FOR VIN > 5V)
BOOST2
16.5k
22.6k
20k
220pF
FB1
FB2
VC1
VC2
RUN/SS1
PG1
RUN/SS2
PG2
VOUT1
2V/DIV
15k
POWER GOOD
2V/DIV
330pF
GND
1nF
VOUT2
2V/DIV
C2
10µF
10.0k
100k
50µs/DIV
1940 TA01
D1, D2: MICROSEMI UPS120
D3, D4: CENTRAL CMDSH-3
L1: SUMIDA CR43-2R2
L2: SUMIDA CR43-3R3
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN JMK316BJ106ML
C3: TAIYO YUDEN EMK316BJ475ML
1940 TA01b
POWER
GOOD
5V/3.3V with Tantalum Output Capacitors
VIN
7V TO 25V
C3
4.7µF
BOOST1
L1
3.3µH
OUT1
3.3V
1.2A
D4
VIN
D3
LT1940
0.1µF
L2
4.7µH
0.1µF
SW2
SW1
D1
D2
16.5k
C1 +
100µF
6.3V
10.0k
OUT2
5V
1.2A
BOOST2
30.1k
FB1
FB2
VC1
VC2
PG1
PG2
+
10.0k
RUN/SS1 RUN/SS2
220pF
100k
20k
1nF
GND
10.0k
C2
47µF
10V
100pF
1nF
3V3
GOOD
100k
5 GOOD
1940 TA03
C1: AVX TPSC107M010R0150
C2: AVX TPSC476M010R0350
C3: TAIYO YUDEN TMK325BJ475ML
D1, D2: MICROSEMI UPS140 OR ON SEMI MBRM140
D3, D4: CENTRAL CMDSH-3
L1: SUMIDA CDRH4D28-3R3
L2: SUMIDA CDRH4D28-4R7
1940fa
17
LT1940/LT1940L
U
TYPICAL APPLICATIO S
3.3V, ±5V
D5
D3B
VIN
10V TO 25V
C3
4.7µF
C4
10µF
VIN
D3A
L1
3.3µH
LT1940
0.1µF
47k
BOOST2
BOOST1
OUT1
3.3V
1.4A
OUT3
–5V
300mA
0.1µF
1µF
L2
4.7µH
OUT2
5V
600mA
SW2
SW1
D1
D2
16.5k
C1
10µF
30.1k
15k
10.0k
FB1
FB2
VC1
VC2
PG1
PG2
20k
RUN/SS1 RUN/SS2
330pF
100k
GND
1nF
C2
10µF
10.0k
220pF
2.2nF
1940 TA05
PGOOD
–5V LOAD SHOULD BE
LESS THAN 1/2 5V LOAD
(SEE DESIGN NOTE 100).
D5: ON SEMI MBR0530
L1: SUMIDA CR43-3R3
L2: COILTRONICS CTX5-1A
C1, C2, C4: TAIYO YUDEN JMK316BJ106ML
C3: TAIYO YUDEN TMK325BJ475ML
D1, D2: MICROSEMI UPS140 OR ON SEMI MBRM140
D3: BAT-54A
Low Ripple, Low Profile 12V to 3.3V/2.4A
Maximum Height = 2.1mm
VIN
6V TO 16V
VIN
C3
4.7µF
RUN/SS1
1nF
D3A
BOOST1
0.1µF
RUN/SS2
SW1
PG1
100k
D1
PG2
PGOOD
L1
4.1µH
OUT2
3.3V
2.4A
LT1940
680pF
VC1
6.8k
D3B
BOOST2
VC2
0.1µF
FB1
L2
4.1µH
330pF
SW2
FB2
16.5k
GND
10.0k
D2
C1
22µF
1940 TA06
D1, D2: MICROSEMI UPS120
D3: BAT-54A
L1, L2: SUMIDA CDRH5D18-4R1
C1: TAIYO YUDEN JMK316BJ226ML
C3: TAIYO YUDEN EMK325BJ475MN
1940fa
18
LT1940/LT1940L
U
TYPICAL APPLICATIO S
TFT LCD Supply
D4B
19V REGULATED
INPUT
4.7µF
22µF
D4A
25V
3mA
C10
1µF
D6
VIN
D3
BOOST2
5V
1.2A
22Ω
+
L2
4.7µH
BOOST1
LT1940
0.1µF
0.1µF
0.022µF
L1
15µH
SW1
SW2
D5B
D1
D2
22Ω
13V
300mA
100k
30.1k
10k
FB2
FB1
VC2
VC1
D5A
10.7k
10k
PG2
20k
PG1
22Ω
10µF
10µF
220pF
RUN/SS2 RUN/SS1
1.5nF
GND
1nF
1nF 220pF
–5V
3mA
D7
1µF
1nF
1µF
1940 TA08
D1, D2: ON SEMI MBRM140T3
D3: CENTRAL CMPD914
D4, D5: BAT-54S
D6, D7: 6.2V ZENER
L1: SUMIDA CDRH4D28-4R7
L2: SUMIDA CDRH5D28-150
U
PACKAGE DESCRIPTIO
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BA
4.90 – 5.10*
(.193 – .201)
2.74
(.108)
2.74
(.108)
16 1514 13 12 1110
6.60 ±0.10
9
2.74
(.108)
4.50 ±0.10
SEE NOTE 4
2.74 6.40
(.108) BSC
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8
RECOMMENDED SOLDER PAD LAYOUT
1.10
(.0433)
MAX
4.30 – 4.50*
(.169 – .177)
0° – 8°
0.09 – 0.20
(.0036 – .0079)
0.45 – 0.75
(.018 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
0.05 – 0.15
(.002 – .006)
FE16 (BA) TSSOP 0203
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
1940fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT1940/LT1940L
U
TYPICAL APPLICATIO
Low Ripple, Low Profile 5V to 3.3V/2.4A Maximum Height = 1.4mm
0.47µF
VIN
4.8V TO 7V
VIN
C3
2.2µF
RUN/SS1
1nF
D3B
0.1µF
RUN/SS2
L1
3.3µH
SW1
PG1
100k
D1
PG2
PGOOD
D3A
BOOST1
OUT2
3.3V
2.4A
LT1940L
1500pF
VC1
4.7k
D4B
D4A
BOOST2
VC2
0.1µF
FB1
L2
3.3µH
330pF
SW2
FB2
16.5k
GND
D2
10k
C1
20µF
1940 TA07
D1, D2: MICROSEMI UPS120
D3, D4: BAT-54S
L1, L2: COILCRAFT LPO1704-332M
C1: 2X TAIYO YUDEN JMK212BJ106ML
C3: 2X TAIYO YUDEN EMK212BJ105MN
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1765
25V, 2.75A (IOUT), 1.25MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3V to 25V, VOUT(MIN): 1.20V, IQ: 1mA, ISD: 15µA
S8, TSSOP16E Package
LT1766
60V, 1.2A (IOUT), 200kHz,
High Efficiency Step-Down DC/DC Converter
VIN: 5.5V to 60V, VOUT(MIN): 1.2V, IQ: 2.5mA, ISD: 25µA
TSSOP16/TSSOP16E Package
LT1767
25V, 1.2A (IOUT), 1.25MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3V to 25V, VOUT(MIN): 1.2V, IQ: 1mA, ISD: 6µA
MS8, MS8E Package
LT1944
Dual Output 350mA ISW, Constant Off-Time,
High Efficiency Step-Up DC/DC Converter
VIN: 1.2V to 15V, VOUT(MAX): 34V, IQ: 20µA, ISD:
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