LT3505
1.2A, Step-Down
Switching Regulator in
3mm × 3mm DFN
FEATURES
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DESCRIPTION
Wide Input Range: 3.6V to 36V Operating,
40V Maximum
Up to 1.2A Output Current
Resistor-Programmable Fixed-Frequency Operation
from 200kHz to 3MHz
Output Adjustable Down to 780mV
Short-Circuit Robust
Uses Tiny Capacitors and Inductors
Soft-Start
Low Shutdown Current: 8V
BOOST
VIN
0.1µF 10µH
SW
SHDN
LT3505
36.5k
GND
75.0k
1µF
11.3k
VC
85
80
22pF
FB
RT
90
10µF
69.8k
68pF
3505 TA01
EFFICIENCY (%)
VIN
4.2V TO 36V
Efficiency
75
70
65
60 VIN = 12V
VOUT = 3.3V
55 fSW = 750kHz
L = 10 H
50
0.8
0
0.2
0.4
0.6
LOAD CURRENT (A)
1.0
1.2
3505fc
1
LT3505
ABSOLUTE MAXIMUM RATINGS
(Note 1)
Input Voltage (VIN) ....................................................40V
BOOST Pin Voltage ..................................................50V
BOOST Pin Above SW Pin.........................................25V
SHDN Pin ..................................................................40V
FB Pin .........................................................................6V
VC Pin .........................................................................3V
RT Pin .........................................................................3V
Operating Temperature Range (Note 2)
LT3505E .............................................. –40°C to 85°C
LT3505I ............................................. –40°C to 125°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range................... –65°C to 150°C
PIN CONFIGURATION
TOP VIEW
BOOST 1
SW 2
VIN 3
9
SHDN 4
8
VC
7
FB
6
RT
5
GND
TOP VIEW
BOOST
SW
VIN
SHDN
DD PACKAGE
8-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 5°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
1
2
3
4
9
8
7
6
5
VC
FB
RT
GND
MS8E PACKAGE
8-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/W, θJC = 5°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3505EDD#PBF
LT3505EDD#TRPBF
LCHB
8-Lead (3mm x 3mm) Plastic DFN
–40°C to 85°C
LT3505IDD#PBF
LT3505IDD#TRPBF
LCHC
8-Lead (3mm x 3mm) Plastic DFN
–40°C to 125°C
LT3505EMS8E#PBF
LT3505EMS8E#TRPBF
LTCNX
8-Lead Plastic MSOP
–40°C to 85°C
LT3505IMS8E#PBF
LT3505IMS8E#TRPBF
LTCNY
8-Lead Plastic MSOP
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3505EDD
LT3505EDD#TR
LCHB
8-Lead (3mm x 3mm) Plastic DFN
–40°C to 85°C
LT3505IDD
LT3505IDD#TR
LCHC
8-Lead (3mm x 3mm) Plastic DFN
–40°C to 125°C
LT3505EMS8E
LT3505EMS8E#TR
LTCNX
8-Lead Plastic MSOP
–40°C to 85°C
LT3505IMS8E
LT3505IMS8E#TR
LTCNY
8-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3505fc
2
LT3505
ELECTRICAL CHARACTERISTICS
The " denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2).
PARAMETER
CONDITIONS
MIN
VIN Operating Range
TYP
3.6
Undervoltage Lockout
Feedback Voltage
"
MAX
36
UNITS
V
3.1
3.35
3.6
V
765
780
795
mV
55
150
nA
FB Pin Bias Current
VFB = Measured VREF (Note 4)
Quiescent Current
Not Switching, RT = 75.0k
2.0
2.7
mA
Quiescent Current in Shutdown
VSHDN = 0V
0.01
2
µA
Reference Line Regulation
VIN = 5V to 36V
Switching Frequency
VFB = 0.7V, RT = 13.7k
VFB = 0.7V, RT = 75.0k
VFB = 0.7V, RT = 357k
Maximum Duty Cycle
RT = 75.0k
Error Amp Transconductance
"
0.007
%/V
3.30
825
220
MHz
kHz
kHz
2.70
675
180
3.01
750
200
90
94
%
VFB = 0.78V
200
µA/V
Error Amp Voltage Gain
VFB = 0.78V
400
V/V
VC Source Current
VFB = 0V, VC = 1.5V
10
µA
VC Sink Current
VFB = 1V, VC = 1.5V
14
µA
VC Switching Threshold Voltage
IOUT = 0mA
0.9
V
VC Clamp Voltage
VFB = 0V
1.7
V
RT Bias Voltage
VFB = 0.6V
VFB = 0V, RT = 75.0k
0.5
50
V
mV
Switch Current Limit
(Note 3)
Switch VCESAT
ISW = 1A
"
1.4
1.75
2.2
350
Switch Leakage Current
A
mV
0.1
2
µA
Minimum Boost Voltage Above Switch
ISW = 1A
1.6
2.2
V
BOOST Pin Current
ISW = 1A
24
50
mA
SHDN Input Voltage High
2.3
V
SHDN Input Voltage Low
SHDN Bias Current
VSHDN = 2.3V (Note 5)
VSHDN = 0V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3505E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LT3505I specifications are
guaranteed over the –40°C to 125°C temperature range.
6
0.01
0.3
V
20
0.1
µA
µA
Note 3: Current limit guaranteed by design and/or correlation to static test.
Slope compensation reduces current limit at higher duty cycle.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
3505fc
3
LT3505
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency (VOUT = 3.3V, L = 10µH,
fSW = 750kHz)
Efficiency (VOUT = 5V, L = 10µH,
fSW = 750kHz)
95
85
80
80
70
65
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
65
50
Efficiency (VOUT = 5V, L = 4.7µH,
fSW = 2.2MHz)
95
1.7
0.4
0.6
0.8
LOAD CURRENT (A)
OUTPUT CURRENT (A)
80
75
70
65
VIN = 8V
VIN = 12V
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
0
1.4
*10% DROPOUT
0.9
0.8
TYPICAL, L = 10µH
1.4
1.3
MINIMUM, L = 10µH
1.2
1.1
*10% DROPOUT
1.0
0.9
5
7
9
11
13
15
INPUT VOLTAGE (V)
0.8
19
17
6
8 10 12 14 16 18 20 22 24 26 28 30
INPUT VOLTAGE (V)
3505 G05
Max Load Current (VOUT = 3.3V,
L = 2.2µH, fSW = 2.2MHz)
1.6
1.80
1.4
1.3
MINIMUM
1.2
1.1
1.0
0.8
5
6
9
8
7
10
INPUT VOLTAGE (V)
400
1.50
TYPICAL
1.40
1.30
MINIMUM
1.20
1.10
1.00
*10% DROPOUT
0.9
450
1.60
TYPICAL
1.5
Switch Voltage Drop
500
TA = 25°C
1.70
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Max Load Current (VOUT = 5V,
L = 3.3µH, fSW = 2.2MHz)
TA = 25°C
1.7
3505 G06
12
3505 G07
0.80
7
8
9
10 11 12 14
INPUT VOLTAGE (V)
TA = 85°C
350
TA = 25°C
300
250
TA = –45°C
200
150
100
*10% DROPOUT
0.90
11
VCE(SWITCH) (mV)
1.8
1.2
1.0
TYPICAL, L = 22µH
1.5
1.2
1.1
0.4
0.6
0.8
LOAD CURRENT (A)
TA = 25°C
1.6
MINIMUM
1.3
0.2
Max Load Current (VOUT = 5V,
fSW = 750kHz)
TYPICAL
1.5
1.2
1.0
VIN = 8V
VIN = 12V
1.7
1.0
60
55
50
1.2
1.0
65
55
TA = 25°C
1.6
85
50
0.2
70
Max Load Current (VOUT = 3.3V,
L = 6.8µH, fSW = 750kHz)
TA = 25°C
90
0
75
60
VIN = 8V
VIN = 12V
VIN = 24V
55
1.2
1.0
70
OUTPUT CURRENT (A)
50
75
60
VIN = 8V
VIN = 12V
VIN = 24V
55
EFFICIENCY (%)
85
80
75
TA = 25°C
90
85
60
EFFICIENCY (%)
95
TA = 25°C
90
EFFICIENCY (%)
EFFICIENCY (%)
95
TA = 25°C
90
Efficiency (VOUT = 3.3V,
L = 4.7µH, fSW = 2.2MHz)
50
16
18
3505 G08
0
0
300
900
1200
600
SWITCH CURRENT (mA)
1500
3505 G09
3505fc
4
LT3505
TYPICAL PERFORMANCE CHARACTERISTICS
Switching Frequency
3.90
2.20
SWITCHING FREQUENCY (MHz)
2.40
3.80
UVLO (V)
3.70
3.60
3.50
3.40
3.30
Frequency Foldback, RT = 75.0k
0.6
RT = 21k
2.00
1.80
1.60
RT = 30.1k
1.40
1.20
1.00
3.20
3.00
–50
–25
75
0
25
50
TEMPERATURE (°C)
100
0.60
–50
125
50
75
0
25
TEMPERATURE (°C)
–25
0.4
0.3
0.2
0.1
RT = 75.0k
0.80
3.10
TA = 25°C
0.5
RT PIN BIAS VOLTAGE (V)
Undervoltage Lockout
4.00
0
125
100
0
0.1
0.2
0.3 0.4 0.5 0.6
FB VOLTAGE (V)
3505 G12
3505 G10
Soft-Start
2.0
50
7.2
TA = 25°C
45
6.8
35
6.6
1.2
30
0.8
INPUT VOLTAGE (V)
40
1.4
1.0
25
20
6.2
6.0
15
0.4
10
5.6
0.2
5
5.4
0
0
0.25 0.50 0.75 1 1.25 1.50 1.75
SHDN PIN VOLTAGE (V)
2
0
2
4
6
TO RUN
5.8
5.2
8 10 12 14 16 18 20
VSHDN (V)
TO START
6.4
0.6
0
TA = 25°C
7.0
1.6
ISHDN (µA)
SWITCH CURRENT LIMIT (A)
Typical Minimum Input Voltage,
(VOUT = 5V, fSW = 750kHz)
SHDN Pin Current
TA = 25°C
1.8
1
10
100
LOAD CURRENT (mA)
Typical Minimum Input Voltage,
(VOUT = 3.3V, fSW = 750kHz)
6.8
TO START
INPUT VOLTAGE (V)
4.9
4.7
4.5
4.3
4.1
7.0
TA = 25°C
5.5
TO START
5.1
TO RUN
5.0
4.5
TO START
6.6
TO RUN
4.0
3.9
6.4
TO RUN
6.2
6.0
5.8
5.6
5.4
5.2
3.7
3.5
Typical Minimum Input Voltage,
(VOUT = 5V, fSW = 2.2MHz)
INPUT VOLTAGE (V)
5.3
INPUT VOLTAGE (V)
Typical Minimum Input Voltage,
(VOUT = 3.3V, fSW = 2.2MHz)
TA = 25°C
1
10
100
LOAD CURRENT (mA)
1000
3.5
1000
3505 G15
3505 G14
5.5
0.8
0.7
1
10
100
LOAD CURRENT (mA)
1000
5.0
TA = 25°C
1
10
100
LOAD CURRENT (mA)
1000
3505fc
5
LT3505
TYPICAL PERFORMANCE CHARACTERISTICS
Switch Current Limit
Switch Current Limit, RT = 75.0k
1.8
1.8
SWITCH CURRENT LIMIT (A)
SWITCH CURRENT LIMIT (A)
1.9
1.7
1.6
1.5
1.4
1.3
1.2
Typical Minimum On Time
160
TA = 25°C
140
1.7
MINIMUM ON TIME (ns)
2.0
1.6
1.5
1.4
1.3
–25
0
25
50
75
TEMPERATURE (°C)
100
1.2
125
100
80
60
40
20
1.1
1.0
–50
120
0
0
–50
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
–25
50
75
0
25
TEMPERATURE (°C)
100
125
3505 G20
RT Pin Bias Voltage
Switching Frequency
3.0
VFB = 0.78V
SWITCHING FREQUENCY (MHz)
BIAS VOLTAGE (mV)
500
495
490
485
480
–50
–25
50
25
75
0
TEMPERATURE(°C)
100
125
Switching Frequency
TA = 25°C
TA = 25°C
2.5
SWITCHING FREQUENCY (MHz)
505
2.0
1.5
1.0
0.5
0
0
5
10
15
30
25
20
RT PIN BIAS CURRENT (µA)
35
1
0.1
10
100
RT PIN RESISTANCE (kΩ)
3505 G24
Operating Waveforms,
Discontinuous Mode
Operating Waveforms
VSW
5V/DIV
VSW
5V/DIV
IL
0.5A/DIV
0
VOUT
20mV/DIV
IL
0.5A/DIV
0
VOUT
20mV/DIV
VIN = 12V
VOUT = 3.3V
IOUT = 0.5A
L = 10µH
COUT = 10µF
RT = 75.0k
1µs/DIV
3505 G18
VIN = 12V
VOUT = 3.3V
IOUT = 50mA
L = 10µH
COUT = 10µF
RT = 75.0k
1µs/DIV
3505 F26
3505fc
6
LT3505
PIN FUNCTIONS
BOOST (Pin 1): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 2): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
VIN (Pin 3): The VIN pin supplies current to the LT3505’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
SHDN (Pin 4): The SHDN pin is used to put the LT3505 in
shutdown mode. Tie to ground to shut down the LT3505.
Tie to 2.3V or more for normal operation. If the shutdown
feature is not used, tie this pin to the VIN pin. SHDN also
provides a soft-start function; see the Applications Information section.
GND (Pin 5): Tie the GND pin to a local ground plane
below the LT3505 and the circuit components. Return the
feedback divider to this pin.
RT (Pin 6): The RT pin is used to program the switching
frequency of the LT3505 by connecting a resistor from
this pin to ground. The Applications Information section of
the data sheet includes a table to determine the resistance
value based on the desired switching frequency. Minimize
capacitance at this pin.
FB (Pin 7): The LT3505 regulates its feedback pin to 780mV.
Connect the feedback resistor divider tap to this pin. Set
the output voltage by selecting R1 according to:
V
R1 = R2 OUT – 1
0.78 V
A good value for R2 is 10.0k.
VC (Pin 8): The VC pin is used to compensate the LT3505
control loop by tying an external RC network from this
pin to ground.
Exposed Pad (Pin 9): The Exposed Pad must be soldered
to the PCB and electrically connected to ground. Use a
large ground plane and thermal vias to optimize thermal
performance.
3505fc
7
LT3505
BLOCK DIAGRAM
3
VIN
VIN
C2
INT REG
AND
UVLO
ON OFF
SLOPE
COMP
R3
4
BOOST
Σ
R
Q
S
Q
D2
1
SHDN
C4
C3
DRIVER
Q1
SW
OSC
L1
VOUT
2
D1
C1
FREQUENCY
FOLDBACK
VC
gm
780mV
5
GND
8
VC
7
FB
6
RT
R1
3505 BD
R2
OPERATION
(Refer to Block Diagram)
The LT3505 is a constant frequency, current mode stepdown regulator. A resistor-programmed oscillator enables
an RS flip-flop, turning on the internal 1.4A power switch
Q1. An amplifier and comparator monitor the current
flowing between the VIN and SW pins, turning the switch
off when this current reaches a level determined by the
voltage at the VC pin. An error amplifier measures the
output voltage through an external resistor divider tied to
the FB pin and servos the VC node. If the error amplifier’s
output increases, more current is delivered to the output;
if it decreases, less current is delivered. An active clamp
(not shown) on the VC node provides current limit. The
VC node is also clamped to the voltage on the SHDN pin;
soft-start is implemented by generating a voltage ramp at
the SHDN pin using an external resistor and capacitor.
An internal regulator provides power to the control circuitry.
This regulator includes an undervoltage lockout to prevent
switching when VIN is less than ~3.4V. The SHDN pin is
used to place the LT3505 in shutdown, disconnecting the
output and reducing the input current to less than 2µA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for efficient operation.
When the FB pin is low, the voltage at the RT pin decreases
to reduce the oscillator frequency. This frequency foldback
helps to control the output current during start-up and
overload.
3505fc
8
LT3505
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
V
R1 = R2 OUT – 1
0.78 V
R2 should be 20k or less to avoid bias current errors.
Reference designators refer to the Block Diagram.
Input Voltage Range
The input voltage range for LT3505 applications depends
on the output voltage, on the absolute maximum ratings
of the VIN and BOOST pins, and on the programmed
switching frequency.
The minimum input voltage is determined by either the
LT3505’s minimum operating voltage of 3.6V, or by its
maximum duty cycle. The duty cycle is the fraction of
time that the internal switch is on and is determined by
the input and output voltages:
VOUT + VD
DC =
VIN – VSW + VD
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
VIN(MIN) =
VOUT + VD
– VD + VSW
DCMAX
with DCMAX = 1 – fSW/8.33, where fSW is in MHz.
The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins. For
constant-frequency operation, the maximum input voltage
is determined by the minimum duty cycle requirement.
As the input voltage increases, the required duty cycle
to regulate the output voltage decreases. The minimum
duty-cycle is:
DCMIN = fSW • tON(MIN)
where fSW is the switching frequency in hertz and tON(MIN) is
the worst-case minimum on-time in seconds. The minimum
on-time of the LT3505 is a strong function of temperature.
The typical performance characteristics section of the
datasheet contains a graph of minimum on-time versus
temperature to help determine the worst-case minimum
on-time for the intended application.
If the input voltage is high enough that the duty-cycle
requirement is lower than DCMIN, the part enters pulseskipping mode. Specifically, the onset of pulse-skipping
occurs at:
VIN(PS) = (VOUT + VD) / DCMIN – VD + VSW
Above VIN(PS) the part turns on for brief periods of time
to control the inductor current and regulate the output
voltage, possibly producing a spectrum of frequencies
below the programmed switching frequency. To remain
in constant-frequency operation the input voltage should
remain below VIN(PS). See the “Minimum On Time” section of the data sheet for more information on operating
above VIN(PS).
Note that this is a restriction on the operating input voltage
to remain in constant-frequency operation; the circuit will
tolerate brief transient inputs up to the absolute maximum
ratings of the VIN and BOOST pins when the output is in
regulation. The input voltage should be limited to VIN(PS)
during overload conditions (short-circuit or start-up).
Minimum On Time
For switching frequencies less than 750kHz, the part
will still regulate the output at input voltages that exceed
VIN(PS) (up to 40V), however, the output voltage ripple
increases as the input voltage is increased. Figure 1 illustrates switching waveforms in continuous mode for a
3V output application near VIN(PS) = 33V.
As the input voltage is increased, the part is required to
switch for shorter periods of time. Delays associated with
turning off the power switch determine the minimum on
time of the part. The worst-case typical minimum on-time
is 130ns. Figure 2 illustrates the switching waveforms
when the input voltage is increased to VIN = 35V.
3505fc
9
LT3505
APPLICATIONS INFORMATION
Now the required on time has decreased below the
minimum on time of 130ns. Instead of the switch pulse
width becoming narrower to accommodate the lower duty
cycle requirement, the switch pulse width remains fixed
at 130ns. In Figure 2 the inductor current ramps up to a
value exceeding the load current and the output ripple
increases to ~200mV. The part then remains off until the
output voltage dips below 100% of the programmed value
before it begins switching again.
VSW
20V/DIV
IL
0.5A/DIV
VOUT
200mV/DIV
AC COUPLED
2 s/DIV
COUT = 10 F ILOAD = 0.75A
VOUT = 3V
L = 10 H
RT = 75.0k
VIN = 30V
3505 F01
Figure 1
VSW
20V/DIV
IL
0.5A/DIV
VOUT
200mV/DIV
AC COUPLED
2 s/DIV
COUT = 10 F
VOUT = 3V
VIN = 35V
3505 F02
ILOAD = 0.75A
L = 10 H
RT = 75.0k
Figure 2
For switching frequencies above 750kHz, the input voltage
must not exceed VIN(PS). See the “Input Voltage Frequency
Foldback” section of the datasheet for a circuit solution
that provides safe operation above VIN(PS) at switching
frequencies exceeding 750kHz. For switching frequencies
below 750kHz, operation above VIN(PS) is safe and will
not damage the part as long as the output voltage stays
in regulation and the inductor does not saturate. Figure
3 shows the switching waveforms of a 750kHz application when the input voltage is increased to its absolute
maximum rating of 40V.
As the input voltage increases, the inductor current ramp
rate increases, the number of skipped pulses increases
and the output voltage ripple increases. The part is robust
enough to survive prolonged operation under these conditions as long as the programmed switching frequency is
less than 750kHz and the peak inductor current does not
exceed 2.2A. Inductor current saturation may further limit
performance in this operating regime.
Frequency Selection
The maximum frequency that the LT3505 can be programmed to is 3MHz. The minimum frequency that the
LT3505 can be programmed to is 200kHz. The switching
frequency is programmed by tying a 1% resistor from the RT
pin to ground. Table 1 can be used to select the value of RT.
Minimum on-time and edge loss must be taken into consideration when selecting the intended frequency of operation.
Higher switching frequency increases power dissipation
and lowers efficiency.
VSW
20V/DIV
IL
0.5A/DIV
VOUT
200mV/DIV
AC COUPLED
COUT = 10µF
VOUT = 3V
VIN = 40V
2µs/DIV
ILOAD = 0.75A
L = 10µH
RT = 75.0k
3505 F03
Figure 3
3505fc
10
LT3505
APPLICATIONS INFORMATION
Finite transition time results in a small amount of power
dissipation each time the power switch turns on and off
(edge loss). Edge loss increases with frequency, switch
current, and input voltage.
Input Voltage Frequency Foldback
In constant frequency operation (below VIN(PS)) edge
loss only reduces the application efficiency. However, at
high switching frequencies exceeding 750kHz and input
voltages exceeding VIN(PS), the part operates in pulse-skipping mode and the switch current can increase above the
current limit of the part, 1.75A. This further increases the
power dissipated during switch transitions and increases
die temperature. To remedy the situation a single resistor (R5) and a zener diode (D3) can be added to a typical
LT3505 circuit as shown in Figure 4.
Although the circuit can be operated indefinitely above
VZENER, this frequency foldback method is intended to
protect circuits during temporary periods of high input
voltage. For example, in many automotive systems, the
normal operating input range might be 9V to 16V, and
the LT3505 can be programmed to operate above the
AM band (>1.8MHz). At the same time, the circuit must
be able to withstand higher input voltages due to load
dump or double-battery jump starts. During these brief
periods, it is usually acceptable to switch at a frequency
within the AM band.
If the output is shorted while the input voltage is greater than
VZENER, the switching frequency will be reduced to 30kHz
and the part will not be able to recover from the short until
the input voltage is reduced below VZENER (see the following
discussion).
2.50
D2
1N4148
VIN
6.7V TO 36V
VIN
BOOST
LT3505
16V
R5
806k
GND
R4
20.0k
C2
1µF
L1
6.8µH
R1
61.9k
D1
MBRM140
R2
11.5k
FB
RT
VOUT
5V
C3
0.1µF
SW
SHDN
ON OFF
D3
BZT52C16T
When the input voltage is below 16V, the zener diode
path conducts no current and the current flowing out
of the RT pin (and through R4) is nominally 0.5V/20k =
25µA, which programs a 2.2MHz switching frequency.
As the input voltage is increased above 16V, the zener
diode begins to conduct and gradually reduces the current flowing out of the RT pin. This mechanism reduces
the switching frequency as the input voltage is increased
above 16V (up to 36V) to ensure that the part constantly
operates in continuous mode without skipping pulses,
thereby preventing the excessive die temperature rise
encountered in pulse-skipping mode.
VC
C5
22pF
C1
10µF
R3
100k
C4
22pF
Frequency [MHz] / Load Current [A]
Finite transistor bandwidth limits the speed at which the
power switch can be turned on and off, effectively setting
the minimum on-time of the LT3505. For a given output voltage, the minimum on-time determines the maximum input
voltage to remain in continuous mode operation, VIN(PS).
See the “Input Voltage Range” section of the datasheet for
more information on determining VIN(PS). For switching
frequencies below 750kHz, operation above VIN(PS) (up
to 40V) is safe provided that the system will tolerate the
pulse-skipping behavior outlined in the “Minimum On
Time” section of the datasheet. At switching frequencies
exceeding 750kHz, edge loss limits operation to input
voltages below VIN(PS).
2.25
2.00
1.75
1.50
1.25
1.00
0.75
Switching
Frequency
0.50
Maximum
Load Current
0.25
0
0
5
3505 F04
10
15 20 25 30
Input Voltage [V]
35
40
LTC3505 • F04b
Figure 4. 2.2MHz, 5V Application with Input Voltage Frequency Foldback Circuit
3505fc
11
LT3505
APPLICATIONS INFORMATION
Component Selection for Input Voltage Frequency
Foldback Circuit
To determine the values of R4, R5, and D3 for a specific
application follow the procedure outlined in this section.
First select the value of R4 from Table 1.
Table 1. RT Pin Resistance
RT PIN RESISTANCE (kΩ)
357
237
165
124
100
84.5
71.5
61.9
54.9
48.7
44.2
40.2
37.4
34.0
31.6
29.4
27.4
25.5
23.7
22.6
21.0
20.0
19.1
18.2
16.9
16.2
15.4
14.7
13.7
SWITCHING FREQUENCY (MHz)
0.20
0.30
0.40
0.50
0.60
0.69
0.80
0.91
1.00
1.11
1.21
1.31
1.39
1.50
1.60
1.70
1.80
1.90
2.02
2.10
2.22
2.31
2.39
2.48
2.62
2.71
2.81
2.90
3.01
Second, determine the value of VIN(PS) from the equation
in the “Input Voltage Range” section of the data sheet.
Select the zener diode, D3, to have a breakdown voltage
(VZENER) below VIN(PS). Next determine the desired foldback
frequency from the following equation:
fSW(MIN) = (VOUT + VD)/[tON(MIN) • (VIN(MAX) + VD – VSW)]
where VD is the forward drop of the catch diode (~0.4V),
and VSW is the voltage drop of the internal power switch
(~0.4V at maximum load), VIN(MAX) is the maximum input
voltage for the application (must be less than 36V), and
tON(MIN) is the worst-case minimum on-time for the intended application. The worst-case minimum on-time can
be determined from the graphs in the “Typical Performance
Characteristics” section of the datasheet. Next look up the
resistance that corresponds to fSW(MIN) in Table 1. This
resistance is RT(MAX), the effective resistance from the RT
pin to ground at VIN(MAX) that programs the oscillator to
a switching frequency equal to fSW(MIN).
Finally determine R5 from the following equation:
R5 = 2 • (VIN(MAX) – VZENER)/(1/R4 – 1/RT(MAX))
where VZENER is the zener diode breakdown voltage,
and VIN(MAX) is the maximum input voltage that will be
applied to the VIN pin. VIN(MAX) must not exceed 36V, the
maximum operating input voltage of the LT3505. The
equation to determine R5 assumes that R5 will compensate a percentage of the current flowing through R4
equal to R4/RT(MAX). Be careful not to select a value of
R5 much less than that determined by the equation above
because it may become possible for R5 to compensate
100% of the current flowing through R4 reducing the
frequency to 30kHz. In this state the part is not able to
start into large output current loads.
Whenever the voltage at the FB pin is below 600mV, the
LT3505 folds back the switching frequency by reducing
the bias voltage at the RT pin. If the input voltage is higher
than the zener voltage, the reduced voltage at the RT pin
results in a larger voltage drop across R5, and a reduced
voltage drop across R4. The current carried by R5 may
be large enough to completely compensate the current
flowing through R4, reducing the frequency to 30kHz. In
this situation the input voltage will have to be reduced until
the input voltage is less than the zener voltage.
Note that when VIN is above VZENER and the frequency is
reduced, the inductor ripple current will be higher and the
maximum load that the LT3505 can regulate will be lower.
See the Inductor Selection and Maximum Output Current
section of this data sheet for more information.
3505fc
12
LT3505
APPLICATIONS INFORMATION
Inductor Selection and Maximum Output Current
Catch Diode
A good first choice for the inductor value is:
Depending on load current, a 1A to 2A Schottky diode is
recommended for the catch diode, D1. The diode must
have a reverse voltage rating equal to or greater than the
maximum input voltage. The ON Semiconductor MBRM140
is a good choice; it is rated for 1A continuous forward
current and a maximum reverse voltage of 40V.
L = 1.2 (VOUT + VD)/fSW
where VD is the voltage drop of the catch diode (~0.4V),
L is in µH and fSW is in MHz. With this value there will
be no subharmonic oscillation for applications with 50%
or greater duty cycle. The inductor’s RMS current rating
must be greater than your maximum load current and
its saturation current should be about 30% higher. For
robust operation in fault conditions, the saturation current
should be above 2.2A. To keep efficiency high, the series
resistance (DCR) should be less than 0.1 . Table 2 lists
several vendors and types that are suitable.
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A
larger value provides a higher maximum load current and
reduces output voltage ripple at the expense of slower
transient response. If your load is lower than 1.2A, then
you can decrease the value of the inductor and operate
with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting
in higher efficiency. There are several graphs in the Typical
Performance Characteristics section of this data sheet that
show the maximum load current as a function of input
voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode
operation, which is okay, but further reduces maximum
load current. For details on maximum output current and
discontinuous mode operation, see Linear Technology
Application Note 44.
Input Capacitor
The input of the LT3505 circuit must be bypassed with a
X7R or X5R type ceramic capacitor. Y5V types have poor
performance over temperature and applied voltage and
should not be used. For switching frequencies higher than
750kHz, bypass the input with a 1µF or higher value ceramic
capacitor. For switching frequencies below 750kHz, bypass
the input with a 2.2µF or higher value ceramic capacitor.
If the input power source has high impedance, or there is
significant inductance due to long wires or cables, additional
bulk capacitance may be necessary. This can be provided
with a low performance electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3505 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
To accomplish this task, the input bypass capacitor must
be placed close to the LT3505 and the catch diode; see
the PCB Layout section. A second precaution regarding
the ceramic input capacitor concerns the maximum input
voltage rating of the LT3505. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (underdamped) tank circuit. If the LT3505 circuit
is plugged into a live supply, the input voltage can ring to
Table 2. Inductor Vendors
VENDOR
URL
PART SERIES
INDUCTANCE RANGE (µH)
Size (mm)
Sumida
www.sumida.com
CDRH4D28
CDRH5D28
CDRH5D28
1.2 to 4.7
2.5 to 10
2.5 to 33
4.5 × 4.5
5.5 × 5.5
8.3 × 8.3
Toko
www.toko.com
A916CY
D585LC
2 to 12
1.1 to 39
6.3 × 6.2
8.1 × 8.0
Würth Elektronik
www.we-online.com
WE-TPC(M)
WE-PD2(M)
WE-PD(S)
1 to 10
2.2 to 22
1 to 27
4.8 × 4.8
5.2 × 5.8
7.3 × 7.3
3505fc
13
LT3505
APPLICATIONS INFORMATION
twice its nominal value, possibly exceeding the LT3505’s
voltage rating. This situation can be easily avoided; see
the Hot Plugging Safely section.
Output Capacitor
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by
the LT3505 to produce the DC output. In this role it determines the output ripple so low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3505’s control loop.
Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance.
A good value is:
COUT = 49/(VOUT • fSW)
where COUT is in µF and fSW is in MHz. Use X5R or X7R
types and keep in mind that a ceramic capacitor biased
with VOUT will have less than its nominal capacitance. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with a
high value capacitor, if the compensation network is also
adjusted to maintain the loop bandwidth.
A lower value of output capacitor can be used, but transient performance will suffer unless the compensation
network is adjusted to reduce the loop gain. Also, a lower
value output capacitor may result in increased sensitivity
to noise which can be alleviated by adding a 22pF phase
lead capacitor from FB to VOUT.
High performance electrolytic capacitors can be used for
the output capacitor. Low ESR is important, so choose one
that is intended for use in switching regulators. The ESR
should be specified by the supplier and should be 0.1Ω
or less. Such a capacitor will be larger than a ceramic
capacitor and will have a larger capacitance, because the
capacitor must be large to achieve low ESR. Table 3 lists
several capacitor vendors.
Figure 5 shows the transient response of the LT3505 with
several output capacitor choices. The output is 3.3V. The
load current is stepped from 500mA to 1.2A and back
to 500mA and the oscilloscope traces show the output
voltage. The upper photo shows the recommended value.
The second photo shows the improved response (less
voltage drop) resulting from a larger output capacitor
and a larger phase lead capacitor. The last photo shows
the response to a high performance electrolytic capacitor. Transient performance is improved due to the large
output capacitance.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 6 shows two
ways to arrange the boost circuit. The BOOST pin must
be at least 2.3V above the SW pin for best efficiency. For
outputs of 3.3V and above, the standard circuit (Figure 6a)
is best. For outputs between 3V and 3.3V, use a 0.22µF
capacitor. For outputs between 2.5V and 3V, use a 0.47µF
Table 3. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
COMMENTS
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
Tantalum
EEF Series
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
T494, T495
Sanyo
(408) 749-9714
www.sanyovideo.com
Ceramic,
Polymer,
Tantalum
POSCAP
Murata
(404) 436-1300
AVX
Taiyo Yuden
(864) 963-6300
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
www.taiyo-yuden.com
Ceramic
TPS Series
3505fc
14
LT3505
APPLICATIONS INFORMATION
ILOAD
1A/DIV
VOUT
32.4k
22pF
FB
10µF
VC
VOUT
20mV/DIV
AC COUPLED
10.0k
100k
22pF
100k
10µs/DIV
3505 F05b
10µs/DIV
3505 F05c
44pF
10µF
×2
FB
VC
3505 F05a
ILOAD
1A/DIV
VOUT
32.4k
10µs/DIV
VOUT
20mV/DIV
AC COUPLED
10.0k
22pF
ILOAD
1A/DIV
VOUT
32.4k
66pF
+
FB
VC
301k
120µF
10.0k
KEMET
A700D127M006ATE015
22pF
VOUT
20mV/DIV
AC COUPLED
Figure 5. Transient Load Response of the LT3505 with Different Output Capacitors as the
Load Current is Stepped from 500mA to 1.2A. VIN = 12V, VOUT = 3.3V, L = 2µH, RT = 20.0k
D2
D2
C3
BOOST
VIN
VIN
C3
BOOST
LT3505
LT3505
SW
VOUT
VIN
VIN
SW
VOUT
GND
GND
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
3505 F06a
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
3505 F06b
(6b)
(6a)
Figure 6. Two Circuits for Generating the Boost Voltage
3505fc
15
LT3505
APPLICATIONS INFORMATION
capacitor and a small Schottky diode (such as the BAT-54).
For lower output voltages tie a Schottky diode to the input
(Figure 6b). The circuit in Figure 6a is more efficient because
the BOOST pin current comes from a lower voltage source.
You must also be sure that the maximum voltage rating
of the BOOST pin is not exceeded.
The minimum operating voltage of an LT3505 application is limited by the undervoltage lockout (3.6V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by
the boost circuit. If the input voltage is ramped slowly,
or the LT3505 is turned on with its SHDN pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on the input and output voltages and on the arrangement
of the boost circuit. The minimum load generally goes to
zero once the circuit has started. Figure 7 shows a plot of
minimum load to start and to run as a function of input
voltage. In many cases the discharged output capacitor
will present a load to the switcher which will allow it to
start. The plots show the worst-case situation where VIN
is ramping verly slowly. For lower start-up voltage, the
boost diode can be tied to VIN; however this restricts the
input range to one-half of the absolute maximum rating
7.2
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
400mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3505, requiring a higher
input voltage to maintain regulation.
Soft-Start
The SHDN pin can be used to soft-start the LT3505, reducing
the maximum input current during start-up. The SHDN pin
is driven through an external RC filter to create a voltage
ramp at this pin. Figure 8 shows the start-up waveforms
with and without the soft-start circuit. By choosing a large
RC time constant, the peak start up current can be reduced
to the current that is required to regulate the output, with
no overshoot. Choose the value of the resistor so that it
can supply 20µA when the SHDN pin reaches 2.3V.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT3505 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3505 is absent. This may occur in battery charging ap-
5.5
TA = 25°C
7.0
INPUT VOLTAGE (V)
6.4
6.2
6.0
TO RUN
4.9
4.7
4.5
4.3
4.1
5.6
3.9
5.4
3.7
5.2
1
TO START
5.1
6.6
5.8
TA = 25°C
5.3
TO START
6.8
INPUT VOLTAGE (V)
of the BOOST pin.
10
100
LOAD CURRENT (mA)
3.5
1000
TO RUN
1
10
100
LOAD CURRENT (mA)
1000
3505 G15
(7a) Typical Minimum Input Voltage, VOUT = 5V, fSW = 750kHz
(7b) Typical Minimum Input Voltage, VOUT = 3.3V, fSW = 750kHz
Figure 7
3505fc
16
LT3505
APPLICATIONS INFORMATION
RUN
plications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT3505’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied
to VIN), then the LT3505’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT3505 can
pull large currents from the output through the SW pin
and the VIN pin. Figure 9 shows a circuit that will run only
when the input voltage is present and that protects against
a shorted or reversed input.
SHDN
GND
VSW
5V/DIV
IL
1A/DIV
VOUT
2V/DIV
VIN = 12V
VOUT = 3.3V
L = 2.5 H
COUT = 10 F
RT = 20.0k
10 s/DIV
3505 F08a
D4
VIN
VOUT
VIN BOOST SW
LT3505
RUN
SHDN
15k
RT
FB
GND
VC
BACKUP
SHDN
0.068 F
GND
3505 F09
Figure 9. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT3505 Runs Only When the Input
is Present
VSW
5V/DIV
Hot Plugging Safely
IL
1A/DIV
VOUT
2V/DIV
VIN = 12V
VOUT = 3.3V
L = 2.5 H
COUT = 10 F
RT = 20.0k
10 s/DIV
3505 F08b
Figure 8. To Soft-Start the LT3505, Add a Resistor and Capacitor
to the SHDN pin. VIN = 12V, VOUT = 3.3V, COUT = 10µF, RLOAD =
5Ω, RT = 20.0k, L = 2.5µH
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3505 circuits. However, these capacitors can cause problems if the LT3505 is plugged into
a live supply (see Linear Technology Application Note 88
for a complete discussion). The low loss ceramic capacitor
combined with stray inductance in series with the power
source forms an underdamped tank circuit and the voltage
at the VIN pin of the LT3505 can ring to twice the nominal
input voltage, possibly exceeding the LT3505’s rating and
3505fc
17
LT3505
APPLICATIONS INFORMATION
damaging the part. If the input supply is poorly controlled
or the user will be plugging the LT3505 into an energized
supply, the input network should be designed to prevent
this overshoot.
Figure 10 shows the waveforms that result when an LT3505
circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The first plot is the response with
a 2.2µF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 9b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and
can slightly improve the efficiency of the circuit, though
it is likely to be the largest component in the circuit. An
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
+
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
+
LT3505
alternative solution is shown in Figure 9c. A 1Ω resistor
is added in series with the input to eliminate the voltage
overshoot (it also reduces the peak input current). A 0.1µF
capacitor improves high frequency filtering. This solution is
smaller and less expensive than the electrolytic capacitor.
For high input voltages its impact on efficiency is minor,
reducing efficiency only one percent for a 5V output at full
load operating from 24V.
Frequency Compensation
The LT3505 uses current mode control to regulate the
output. This simplifies loop compensation. In particular,
the LT3505 does not require the ESR of the output capacitor for stability allowing the use of ceramic capacitors to
achieve low output ripple and small circuit size.
Frequency compensation is provided by the components
tied to the VC pin, as shown in Figure 10. Generally a
VIN
20V/DIV
2.2µF
20µs/DIV
(9a)
LT3505
+
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM
RATING OF THE LT3505
IIN
5A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
10µF
35V
AI.EI.
DANGER!
VIN
20V/DIV
2.2µF
IIN
5A/DIV
(9b)
1Ω
+
0.1µF
LT3505
20µs/DIV
VIN
20V/DIV
2.2µF
IIN
5A/DIV
(9c)
20µs/DIV
3505 F10
Figure 10. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT3505 is Connected to a Live Supply
3505fc
18
LT3505
APPLICATIONS INFORMATION
Figure 11 shows an equivalent circuit for the LT3505 control
loop. The error amp is a transconductance amplifier with
finite output impedance. The power section, consisting of
the modulator, power switch and inductor, is modeled as
a transconductance amplifier generating an output current proportional to the voltage at the VC node. Note that
the output capacitor integrates this current and that the
capacitor on the VC node (CC) integrates the error amplifier output current, resulting in two poles in the loop. RC
provides a zero. With the recommended output capacitor,
the loop crossover occurs above the RCCC zero. This simple
model works well as long as the value of the inductor is
not too high and the loop crossover frequency is much
lower than the switching frequency. With a larger ceramic
capacitor (very low ESR), crossover may be lower and a
phase lead capacitor (CPL) across the feedback divider may
improve the phase margin and transient response. Large
electrolytic capacitors may have an ESR large enough to
create an additional zero and the phase lead may not be
necessary.
If the output capacitor is different than the recommended
capacitor, stability should be checked across all operating conditions, including load current, input voltage and
temperature.
–
0.8V
OUT
CPL
R1
FB
–
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stability
using a transient load.
CURRENT MODE
POWER STAGE
SW
gm =
+1.1A/V
LT3505
gm =
200µA/V
VC
ERROR
AMPLIFIER
2M
+
capacitor (CC) and a resistor (RC) in series to ground are
used. In addition, a lower value filter capacitor (CF) may be
added in parallel. The filter capacitor is not a part of the loop
compensation but is used to filter noise at the switching
frequency, and is required only if a phase-lead capacitor
is used or if the output capacitor has high ESR.
ESR
780mV
C1
C1
+
R2
RC
CF
CC
3505 F11
Figure 11. Model for Loop Response
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 12 shows the
recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT3505’s VIN and SW pins, the catch diode (D1)
and the input capacitor (C2). The loop formed by these
components should be as small as possible and tied to
SYSTEM
GROUND VOUT
: VIAS TO LOCAL GROUND PLANE
: OUTLINE OF LOCAL GROUND PLANE
C1
VOUT
BOOST
SW
1
D1
C2
2
8 VC
7
3
6
4
5
POWER
GROUND
FB
RT
SIGNAL
GROUND
3505 F12
VIN SHUTDOWN
Figure 12. A Good PCB Layout Ensures Proper, Low EMI Operation
3505fc
19
LT3505
APPLICATIONS INFORMATION
system ground in only one place. These components, along
with the inductor and output capacitor, should be placed
on the same side of the circuit board and their connections
should be made on that layer. Place a local, unbroken ground
plane below these components and tie this ground plane
to system ground at one location, ideally at the ground
terminal of the output capacitor C1. The SW and BOOST
nodes should be as small as possible. Finally, keep the
FB node small so that the ground pin and ground traces
will shield it from the SW and BOOST nodes. Include vias
near the exposed GND pad of the LT3505 to help remove
heat from the LT3505 to the ground plane.
High Temperature Considerations
The die temperature of the LT3505 must be lower than the
maximum rating of 125°C. This is generally not a concern
unless the ambient temperature is above 85°C. For higher
temperatures, care should be taken in the layout of the
circuit to ensure good heat sinking of the LT3505. The
maximum load current should be derated as the ambient
temperature approaches 125°C. The die temperature is
calculated by multiplying the LT3505 power dissipation
by the thermal resistance from junction to ambient. Power
dissipation within the LT3505 can be estimated by calculat-
ing the total power loss from an efficiency measurement
and subtracting the catch diode loss. Thermal resistance
depends on the layout of the circuit board, but 43°C/W is
typical for the (3mm × 3mm) DFN (DD) package.
Outputs Greater Than 6V
For outputs greater than 6V, add a 1k to 2.5k resistor
across the inductor to damp the discontinuous ringing
of the SW node, preventing unintended SW current. The
12V Step-Down Converter circuit in the Typical Applications section shows the location of this resistor. Also note
that for outputs above 10V, the input voltage range will
be limited by the maximum rating of the BOOST pin. The
12V circuit shows how to overcome this limitation using
an additional zener diode.
Other Linear Technology Publications
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for Buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
Note DN100 shows how to generate a bipolar output
supply using a Buck regulator.
3505fc
20
LT3505
TYPICAL APPLICATIONS
2.2MHz, 3.3V Step-Down Converter
1N4148
VIN
0.1µF 3.3µH
SW
SHDN
ON OFF
LT3505
10V
CMPZ5240B
36.5k
FB
698k
RT
GND
MBRM140
VC
20.0k
22pF
10µF
11.3k
100k
1µF
2.50
VOUT
3.3V
BOOST
Frequency [MHz] / Load Current [A]
VIN
6V TO 36V
22pF
Switching
Frequency
2.25
2.00
Maximum
Load Current
1.75
1.50
1.25
1.0
0.75
0.50
0.25
0.00
3505 TA02
5
15
10
25
20
30
Input Voltage [V]
35
40
LTC3505 • TA02b
1.2MHz, 1.8V Step-Down Converter
BAT54
BOOST
VIN
0.1µF 4.7µH
SW
SHDN
ON OFF
LT3505
12V
CMPZ5242B
26.1k
68pF
VOUT
1.8V
1.2A
FB
1.5M
RT
GND
44.2k
2.2µF
MBRM140
VC
20.0k
22µF
60.4k
120pF
Frequency [MHz] / Load Current [A]
1.60
VIN
3.6V TO 25V
1.40
1.20
1.00
0.80
0.60
0.40
Switching
Frequency
0.20
Maximum
Load Current
0.00
3505 TA03
0
5
10
15
INPUT VOLTAGE (V)
20
25
LT3505 • TA03b
3505fc
21
LT3505
TYPICAL APPLICATIONS
750kHz, 3.3V Step-Down Converter
1N4148
VIN
4.2V TO 36V
VIN
VOUT
3.3V
1.1A, VIN > 5V
1.2A, VIN > 8V
BOOST
0.1µF 10µH
SW
SHDN
ON OFF
LT3505
36.5k
68pF
FB
RT
GND
11.3k
VC
10µF
MBRM140
75.0k
69.8k
70pF
1µF
3505 TA04
1MHz, 12V Step-Down Converter
CMDZ5235B
6V
0.1µF
1k*
0.25W
BOOST
VIN
13.5V TO 36V
ON OFF
15µH
SW
VIN
LT3505
SHDN
54.9k
71.5k
22pF
VOUT
12V
1A, VIN > 16.5V
1.1A, VIN > 20.5V
FB
GND
RT
3.3µF
1N4148
MBRM140
VC
4.99k
10µF
100k
22pF
*FOR CONTINUOUS OPERATION ABOVE 30V,
USE TWO 2k, 0.25W RESISTORS IN PARALLEL
3505 TA05
3505fc
22
LT3505
PACKAGE DESCRIPTION
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
R = 0.115
TYP
5
0.38 ± 0.10
8
0.675 ±0.05
3.5 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
3.00 ±0.10
(4 SIDES)
PIN 1
TOP MARK
(NOTE 6)
4
0.25 ± 0.05
0.75 ±0.05
0.200 REF
0.50
BSC
2.38 ±0.05
(2 SIDES)
1.65 ± 0.10
(2 SIDES)
1
(DD) DFN 1203
0.50 BSC
2.38 ±0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE
OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
MS8E Package
8-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1662)
0.889 ± 0.127
(.035 ± .005)
2.794 ± 0.102
(.110 ± .004)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.06 ± 0.102
(.081 ± .004)
1
5.23
(.206)
MIN
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
8
7 6 5
1.83 ± 0.102
(.072 ± .004)
2.083 ± 0.102 3.20 – 3.45
(.082 ± .004) (.126 – .136)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
0.65
(.0256)
BSC
0.42 ± 0.038
(.0165 ± .0015)
TYP
0.254
(.010)
1
8
RECOMMENDED SOLDER PAD LAYOUT
1.10
(.043)
MAX
DETAIL “A”
DETAIL “A”
0° – 6° TYP
0.52
(.0205)
REF
2 3
4
0.86
(.034)
REF
0.18
(.007)
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.65
(.0256)
BSC
0.127 ± 0.076
(.005 ± .003)
MSOP (MS8E) 0603
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
3505fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3505
TYPICAL APPLICATIONS
300kHz, 3.3V Step-Down Converter
VIN
4V TO 36V
ON OFF
1N4148
VIN
VOUT
3.3V
1A, VIN > 5V
1.2A, VIN > 8.5V
BOOST
0.47µF 22µH
SW
SHDN
LT3505
36.5k
100pF
FB
GND
RT
11.3k
VC
MBRM140
226k
2.2µF
68µF
KEMET
A700D686M010ATE015
100k
150pF
3505 TA06
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