LT3506/LT3506A
Dual Monolithic 1.6A
Step-Down Switching Regulator
FEATURES
DESCRIPTION
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The LT®3506 is a dual current mode PWM step-down DC/DC
converter with internal 2A power switches. Both converters are synchronized to a single oscillator and run with
opposite phases, reducing input ripple current. The output
voltages are set with external resistor dividers, and each
regulator has independent shutdown and soft-start circuits.
Each regulator generates a power-good signal when its
output is in regulation, easing power supply sequencing
and interfacing with microcontrollers and DSPs.
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Wide Input Voltage Range, 3.6V to 25V
Two 1.6A Output Switching Regulators with Internal
Power Switches
Constant Switching Frequency
LT3506: 575kHz
LT3506A: 1.1MHz
Anti-Phase Switching Reduces Ripple
Accurate 0.8V Reference, ±1%
Independent Shutdown/Soft-Start Pins
Independent Power Good Indicators Ease Supply
Sequencing
Uses Small Inductors and Ceramic Capacitors
Small 16-Lead Thermally Enhanced 5mm × 4mm
DFN and TSSOP Surface Mount Packages
APPLICATIONS
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Disk Drives
DSP Power Supplies
Wall Transformer Regulation
Distributed Power Regulation
DSL Modems
Cable Modems
The LT3506 switching frequency is 575kHz and the LT3506A
is 1.1MHz. These high switching frequencies allow the
use of tiny inductors and capacitors, resulting in a very
small dual 1.6A output solution. Constant frequency and
ceramic capacitors combine to produce low, predictable
output ripple voltage. With its wide input range of 3.6V to
25V, the LT3506 regulates a wide variety of power sources,
from 4-cell batteries and 5V logic rails to unregulated wall
transformers, lead acid batteries and distributed-power
supplies. Current mode PWM architecture provides fast
transient response with simple compensation components
and cycle-by-cycle current limiting. Frequency foldback
and thermal shutdown provide additional protection.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
TYPICAL APPLICATION
Efficiency
VIN
4.5V TO 25V
1/2 BAT-54A
BOOST1
VOUT1
1.8V
1.6A
BOOST2
90
0.22μF
4.7μH
0.22μF
SW1
SW2
FB1
FB2
VC1
VC2
6.4μH
18.7k
33.2k
1000pF
D1
47μF
15k
15k
VIN = 5V
1/2 BAT-54A
VIN1 VIN2
LT3506
VOUT2
3.3V
1.6A
2200pF
D2
10k
10.7k
EFFICIENCY (%)
22μF
100
VOUT = 3.3V
80
VOUT = 1.8V
70
22μF
60
RUN/SS1 RUN/SS2
100k
1.5nF
PGOOD1
1.5nF
50
0
100k
PGOOD2
PGOOD1
PGOOD2
3506 TA01a
GND
D1, D2: ON SEMI MBR5230LT3
0.5
1.0
IOUT (A)
1.5
2.0
3506 TA01b
3506afc
1
LT3506/LT3506A
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN Voltage ................................................. –0.3V to 25V
BOOST Pin Voltage ...................................................50V
BOOST Pin Above SW Pin.........................................25V
PG Pin Voltage ..........................................................25V
RUN/SS, FB, VC Pins ................................................5.5V
Maximum Junction Temperature .......................... 125°C
Operating Temperature Range (Note 2)
E Grade ................................................–40°C to 85°C
I Grade ............................................... –40°C to 125°C
Storage Temperature Range .................. –65°C to 125°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
BOOST1
1
16 FB1
BOOST1
1
16 FB1
SW1
2
15 VC1
SW1
2
15 VC1
14 PG1
VIN1
3
14 PG1
13 RUN/SS1
VIN1
4
12 RUN/SS2
VIN2
5
12 RUN/SS2
VIN1
3
VIN1
4
VIN2
5
VIN2
6
11 PG2
VIN2
6
11 PG2
SW2
7
10 VC2
SW2
7
10 VC2
BOOST2
8
9
BOOST2
8
9
17
FB2
DHD PACKAGE
16-LEAD PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 4.3°C/W
EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB
17
13 RUN/SS1
FB2
FE PACKAGE
16-LEAD PLASTIC TSSOP NARROW
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3506EDHD#PBF
LT3506EDHD#TRPBF
3506
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 85°C
LT3506AEDHD#PBF
LT3506AEDHD#TRPBF
3506A
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 85°C
LT3506IDHD#PBF
LT3506IDHD#TRPBF
3506
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 125°C
LT3506AIDHD#PBF
LT3506AIDHD#TRPBF
3506A
LT3506EFE#PBF
LT3506AEFE#PBF
LT3506IFE#PBF
LT3506AIFE#PBF
LT3506EFE#TRPBF
LT3506AEFE#TRPBF
LT3506IFE#TRPBF
LT3506AIFE#TRPBF
3506EFE
3506AEFE
3506IFE
3506AIEFE
16-Lead (5mm × 4mm) Plastic DFN
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
–40°C to 125°C
–40°C to 85°C
–40°C to 85°C
–40°C to 125°C
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3506EDHD
LT3506EDHD#TR
3506
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 85°C
LT3506AEDHD
LT3506AEDHD#TR
3506A
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 85°C
LT3506IDHD
LT3506IDHD#TR
3506
16-Lead (5mm × 4mm) Plastic DFN
–40°C to 125°C
16-Lead (5mm × 4mm) Plastic DFN
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
–40°C to 125°C
LT3506AIDHD
LT3506AIDHD#TR
3506A
LT3506EFE
LT3506AEFE
LT3506IFE
LT3506AIFE
LT3506EFE#TR
LT3506AEFE#TR
LT3506IFE#TR
LT3506AIFE#TR
3506EFE
3506AEFE
3506IFE
3506AIEFE
–40°C to 85°C
–40°C to 85°C
–40°C to 125°C
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3506afc
2
LT3506/LT3506A
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VBOOST = 8V, unless otherwise noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
VIN(MIN)
Undervoltage Lockout
3.4
3.6
V
IINQ
Quiescent Current
IINSD
Shutdown Current
Not Switching
3.8
4.8
mA
VRUNSS = 0V
30
45
μA
VFB
Feedback Voltage
–40°C to 85°C, DHD
–40°C to 85°C, EFE
–40°C to 125°C, IFE
l
l
l
800
800
800
808
816
816
mV
mV
mV
IFB
FB Pin Bias Current
VFB = 800mV, VC = 0.4V
l
40
100
nA
VIN = 5V to 25V
l
792
784
784
VFB(REG)
Reference Line Regulation
gmEA
Error Amp GM
0.005
350
AV
Error Amp Voltage Gain
400
IVC
VC Source Current
VC Sink Current
VVC(THRESH)
VFB = 0.6V, VC = 0V
VFB = 1.2V, VC = 1100mV
UNITS
%/V
μMhos
30
30
μA
μA
VC Switching Threshold
0.7
V
VVC(CLAMP)
VC Clamp Voltage
1.9
V
fSW
Switching Frequency
LT3506
LT3506A
Switching Phase
(Note 5)
DC
Maximum Duty Cycle
LT3506
LT3506A
VFB(SWTHRESH)
Frequency Shift Threshold on FB
fFOLD
Foldback Frequency
VFB = 0V
ISW
Switch Current Limit
(Note 3)
VSW(SAT)
Switch VCESAT (Note 4)
ISW = 1A
ILSW
Switch Leakage Current
VBOOST(MIN)
Minimum Boost Voltage Above Switch
ISW = 1A
IBOOST
BOOST Pin Current
ISW = 1A
IRUN/SS
RUN/SS Current
VRUN/SS(THRESH)
RUN/SS Threshold
VFB(PGTHRESH)
VFB PG Threshold
VPG(LOW)
ILPG
500
1
89
78
575
1.1
650
1.2
180
Deg
93
88
%
%
0.4
V
170
2.0
kHz
MHz
2.6
kHz
3.6
210
A
mV
10
μA
1.5
2.5
V
20
30
mA
2.1
μA
0.8
V
VFB Rising
720
mV
PG Voltage Output Low
VFB = 640mV, IPG = 250μA
0.22
0.4
V
PG Pin Leakage
VPG = 2V
0.1
1
μA
0.3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3506E/LT3506AE are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3506I/LT3506AI are
guaranteed and tested over the full –40°C to 125°C operating temperature
range.
Note 3: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at high duty cycle.
Note 4: Switch VCESAT guaranteed by design.
Note 5: Switching phase is guaranteed by design.
3506afc
3
LT3506/LT3506A
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency, VOUT = 1.8V (LT3506A)
Efficiency, VOUT = 3.3V (LT3506)
VOUT = 1.8V
L = 2.2μH (COILCRAFT LPS4012-222)
85
TA = 25°C
VOUT = 3.3V
95 L = 6.4μH (SUMIDA CR54-6R4)
TA = 25°C
90
70
VIN = 10V
65
60
VIN = 5V
85
80
75
VIN = 12V
70
65
VIN = 7V
80
70
60
55
55
50
50
0.6
0.8
1.0
1.2
1.4
1.6
0
0.4
IOUT (A)
0.8
IOUT (A)
0
1.8
L = 2.2μH
1.6
LOAD CURRENT (A)
1.6
L = 1μH
Switch VCESAT
TA = 25°C
L = 4.7μH
L = 3.3μH
1.4
L = 2.2μH
1.2
1.2
1.0
0
2
4
8
10 12
6
INPUT VOLTAGE (V)*
14
16
5
0
15
20
10
INPUT VOLTAGE (V)*
Boost Pin Current
200
100
0
25
0
1.0
0.5
1.5
2.0
SW CURRENT (A)
3506 G06
Current Limit vs Duty Cycle
3.0
TA = 25°C
TA = 25°C
TYPICAL
2.5
30
CURRENT LIMIT (A)
BOOST CURRENT (mA)
300
3506 G05
3506 G04
40
1.6
400
SLOPE COMPENSATION REQUIRES
L > 2.2μH FOR VIN < 7 WITH VOUT = 3.3V
TA = 25°C
SWITCH VOLTAGE (mV)
TA = 25°C
1.4
1.2
3506 G03
Maximum Load Current,
VOUT = 3.3V (LT3506A)
L = 1.5μH
0.8
IOUT (A)
0.4
3506 G02
Maximum Load Current,
VOUT = 1.8V (LT3506A)
1.0
1.6
1.2
3506 G01
1.8
VIN = 25V
65
50
0.4
VIN = 15V
75
55
0.2
VIN = 8V
85
VIN = 25V
60
0
LOAD CURRENT (A)
VOUT = 5V
95 L = 10μH (COOPER UP1B-100)
TA = 25°C
90
EFFICIENCY (%)
VIN = 4.5V
EFFICIENCY (%)
EFFICIENCY (%)
80
75
Efficiency, VOUT = 5V (LT3506)
100
100
90
20
2.0
MINIMUM
1.5
1.0
10
0.5
0
0
1.0
1.5
0.5
SWITCH CURRENT (A)
0
2.0
3506 G07
0
20
60
40
DUTY CYCLE (%)
80
100
3506 G08
3506afc
4
LT3506/LT3506A
TYPICAL PERFORMANCE CHARACTERISTICS
RUN/SS Thresholds
vs Temperature
IRUN/SS vs Temperature
Frequency vs Temperature
700
1.20
650
1.15
1.4
3.0
1.2
1.10
LT3506
550
1.05
RUNN/SS THRESHOLDS (V)
600
RUN/SS CURRENT (μA)
LT3506A
FREQUENCY (MHz)
FREQUENCY (kHz)
2.5
2.0
1.5
1.0
0.5
500
–50 –25
0
25
50
75
TEMPERATURE (°C)
100
1.00
125
0
–50
1.0
TO SWITCH
0.8
0.6
TO RUN
0.4
0.2
–25
75
0
25
50
TEMPERATURE (°C)
100
125
3506 G10
3506 G12
0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
3506 G13
PIN FUNCTIONS
BOOST1 (Pin 1), BOOST2 (Pin 8): The BOOST pins are
used to provide drive voltages, higher than the input
voltage, to the internal bipolar NPN power switches. Tie
through a diode from VOUT or from VIN.
SW1 (Pin 2), SW2 (Pin 7): The SW pins are the outputs
of the internal power switches. Connect these pins to the
inductors, catch diodes and boost capacitors.
VIN1 (Pins 3, 4): The VIN1 pins supply current to the LT3506’s
internal regulator and to the internal power switch connected to SW1. These pins must be locally bypassed.
VIN2 (Pins 5, 6): The VIN2 pins supply current to the internal power switch connected to SW2 and must be locally
bypassed. Connect these pins directly to VIN1 unless power
for channel 2 is coming from a different source.
RUN/SS1 (Pin 13), RUN/SS2 (Pin 12): The RUN/SS pins
are used to shut down the individual switching regulators and the internal bias circuits. They also provide a
soft-start function. To shut down either regulator, pull the
RUN/SS pin to ground with an open drain or collector.
Tie a capacitor from these pins to ground to limit switch
current during start-up. If neither feature is used, leave
these pins unconnected.
PG1 (Pin 14), PG2 (Pin 11): The Power Good pins are
the open collector outputs of an internal comparator. PG
remains low until the FB pin is within 10% of the final
regulation voltage. As well as indicating output regulation,
the PG pins can be used to sequence the two switching
regulators. These pins can be left unconnected. The PG
outputs are valid when VIN is greater than 3.4V and either
of the RUN/SS pins is high. The PG comparators are
disabled in shutdown.
VC1 (Pin 15), VC2 (Pin 10): The VC pins are the outputs of
the internal error amps. The voltages on these pins control
the peak switch currents. These pins are normally used
to compensate the control loops, but can also be used to
override the loops. Pull these pins to ground with an open
drain to shut down each switching regulator.
FB1 (Pin 16), FB2 (Pin 9): The LT3506 regulates each
feedback pin to 800mV. Connect the feedback resistor
divider taps to these pins.
Exposed Pad (Pin 17): The Exposed Pad of the package
provides both electrical contact to ground and good thermal
contact to the printed circuit board. The Exposed Pad must
be soldered to the circuit board for proper operation.
3506afc
5
LT3506/LT3506A
BLOCK DIAGRAM
VIN
2μA
RUN/SS2
INT REG
AND REF
RUN/SS1
MASTER
OSC
CLK1
CLK2
2μA
VIN
VIN
CIN
0.75V
¤
SLOPE
R
C1
S
CLK
BOOST
D2
Q
C3
FOLDBACK
LOGIC
SW
L1
OUT
C1
D1
FB
–
VC
ERROR
AMP
CC
RUN/SS
+
R2
+
CF
–
RC
R1
800mV
ILIMIT
CLAMP
80mV
PG
+
GND
–
3506 F02
Figure 2. Block Diagram of the LT3506 with Associated External Components (1 of 2 Regulators Shown)
3506afc
6
LT3506/LT3506A
OPERATION
(Refer to the Block Diagram)
The LT3506 is a dual, constant frequency, current mode
buck regulator with internal 2A power switches. The two
regulators share common circuitry including voltage
reference and oscillator. In addition, the analog blocks
on both regulators share the VIN1 supply voltage, but are
otherwise independent. This section describes the operation of the LT3506.
If the RUN/SS (run/soft-start) pins are both tied to ground,
the LT3506 is shut down and draws 30μA from VIN1.
Internal 2μA current sources charge external soft-start
capacitors, generating voltage ramps at these pins. If either
RUN/SS pin exceeds 0.6V, the internal bias circuits turn
on, including the internal regulator, 800mV reference and
575kHz master oscillator. In this state, the LT3506 draws
3.8mA from VIN1, whether one or both RUN/SS pins are
high. Neither switching regulator will begin to operate
until its RUN/SS pin reaches ~0.8V. The master oscillator
generates two clock signals of opposite phase.
The two switchers are current mode, step-down regulators.
This means that instead of directly modulating the duty
cycle of the power switch, the feedback loop controls the
peak current in the switch during each cycle. This current mode control improves loop dynamics and provides
cycle-by-cycle current limit.
The Block Diagram in Figure 2 shows only one of the two
switching regulators. A pulse from the slave oscillator
sets the RS flip-flop and turns on the internal NPN bipolar
power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level
determined by the voltage at VC, current comparator C1
resets the flip-flop, turning off the switch. The current in
the inductor flows through the external Schottky diode,
and begins to decrease. The cycle begins again at the next
pulse from the oscillator. In this way the voltage on the VC
pin controls the current through the inductor to the output.
The internal error amplifier regulates the output voltage
by continually adjusting the VC pin voltage.
The threshold for switching on the VC pin is 0.75V, and an
active clamp of 1.9V limits the output current. The VC pin
is also clamped to the RUN/SS pin voltage. As the internal
current source charges the external soft-start capacitor,
the current limit increases slowly. Each switcher contains
an independent oscillator. This slave oscillator is normally
synchronized to the master oscillator. However, during
start-up, short-circuit or overload conditions, the FB pin
voltage will be near zero and an internal comparator gates
the master oscillator clock signal. This allows the slave
oscillator to run the regulator at a lower frequency. This
frequency foldback behavior helps to limit switch current
and power dissipation under fault conditions.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for efficient operation.
A power good comparator trips when the FB pin is at 90%
of its regulated value. The PG output is an open collector
transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power
good is valid when the LT3506 is enabled (either RUN/SS
pin is high) and VIN is greater than ~3.4V.
3506afc
7
LT3506/LT3506A
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
R1 = R2(VOUT/0.8 – 1)
maximum input voltage is ~8V with VOUT = 0.8V. Note
that this is a restriction on the operating input voltage;
the circuit will tolerate transient inputs up to the absolute
maximum rating.
Inductor Selection and Maximum Output Current
The parallel combination of R1 and R2 should be 10k or
less to avoid bias current errors. Reference designators
refer to the Block Diagram in Figure 2.
A good first choice for the inductor value is:
Input Voltage Range
where VD is the voltage drop of the catch diode (~0.4V)
and L is in μH. With this value the maximum load current
will be ~1.6A, independent of input voltage. The inductor’s
RMS current rating must be greater than your maximum
load current and its saturation current should be about 30%
higher. To keep efficiency high, the series resistance (DCR)
should be less than 0.1Ω. Table 1 lists several vendors and
types that are suitable. Of course, such a simple design
guide will not always result in the optimum inductor for
your application. A larger value provides a slightly higher
maximum load current, and will reduce the output voltage ripple. If your load is lower than 1.6A, then you can
decrease the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the
simple rule above, then the maximum load current will
depend on input voltage. There are several graphs in the
Typical Performance Characteristics section of this data
sheet that show the maximum load current as a function
of input voltage and inductor value for several popular
output voltages. Also, low inductance may result in discontinuous mode operation, which may be acceptable,
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50%(VOUT/VIN < 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See Application Note 19 for detailed information on subharmonic oscillations. The following discussion
assumes continuous inductor current.
The minimum input voltage is determined by either the
LT3506’s minimum operating voltage of ~3.6V, or by its
maximum duty cycle. The duty cycle is the fraction of
time that the internal switch is on and is determined by
the input and output voltages:
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.3V at maximum load). This leads to a minimum input
voltage of:
VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW
with DCMAX = 0.89 (0.78 for the LT3506A).
A more detailed analysis includes inductor loss and the
dependence of the diode and switch drop on operating
current. A common application where the maximum duty
cycle limits the input voltage range is the conversion of 5V
to 3.3V. The maximum load current that the LT3506 can
deliver at 3.3V depends on the accuracy of the 5V input
supply. With a low loss inductor (DCR less than 80mΩ),
the LT3506 can deliver 1.2A for VIN > 4.7V and 1.6A for
VIN > 4.85V. The maximum input voltage is determined
by the absolute maximum ratings of the VIN and BOOST
pins and by the minimum duty cycle DCMIN = 0.08 (0.15
for the LT3506A):
VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW.
This limits the maximum input voltage to ~21V with VOUT
= 1.2V and ~15V with VOUT = 0.8V. For the LT3506A the
L = 2 • (VOUT + VD) for the LT3506
L = (VOUT + VD) for the LT3506A
3506afc
8
LT3506/LT3506A
APPLICATIONS INFORMATION
The current in the inductor is a triangle wave with an
average value equal to the load current. The peak switch
current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3506 limits its switch
current in order to protect itself and the system from
overload faults. Therefore, the maximum output current
that the LT3506 will deliver depends on the current limit,
the inductor value and the input and output voltages. L
is chosen based on output current requirements, output
voltage ripple requirements, size restrictions and efficiency
goals. When the switch is off, the inductor sees the output
voltage plus the catch diode drop. This gives the peak-topeak ripple current in the inductor:
ΔIL = (1 – DC)(VOUT + VD)/(L • f)
where f is the switching frequency of the LT3506 and L
is the value of the inductor. The peak inductor and switch
current is
ISWPK = ILPK = IOUT + ΔIL/2.
To maintain output regulation, this peak current must be
less than the LT3506’s switch current limit ILIM. ILIM is at
least 2A at low duty cycle and decreases linearly to 1.7A
at DC = 0.8. The maximum output current is a function of
the chosen inductor value:
IOUT(MAX) = ILIM – ΔIL/2 = 2A•(1 – 0.21•DC) – ΔIL/2
If the inductor value is chosen so that the ripple current
is small, then the available output current will be near
the switch current limit. One approach to choosing the
inductor is to start with the simple rule given above, look
at the available inductors, and choose one to meet cost or
space goals. Then use these equations to check that the
LT3506 will be able to deliver the required output current.
Note again that these equations assume that the inductor
current is continuous. Discontinuous operation occurs
when IOUT is less than ΔIL/2 as calculated above.
Table 1. Inductors
VALUE
(μH)
ISAT (A)
DCR (Ω)
HEIGHT
(mm)
CR43-3R3
3.3
1.44
0.086
3.5
CR43-4R7
4.7
1.15
0.109
3.5
CDC5d23-2R2
2.2
2.16
0.030
2.5
CDRH5D28-2R6
2.6
2.60
0.013
3.0
CDRH6D26-5R6
5.6
2.00
0.027
2.8
CDH113-100
10
2.00
0.047
3.7
DO1606T-152
1.5
2.10
0.060
2.0
DO1606T-222
2.2
1.70
0.070
2.0
DO1608C-332
3.3
2.00
0.080
2.9
DO1608C-472
4.7
1.50
0.090
2.9
DO1813P-682HC
6.8
2.20
0.080
5.0
SD414-2R2
2.2
2.73
0.061
1.35
SD414-6R8
6.8
1.64
0.135
1.35
UP1B-100
10
1.90
0.111
5.0
(D62F)847FY-2R4M
2.4
2.5
0.037
2.7
(D73LF)817FY-2R2M
2.2
2.7
0.03
3.0
PART NUMBER
Sumida
Coilcraft
Cooper
Toko
Input Capacitor Selection
Bypass the input of the LT3506 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type can be used if there is
additional bypassing provided by bulk electrolytic or tantalum capacitors. The following paragraphs describe the
input capacitor considerations in more detail. Step-down
regulators draw current from the input supply in pulses
with very fast rise and fall times. The input capacitor is
required to reduce the resulting voltage ripple at the LT3506
and to force this very high frequency switching current
into a tight local loop, minimizing EMI. The input capaci-
3506afc
9
LT3506/LT3506A
APPLICATIONS INFORMATION
tor must have low impedance at the switching frequency
to do this effectively, and it must have an adequate ripple
current rating. With two switchers operating at the same
frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple.
However, a conservative value is the RMS input current for
the channel that is delivering most power (VOUT • IOUT).
This is given by:
22μF (10μF for the LT3506A) will be required to meet the
ESR and ripple current requirements. Because the input
capacitor is likely to see high surge currents when the input
source is applied, tantalum capacitors should be surge
rated. The manufacturer may also recommend operation
below the rated voltage of the capacitor. Be sure to place
the 1μF ceramic as close as possible to the VIN and GND
pins on the IC for optimal noise immunity.
VOUT • ( VIN − VOUT )
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3506. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
IINRMS = IOUT
VIN
I
< OUT
2
and is largest when VIN = 2VOUT (50% duty cycle). As
the second, lower power channel draws input current,
the input capacitor’s RMS current actually decreases as
the out-of-phase current cancels the current drawn by
the higher power channel. Considering that the maximum
load current from a single channel is ~1.6A, RMS ripple
current will always be less than 0.8A.
The high frequency of the LT3506 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 22μF (less than 10μF
for the LT3506A). The combination of small size and low
impedance (low equivalent series resistance or ESR) of
ceramic capacitors makes them the preferred choice.
The low ESR results in very low voltage ripple and the
capacitors can handle plenty of ripple current. They are also
comparatively robust and can be used in this application
at their rated voltage. X5R and X7R types are stable over
temperature and applied voltage, and give dependable
service. Other types (Y5V and Z5U) have very large temperature and voltage coefficients of capacitance, so they
may have only a small fraction of their nominal capacitance
in your application. While they will still handle the RMS
ripple current, the input voltage ripple may become fairly
large, and the ripple current may end up flowing from
your input supply or from other bypass capacitors in your
system, as opposed to being fully sourced from the local
input capacitor.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example
a 1μF ceramic capacitor in parallel with a low ESR tantalum
capacitor. For the electrolytic capacitor, a value larger than
Output Capacitor Selection
The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores
energy in order satisfy transient loads and to stabilize the
LT3506’s control loop. Because the LT3506 operates at a
high frequency, you don’t need much output capacitance.
Also, the current mode control loop doesn’t require the
presence of output capacitor series resistance (ESR). For
these reasons, you are free to use ceramic capacitors to
achieve very low output ripple and small circuit size.
Estimate output ripple with the following equations:
VRIPPLE = ΔIL/(8 • f • COUT) for ceramic capacitors
VRIPPLE = ΔIL • ESR for electrolytic capacitors (tantalum
and aluminum)
where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low, and the
RMS current rating of the output capacitor is usually not
of concern.
Another constraint on the output capacitor is that it must
have greater energy storage than the inductor; if the stored
energy in the inductor is transferred to the output, you
would like the resulting voltage step to be small compared
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LT3506/LT3506A
APPLICATIONS INFORMATION
to the regulation voltage. For a 5% overshoot, this requirement becomes
COUT > 10L(ILIM/VOUT)2.
Finally, there must be enough capacitance for good transient
performance. The last equation gives a good starting point.
Alternatively, you can start with one of the designs in this
data sheet and experiment to get the desired performance.
This topic is covered more thoroughly in the section on
loop compensation.
For 5V and 3.3V outputs with greater than 1A output, a
22μF 6.3V ceramic capacitor (X5R or X7R) at the output
results in very low output voltage ripple and good transient response. For lower voltages, 22μF is adequate but
increasing COUT will improve transient performance. For
the LT3506A, 10μF of output capacitance is sufficient at
VOUT between 3.3V and 5V. Other types and values can be
used. The following discusses tradeoffs in output ripple
and transient performance.
The high performance (low ESR), small size and robustness of ceramic capacitors make them the preferred type
for LT3506 applications. However, all ceramic capacitors
are not the same. As mentioned above, many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular, Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and temperature extremes. Because
the loop stability and transient response depend on the
value of COUT, you may not be able to tolerate this loss.
Use X7R and X5R types.
You can also use electrolytic capacitors. The ESRs of most
aluminum electrolytics are too large to deliver low output
ripple. Tantalum and newer, lower ESR organic electrolytic
capacitors intended for power supply use are suitable,
and the manufacturers will specify the ESR. The choice of
capacitor value will be based on the ESR required for low
ripple. Because the volume of the capacitor determines
its ESR, both the size and the value will be larger than a
ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give
better transient response for large changes in load current.
Table 2 lists several capacitor vendors.
Table 2. Low-ESR Surface Mount Capacitors
VENDOR
TYPE
Taiyo-Yuden
Ceramic
SERIES
AVX
Ceramic
Tantalum
Kemet
Tantalum
Tantalum
Organic
Aluminum
Organic
T491, T494, T495, T520
Sanyo
Tantalum or Aluminum
Organic
POSCAP
Panasonic
Aluminum
Organic
SP
CAP
TDK
Ceramic
TPS
A700
Catch Diode
The catch diode (D1 in Figure 2) must have a reverse voltage rating greater than the maximum input voltage. The
average current of the catch diode is given by:
IDAVE = IOUT (1 – DCMIN)
A Schottky diode with a 1A average forward current rating
will suffice for most applications. The ON Semiconductor
MBRM120LT3 (20V) and MBRM130LT3 (30V) are good
choices; they have a tiny package with good thermal properties. Many vendors have suitable surface mount versions of
the 1N5817 (20V) and 1N5818 (30V) 1A Schottky diodes
such as the Microsemi UPS120.
Applications with large step down ratios and high output
currents may have more than 1A of average diode current.
The ON Semiconductor MBRS230LT3 or International Rectifier 20BQ030 (both 2A, 30V) would be good choices.
3506afc
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APPLICATIONS INFORMATION
BOOST Pin Considerations
The capacitor and diode tied to the BOOST pin generate
a voltage that is higher than the input voltage. In most
cases a 0.1μF capacitor and fast switching diode (such
as the CMDSH-3 or FMMD914) will work well. Figure 3
shows three ways to arrange the boost circuit. The BOOST
pin must be more than 2.5V above the SW pin for full
efficiency. For outputs of 3.3V and higher the standard
circuit (Figure 3a) is best. For outputs between 2.8V and
3.3V, use a small Schottky diode (such as the BAT-54).
For lower output voltages the boost diode can be tied to
the input (Figure 3b). The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower
voltage source. Finally, as shown in Figure 3c, the anode
of the boost diode can be tied to another source that is
at least 3V. For example, if you are generating 3.3V and
1.8V and the 3.3V is on whenever the 1.8V is on, the 1.8V
boost diode can be connected to the 3.3V output. In any
case, you must also be sure that the maximum voltage at
the BOOST pin is less than the maximum specified in the
Absolute Maximum Ratings section.
The boost circuit can also run directly from a DC voltage
that is higher than the input voltage by more than 3V,
as in Figure 3d. The diode is used to prevent damage to
the LT3506 in case VINB is held low while VIN is present.
The circuit saves several components (both BOOST pins
can be tied to D2). However, efficiency may be lower and
dissipation in the LT3506 may be higher. Also, if VINB is
absent, the LT3506 will still attempt to regulate the output,
but will do so with very low efficiency and high dissipation
because the switch will not be able to saturate, dropping
1.5V to 2V in conduction.
D2
D2
C3
BOOST
LT3506
LT3506
VIN
VIN
VOUT
SW
VIN
VIN
SW
VBOOST – VSW VIN
MAX VBOOST 2VIN
VBOOST – VSW VOUT
MAX VBOOST VIN + VOUT
(3b)
(3a)
D2
D2
VINB
>VIN + 3V
VINB > 3V
BOOST
BOOST
C3
LT3506
VIN
VOUT
GND
GND
VIN
C3
BOOST
LT3506
SW
VOUT
VIN
VIN
GND
SW
VOUT
GND
VBOOST – VSW VINB
MAX VBOOST VINB + VIN
MINIMUM VALUE FOR VINB = 3V
MAX VBOOST – VSW VINB
MAX VBOOST VINB
MINIMUM VALUE FOR VINB = VIN + 3V
(3c)
3506 F03
(3d)
Figure 3. Generating the Boost Voltage
3506afc
12
LT3506/LT3506A
APPLICATIONS INFORMATION
The minimum input voltage of an LT3506 application is
limited by the minimum operating voltage (