LT3573
Isolated Flyback Converter
without an Opto-Coupler
FEATURES
DESCRIPTION
3V to 40V Input Voltage Range
1.25A, 60V Integrated NPN Power Switch
Boundary Mode Operation
No Transformer Third Winding or
Opto-Isolator Required for Regulation
n Improved Primary-Side Winding Feedback
Load Regulation
n V
OUT Set with Two External Resistors
n BIAS Pin for Internal Bias Supply and Power
NPN Driver
n Programmable Soft-Start
n Programmable Power Switch Current Limit
n Thermally Enhanced 16-Lead MSOP
The LT®3573 is a monolithic switching regulator specifically designed for the isolated flyback topology. No third
winding or opto-isolator is required for regulation. The
part senses the isolated output voltage directly from the
primary-side flyback waveform. A 1.25A, 60V NPN power
switch is integrated along with all control logic into a
16‑lead MSOP package.
n
n
n
n
The LT3573 operates with input supply voltages from
3V to 40V, and can deliver output power up to 7W with
no external power devices.The LT3573 utilizes boundary
mode operation to provide a small magnetic solution with
improved load regulation.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and No RSENSE and ThinSOT are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners. Protected by U.S. Patents, including
5438499 and 7471522.
APPLICATIONS
Industrial, Automotive and Medical Isolated
Power Supplies
n
TYPICAL APPLICATION
5V Isolated Flyback Converter
VIN
12V TO 24V
357k
0.22µF
VIN
2k
24µH
SHDN/UVLO
PMEG6010
51.1k
LT3573
RFB
3:1
VOUT+
5V, 0.7A
2.6µH
47µF
VOUT–
80.6k
RREF
TC
6.04k
RILIM
SS
SW
VC
28.7k
GND
10nF
2
1
0
VIN = 24V
VIN = 12V
–1
–2
TEST BIAS
–3
20k
10k
3
OUTPUT VOLTAGE ERROR (%)
10µF
Load Regulation
B340A
200
400
600
800 1000 1200 1400
IOUT (mA)
4.7µF
1nF
0
3573 TA01b
3573 TA01
3573fd
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1
LT3573
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
SW.............................................................................60V
VIN, SHDN/UVLO, RFB, BIAS......................................40V
SS, VC, TC, RREF , RILIM................................................5V
Maximum Junction Temperature........................... 125°C
Operating Junction Temperature Range (Note 2)
LT3573E............................................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
TOP VIEW
GND
TEST
GND
SW
VIN
BIAS
SHDN/UVLO
GND
1
2
3
4
5
6
7
8
17
16
15
14
13
12
11
10
9
GND
TC
RREF
RFB
VC
RILIM
SS
GND
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 50°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE CONNECTED TO GND
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3573EMSE#PBF
LT3573EMSE#TRPBF
3573
16-Lead Plastic MSOP
–40°C to 125°C
LT3573IMSE#PBF
LT3573IMSE#TRPBF
3573
16-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted.
PARAMETER
CONDITIONS
Input Voltage Range
MIN
l
Quiescent Current
SS = 0V
VSHDN/UVLO = 0V
Soft-Start Current
SS = 0.4V
SHDN/UVLO Pin Threshold
UVLO Pin Voltage Rising
SHDN/UVLO Pin Hysteresis Current
VUVLO = 1V
TYP
3
3.5
0
MAX
40
V
1
mA
µA
7
l
1.15
2
Soft-Start Threshold
µA
1.22
1.29
V
2.5
3
µA
0.7
Maximum Switching Frequency
V
1000
Switch Current Limit
RILIM = 10k
Minimum Current Limit
VC = 0V
Switch VCESAT
ISW = 0.5A
RREF Voltage
VIN = 3V
1.25
l
RREF Voltage Line Regulation
3V < VIN < 40V
RREF Pin Bias Current
(Note 3)
IREF Reference Current
Measured at RFB Pin with RREF = 6.49k
1.55
kHz
1.85
200
l
1.21
1.20
UNITS
A
mA
150
250
mV
1.23
1.25
1.25
V
0.01
0.03
%/ V
100
600
nA
190
µA
3573fd
2
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LT3573
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted.
PARAMETER
CONDITIONS
Error Amplifier Voltage Gain
VIN = 3V
MIN
150
V/V
Error Amplifier Transconductance
DI = 10µA, VIN = 3V
150
µmhos
Minimum Switching Frequency
VC = 0.35V
40
kHz
TC Current into RREF
RTC = 20.1k
27.5
µA
BIAS Pin Voltage
IBIAS = 30mA
2.9
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3573E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
Output Voltage
5.15
6
5.00
4.95
VIN = 5V
3
2
4.90
–25
50
0
25
75
TEMPERATURE (°C)
100
125
3573 G01
0
–50
VIN = 12V
2.8
2.6
2.4
2.2
1
4.85
4.80
–50
4
V
VIN = 40V
3.0
VIN = 40V
BIAS VOLTAGE (V)
5.05
3.1
Bias Pin Voltage
3.2
5
IQ (mA)
VOUT (V)
5.10
UNITS
TA = 25°C, unless otherwise noted.
Quiescent Current
7
3
MAX
to 125°C operating junction temperature range are assured by design
characterization and correlation with statistical process controls. The
LT3573I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 3: Current flows out of the RREF pin.
TYPICAL PERFORMANCE CHARACTERISTICS
5.20
TYP
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3573 G02
2.0
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3573 G03
3573fd
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3
LT3573
TYPICAL PERFORMANCE CHARACTERISTICS
Switch Current Limit
400
1.8
350
1.6
25°C
125°C
250
–50°C
200
150
100
0.8
RILIM = 10k
0.6
0.25
0.50 0.75 1.00
SWITCH CURRENT (A)
1.25
1.50
0
–50
–25
0
25
50
75
1.0
0.8
0.6
0.2
100
125
TEMPERATURE (°C)
0
0
10
20
30
40
50
RILIM RESISTANCE (k)
3573 G05
3573 G04
3573 G06
SS Pin Current
SHDN/UVLO Falling Threshold
1.28
12
10
1.26
SS PIN CURRENT (µA)
SHDN/UVLO VOLTAGE (V)
1.2
0.4
MINIMUM CURRENT LIMIT
0.2
0
1.6
1.2
1.0
Switch Current Limit vs RILIM
1.4
0.4
50
0
MAXIMUM CURRENT LIMIT
CURRENT LIMIT (A)
300
1.8
1.4
CURRENT LIMIT (A)
SWITCH VCESAT VOLTAGE (mV)
Switch VCESAT
TA = 25°C, unless otherwise noted.
1.24
1.22
1.20
8
6
4
2
1.18
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
0
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
3573 G07
3573 G08
3573fd
4
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LT3573
PIN FUNCTIONS
GND: Ground.
TEST: This pin is used for testing purposes only and must
be connected to ground for the part to operate properly.
SW: Collector Node of the Output Switch. This pin has
large currents flowing through it. Keep the traces to the
switching components as short as possible to minimize
electromagnetic radiation and voltage spikes.
VIN : Input Voltage. This pin supplies current to the internal
start-up circuitry and as a reference voltage for the DCM
comparator and feedback circuitry. This pin must be locally
bypassed with a capacitor.
BIAS: Bias Voltage. This pin supplies current to the switch
driver and internal circuitry of the LT3573. This pin must
be locally bypassed with a capacitor. This pin may also be
connected to VIN if a third winding is not used and if VIN
≤ 15V. If a third winding is used, the BIAS voltage should
be lower than the input voltage for proper operation.
SHDN/UVLO: Shutdown/Undervoltage Lockout. A resistor
divider connected to VIN is tied to this pin to program the
minimum input voltage at which the LT3573 will operate.
At a voltage below ~0.7V, the part draws no quiescent
current. When below 1.22V and above ~0.7V, the part will
draw 10µA of current, but internal circuitry will remain off.
Above 1.25V, the internal circuitry will start and a 10µA
current will be fed into the SS pin. When this pin falls
below 1.25V, 2.5µA will be pulled from the pin to provide
programmable hysteresis for UVLO.
RILIM: Maximum Current Limit Adjust Pin. A resistor
should be tied to this pin to ground to set the current
limit. Use a 10k resistor for the full current capabilities
of the switch.
SS: Soft-Start Pin. Place a soft-start capacitor here to
limit start-up inrush current and output voltage ramp
rate. Switching starts when the voltage at this pin reaches
~0.7V.
VC: Compensation Pin for Internal Error Amplifier. Connect
a series RC from this pin to ground to compensate the
switching regulator. A 100pF capacitor in parallel helps
eliminate noise.
RFB: Input Pin for External Feedback Resistor. This pin is
connected to the transformer primary (VSW). The ratio
of this resistor to the RREF resistor, times the internal
bandgap reference, determines the output voltage (plus
the effect of any non-unity transformer turns ratio). The
average current through this resistor during the flyback
period should be approximately 200µA. For nonisolated
applications, this pin should be connected to VIN.
RREF : Input Pin for External Ground-Referred Reference
Resistor. This resistor should be in the range of 6k, but
for convenience, need not be precisely this value. For
nonisolated applications, a traditional resistor voltage
divider may be connected to this pin.
TC: Output Voltage Temperature Compensation. Connect
a resistor to ground to produce a current proportional to
absolute temperature to be sourced into the RREF node.
ITC = 0.55V/RTC .
Exposed Pad: Ground. The Exposed Pad of the package
provides both electrical contact to ground and good thermal
contact to the printed circuit board. The Exposed Pad must
be soldered to the circuit board for proper operation and
should be well connected with many vias to an internal
ground plane.
3573fd
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5
LT3573
BLOCK DIAGRAM
D1
T1
VIN
C1
L1A
VOUT +
L1B
C2
R3
VOUT –
N:1
TC
CURRENT
Q3
TC
R6
RFB
VIN
SW
FLYBACK
ERROR
AMP
Q2
I2
20µA
1.23V
–g
m
+
–
+
ONE
SHOT
CURRENT
COMPARATOR
A2
–
A1
+
S
S
BIAS
R
DRIVER
BIAS
MASTER
LATCH
SHDN/UVLO
R2
1.22V
+
A5
–
2.5µA
INTERNAL
REFERENCE
AND
REGULATORS
I1
7µA
Q1
Q
C5
R1
–
VIN
RREF
R4
+
V1
120mV
+
–
A4
RSENSE
0.02Ω GND
OSCILLATOR
VC
Q4
R7
SS
RILIM
C3
C4
3573 BD
R5
3573fd
6
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LT3573
OPERATION
The LT3573 is a current mode switching regulator IC
designed specifically for the isolated flyback topology.
The special problem normally encountered in such circuits is that information relating to the output voltage on
the isolated secondary side of the transformer must be
communicated to the primary side in order to maintain
regulation. Historically, this has been done with optoisolators or extra transformer windings. Opto-isolator
circuits waste output power and the extra components
increase the cost and physical size of the power supply.
Opto-isolators can also exhibit trouble due to limited
dynamic response, nonlinearity, unit-to-unit variation
and aging over life. Circuits employing extra transformer
windings also exhibit deficiencies. Using an extra winding adds to the transformer’s physical size and cost, and
dynamic response is often mediocre.
The LT3573 derives its information about the isolated
output voltage by examining the primary-side flyback
pulse waveform. In this manner, no opto-isolator nor extra
transformer winding is required for regulation. The output
voltage is easily programmed with two resistors. Since this
IC operates in boundary control mode, the output voltage
is calculated from the switch pin when the secondary current is almost zero. This method improves load regulation
without external resistors and capacitors.
The Block Diagram shows an overall view of the system.
Many of the blocks are similar to those found in traditional
switching regulators including: internal bias regulator,
oscillator, logic, current amplifier and comparator, driver,
and output switch. The novel sections include a special
flyback error amplifier and a temperature compensation
circuit. In addition, the logic system contains additional
logic for boundary mode operation, and the sampling
error amplifier.
The LT3573 features a boundary mode control method,
where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode.
The VC pin controls the current level just as it does in normal
current mode operation, but instead of turning the switch
on at the start of the oscillator period, the part detects
when the secondary side winding current is zero.
Boundary Mode Operation
Boundary mode is a variable frequency, current-mode
switching scheme. The switch turns on and the inductor
current increases until a VC pin controlled current limit. The
voltage on the SW pin rises to the output voltage divided
by the secondary-to-primary transformer turns ratio plus
the input voltage. When the secondary current through
the diode falls to zero, the SW pin voltage falls below VIN .
A discontinuous conduction mode (DCM) comparator
detects this event and turns the switch back on.
Boundary mode returns the secondary current to zero
every cycle, so the parasitic resistive voltage drops do not
cause load regulation errors. Boundary mode also allows
the use of a smaller transformer compared to continuous
conduction mode and no subharmonic oscillation.
At low output currents the LT3573 delays turning on the
switch, and thus operates in discontinuous mode. Unlike
a traditional flyback converter, the switch has to turn on
to update the output voltage information. Below 0.6V on
the VC pin, the current comparator level decreases to
its minimum value, and the internal oscillator frequency
decreases in frequency. With the decrease of the internal
oscillator, the part starts to operate in DCM. The output
current is able to decrease while still allowing a minimum
switch off-time for the error amp sampling circuitry. The
typical minimum internal oscillator frequency with VC
equal to 0V is 40kHz.
3573fd
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7
LT3573
APPLICATIONS INFORMATION
ERROR AMPLIFIER—PSEUDO DC THEORY
In the Block Diagram, the RREF (R4) and RFB (R3) resistors
can be found. They are external resistors used to program
the output voltage. The LT3573 operates much the same
way as traditional current mode switchers, the major
difference being a different type of error amplifier which
derives its feedback information from the flyback pulse.
In combination with the previous VFLBK expression yields
an expression for VOUT, in terms of the internal reference,
programming resistors, transformer turns ratio and diode
forward voltage drop:
R 1
VOUT = VBG FB
− VF −ISEC (ESR)
RREF a NPS
Operation is as follows: when the output switch, Q1, turns
off, its collector voltage rises above the VIN rail. The amplitude of this flyback pulse, i.e., the difference between
it and VIN, is given as:
Additionally, it includes the effect of nonzero secondary
output impedance (ESR). This term can be assumed to
be zero in boundary control mode. More details will be
discussed in the next section.
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
Temperature Compensation
VF = D1 forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current by
the action of RFB and Q2. Nearly all of this current flows
through resistor RREF to form a ground-referred voltage. This voltage is fed into the flyback error amplifier.
The flyback error amplifier samples this output voltage
information when the secondary side winding current is
zero. The error amplifier uses a bandgap voltage, 1.23V,
as the reference voltage.
The relatively high gain in the overall loop will then cause
the voltage at the RREF resistor to be nearly equal to the
bandgap reference voltage VBG. The relationship between
VFLBK and VBG may then be expressed as:
V
V
a FLBK = BG
RFB RREF
The first term in the VOUT equation does not have a temperature dependence, but the diode forward drop has a
significant negative temperature coefficient. To compensate for this, a positive temperature coefficient current
source is connected to the RREF pin. The current is set by
a resistor to ground connected to the TC pin. To cancel the
temperature coefficient, the following equation is used:
dVTC
dVF
R
1
= − FB •
•
dT
R TC
NPS
dT
R TC =
or,
dV
−RFB
1
R
•
• TC ≈ FB
NPS dVF / dT dT NPS
(dVF /dT) = Diode’s forward voltage temperature
coefficient
(dVTC /dT) = 2mV
VTC = 0.55V
The resistor value given by this equation should also be
verified experimentally, and adjusted if necessary to achieve
optimal regulation over temperature.
or,
R 1
VFLBK = VBG FB
RREF a
The revised output voltage is as follows:
R 1
VOUT = VBG FB
− VF
RREF NPS a
a = Ratio of Q1 IC to IE, typically ≈ 0.986
VBG = Internal bandgap reference
V R
− TC • FB –ISEC (ESR)
R TC NPS a
3573fd
8
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LT3573
APPLICATIONS INFORMATION
ERROR AMPLIFIER—DYNAMIC THEORY
Selecting RFB and RREF Resistor Values
Due to the sampling nature of the feedback loop, there
are several timing signals and other constraints that are
required for proper LT3573 operation.
The expression for VOUT, developed in the Operation section, can be rearranged to yield the following expression
for RFB:
Minimum Current Limit
The LT3573 obtains output voltage information from the
SW pin when the secondary winding conducts current.
The sampling circuitry needs a minimum amount of time
to sample the output voltage. To guarantee enough time,
a minimum inductance value must be maintained. The
primary-side magnetizing inductance must be chosen
above the following value:
LPRI ≥ VOUT •
tMIN
1.4µH
•NPS = VOUT •NPS •
V
IMIN
RFB =
RREF •NPS ( VOUT + VF ) a + VTC
VBG
where,
VOUT = Output voltage
VF = Switching diode forward voltage
a = Ratio of Q1, IC to IE, typically 0.986
NPS = Effective primary-to-secondary turns ratio
VTC = 0.55V
The equation assumes the temperature coefficients of
the diode and VTC are equal, which is a good first-order
approximation.
tMIN = minimum off-time, 350ns
IMIN = minimum current limit, 250mA
The minimum current limit is higher than that on the Electrical Characteristics table due to the overshoot caused by
the comparator delay.
Leakage Inductance Blanking
When the output switch first turns off, the flyback pulse
appears. However, it takes a finite time until the transformer
primary-side voltage waveform approximately represents
the output voltage. This is partly due to the rise time on
the SW node, but more importantly due to the transformer leakage inductance. The latter causes a very fast
voltage spike on the primary-side of the transformer that
is not directly related to output voltage (some time is also
required for internal settling of the feedback amplifier
circuitry). The leakage inductance spike is largest when
the power switch current is highest.
In order to maintain immunity to these phenomena, a fixed
delay is introduced between the switch turn-off command
and the beginning of the sampling. The blanking is internally
set to 150ns. In certain cases, the leakage inductance may
not be settled by the end of the blanking period, but will
not significantly affect output regulation.
Strictly speaking, the above equation defines RFB not as an
absolute value, but as a ratio of RREF. So, the next question is, “What is the proper value for RREF?” The answer
is that RREF should be approximately 6.04k. The LT3573
is trimmed and specified using this value of RREF. If the
impedance of RREF varies considerably from 6.04k, additional errors will result. However, a variation in RREF of
several percent is acceptable. This yields a bit of freedom
in selecting standard 1% resistor values to yield nominal
RFB /RREF ratios.
Tables 1-4 are useful for selecting the resistor values for
RREF and RFB with no equations. The tables provide RFB,
RREF and RTC values for common output voltages and
common winding ratios.
Table 1. Common Resistor Values for 1:1 Transformers
VOUT (V)
NPS
RFB (kΩ)
RREF (kΩ)
RTC (kΩ)
3.3
1.00
18.7
6.04
19.1
5
1.00
27.4
6.04
28
12
1.00
64.9
6.04
66.5
15
1.00
80.6
6.04
80.6
20
1.00
107
6.04
105
3573fd
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9
LT3573
APPLICATIONS INFORMATION
relatively constant maximum output current regardless of
input voltage. This is due to the continuous nonswitching
behavior of the two currents. A flyback converter has both
discontinuous input and output currents which makes it
similar to a nonisolated buck-boost. The duty cycle will
affect the input and output currents, making it hard to
predict output power. In addition, the winding ratio can
be changed to multiply the output current at the expense
of a higher switch voltage.
Table 2. Common Resistor Values for 2:1 Transformers
NPS
RFB (kΩ)
RREF (kΩ)
RTC (kΩ)
3.3
2.00
5
2.00
37.4
6.04
18.7
56
6.04
28
12
2.00
130
6.04
66.5
15
2.00
162
6.04
80.6
Table 3. Common Resistor Values for 3:1 Transformers
VOUT (V)
NPS
RFB (kΩ)
RREF (kΩ)
RTC (kΩ)
3.3
3.00
56.2
6.04
20
5
3.00
80.6
6.04
28.7
10
3.00
165
6.04
54.9
The graphs in Figures 1-3 show the maximum output
power possible for the output voltages 3.3V, 5V, and 12V.
The maximum power output curve is the calculated output
power if the switch voltage is 50V during the off-time. To
achieve this power level at a given input, a winding ratio
value must be calculated to stress the switch to 50V,
resulting in some odd ratio values. The curves below are
examples of common winding ratio values and the amount
of output power at given input voltages.
Table 4. Common Resistor Values for 4:1 Transformers
VOUT (V)
NPS
RFB (kΩ)
RREF (kΩ)
RTC (kΩ)
3.3
4.00
5
4.00
76.8
6.04
19.1
113
6.04
28
Output Power
One design example would be a 5V output converter with
a minimum input voltage of 20V and a maximum input
voltage of 30V. A three-to-one winding ratio fits this design
example perfectly and outputs close to six watts at 30V
but lowers to five watts at 20V.
A flyback converter has a complicated relationship between the input and output current compared to a buck
or a boost. A boost has a relatively constant maximum
input current regardless of input voltage and a buck has a
8
7
6
N = 7:1
N = 10:1
5
N = 4:1
4
N = 3:1
3
2
1
0
8
MAXIMUM
OUTPUT
POWER
N = 5:1
7
OUTPUT POWER (W)
OUTPUT POWER (W)
8
MAXIMUM
OUTPUT
POWER
6
5 N = 7:1
N = 3:1
4
N = 2:1
3
2
5
10
15
20
25
INPUT VOLTAGE (V)
30
35
40
3573 F01
Figure 1. Output Power for 3.3V Output
0
6
N = 3:1
5
N = 1:1
4
3
2
1
1
0
MAXIMUM
OUTPUT
POWER
N = 2:1
7
OUTPUT POWER (W)
VOUT (V)
0
5
10
15
20
25
30
INPUT VOLTAGE (V)
35
40
45
3573 F02
Figure 2. Output Power for 5V Output
0
0
5
10
15
20
25
30
INPUT VOLTAGE (V)
35
40
45
3573 F03
Figure 3. Output Power for 12V Output
3573fd
10
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LT3573
APPLICATIONS INFORMATION
TRANSFORMER DESIGN CONSIDERATIONS
Transformer specification and design is perhaps the most
critical part of successfully applying the LT3573. In addition
to the usual list of caveats dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed
flyback transformers for use with the LT3573. Table 5 shows
the details of several of these transformers.
Table 5. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER
PART NUMBER
PA2362NL
SIZE (W × L × H)
(mm)
LPRI
(µH)
LLEAKAGE
(nH)
NP:NS:NB
RPRI
(mΩ)
RSEC
(mΩ)
VENDOR
TARGET
APPLICATIONS
15.24 × 13.1 × 11.45
24
550
4:1:1
117
9.5
Pulse Engineering
24V→3.3V, 1.5A
PA2454NL
15.24 × 13.1 × 11.45
24
430
3:1:1
82
11
Pulse Engineering
24V→5V, 1A
PA2455NL
15.24 × 13.1 × 11.45
25
450
2:1:1
82
22
Pulse Engineering
24V→12V, 0.5A
PA2456NL
15.24 × 13.1 × 11.45
25
390
1:1:1
82
84
Pulse Engineering
12V→12V, 0.3A
24V→12V, 0.4A
36V→5V, 0.6A
PA2617NL
12.70 × 10.67 × 9.14
21
245
1:1:0.33
164
166
Pulse Engineering
24V→15V, 0.4A
PA2626NL
12.70 × 10.67 × 9.14
30
403
3:1:1
240
66
Pulse Engineering
24V→5V, 1A
PA2627NL
15.24 × 13.1 × 11.45
50
766
3:1:1
420
44
Pulse Engineering
24V→5V, 1A
GA3429-BL
15.24 × 12.7 × 11.43
25
566
4:1:1
95
7.5
Coilcraft
24V→3.3V, 1.5A
750310471
15.24 × 13.3 × 11.43
25
350
3:1:1
57
11
Würth Elektronik
24V→5V, 1A
750310559
15.24 × 13.3 × 11.43
24
400
4:1:1
51
16
Würth Elektronik
24V→3.3V, 1.5A
750310562
15.24 × 13.3 × 11.43
25
330
2:1:1
60
20
Würth Elektronik
24V→12V, 0.5A
750310563
15.24 × 13.3 × 11.43
25
325
1:1:0.5
60
60
Würth Elektronik
12V→12V, 0.3A
24V→12V, 0.4A
36V→5V, 0.6A
750310564
15.24 × 13.3 × 11.43
63
450
3:1:1
115
50
Würth Elektronik
24V→±5V, 0.5A
750310799
9.14 × 9.78 × 10.54
25
125
1:1:0.33
60
74
Würth Elektronik
24V→15V, 0.4A
750370040
9.14 × 9.78 × 10.54
30
150
3:1:1
60
12.5
Würth Elektronik
24V→5V, 1A
750370041
9.14 × 9.78 × 10.54
50
450
3:1:1
190
26
Würth Elektronik
24V→5V, 1A
750370047
13.35 × 10.8 × 9.14
30
150
3:1:1
60
12.5
Würth Elektronik
24V→5V, 1A
750311681
17.75 × 13.46 × 12.70
100
3000
1:10
220
28500
Würth Elektronik
12V→300V, 5mA
L11-0059
9.52 × 9.52 × 4.95
24
3:1
266
266
BH Electronics
24V→5V, 1A
L10-1019
9.52 × 9.52 × 4.95
18
1:1
90
90
BH Electronics
5V→5V, 0.2A
3573fd
For more information www.linear.com/LT3573
11
LT3573
APPLICATIONS INFORMATION
Turns Ratio
Leakage Inductance
Note that when using an RFB /RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, simpler ratios of small integers, e.g., 1:1, 2:1,
3:2, etc., can be employed to provide more freedom in
setting total turns and mutual inductance.
Transformer leakage inductance (on either the primary or
secondary) causes a voltage spike to appear at the primary
after the output switch turns off. This spike is increasingly
prominent at higher load currents where more stored
energy must be dissipated. In most cases, a snubber
circuit will be required to avoid overvoltage breakdown at
the output switch node. Transformer leakage inductance
should be minimized.
Typically, the transformer turns ratio is chosen to maximize
available output power. For low output voltages (3.3V or 5V),
a N:1 turns ratio can be used with multiple primary windings
relative to the secondary to maximize the transformer’s
current gain (and output power). However, remember that
the SW pin sees a voltage that is equal to the maximum
input supply voltage plus the output voltage multiplied by
the turns ratio. This quantity needs to remain below the
ABS MAX rating of the SW pin to prevent breakdown of
the internal power switch. Together these conditions place
an upper limit on the turns ratio, N, for a given application.
Choose a turns ratio low enough to ensure:
N<
50V – VIN(MAX)
VOUT + VF
For larger N:1 values, a transformer with a larger physical
size is needed to deliver additional current and provide a
large enough inductance value to ensure that the off-time is
long enough to accurately measure the output voltage.
For lower output power levels, a 1:1 or 1:N transformer
can be chosen for the absolute smallest transformer size.
A 1:N transformer will minimize the magnetizing inductance (and minimize size), but will also limit the available
output power. A higher 1:N turns ratio makes it possible
to have very high output voltages without exceeding the
breakdown voltage of the internal power switch.
Linear Technology has worked with several magnetic
component manufacturers to produce predesigned flyback
transformers for use with the LT3573. Table 5 shows the
details of several of these transformers.
An RCD (resistor capacitor diode) clamp, shown in
Figure 4, is required for most designs to prevent the
leakage inductance spike from exceeding the breakdown
voltage of the power device. The flyback waveform is
depicted in Figure 5. In most applications, there will be a
very fast voltage spike caused by a slow clamp diode that
may not exceed 60V. Once the diode clamps, the leakage
inductance current is absorbed by the clamp capacitor.
This period should not last longer than 150ns so as not to
interfere with the output regulation, and the voltage during
this clamp period must not exceed 55V. The clamp diode
turns off after the leakage inductance energy is absorbed
and the switch voltage is then equal to:
VSW(MAX) = VIN(MAX) + N(VOUT + VF)
This voltage must not exceed 50V. This same equation
also determines the maximum turns ratio.
When choosing the snubber network diode, careful attention must be paid to maximum voltage seen by the SW
pin. Schottky diodes are typically the best choice to be
used in the snubber, but some PN diodes can be used if
they turn on fast enough to limit the leakage inductance
spike. The leakage spike must always be kept below 60V.
Figures 6 and 7 show the SW pin waveform for a 24VIN,
5VOUT application at a 1A load current. Notice that the
leakage spike is very high (more than 65V) with the “bad”
diode, while the “good” diode effectively limits the spike
to less than 55V.
3573fd
12
For more information www.linear.com/LT3573
LT3573
APPLICATIONS INFORMATION
LS
–
+
VSW
< 60V
C
R
< 55V
< 50V
D
t OFF > 350ns
tSP
3573 F04
Figure 4. RCD Clamp
3573 F05
TIME
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
10V/DIV
10V/DIV
100ns/DIV
3573 F06
Figure 6. Good Snubber Diode Limits SW Pin Voltage
100ns/DIV
3573 F07
Figure 7. Bad Snubber Diode Does Not Limit SW Pin Voltage
3573fd
For more information www.linear.com/LT3573
13
LT3573
APPLICATIONS INFORMATION
Secondary Leakage Inductance
In addition to the previously described effects of leakage
inductance in general, leakage inductance on the secondary in particular exhibits an additional phenomenon. It
forms an inductive divider on the transformer secondary
that effectively reduces the size of the primary-referred
flyback pulse used for feedback. This will increase the
output voltage target by a similar percentage. Note that
unlike leakage spike behavior, this phenomenon is load
independent. To the extent that the secondary leakage
inductance is a constant percentage of mutual inductance
(over manufacturing variations), this can be accommodated
by adjusting the RFB /RREF resistor ratio.
The Switch Current Limit vs RILIM plot in the Typical Performance Characteristics section depicts a more accurate
current limit.
Undervoltage Lockout (UVLO)
The SHDN/UVLO pin is connected to a resistive voltage
divider connected to VIN as shown in Figure 8. The voltage
threshold on the SHDN/UVLO pin for VIN rising is 1.22V.
To introduce hysteresis, the LT3573 draws 2.5µA from the
SHDN/UVLO pin when the pin is below 1.22V. The hysteresis
is therefore user-adjustable and depends on the value of
R1. The UVLO threshold for VIN rising is:
VIN(UVLO,RISING) =
1.22V •(R1+R2)
+ 2.5µA •R1
R2
Winding Resistance Effects
Resistance in either the primary or secondary will reduce
overall efficiency (POUT /PIN). Good output voltage regulation will be maintained independent of winding resistance
due to the boundary mode operation of the LT3573.
The UVLO threshold for VIN falling is:
Bifilar Winding
A bifilar, or similar winding technique, is a good way to
minimize troublesome leakage inductances. However, remember that this will also increase primary-to-secondary
capacitance and limit the primary-to-secondary breakdown
voltage, so, bifilar winding is not always practical. The
Linear Technology applications group is available and
extremely qualified to assist in the selection and/or design
of the transformer.
VIN(UVLO,FALLING) =
To implement external run/stop control, connect a small
NMOS to the UVLO pin, as shown in Figure 8. Turning the
NMOS on grounds the UVLO pin and prevents the LT3573
from operating, and the part will draw less than a 1µA of
quiescent current.
VIN
R1
SHDN/UVLO
Setting the Current Limit Resistor
R2
LT3573
The maximum current limit can be set by placing a resistor
between the RILIM pin and ground. This provides some
flexibility in picking standard off-the-shelf transformers that
may be rated for less current than the LT3573’s internal
power switch current limit. If the maximum current limit
is needed, use a 10k resistor. For lower current limits,
the following equation sets the approximate current limit:
1.22V •(R1+R2)
R2
RUN/STOP
CONTROL
(OPTIONAL)
GND
3573 F08
Figure 8. Undervoltage Lockout (UVLO)
3
RILIM = 65 •10 (1.6A −ILIM )+10k
3573fd
14
For more information www.linear.com/LT3573
LT3573
APPLICATIONS INFORMATION
Minimum Load Requirement
The LT3573 obtains output voltage information through
the transformer while the secondary winding is conducting
current. During this time, the output voltage (multiplied
times the turns ratio) is presented to the primary side of
the transformer. The LT3573 uses this reflected signal to
regulate the output voltage. This means that the LT3573
must turn on every so often to sample the output voltage,
which delivers a small amount of energy to the output.
This sampling places a minimum load requirement on the
output of 1% to 2% of the maximum load.
BIAS Pin Considerations
For applications with an input voltage less than 15V, the
BIAS pin is typically connected directly to the VIN pin. For
input voltages greater than 15V, it is preferred to leave the
BIAS pin separate form the VIN pin. In this condition, the
BIAS pin is regulated with an internal LDO to a voltage of
3V. By keeping the BIAS pin separate from the input voltage
at high input voltages, the physical size of the capacitors
can be minimized (the BIAS pin can then use a 6.3V or
10V rated capacitor).
If too large of an RC value is used, the part will be more
susceptible to high frequency noise and jitter. If too small
of an RC value is used, the transient performance will
suffer. The value choice for CC is somewhat the inverse
of the RC choice: if too small a CC value is used, the loop
may be unstable, and if too large a CC value is used, the
transient performance will also suffer. Transient response
plays an important role for any DC/DC converter.
Design Example
The following example illustrates the converter design
process using LT3573.
Given the input voltage of 20V to 28V, the required output
is 5V, 1A.
VIN(MIN) = 20V, VIN(MAX) = 28V, VOUT = 5V, VF = 0.5V
and IOUT = 1A
1. Select the transformer turns ratio to accommodate
the output.
The output voltage is reflected to the primary side by a
factor of turns ratio N. The switch voltage stress VSW is
expressed as:
Overdriving the BIAS Pin with a Third Winding
The LT3573 provides excellent output voltage regulation
without the need for an opto-coupler, or third winding, but
for some applications with higher input voltages (>20V),
it may be desirable to add an additional winding (often
called a third winding) to improve the system efficiency.
For proper operation of the LT3573, if a winding is used as
a supply for the BIAS pin, ensure that the BIAS pin voltage
is at least 3.15V and always less than the input voltage.
For a typical 24VIN application, overdriving the BIAS pin
will improve the efficiency gain 4-5%.
N=
VSW(MAX) = VIN +N(VOUT + VF ) < 50V
Or rearranged to:
N<
50 − VIN(MAX)
(VOUT + VF )
On the other hand, the primary-side current is multiplied by
the same factor of N. The converter output capability is:
Loop Compensation
The LT3573 is compensated using an external resistor-capacitor network on the VC pin. Typical values are in the range
of RC = 50k and CC = 1nF (see the numerous schematics in
the Typical Applications section for other possible values).
NP
1
NS
IOUT(MAX) = 0.8 •(1−D)•
D=
1
NIPK
2
N(VOUT + VF )
VIN +N(VOUT + VF )
3573fd
For more information www.linear.com/LT3573
15
LT3573
APPLICATIONS INFORMATION
The transformer turns ratio is selected such that the converter has adequate current capability and a switch stress
below 50V. Table 6 shows the switch voltage stress and
output current capability at different transformer turns
ratio.
Table 6. Switch Voltage Stress and Output Current Capability vs
Turns-Ratio
N
VSW(MAX) AT VIN(MAX)
(V)
IOUT(MAX) AT VIN(MIN)
(A)
DUTY CYCLE
(%)
1:1
33.5
0.39
16~22
2:1
39
0.65
28~35
3:1
44.5
0.825
37~45
4:1
50
0.96
44~52
BIAS winding turns ratio is selected to program the BIAS
voltage to 3V~5V. The BIAS voltage shall not exceed the
input voltage.
The turns ratio is then selected as primary: secondary:
BIAS = 3:1:1.
2. Select the transformer primary inductance for target
switching frequency.
The LT3573 requires a minimum amount of time to sample
the output voltage during the off-time. This off-time,
tOFF(MIN), shall be greater than 350ns over all operating
conditions. The converter also has a minimum current limit,
LMIN, of 250mA to help create this off-time. This defines
the minimum required inductance as defined as:
LMIN =
N(VOUT + VF )
• tOFF(MIN)
IMIN
The transformer primary inductance also affects the
switching frequency which is related to the output ripple. If
above the minimum inductance, the transformer’s primary
inductance may be selected for a target switching frequency
range in order to minimize the output ripple.
The following equation estimates the switching frequency.
fSW =
1
1
=
I
IPK
tON + tOFF
PK
+
VIN
NPS (VOUT + VF )
L
L
Table 7.Switching Frequency at Different Primary
Inductance at IPK
L (µH)
fSW AT VIN(MIN)
(kHz)
fSW AT VIN(MAX)
(kHz)
25
236
305
50
121
157
100
61
80
Note: The switching frequency is calculated at maximum output.
In this design example, the minimum primary inductance is
used to achieve a nominal switching frequency of 275kHz
at full load. The PA2454NL from Pulse Engineering is
chosen as the flyback transformer.
Given the turns ratio and primary inductance, a customized transformer can be designed by magnetic component
manufacturer or a multi-winding transformer such as a
Coiltronics Versa-Pac may be used.
3. Select the output diodes and output capacitor.
The output diode voltage stress VD is the summation of
the output voltage and reflection of input voltage to the
secondary side. The average diode current is the load
current.
VD = VOUT +
VIN
N
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in
size and cost of a larger capacitor. The following equation
calculates the output voltage ripple.
LI 2PK
DVMAX =
2CVOUT
4. Select the snubber circuit to clamp the switch
voltage spike.
A flyback converter generates a voltage spike during switch
turn-off due to the leakage inductance of the transformer.
In order to clamp the voltage spike below the maximum
rating of the switch, a snubber circuit is used. There are
many types of snubber circuits, for example R-C, R-C-D and
3573fd
16
For more information www.linear.com/LT3573
LT3573
APPLICATIONS INFORMATION
Zener clamps. Among them, RCD is widely used. Figure 9
shows the RCD snubber in a flyback converter.
RTC resistor for temperature compensation of the output
voltage. RREF is selected as 6.04k.
A typical switch node waveform is shown in Figure 10.
A small capacitor in parallel with RREF filters out the noise
during the voltage spike, however, the capacitor should
limit to 10pF. A large capacitor causes distortion on voltage sensing.
During switch turn-off, the energy stored in the leakage
inductance is transferred to the snubber capacitor, and
eventually dissipated in the snubber resistor.
V (V −N• VOUT )
1 2
LS I PK fSW = C C
R
2
The snubber resistor affects the spike amplitude VC and
duration tSP, the snubber resistor is adjusted such that
tSP is about 150ns. Prolonged tSP may cause distortion
to the output voltage sensing.
The previous steps finish the flyback power stage design.
5. Select the feedback resistor for proper output
voltage.
Using the resistor Tables 1-4, select the feedback resistor RFB, and program the output voltage to 5V. Adjust the
6. Optimize the compensation network to improve the
transient performance.
The transient performance is optimized by adjusting the
compensation network. For best ripple performance, select
a compensation capacitor not less than 1nF, and select a
compensation resistor not greater than 50k.
7. Current limit resistor, soft-start capacitor and UVLO
resistor divider
Use the current limit resistor RLIM to lower the current
limit if a compact transformer design is required. Soft-start
capacitor helps during the start-up of the flyback converter.
Select the UVLO resistor divider for intended input operation range. These equations are aforementioned.
LS
–
+
C
R
VC
NVOUT
D
VIN
tSP
3573 F10
3573 F09
Figure 9. RCD Snubber in a Flyback Converter
Figure 10. Typical Switch Node Waveform
3573fd
For more information www.linear.com/LT3573
17
LT3573
TYPICAL APPLICATIONS
Low Input Voltage 5V Isolated Flyback Converter
VIN
5V
D1
3:1
C1
10µF
R1
200k
C6
0.22µF
VIN
R8 T1
2k 24µH
2.6µH
SHDN/UVLO
R2
90.9k
LT3573
RFB
RREF
R3
80.6k
VOUT+
5V, 350mA
C5
47µF
VOUT–
D2
R4
6.04k
TC
SW
RILIM
SS
VC
R6
28.7k
R5
10k
GND TEST BIAS
R7
57.6k
C3
1000pF
C2
10nF
VIN
3573 TA02
T1: PULSE PA2454NL OR WÜRTH ELEKTRONIK 750310471
D1: B340A
D2: PMEG6010
C5: MURATA, GRM32ER71A476K
±12V Isolated Flyback Converter
VIN
5V
D1
2:1:1
C1
10µF
R1
200k
C6
0.22µF
VIN
R8
T1
2k 43.6µH
10.9µH
SHDN/UVLO
R2
90.9k
LT3573
RFB
RREF
R3
118k
D2
D3
10.9µH
R4
6.04k
TC
VOUT1+
12V, 100mA
C5
47µF
VOUT 1–
VOUT2+
C6
47µF
VOUT 2–
–12V, 100mA
SW
RILIM
SS
VC
R6
59k
R5
10k
C2
10nF
GND TEST BIAS
R7
56.2k
C3
3300pF
VIN
T1: COILTRONICS VPH1-0076-R
D1, D2: B240A
D3: PMEG6010
C5, C6: MURATA, GRM32ER71A476K
3573 TA03
3573fd
18
For more information www.linear.com/LT3573
LT3573
TYPICAL APPLICATIONS
5V Isolated Flyback Converter
VIN
12V TO
24V
(*30V)
3:1:1
C1
10µF
R1
499k
C6
0.22µF
VIN
R2
71.5k
R8 T1
2k 24µH
D1
2.6µH
VOUT +
5V, 700mA
C5
47µF
SHDN/UVLO
LT3573
R3
80.6k
RFB
RREF
VOUT –
D3
R4
6.04k
TC
RILIM
SW
SS
VC
R6
28.7k
R5
10k
GND TEST BIAS
D2
R7
45.3k
C2
10nF
C4
4.7µF
C3
1000pF
L1C
2.6µH
*OPTIONAL THIRD
WINDING FOR
30V OPERATION
3573 TA04
T1: PULSE PA2454NL
OR WÜRTH ELEKTRONIK 750370047
D1: B340A
D3: PMEG6010
C5: MURATA, GRM32ER71A476K
Efficiency
90
VIN = 24V
80
EFFICIENCY (%)
70
VIN = 12V
60
50
40
30
20
10
0
0
200
400
600
800 1000 1200 1400
IOUT (mA)
3573 TA04b
3573fd
For more information www.linear.com/LT3573
19
LT3573
TYPICAL APPLICATIONS
3.3V Isolated Flyback Converter
VIN
12V TO 24V
(*30V)
4:1:1
C1
10µF
R1
499k
C6
0.22µF
VIN
R8 T1
2k 24µH
D1
1.5µH
VOUT +
3.3V, 1A
C5
47µF
SHDN/UVLO
R2
71.5k
LT3573
R3
76.8k
RFB
RREF
VOUT –
D3
R4
6.04k
TC
RILIM
SW
SS
GND TEST BIAS
VC
R6
19.1k
R5
10k
D2
R7
25.5k
C3
1500pF
C2
10nF
C4
4.7µF
L1C
1.5µH
*OPTIONAL THIRD
WINDING FOR
30V OPERATION
3573 TA05
T1: PULSE PA2362NL
OR COILCRAFT GA3429-BL
D1: B340A
D3: PMEG6010
12V Isolated Flyback Converter
VIN 5V
3:1
C1
10µF
R1
499k
R2
71.5k
SHDN/UVLO
C6
0.22µF
VIN
LT3573
RFB
RREF
TC
RILIM
SS
VC
R6
59k
R5
10k
C2
10nF
3573 TA06
R3
178k
R8
T1
2k 58.5µH
D2
D1
6.5µH
VOUT
12V, 400mA
C5
47µF
VOUT–
R4
6.04k
SW
GND TEST BIAS
R7
40.2k
C3
4700pF
VIN
T1: COILTRONICS VP1-0102-R
D1: B340A
D2: PMEG6010
3573fd
20
For more information www.linear.com/LT3573
LT3573
TYPICAL APPLICATIONS
Four Output 12V Isolated Flyback Converter
VIN
12V TO
24V
D1
2:1:1:1:1
C1
10µF
R1
499k
C6
0.22µF
VIN
LT3573
RFB
RREF
TC
VC
C2
10nF
R3
118k
VOUT2+
12V, 60mA
VOUT 2–
D3
R4
6.04k
VOUT3+
12V, 60mA
C7
47µF
VOUT 3–
10.9µH
GND TEST BIAS
R7
20k
C3
0.01µF
C6
47µF
10.9µH
SW
SS
R5
10k
VOUT 1–
D5
RILIM
R6
59k
C5
47µF
10.9µH
D2
SHDN/UVLO
R2
71.5k
T1
R8
2k 43.6µH
VOUT1+
12V, 60mA
D4
VIN
VOUT4+
12V, 60mA
C8
47µF
10.9µH
VOUT 4–
T1: COILTRONICS VPH1-0076-R
D1-D4: B240A
D5: PMEG6010
3573 TA07
5V Isolated Flyback Converter Using a Tiny Transformer
VIN 12V
3:1
C1
10µF
R1
200k
R2
90.9k
SHDN/UVLO
C6
0.22µF
VIN
LT3573
RFB
RREF
TC
RILIM
SS
VC
R6
28.7k
R5
30k
C2
10nF
3573 TA08
R8
2k
R3
80.6k
D2
T1
20µH
D1
2.2µH
VOUT
5V, 600mA
C5
47µF
VOUT–
R4
6.04k
SW
GND TEST BIAS
R7
47.5k
C3
1000pF
VIN
T1: BH ELECTRONICS L11-0059
D1: B340A
D2: PMEG6010
3573fd
For more information www.linear.com/LT3573
21
LT3573
TYPICAL APPLICATIONS
5V Isolated Flyback Converter Using Coupling Inductor
VIN
5V
1:1
C1
10µF
R1
200k
R2
90.9k
C6
0.22µF
VIN
SHDN/UVLO
LT3573
RFB
RREF
TC
RILIM
SS
VC
R6
26.1k
R5
10k
C2
10nF
3573 TA09
R8
T1
2k 23.6µH
R3
26.1k
D2
D1
23.6µH
VOUT+
5V, 0.2A
C5
47µF
VOUT–
R4
6.04k
SW
GND TEST BIAS
R7
56.2k
C3
1500pF
VIN
T1: BH ELECTRONICS, L10-1022
D1: B220A
D2: CMD5H-3
3573fd
22
For more information www.linear.com/LT3573
LT3573
TYPICAL APPLICATIONS
300V Isolated Flyback Converter
VIN
6V TO 15V
1:10
C1
10µF
R1
100k
C6
0.22µF
VIN
SHDN/UVLO
R2
36k
LT3573
RFB
RREF
TC
RILIM
SS
VC
R6
20.5k
R5
10k
C2
10nF
3573 TA10
R8
1k
C8
100pF
R3
150k
D2
T1
100µH
D1
VOUT+
300V, 5mA
1M
C5, 500V,
0.056µF
×4
VOUT–
R4
6.04k
SW
GND TEST BIAS
R7
25k
C3
2200pF
VIN
T1: WÜRTH ELEKTRONIK 750311681
D1: CMMRIF-06
D2: CMMSHI-60
3573fd
For more information www.linear.com/LT3573
23
LT3573
PACKAGE DESCRIPTION
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev A)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ± 0.102
(.112 ± .004)
5.23
(.206)
MIN
2.845 ± 0.102
(.112 ± .004)
0.889 ± 0.127
(.035 ± .005)
8
1
1.651 ± 0.102
(.065 ± .004)
1.651 ± 0.102 3.20 – 3.45
(.065 ± .004) (.126 – .136)
0.305 ± 0.038
(.0120 ± .0015)
TYP
16
0.50
(.0197)
BSC
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ± 0.076
(.011 ± .003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.86
(.034)
REF
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE16) 0608 REV A
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24
For more information www.linear.com/LT3573
LT3573
REVISION HISTORY
REV
DATE
DESCRIPTION
B
10/09
Replaced Figure 1
(Revision history begins at Rev B)
PAGE NUMBER
10
Updated Typical Applications drawings
C
D
07/10
12/13
18, 21, 22
Added patent numbers and revised Typical Application drawing
1
Revised D1 on Block Diagram
6
Revised Table 5
11
Revised Figure 4 in Applications Information section
13
Revised all drawings in Typical Applications section
18-23
Replaced Related Parts list
26
Added reference to Note 1
2
Changed flyback error amp input to 1.23V
6
Changed Table 6 IOUT(MAX)
16
Changed VIN(MAX) from 40V to 30V
20
3573fd
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT3573
25
LT3573
TYPICAL APPLICATION
9V to 30VIN, +5V/–5VOUT Isolated Flyback Converter
T1
3:1:1:1
VIN
9V TO 30V
C1
10µF
R1
357k
SHDN/UVLO
R2
51.1k
LT3573
RFB
C5
47µF
COM
C6
47µF
D2
VOUT –
–5V, 350mA
R4
6.04k
TC
SW
SS
VC
C2
10nF
VOUT +
+5V, 350mA
L1B
7µH
L1C
7µH
R3
80.6k
RILIM
R6
10k
L1A
63µH
D4
RREF
R5
28.7k
R8
2k
C6
0.22µF
VIN
D1
GND TEST BIAS
R7
23.7k
C3
2700pF
D3
C4
4.7µF
*OPTIONAL THIRD
WINDING FOR
>24V OPERATION
L1D
7µH
T1: WÜRTH ELEKTRONIK 750310564
3573 TA11
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT8300
100VIN Micropower Isolated Flyback Converter with
150V/260mA Switch
Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT23
LT3574/LT3575
40V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 0.65A/2.5A Switch
LT3511/LT3512
100V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 240mA/420mA
Switch, MSOP-16(12)
LT3748
100V Isolated Flyback Controller
5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12)
LT3798
Off-Line Isolated No-Opto Flyback Controller with Active PFC
VIN and VOUT Limited Only by External Components
LT3757A/LT3759
LT3758
40V/100V Flyback/Boost Controllers
Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958
40V/80V Boost/Flyback Converters
Monolithic with Integrated 5A/3.3A Switch
LTC®3803/LTC3803-3 200kHz/300kHz Flyback Controller in SOT-23
LTC3803-5
VIN and VOUT Limited Only by External Components
LTC3805/LTC3805-5
VIN and VOUT Limited Only by External Components
Adjustable Frequency Flyback Controllers
3573fd
26
Linear Technology Corporation
LT 1213 REV D • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
For more information www.linear.com/LT3573
www.linear.com/LT3573
LINEAR TECHNOLOGY CORPORATION 2008
Mouser Electronics
Authorized Distributor
Click to View Pricing, Inventory, Delivery & Lifecycle Information:
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