LT3791-1
60V 4-Switch Synchronous
Buck-Boost Controller
Features
Description
4-Switch Single Inductor Architecture Allows VIN
Above, Below or Equal to VOUT
n Synchronous Switching: Up to 98.5% Efficiency
n Wide V Range: 4.7V to 60V
IN
n 2% Output Voltage Accuracy: 1.2V ≤ V
OUT < 60V
n 6% Output Current Accuracy: 0V ≤ V
OUT < 60V
n Input and Output Current Regulation with Current
Monitor Outputs
n No Top FET Refresh in Buck or Boost
n V
OUT Disconnected from VIN During Shutdown
n C/10 Charge Termination and Output Shorted Flags
n Capable of 100W or greater per IC
n 38-Lead TSSOP with Exposed Pad
The LT®3791-1 is a synchronous 4-switch buck-boost voltage/current regulator controller. The controller can regulate
output voltage, output current, or input current with input
voltages above, below, or equal to the output voltage. The
constant-frequency, current mode architecture allows its
frequency to be adjusted or synchronized from 200kHz
to 700kHz. No top FET refresh switching cycle is needed
in buck or boost operation. With 60V input, 60V output
capability and seamless transitions between operating
regions, the LT3791-1 is ideal for voltage regulator, battery/super-capacitor charger applications in automotive,
industrial, telecom, and even battery-powered systems.
n
For new designs we recommend the LT8390: 60V synchronous 4-switch buck-boost controller due to its numerous
performance improvements over the LT3791-1.
Applications
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Analog
Devices, Inc. All other trademarks are the property of their respective owners.
Automotive, Telecom, Industrial Systems
n High Power Battery-Powered System
n
Typical Application
120W (24V 5A) Buck-Boost Voltage Regulator
0.003Ω
VIN
1µF
50Ω
IVINP
TEST1
499k
0.1µF
TG1
0.1µF
SWI
OVLO
56.2k
BG1
INTVCC
27.4k
100k
LT3791-1
200k
47µF
80V
220µF
35V
0.015Ω
4.7µF
50V
×2
10µH
3.83k
SNSN
PGND
CTRL SS SYNC V
C
RT
5.1k
33nF
10nF
SGND
100
95
90
85
80
75
70
VIN = 12V
VIN = 24V
VIN = 54V
65
BG2
SW2
TG2
ISP
ISN
FB
Efficiency vs Load Current
VOUT
24V
5A
73.2k
0.004Ω
PWM
100k
+
SNSP
SHORT
C/10
CCM
IVINMON
ISMON
CLKOUT
PWMOUT
VREF
0.1µF
+
BST1
EN/UVLO
4.7µF
100V
4.7µF
BST2
IVINN
470nF
499k
INTVCC
EFFICIENCY (%)
VIN
12V TO
58V
60
0
1
2
3
LOAD CURRENT (A)
4
5
37911 TA01b
37911 TA01a
147k
200kHz
37911fb
For more information www.linear.com/LT3791-1
1
LT3791-1
Absolute Maximum Ratings
(Note 1)
Supply Voltages
Input Supply (VIN)......................................................60V
SW1, SW2.......................................................–1V to 60V
C/10, SHORT..............................................................15V
EN/UVLO, IVINP, IVINN, ISP, ISN...............................60V
INTVCC, (BST1-SW1), (BST2-SW2)..............................6V
CCM, SYNC, RT, CTRL, OVLO, PWM...........................6V
IVINMON, ISMON, FB, SS, VC, VREF............................6V
IVINP-IVINN, ISP-ISN, SNSP-SNSN........................±0.5V
SNSP, SNSN............................................................±0.3V
Operating Junction Temperature (Notes 2, 3)
LT3791E-1/LT3791I-1.......................... –40°C to 125°C
LT3791H-1.......................................... –40°C to 150°C
LT3791MP-1....................................... –55°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................... 300°C
Pin Configuration
TOP VIEW
CTRL
1
38 OVLO
SS
2
37 FB
PWM
3
36 VC
C/10
4
35 RT
SHORT
5
34 SYNC
VREF
6
33 CLKOUT
ISMON
7
32 CCM
IVINMON
8
31 PWMOUT
EN/UVLO
9
IVINP 10
30 SGND
39
SGND
29 TEST1
IVINN 11
28 SNSN
VIN 12
27 SNSP
INTVCC 13
26 ISN
TG1 14
25 ISP
BST1 15
24 TG2
SW1 16
23 NC
PGND 17
22 BST2
BG1 18
21 SW2
BG2 19
20 PGND
FE PACKAGE
38-LEAD PLASTIC TSSOP
TJMAX = 150°C, θJA = 28°C/W
EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB
Order Information
http://www.linear.com/product/LT3791-1#orderinfo
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3791EFE-1#PBF
LT3791EFE-1#TRPBF
LT3791FE-1
38-Lead Plastic TSSOP
–40°C to 125°C
LT3791IFE-1#PBF
LT3791IFE-1#TRPBF
LT3791FE-1
38-Lead Plastic TSSOP
–40°C to 125°C
LT3791HFE-1#PBF
LT3791HFE-1#TRPBF
LT3791FE-1
38-Lead Plastic TSSOP
–40°C to 150°C
LT3791MPFE-1#PBF
LT3791MPFE-1#TRPBF
LT3791FE-1
38-Lead Plastic TSSOP
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
Electrical
Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input
VIN Operating Voltage
4.7
60
V
VIN Shutdown IQ
VEN/UVLO = 0V
0.1
1
µA
VIN Operating IQ (Not Switching)
FB = 1.3V, RT = 59.0k
3.0
4
mA
37911fb
2
For more information www.linear.com/LT3791-1
LT3791-1
Electrical
Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
1.16
1.2
1.24
UNITS
Logic Inputs
EN/UVLO Falling Threshold
l
EN/UVLO Rising Hysteresis
15
EN/UVLO Input Low Voltage
IVIN Drops Below 1µA
EN/UVLO Pin Bias Current Low
VEN/UVLO = 1V
EN/UVLO Pin Bias Current High
VEN/UVLO = 1.6V
2
CCM Threshold Voltage
CTRL Input Bias Current
0.3
V
3
4
µA
10
100
nA
1.5
V
0.3
VCTRL = 1V
20
CTRL Latch-Off Threshold
50
175
OVLO Rising Shutdown Voltage
l
2.85
OVLO Falling Hysteresis
V
mV
3
nA
mV
3.15
75
V
mV
Regulation
VREF Voltage
l
VREF Line Regulation
4.7V < VIN < 60V
V(ISP-ISN) Threshold
VCTRL = 2V
1.96
2.00
2.04
V
0.002
0.04
%/V
l
97.5
94
100
100
102.5
106
mV
mV
l
87
84
90
90
93
96
mV
mV
l
47.5
46
50
50
52.5
54
mV
mV
l
6.5
5
10
10
13.5
15
mV
mV
VCTRL = 1100mV
VCTRL = 700mV
VCTRL = 300mV
ISP Bias Current
110
µA
ISN Bias Current
20
µA
Output Current Sense Common Mode Range
0
Output Current Sense Amplifier gm
60
890
ISMON Monitor Voltage
V(ISP-ISN) = 100mV
l
0.96
1
1.04
Input Current Sense Threshold V(IVINP-IVINN)
3V ≤ VIVINP ≤ 60V
l
46.5
50
54
IVINP Bias Current
90
IVINN Bias Current
20
Input Current Sense Common Mode Range
3
Input Current Sense Amplifier gm
IVINMON Monitor Voltage
µA
60
mS
0.96
1
1.04
V
l
1.194
1.176
1.2
1.2
1.206
1.220
V
V
0.002
0.025
%/V
4.7V < VIN < 60V
FB Amplifier gm
565
FB Pin Input Bias Current
FB in Regulation
VC Standby Input Bias Current
PWM = 0V
VSENSE(MAX) (VSNSP-SNSN)
Boost
Buck
V
l
FB Regulation Voltage
FB Line Regulation
V
mV
µA
2.12
V(IVINP-IVINN) = 50mV
V
µS
100
–20
l
l
42
–56
51
–47.5
µS
200
nA
20
nA
60
–39
mV
mV
Fault
SS Pull-Up Current
VSS = 0V
SS Discharge Current
14
µA
1.4
µA
37911fb
For more information www.linear.com/LT3791-1
3
LT3791-1
Electrical
Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
C/10 Rising Threshold (VFB)
V(ISP-ISN) = 0V
C/10 Falling Threshold (VFB)
C/10 Falling Threshold (V(ISP-ISN))
VFB = 1.2V
MIN
TYP
MAX
UNITS
l
1.127
1.15
1.173
V
l
1.078
1.1
1.122
V
5
10
15
mV
380
400
450
mV
C/10 Pin Output Impedance
1.1
2.0
kΩ
SHORT Pin Output Impedance
1.1
2.0
kΩ
SS Latch-Off Threshold
1.75
V
SS Reset Threshold
0.2
V
SHORT Falling Threshold (VFB)
Oscillator
Switching Frequency
RT = 147k
RT = 59.0k
RT = 29.1k
190
380
665
SYNC Frequency
200
400
700
200
SYNC Pin Resistance to GND
210
420
735
kHz
kHz
kHz
700
kHz
90
SYNC Threshold Voltage
0.3
kΩ
1.5
V
5
5.2
V
240
350
mV
3.5
3.9
V
Internal VCC Regulator
INTVCC Regulation Voltage
Dropout (VIN – INTVCC)
4.8
IINTVCC = –10mA, VIN = 5V
INTVCC Undervoltage Lockout
INTVCC Current Limit
3.1
VINTVCC = 4V
67
mA
PWM
PWM Threshold Voltage
0.3
1.5
V
PWM Pin Resistance to GND
90
kΩ
PWMOUT Pull-Up Resistance
10
20
Ω
PWMOUT Pull-Down Resistance
5
10
Ω
NMOS Drivers
TG1, TG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
VBST – VSW = 5V
BG1, BG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
VINTVCC = 5V
TG Off to BG On Delay
BG Off to TG On Delay
TG1, TG2, tOFF(MIN)
2.6
1.7
Ω
Ω
3
1.2
Ω
Ω
CL = 3300pF
60
ns
CL = 3300pF
60
ns
RT = 59.0k
240
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3791E-1 is guaranteed to meet performance from
0°C to 125°C junction temperature. Specification over the -40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls.
The LT3791I-1 is guaranteed to meet performance specifications over the
–40°C to 125°C operating junction temperature range. The LT3791H-1 is
guaranteed to meet performance specifications over the –40°C to 150°C
320
ns
operating junction temperature range. The LT3791MP-1 is guaranteed
to meet performance specifications over the –55°C to 150°C operating
junction temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated for junction temperatures greater
than 125°C.
Note 3: The LT3791-1 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified absolute maximum operating junction temperature may
impair device reliability.
37911fb
4
For more information www.linear.com/LT3791-1
LT3791-1
Typical Performance Characteristics
INTVCC Dropout Voltage
vs Current, Temperature
80
5.15
5.10
1.5
INTVCC (V)
VIN-VINTVCC (V)
90
5.20
TA = 150°C
TA = 25°C
TA = –50°C
2.0
INTVCC Current Limit
vs Temperature
INTVCC Voltage vs Temperature
INTVCC CURRENT LIMIT (mA)
2.5
TA = 25°C, unless otherwise noted.
1.0
5.05
VIN = 60V
5.00
VIN = 12V
4.95
4.90
0.5
4.85
0
20
10
0
30
LDO CURRENT (mA)
0
37911 G01
50
40
30
20
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
VREF Voltage vs Temperature
VREF Load Regulation
5.75
2.03
2.15
5.50
2.02
2.10
5.25
2.01
2.05
2.00
4.75
1.99
4.50
1.98
4.25
10
20
30
40
50
60
1.96
–50 –25
70
1.80
25 50 75 100 125 150
TEMPERATURE (°C)
80
V(ISP-ISN) (mV)
70
60
50
40
30
20
10
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
VCTRL (V)
37911 G07
0
50
V(ISP-ISN) Threshold
vs Temperature
108
108
106
106
104
104
102
100
98
100
98
96
94
94
0
10
20
40
30
VISP (V)
50
60
37911 G08
VIN = 12V
102
96
92
100 150 200 250 300 350 400
IREF (µA)
37911 G06
V(ISP-ISN) Threshold vs VISP
90
V(ISP-ISN) (mV)
1.85
37911 G05
V(ISP-ISN) Threshold vs VCTRL
0
0
37911 G04
100
0
1.90
VIN = 60V
VIN = 12V
VIN = 4.7V
V(ISP-ISN) (mV)
0
2.00
1.95
1.97
ILOAD (mA)
110
VREF (V)
2.20
VREF (V)
2.04
4.00
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G03
6.00
5.00
0
37911 G02
INTVCC Load Regulation
INTVCC (V)
60
10
4.80
–50 –25
40
70
92
–50 –25
VISP = 60V
VISP = 12V
VISP = 0V
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G09
37911fb
For more information www.linear.com/LT3791-1
5
LT3791-1
Typical Performance Characteristics
120
V(ISP-ISN) Threshold vs VFB
TA = 25°C, unless otherwise noted.
ISMON Voltage vs Temperature
1.04
VIN = 12V
V(ISP-ISN) = 100mV
1.03
100
1.0
0.9
0.8
60
40
0.7
1.01
VISMON (V)
VISMON (V)
V(ISP-ISN) (mV)
1.02
80
1.00
0.99
1.19
1.18
1.20 1.21
VFB (V)
1.22
0
25 50 75 100 125 150
TEMPERATURE (°C)
V(IVINP-IVINN) Threshold
vs VIVINP
1.04
54
51.5
1.03
51.0
1.02
50.5
1.01
50
VIVINP = 3V
48
46
44
42
–50 –25
0
50.0
49.5
48.5
0.97
10
20
30
40
VIVINP (V)
37911 G13
50
50
40
0.500
1.23
0.475
1.22
0.450
VFB (V)
1.20
1.18
10
1.20 1.21
VFB (V)
1.22
1.23
37911 G16
1.16
–50 –25
RISING
0.425
0.400
FALLING
0.375
0.350
VIN = 60V
VIN = 12V
VIN = 4.7V
1.17
1.19
37911 G15
1.24
1.19
20
25 50 75 100 125 150
TEMPERATURE (°C)
SHORT Threshold
vs Temperature
1.21
30
0
37911 G14
FB VOLTAGE (V)
60
1.18
0.96
–50 –25
60
FB Regulation Voltage
vs Temperature
V(IVINP-IVINN) Threshold vs VFB
0
1.17
0.99
0.98
0
VIVINP = 12V
V(IVINP-VINN) = 50mV
1.00
49.0
48.0
25 50 75 100 125 150
TEMPERATURE (°C)
VIVINMON (V)
52.0
VIVINP = 60V
10 20 30 40 50 60 70 80 90 100
V(ISP-ISN) (mV)
IVINMON Voltage vs Temperature
56
52
0
37911 G12
37911 G11
V(IVINP-IVINN) (mV)
V(IVINP-IVINN) (mV)
0
37911 G10
V(IVINP-IVINN) Threshold
vs Temperature
V(IVINP-IVINSN) (mV)
0.4
0.1
0.96
–50 –25
1.23
0.5
0.2
0.97
0
1.17
0.6
0.3
0.98
20
ISMON Voltage vs V(ISP-ISN)
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G17
0.325
0.300
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G18
37911fb
6
For more information www.linear.com/LT3791-1
LT3791-1
Typical Performance Characteristics
C/10 Threshold vs Temperature
OVLO Threshold vs Temperature
1.125
FALLING
1.100
1.075
1.050
1.025
1.000
–50 –25
0
16
3.2
14
3.1
12
RISING
3.0
2.9
FALLING
2.7
4
2.6
2
0
Supply Current vs Input Voltage
7
IQ (mA)
2.0
1.5
1.0
TA = 150°C
TA = 25°C
TA = –50°C
0
0
10
20
30
VIN (V)
40
50
60
5
4
3
2
RT = 59.0k
RT = 147k
200
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G25
RISING
1.20
1.18
FALLING
1.16
1.14
1.12
0
–50 –25
1.10
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G23
TG1, TG2 MINIMUM ON-TIME (ns)
SWITCHING FREQUENCY (kHz)
600
300
1.22
37911 G24
TG1, TG2 Minimum On-Time
vs Temperature
RT = 29.1k
400
1.26
1.24
1
100
500
1.28
37911 G22
800
0
–50 –25
VEN/ULO = 1V
6
Oscillator Frequency
vs Temperature
700
EN/UVLO Threshold Voltage
1.30
EN/UVLO THRESHOLD (V)
EN/UVLO PIN CURRENT (µA)
3.5
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G21
EN/UVLO Pin Current
8
2.5
0
37911 G20
4.0
3.0
DISCHARGING
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G19
0.5
8
6
2.5
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
CHARGING
10
2.8
TG1, TG2 Minimum Off-Time
vs Temperature
350
TG2
90
TG1, TG2 MINIMUM OFF-TIME (ns)
FB VOLTAGE (V)
1.150
OVLO THRESHOLD (V)
RISING
Soft-Start Current vs Temperature
3.3
ISS (µA)
1.200
1.175
TA = 25°C, unless otherwise noted.
TG1
80
70
60
50
40
30
20
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G26
300
fSW = 200kHz
250
fSW = 400kHz
200
fSW = 700kHz
150
100
50
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G27
37911fb
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7
LT3791-1
Typical Performance Characteristics
BG1, BG2 Driver On-Resistance
vs Temperature
4.5
3.8
4.0
RISING
3.7
3.6
3.5
3.4
FALLING
3.3
3.2
4.0
3.5
PULL-UP
3.5
3.0
2.5
2.0
PULL-DOWN
1.5
1.0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
37911 G28
PWMOUT RESISTANCE (Ω)
14
1.6
12
1.4
1.0
25 50 75 100 125 150
TEMPERATURE (°C)
V(SNSP-SNSN) Buck Threshold
vs VC
VC (V)
60
40
BG2
1.0
PULL-DOWN
0
37911 G30
V(SNSP-SNSN) = 0V
1.2
PULL-UP
6
1.5
VC Voltage vs Duty Cycle
8
PULL-DOWN
2.0
37911 G29
PWMOUT On-Resistance
vs Temperature
10
2.5
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
V(SNSP-SNSN) (mV)
0
PULL-UP
3.0
0.5
0.5
3.1
–50 –25
BG1
0.8
0.6
4
20
0
–20
0.4
2
–40
0.2
0
–50 –25
0
25
50
0
75
100 125 150
TEMPERATURE (°C)
0
20
40
60
DUTY CYCLE (%)
80
V(SNSP-SNSN) Buck Threshold
vs Temperature
V(SNSP-SNSN) (mV)
20
0
–20
VC(MAX)
–60
–50 –25
0
60
60
40
40
20
0
–20
–40
–60
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G34
–80
0.6
0.8
1.0
1.2
VC (V)
1.4
1.0
0.8
1.2
VC (V)
1.4
1.6
1.8
37911 G33
V(SNSP-SNSN) Boost Threshold
vs Temperature
V(SNSP-SNSN) THRESHOLD (mV)
VC(MIN)
–40
0.6
V(SNSP-SNSN) Boost Threshold
vs VC
60
40
–60
100
37911 G32
37911 G31
V(SNSP-SNSN) THRESHOLD (mV)
TG1, TG2 Driver On-Resistance
vs Temperature
TG1, TG2 RESISTANCE (Ω)
3.9
BG1, BG2 RESISTANCE (Ω)
V(BST1-SW1), V(BST2,SW2) (V)
V(BST1-SW1), V(BST2-SW2) UVLO
vs Temperature
TA = 25°C, unless otherwise noted.
1.6
1.8
37911 G35
VC(MAX)
20
0
–20
–40
–60
–80
–50 –25
VC(MIN)
0
25 50 75 100 125 150
TEMPERATURE (°C)
37911 G36
37911fb
8
For more information www.linear.com/LT3791-1
LT3791-1
Pin Functions
CTRL (Pin 1): Output Current Sense Threshold Adjustment
Pin. Regulating threshold V(ISP-ISN) is 1/10th of (VCTRL –
200mV). CTRL linear range is from 200mV to 1.1V. For
VCTRL > 1.3V, the current sense threshold is constant at
the full-scale value of 100mV. For 1.1V < VCTRL < 1.3V, the
dependence of the current sense threshold upon VCTRL transitions from a linear function to a constant value, reaching
98% of full scale by VCTRL = 1.2V. Connect CTRL to VREF
for the 100mV default threshold. Force less than 175mV
(typical) to stop switching. Do not leave this pin open.
EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate
1.2V falling threshold with an externally programmable
hysteresis is generated by the external resistor divider
and a 3µA pull-down current. Above the 1.2V (typical)
threshold (but below 6V), EN/UVLO input bias current is
sub-µA. Below the falling threshold, a 3µA pull-down current is enabled so the user can define the hysteresis with
the external resistor selection. An undervoltage condition
resets soft-start. Tie to 0.3V, or less, to disable the device
and reduce VIN quiescent current below 1µA.
SS (Pin 2): Soft-start reduces the input power sources
surge current by gradually increasing the controller’s current limit. A minimum value of 22nF is recommended on
this pin. A 100k resistor must be placed between SS and
VREF for the LT3791-1.
IVINP (Pin 10): Positive Input for the Input Current Limit
and Monitor. Input bias current for this pin is typically 90µA.
PWM (Pin 3): A signal low turns off switches, idles switching and disconnects the VC pin from all external loads. The
PWMOUT pin follows the PWM pin. PWM has an internal
90k pull-down resistor. If not used, connect to INTVCC.
VIN (Pin 12): Main Input Supply. Bypass this pin to PGND
with a capacitor.
C/10 (Pin 4): C/10 Charge Termination Pin. An open-drain
pull-down on C/10 asserts if FB is greater than 1.15V (typical) and V(ISP-ISN) is less than 10mV (typical). To function,
the pin requires an external pull-up resistor.
SHORT (Pin 5): Output Shorted Pin. An open-drain pulldown on SHORT asserts if FB is less than 400mV (typical).
To function, the pin requires an external pull-up resistor.
VREF (Pin 6): Voltage Reference Output Pin, Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for output current adjustment or for temperature limit/
compensation of the output load. Can supply up to 200µA
of current.
ISMON (Pin 7): Monitor pin that produces a voltage that
is ten times the voltage V(ISP-ISN). ISMON will equal 1V
when V(ISP-ISN) = 100mV.
IVINMON (Pin 8): Monitor pin that produces a voltage
that is twenty times the voltage V(IVINP-IVINN). IVINMON
will equal 1V when V(IVINP-IVINN) = 50mV.
IVINN (Pin 11): Negative Input for the Input Current Limit
and Monitor. The input bias current for this pin is typically
20µA.
INTVCC (Pin 13): Internal 5V Regulator Output. The driver
and control circuits are powered from this voltage. Bypass
this pin to PGND with a minimum 4.7µF ceramic capacitor.
TG1 (Pin 14): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage equal to INTVCC superimposed on
the switch node voltage SW1.
BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin
swings from a diode voltage below INTVCC up to a diode
voltage below VIN + INTVCC.
SW1 (Pin 16): Switch Node. SW1 pin swings from a diode
voltage drop below ground up to VIN.
PGND (Pins 17, 20): Power Ground. Connect these pins
closely to the source of the bottom N-channel MOSFET.
BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
BG2 (Pin 19): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
SW2 (Pin 21): Switch Node. SW2 pin swings from a diode
voltage drop below ground up to VOUT.
37911fb
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9
LT3791-1
Pin Functions
BST2 (Pin 22): Bootstrapped Driver Supply. The BST2 pin
swings from a diode voltage below INTVCC up to a diode
voltage below VOUT + INTVCC.
NC (Pin 23): No Connect Pin. Leave this pin floating.
TG2 (Pin 24): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage equal to INTVCC superimposed on
the switch node voltage SW2.
ISP (Pin 25): Connection Point for the Positive Terminal
of the Output Current Feedback Resistor.
ISN (Pin 26): Connection Point for the Negative Terminal
of the Output Current Feedback Resistor.
SNSP (Pin 27): The Positive Input to the Current Sense
Comparator. The VC pin voltage and controlled offsets
between the SNSP and SNSN pins, in conjunction with a
resistor, set the current trip threshold.
SNSN (Pin 28): The Negative Input to the Current Sense
Comparator.
TEST1 (Pin 29): This pin is used for testing purposes only
and must be connected to SGND for the part to operate
properly.
SGND (Pin 30, Exposed Pad Pin 39): Signal Ground.
All small-signal components and compensation should
connect to this ground, which should be connected to
PGND at a single point. Solder the exposed pad directly
to the ground plane.
PWMOUT (Pin 31): Buffered Version of PWM Signal for
Driving Output Load Disconnect N-Channel MOSFET. The
PWMOUT pin is driven from INTVCC. Use of a MOSFET
with a gate cutoff voltage higher than 1V is recommended.
CCM (Pin 32): Continuous Conduction Mode Pin. When
the pin voltage is higher than 1.5V, the part runs in fixed
frequency forced continuous conduction mode and allows the inductor current to flow negative. When the pin
voltage is less than 0.3V, the part runs in discontinuous
conduction mode and does not allow the inductor current
to flow backward. This pin is only meant to block inductor
reverse current, and should only be pulled low when the
output current is low. This pin must be either connected to
INTVCC (pin 13) for continuous conduction mode across
all loads, or it must be connected to the C/10 (pin 4) with
a pull-up resistor to INTVCC for continuous conduction
mode at heavy load and for discontinuous conduction
mode at light load.
CLKOUT (Pin 33): Clock Output Pin. A 180° out-of-phase
clock is provided at the oscillator frequency to allow for paralleling two devices for extending output power capability.
SYNC (Pin 34): External Synchronization Input Pin. This
pin is internally terminated to GND with a 90k resistor.
The internal buck clock is synchronized to the rising edge
of the SYNC signal while the internal boost clock is 180°
phase shifted.
RT (Pin 35): Frequency Set Pin. Place a resistor to GND
to set the internal frequency. The range of oscillation is
200kHz to 700kHz.
VC (Pin 36): Current Control Threshold and Error Amplifier
Compensation Point. The current comparator threshold
increases with this control voltage. The voltage ranges
from 0.7V to 1.9V.
FB (Pin 37): Voltage Loop Feedback Pin. FB is intended for
constant-voltage regulation. The internal transconductance
amplifier with output VC will regulate FB to 1.2V (typical)
through the DC/DC converter. If the FB input is regulating the loop and V(ISP-ISN) < 10mV, the C/10 pull-down
is asserted. If the FB pin is less than 400mV, the SHORT
pull-down is asserted.
OVLO (Pin 38): Overvoltage Input Pin. This pin is used for
OVLO, if OVLO > 3V then SS is pulled low, the part stops
switching and resets. Do not leave this pin open.
37911fb
10
For more information www.linear.com/LT3791-1
LT3791-1
Block Diagram
26
–
+
A = 10
7
8
9
A1
A = 24
ISMON_INT
33
13
VREF
INTVCC
REGS
SHDN_INT
TSD
IVINMON_INT
BST1
+
IVINMON
A13
A3
EN/UVLO
SW1
–
A4
SHDN_INT
SHDN_INT
SS_RESET
SS LATCH
PWM
+
A14
BG1
PGND
A15
16
18
17
19
INTVCC
RT
BOOST
LOGIC
SYNC
CLKOUT
SW2
+
A16
TG2
BST2
–
SHORT
+
SNSP
–
SNSN
A10
+
0.4V
–
A11
FB
+
C/10
+
Q
R
S
VREF
0.2V
A8
A6
–
1.15V
+
SS LATCH
IVINMON_INT
FB
21
24
22
27
28
37
1.2V
A12
SS RESET
–
–
+
14µA
–
+
+
–
ISMON_INT
+
3V
CTRL
1
A9
–
1.75V
INTVCC
PWM
BG2
SLOPE_COMP_BOOST
A5
3
14
INTVCC
CCM
SLOPE_COMP_BUCK
4
15
BUCK
LOGIC
A7
5
TG1
–
OSC
34
6
VIN
ISMON
1.2V
35
12
IVINN
A = 10 A = 20
3µA
32
11
IVINP
+
A2
10
ISN
ISP
–
25
A18
A17
PWMOUT
SGND
30, 39
1.4µA
31
–
38
VC
SS
2
OVLO
36
37911 BD
37911fb
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11
LT3791-1
Operation
The LT3791-1 is a current mode controller that provides an
output voltage above, equal to or below the input voltage.
The LTC proprietary topology and control architecture uses
a current sensing resistor in buck or boost operation. The
sensed inductor current is controlled by the voltage on
the VC pin, which is the output of the feedback amplifiers
A11 and A12. The VC pin is controlled by three inputs, one
input from the output current loop, one input from the
input current loop, and the third input from the feedback
loop. Whichever feedback input is higher takes precedence,
forcing the converter into either a constant-current or a
constant-voltage mode.
The LT3791-1 is designed to transition cleanly between
the two modes of operation. Current sense amplifier A1
senses the voltage between the IVINP and IVINN pins and
provides a pre-gain to amplifier A11. When the voltage
between IVINP and IVINN reaches 50mV, the output of A1
provides IVINMON_INT to the inverting input of A11 and
the converter is in constant-current mode. If the current
sense voltage exceeds 50mV, the output of A1 increases
causing the output of A11 to decrease, thus reducing the
amount of current delivered to the output. In this manner
the current sense voltage is regulated to 50mV.
The output current amplifier works similar to the input
current amplifier but with a 100mV voltage instead of
50mV. The output current sense level is also adjustable
by the CTRL pin. Forcing CTRL to less than 1.2V forces
ISMON_INT to the same level as CTRL, thus providing
current-level control. The output current amplifier provides
rail-to-rail operation. Similarly if the FB pin goes above
1.2V the output of A11 decreases to reduce the current
level and regulate the output (constant-voltage mode).
The LT3791-1 provides monitoring pins IVINMON and
ISMON that are proportional to the voltage across the
input and output current amplifiers respectively.
The main control loop is shut down by pulling the EN/
UVLO pin low. When the EN/UVLO pin is higher than 1.2V,
an internal 14µA current source charges soft-start capacitor CSS at the SS pin. The VC voltage is then clamped a
diode voltage higher than the SS voltage while the CSS is
slowly charged during start-up. This soft-start clamping
prevents abrupt current from being drawn from the input
power supply.
The top MOSFET drivers are biased from floating bootstrap capacitors C1 and C2, which are normally recharged
through an external diode when the top MOSFET is turned
off. A unique charge sharing technique eliminates top FET
refresh switching cycle in buck or boost operation.Schottky
diodes across the synchronous switch M4 and synchronous
switch M2 are not required, but they do provide a lower drop
during the dead time. The addition of the Schottky diode
typically improves peak efficiency by 1% to 2% at 500kHz.
Power Switch Control
Figure 1 shows a simplified diagram of how the four
power switches are connected to the inductor, VIN, VOUT
and GND. Figure 2 shows the regions of operation for
the LT3791-1 as a function of duty cycle D. The power
switches are properly controlled so the transfer between
regions is continuous. When VIN approaches VOUT, the
buck-boost region is reached.
VOUT
VIN
TG1
M1
SW1
BG1
TG2
M4
L1
M2
SW2
M3
BG2
RSENSE
37911 F01
Figure 1. Simplified Diagram of the Output Switches
DMAX
BOOST
(BG2)
DMIN
BOOST
DMAX
BUCK
(TG1)
DMIN
BUCK
BOOST REGION
BUCK-BOOST REGION
BUCK REGION
M1 ON, M2 OFF
PWM M3, M4 SWITCHES
4-SWITCH PWM
M4 ON, M3 OFF
PWM M2, M1 SWITCHES
37911 F02
Figure 2. Operating Regions vs Duty Cycle
37911fb
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LT3791-1
Operation
Buck Region (VIN > VOUT)
Switch M4 is always on and switch M3 is always off during
this mode. At the start of every cycle, synchronous switch
M2 is turned on first. Inductor current is sensed when
synchronous switch M2 is turned on. After the sensed
inductor current falls below the reference voltage, which
is proportional to VC, synchronous switch M2 is turned off
and switch M1 is turned on for the remainder of the cycle.
Switches M1 and M2 will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch M1
increases until the maximum duty cycle of the converter
in buck operation reaches DMAX(BUCK, TG1), given by:
DMAX(BUCK,TG1) = 100% – D(BUCK-BOOST)
where D(BUCK-BOOST) is the duty cycle of the buck-boost
switch range:
D(BUCK-BOOST) = 8%
Figure 3 shows typical buck operation waveforms. If VIN
approaches VOUT, the buck-boost region is reached.
Buck-Boost Region (VIN ~ VOUT)
When VIN is close to VOUT, the controller is in buck-boost
operation. Figure 4 and Figure 5 show typical waveforms in
this operation. Every cycle the controller turns on switches
M2 and M4, then M1 and M4 are turned on until 180° later
when switches M1 and M3 turn on, and then switches
M1 and M4 are turned on for the remainder of the cycle.
Boost Region (VIN < VOUT)
Switch M1 is always on and synchronous switch M2 is
always off in boost operation. Every cycle switch M3 is
turned on first. Inductor current is sensed when synchronous switch M3 is turned on. After the sensed inductor
current exceeds the reference voltage which is proportional
to VC, switch M3 turns off and synchronous switch M4
is turned on for the remainder of the cycle. Switches M3
and M4 alternate, behaving like a typical synchronous
boost regulator.
The duty cycle of switch M3 decreases until the minimum
duty cycle of the converter in boost operation reaches
DMIN(BOOST,BG2), given by:
DMIN(BOOST,BG2) = D(BUCK-BOOST)
where D(BUCK-BOOST) is the duty cycle of the buck-boost
switch range:
D(BUCK-BOOST) = 8%
Figure 6 shows typical boost operation waveforms. If VIN
approaches VOUT, the buck-boost region is reached.
Low Current Operation
The LT3791-1 is recommended to run in forced continuous
conduction mode at heavy load by pulling the CCM pin
higher than 1.5V. In this mode the controller behaves as
a continuous, PWM current mode synchronous switching
regulator. In boost operation, switch M1 is always on,
switch M3 and synchronous switch M4 are alternately
turned on to maintain the output voltage independent
of the direction of inductor current. In buck operation,
synchronous switch M4 is always on, switch M1 and synchronous switch M2 are alternately turned on to maintain
the output voltage independent of the direction of inductor
current. In the forced continuous mode, the output can
source or sink current.
However, reverse inductor current from the output to the
input is not desired for certain applications. For these applications, the CCM pin must be connected to C/10 (pin 4)
with a pull-up resistor to INTVCC (see front page Typical
Application). Therefore, the CCM pin will be pulled lower
than 0.3V for discontinuous conduction mode by the C/10
pin when the output current is low. In this mode, switch
M4 turns off when the inductor current flows negative.
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13
LT3791-1
Operation
M2 + M4
M2 + M4
M1 + M4
M2 + M4
M1 + M4
M1 + M4
37911 F03
Figure 3. Buck Operation (VIN > VOUT)
M1 + M4
M1 + M4
M2 + M4
M1+ M3
M1+ M3
M1 + M4
M1 + M4
M2 + M4
M1+ M3
M2 + M4
M1 + M4
M1 + M4
37911 F04
Figure 4. Buck-Boost Operation (VIN ≤ VOUT)
M1 + M4
M2 + M4
M1 + M3
M1 + M4
M1 + M4
M2 + M4
M1 + M3
M1 + M4
M1 + M4
M2 + M4
M1 + M3
M1 + M4
37911 F05
Figure 5. Buck-Boost Operation (VIN ≥ VOUT)
M1 + M3
M1 + M4
M1 + M3
M1 + M4
M1 + M3
M1 + M4
37911 F06
Figure 6. Boost Operation (VIN < VOUT)
37911fb
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LT3791-1
Applications Information
The Typical Application on the front page is a basic LT3791-1
application circuit. External component selection is driven
by the load requirement, and begins with the selection of
RSENSE and the inductor value. Next, the power MOSFETs
are selected. Finally, CIN and COUT are selected. This circuit
can operate up to an input voltage of 60V.
200
180
∆IL/ISENSE(MAX) (%)
160
Programming The Switching Frequency
The RT frequency adjust pin allows the user to program the
switching frequency from 200kHz to 700kHz to optimize
efficiency/performance or external component size. Higher
frequency operation yields smaller component size but
increases switching losses and gate driving current, and
may not allow sufficiently high or low duty cycle operation.
Lower frequency operation gives better performance at the
cost of larger external component size. For an appropriate
RT resistor value see Table 1. An external resistor from
the RT pin to GND is required; do not leave this pin open.
140
100
80
60
40
RT (kΩ)
200
147
300
84.5
400
59.0
500
45.3
600
35.7
700
29.4
Frequency Synchronization
The LT3791-1 switching frequency can be synchronized
to an external clock using the SYNC pin. Driving SYNC
with a 50% duty cycle waveform is always a good choice,
otherwise maintain the duty cycle between 10% and 90%.
The falling edge of CLKOUT corresponds to the rising edge
of SYNC thus allowing 2-phase paralleling converters. The
rising edge of CLKOUT turns on switch M3 and the falling
edge of CLKOUT turns on switch M2.
Inductor Selection
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. The inductor
value has a direct effect on ripple current. The maximum
inductor current ripple ΔIL can be seen in Figure 7. This
BUCK ∆IL/
ISENSE(MAX) LIMIT
20
0
50 55 60 65 70 75 80 85 90 95 100
BG1, BG2 DUTY CYCLE (%)
37911 F07
Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle
is the maximum ripple that will prevent subharmonic
oscillation and also regulate with zero load. The ripple
should be less than this to allow proper operation over
all load currents. For a given ripple the inductance terms
in continuous mode are as follows:
Table 1. Switching Frequency vs RT Value
fOSC (kHz)
BOOST ∆IL/
ISENSE(MAX) LIMIT
120
LBUCK >
VOUT • ( VIN(MAX) – VOUT ) • 100
f •IOUT(MAX) • %Ripple • VIN(MAX)
LBOOST >
VIN(MIN)2 • ( VOUT – VIN(MIN) ) • 100
f •IOUT(MAX) • %Ripple • VOUT 2
where:
f is operating frequency
% ripple is allowable inductor current ripple
VIN(MIN) is minimum input voltage
VIN(MAX) is maximum input voltage
VOUT is output voltage
IOUT(MAX) is maximum output load current
For high efficiency, choose an inductor with low core
loss. Also, the inductor should have low DC resistance to
reduce the I2R losses, and must be able to handle the peak
inductor current without saturating. To minimize radiated
noise, use a shielded inductor.
RSENSE Selection and Maximum Output Current
RSENSE is chosen based on the required output current. The
current comparator threshold sets the peak of the inductor
37911fb
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15
LT3791-1
Applications Information
current in boost operation and the maximum inductor valley
current in buck operation. In boost operation, the maximum
average load current at VIN(MIN) is:
51mV ∆I VIN(MIN)
IOUT(MAX _ BOOST) =
– L •
2 VOUT
RSENSE
where ΔIL is peak-to-peak inductor ripple current. In buck
operation, the maximum average load current is:
47.5mV ∆I
IOUT(MAX _ BUCK) =
+ L
2
RSENSE
The maximum current sensing RSENSE value for the boost
operation is:
RSENSE(MAX) =
2 • 51mV • VIN(MIN)
2 •ILED • VOUT + ∆IL(BOOST) • VIN(MIN)
The maximum current sensing RSENSE value for the buck
operation is:
2 • 47.5mV
RSENSE(MAX) =
2 •ILED – ∆IL(BUCK)
The final RSENSE value should be lower than the calculated
RSENSE(MAX) in both the boost and buck operation. A 20%
to 30% margin is usually recommended.
CIN and COUT Selection
In boost operation, input current is continuous. In buck
operation, input current is discontinuous. In buck operation, the selection of input capacitor, CIN, is driven by the
need to filter the input square wave current. Use a low ESR
capacitor sized to handle the maximum RMS current. For
buck operation, the input RMS current is given by:
IRMS = ILED2 • D+
∆IL 2
•D
12
The formula has a maximum at VIN = 2VOUT. Note that
ripple current ratings from capacitor manufacturers are
often based on only 2000 hours of life which makes it
advisable to derate the capacitor.
In boost operation, the discontinuous current shifts
from the input to the output, so COUT must be capable
of reducing the output voltage ripple. The effects of ESR
(equivalent series resistance) and the bulk capacitance
must be considered when choosing the right capacitor
for a given output ripple voltage. The steady ripple due to
charging and discharging the bulk capacitance is given by:
∆VRIPPLE (BOOST _ CAP ) =
ILED • ( VOUT – VIN(MIN) )
COUT • VOUT • f
∆IL
∆VRIPPLE (BUCK _ CAP ) ≈
8 • f • COUT
where COUT is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
ΔVBOOST(ESR) = ILED • ESR
ΔVBUCK(ESR) = ILED • ESR
Multiple capacitors placed in parallel may be needed to meet
the ESR and RMS current handling requirements. Output
capacitors are also used for stability for the LT3791‑1. A
good starting point for output capacitors is seen in the
Typical Applications circuits. Ceramic capacitors have
excellent low ESR characteristics but can have a high
voltage coefficient and are recommended for applications
less than 100W. Capacitors available with low ESR and
high ripple current ratings, such as OS-CON and POSCAP
may be needed for applications greater than 100W.
Programming VIN UVLO and OVLO
The falling UVLO value can be accurately set by the resistor
divider R1 and R2. A small 3µA pull-down current is active
when the EN/UVLO is below the threshold. The purpose
of this current is to allow the user to program the rising
hysteresis. The following equations should be used to
determine the resistor values:
VIN(UVLO –) = 1.2 •
R1+R2
R2
VIN(UVLO+ ) = 3µA • R1+ 1.215 •
R1+R2
R2
37911fb
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LT3791-1
Applications Information
The rising OVLO value can be accurately set by the resistor divider R3 and R4. The following equations should be
used to determine the resistor values:
Table 2. V(ISP-ISN) Threshold vs CTRL
VCTRL (V)
R3+R4
R4
R3+R4
VIN(OVLO –) = 2.925 •
R4
VIN(OVLO+ ) = 3 •
R1
R3
R2
R4
OVLO
EN/UVLO
37911 F08
Figure 8. Resistor Connection to Set VIN UVLO and
OVLO Thresholds
Programming Output Current
The output current is programmed by placing an appropriate value current sense resistor, ROUT, in series with
the output load. The voltage drop across ROUT is (Kelvin)
sensed by the ISP and ISN pins. The CTRL pin should
be tied to a voltage higher than 1.2V to get the full-scale
100mV (typical) threshold across the sense resistor. The
CTRL pin can also be used to adjust the output current,
although relative accuracy decreases with the decreasing
sense threshold. When the CTRL pin voltage is less than
1V, the output current is:
IOUT =
1.1
90
1.15
94.5
1.2
98
1.25
99.5
1.3
100
When VCTRL is higher than 1.3V, the output current is
regulated to:
VIN
LT3791-1
V(ISP-ISN) (mV)
VCTRL – 200mV
ROUT • 10
IOUT =
100mV
ROUT
The CTRL pin should not be left open (tie to VREF if not
used). The CTRL pin can also be used in conjunction with
a thermistor to provide overtemperature protection for
the output load, or with a resistor divider to VIN to reduce
output power and switching current when VIN is low.
The presence of a time varying differential voltage signal
(ripple) across ISP and ISN at the switching frequency
is expected. The amplitude of this signal is increased by
high output load current, low switching frequency and/
or a smaller value output filter capacitor. Some level of
ripple signal is acceptable: the compensation capacitor
on the VC pin filters the signal so the average difference
between ISP and ISN is regulated to the user-programmed
value. Ripple voltage amplitude (peak-to-peak) in excess
of 20mV should not cause mis-operation, but may lead
to noticeable offset between the average value and the
user-programmed value.
ISMON
When the CTRL pin voltage is between 1.1V and 1.3V the
output current varies with VCTRL, but departs from the
equation above by an increasing amount as VCTRL voltage increases. Ultimately, when VCTRL > 1.3V the output
current no longer varies. The typical V(ISP-ISN) threshold
vs VCTRL is listed in Table 2.
The ISMON pin provides a linear indication of the current
flowing through the output. The equation for VISMON is
V(ISP–ISN) • 10. This pin is suitable for driving an ADC input,
however, the output impedance of this pin is 12.5kΩ so
care must be taken not to load this pin.
Programming Input Current Limit
The LT3791-1 has a standalone current sense amplifier.
It can be used to limit the input current. The input current
limit is calculated by the following equation:
37911fb
For more information www.linear.com/LT3791-1
17
LT3791-1
Applications Information
IIN =
Dimming Control
50mV
RIN
For loop stability a lowpass RC filter is needed. For
most applications, a 50Ω resistor and 470nF capacitor
is sufficient.
Table 3
RIN (mΩ)
20
15
12
10
6
5
4
3
2
ILIMIT (A)
2.5
3.3
4.2
5.0
8.3
10.0
12.5
16.7
25
IVINMON
The IVINMON pin provides a linear indication of the current
flowing through the input. The equation for VIVINMON is
V(IVINP-IVINN) • 20. This pin is suitable for driving an ADC
input, however, the output impedance of this pin is 12.5kΩ
so care must be taken not to load this pin.
Programming Output Voltage (Constant Voltage
Regulation)
For a voltage regulator, the output voltage can be set by
selecting the values of R5 and R6 (see Figure 9) according
to the following equation:
VOUT = 1.2 •
R5+R6
R6
There are two methods to control the current source for
dimming using the LT3791-1. One method uses the CTRL
pin to adjust the current regulated in the output. A second
method uses the PWM pin to modulate the current source
between zero and full current to achieve a precisely programmed average current. To make PWM dimming more
accurate, the switch demand current is stored on the VC
node during the quiescent phase when PWM is low. This
feature minimizes recovery time when the PWM signal goes
high. To further improve the recovery time a disconnect
switch may be used in the output current path to prevent
the ISP node from discharging during the PWM signal low
phase. The minimum PWM on- or off-time is affected by
choice of operating frequency and external component
selection. The best overall combination of PWM and
analog dimming capabilities is available if the minimum
PWM pulse is at least six switching cycles and the PWM
pulse is synchronized to the SYNC signal.
SHORT Pin
The LT3791-1 provides an open-drain status pin,
SHORT, which pulls low when the FB pin is below 400mV.
The only time the FB pin will be below 400mV is during
start-up or if the output is shorted. During start-up the
LT3791-1 ignores the voltage on the FB pin until the softstart capacitor reaches 1.75V. To prevent false tripping
after startup, a large enough soft-start capacitor must
be used to allow the output to get up to approximately
40% to 50% of the final value.
C/10 Pin
VOUT
LT3791-1
R5
FB
R6
37911 F09
Figure 9. Resistor Connection for Constant Output
Voltage Regulation
The LT3791-1 provides an open-drain status pin, C/10,
which pulls low when the FB pin is above 1.15V and the
voltage across V(ISP-ISN) is less than 10mV. For voltage
regulator applications with both ISP and ISN pins tied
together to the output (i.e., no output current sense and
limit), the C/10 pin provides a power good flag. For battery
charger applications with output current sense and limit,
the C/10 provides a C/10 charge termination flag.
37911fb
18
For more information www.linear.com/LT3791-1
LT3791-1
Applications Information
Soft-Start
Soft-start reduces the input power sources’ surge currents
by gradually increasing the controller’s current limit (proportional to an internally buffered clamped equivalent of
VC). The soft-start interval is set by the soft-start capacitor
selection according to the following equation
tSS =
1.2V
• CSS
14µA
A 100k resistor must be placed between SS and VREF for
the LT3791-1. This 100k resistor also contributes the extra
SS charge current. Make sure CSS is large enough when
there is loading during start-up.
Loop Compensation
The LT3791-1 uses an internal transconductance error
amplifier whose VC output compensates the control loop.
The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability.
In order to select the power MOSFETs, the power dissipated by the device must be known. For switch M1, the
maximum power dissipation happens in boost operation,
when it remains on all the time. Its maximum power dissipation at maximum output current is given by:
ILED • VOUT 2
PM1(BOOST) =
• ρT • RDS(ON)
V
IN
where ρT is a normalization factor (unity at 25°C)
accounting for the significant variation in on-resistance
with temperature, typically 0.4%/°C as shown in Figure 10.
For a maximum junction temperature of 125°C, using a
value of ρT = 1.5 is reasonable.
Switch M2 operates in buck operation as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
PM2(BUCK) =
VIN – VOUT
•ILED2 • ρT • RDS(ON)
VIN
The inductor and output capacitor are chosen based on
performance, size and cost. The compensation resistor and
capacitor at VC are set to optimize control loop response
and stability. For typical applications, a 10nF compensation
capacitor at VC is adequate, and a series resistor should
always be used to increase the slew rate on the VC pin to
maintain tighter regulation of output current during fast
transients on the input supply of the converter.
Switch M3 operates in boost operation as the control
switch. Its power dissipation at maximum current is
given by:
Power MOSFET Selections and Efficiency
Considerations
where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused
by reverse-recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
The LT3791-1 requires four external N-channel power
MOSFETs, two for the top switches (switch M1 and M4,
shown in Figure 1) and two for the bottom switches (switch
M2 and M3 shown in Figure 1). Important parameters for
the power MOSFETs are the breakdown voltage, VBR(DSS),
threshold voltage, VGS(TH), on-resistance, RDS(ON), reverse
transfer capacitance, CRSS, and maximum current, IDS(MAX).
The drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used
in LT3791-1 applications. If the input voltage is expected
to drop below the 5V, then sub-logic threshold MOSFETs
should be considered.
PM3(BOOST) =
( VOUT – VIN ) • VOUT •I
VIN2
+ k • VOUT 3 •
LED
2
• ρT • RDS(ON)
ILED
• CRSS • f
VIN
For switch M4, the maximum power dissipation happens
in boost operation, when its duty cycle is higher than
50%. Its maximum power dissipation at maximum output
current is given by:
PM4(BOOST) =
VIN
VOUT
I • V 2
• LED OUT • ρT • RDS(ON)
VIN
For the same output voltage and current, switch M1 has
the highest power dissipation and switch M2 has the lowest power dissipation unless a short occurs at the output.
37911fb
For more information www.linear.com/LT3791-1
19
LT3791-1
Applications Information
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
TJ = TA + P • RTH(JA)
The RTH(JA) to be used in the equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(JC)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
ρT NORMALIZED ON-RESISTANCE (Ω)
2.0
1.5
1.0
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LT3791-1 to be
exceeded. The system supply current is normally dominated
by the gate charge current. Additional external loading of
the INTVCC also needs to be taken into account for the
power dissipation calculations. Power dissipation for the
IC in this case is VIN • IINTVCC, and overall efficiency is
lowered. The junction temperature can be estimated by
using the equations given
TJ = TA + (PD • θJA)
0.5
0
–50
INTVCC pin regulator can supply a peak current of 67mA
and must be bypassed to ground with a minimum of 4.7µF
ceramic capacitor or low ESR electrolytic capacitor. An
additional 0.1µF ceramic capacitor placed directly adjacent
to the INTVCC and PGND IC pins is highly recommended.
Good bypassing is necessary to supply the high transient
current required by MOSFET gate drivers.
where θJA (in °C/W) is the package thermal impedance.
50
100
0
JUNCTION TEMPERATURE (°C)
150
37911 F10
Figure 10. Normalized RDS(ON) vs Temperature
Optional Schottky Diode (D3, D4) Selection
The Schottky diodes D3 and D4 shown in the Typical Applications section conduct during the dead time between
the conduction of the power MOSFET switches. They
are intended to prevent the body diode of synchronous
switches M2 and M4 from turning on and storing charge
during the dead time. In particular, D4 significantly reduces
reverse-recovery current between switch M4 turn-off and
switch M3 turn-on, which improves converter efficiency
and reduces switch M3 voltage stress. In order for the
diode to be effective, the inductance between it and the
synchronous switch must be as small as possible, mandating that these components be placed adjacently.
INTVCC Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LT3791-1. The
For example, a typical application operating in continuous
current operation might draw 24mA from a 24V supply:
TJ = 70°C + 24mA • 24V • 28°C/W = 86°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
Top Gate (TG) MOSFET Driver Supply (C1, D1, C2, D2)
The external bootstrap capacitors C1 and C2 connected
to the BST1 and BST2 pins supply the gate drive voltage
for the topside MOSFET switches M1 and M4. When the
top MOSFET switch M1 turns on, the switch node SW1
rises to VIN and the BST1 pin rises to approximately VIN +
INTVCC. When the bottom MOSFET switch M2 turns on, the
switch node SW1 drops low and the bootstrap capacitor
C1 is charged through D1 from INTVCC. When the bottom
MOSFET switch M3 turns on, the switch node SW2 drops
low and the bootstrap capacitor C2, is charged through D2
from INTVCC. The bootstrap capacitors C1 and C2 need to
store about 100 times the gate charge required by the top
MOSFET switch M1 and M4. In most applications a 0.1µF
to 0.47µF, X5R or X7R ceramic capacitor is adequate.
37911fb
20
For more information www.linear.com/LT3791-1
LT3791-1
Applications Information
Efficiency Considerations
The power efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuits produce losses, four main sources
account for most of the losses in LT3791-1 circuits:
1. DC I2R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
2. Transition loss. This loss arises from the brief amount
of time switch M1 or switch M3 spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss
is significant at input voltages above 20V and can be
estimated from:
When making adjustments to improve efficiency, the input
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in the input
current, then there is no change in efficiency.
PC Board Layout Checklist
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
The PGND ground plane layer should not have any traces
and it should be as close as possible to the layer with
power MOSFETs.
n
Place CIN, switch M1, switch M2 and D1 in one compact
area. Place COUT, switch M3, switch M4 and D2 in one
compact area.
n
Use immediate vias to connect the components (including the LT3791-1’s SGND and PGND pins) to the
ground plane. Use several large vias for each power
component.
n
Transition Loss ≈ 2.7 • VIN2 • IOUT • CRSS • f
where CRSS is the reverse-transfer capacitance.
3. INTVCC current. This is the sum of the MOSFET driver
and control currents.
4. CIN and COUT loss. The input capacitor has the difficult
job of filtering the large RMS input current to the regulator in buck operation. The output capacitor has the
difficult job of filtering the large RMS output current
in boost operation. Both CIN and COUT are required to
have low ESR to minimize the AC I2R loss and sufficient
capacitance to prevent the RMS current from causing
additional upstream losses in fuses or batteries.
5. Other losses. Schottky diode D3 and D4 are responsible for conduction losses during dead time and light
load conduction periods. Inductor core loss occurs
predominately at light loads. Switch M3 causes reverse
recovery current loss in boost operation.
Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
n
Flood all unused areas on all layers with copper. Flooding
with copper will reduce the temperature rise of power
components. Connect the copper areas to any DC net
(VIN or PGND).
n
Separate the signal and power grounds. All small-signal
components should return to the SGND pin at one point,
which is then tied to the PGND pin close to the sources
of switch M2 and switch M3.
n
Place switch M2 and switch M3 as close to the controller as possible, keeping the PGND, BG and SW traces
short.
n
Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and
TG2 nodes away from sensitive small-signal nodes.
n
37911fb
For more information www.linear.com/LT3791-1
21
LT3791-1
Applications Information
The path formed by switch M1, switch M2, D1 and the
CIN capacitor should have short leads and PC trace
lengths. The path formed by switch M3, switch M4, D2
and the COUT capacitor also should have short leads
and PC trace lengths.
n
The output capacitor (–) terminals should be connected
as close as possible to the (–) terminals of the input
capacitor.
n
Connect the top driver bootstrap capacitor, C1, closely
to the BST1 and SW1 pins. Connect the top driver
bootstrap capacitor, C2, closely to the BST2 and SW2
pins.
n
Connect the input capacitors, CIN, and output capacitors,
COUT, closely to the power MOSFETs. These capacitors carry the MOSFET AC current in boost and buck
operation.
n
Route SNSN and SNSP leads together with minimum
PC trace spacing. Avoid sense lines pass through noisy
areas, such as switch nodes. Ensure accurate current
sensing with Kelvin connections at the SENSE resistor.
n
Connect the VC pin compensation network close to the
IC, between VC and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output
voltage ripple voltage from the compensation loop.
n
Connect the INTVCC bypass capacitor, CVCC, close to the
IC, between the INTVCC and the power ground pins. This
capacitor carries the MOSFET drivers’ current peaks. An
additional 0.1µF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve
noise performance substantially.
n
37911fb
22
For more information www.linear.com/LT3791-1
LT3791-1
Typical Applications
98% Efficient 60W (12V 5A) Voltage Regulator Runs Down to 3V VIN
RIN
0.003Ω
VIN
3V TO 55V
STARTS UP
FROM 5.5V
R7
50Ω
D5
D6
INTVCC
VIN
C3
1µF
BST2
IVINN
IVINP
TG1
TEST1
EN/UVLO
BG1
INTVCC
R9
100k
R10
200k
RFAULT
100k
M2
D4
+
VO
M4
L1, 6.8µH
COUT2
100µF
35V
ROUT
0.015Ω
VOUT
12V
5A
COUT
10µF
25V
×3
R5
73.2k
M3
D3
R6
8.06k
SNSP
RSENSE
0.004Ω
LT3791-1
SNSN
SHORT
C/10
CCM
IVINMON
PGND
BG2
SW2
TG2
ISP
ISN
FB
ISMON
CLKOUT
PWMOUT
VREF
C8
0.1µF
M1
SWI
OVLO
R4
57.6k
C2
0.1µF
C1
0.1µF
BST1
R3
1M
R2
576k
CVCC
4.7µF
D1 D2
C7
470nF
R1
866k
CIN
4.7µF
100V
×4
VO
PWM
CTRL
SS SYNC VC
RT
RC
5.1k
CC
22nF
CSS
33nF
SGND
R8
84.5k
300kHz
37911 TA02a
D1, D2: NXP BAT46WJ
D3: IRF 10BQ060
D4: IRF 10BQ040
D5, D6: DIODES INC. BAT46W
L1: WURTH ELEKTRONIK WE-HCI 7443556680
M1, M2: RENASAS RJK0651DPB 60VDS
M3, M4: VISHAY SiR424DP 40VDS
COUT2: SUNCON 35HVT100M
Efficiency vs Load Current
Maximum Output Current vs VIN
100
6
MAXIMUM OUTPUT CURRENT (A)
95
EFFICIENCY (%)
90
85
80
75
70
VIN = 3V
VIN = 6V
VIN = 12V
VIN = 28V
VIN = 48V
65
60
55
50
0
1
2
3
LOAD CURRENT (A)
4
5
5
4
3
2
1
0
3
37911 TA02b
4
5
6
7 8 9 10 20 30 40 50 60
INPUT VOLTAGE (V)
37911 TA02c
37911fb
For more information www.linear.com/LT3791-1
23
LT3791-1
Package Description
Please refer to http://www.linear.com/product/LT3791-1#packaging for the most recent package drawings.
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev C)
Exposed Pad Variation AA
4.75 REF
38
9.60 – 9.80*
(.378 – .386)
4.75 REF
(.187)
20
6.60 ±0.10
4.50 REF
2.74 REF
SEE NOTE 4
6.40
2.74
REF (.252)
(.108)
BSC
0.315 ±0.05
1.05 ±0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.25
REF
19
1.20
(.047)
MAX
0° – 8°
0.50
(.0196)
BSC
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1
0.17 – 0.27
(.0067 – .0106)
TYP
0.05 – 0.15
(.002 – .006)
FE38 (AA) TSSOP REV C 0910
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
37911fb
24
For more information www.linear.com/LT3791-1
LT3791-1
Revision History
REV
DATE
DESCRIPTION
A
12/13
Clarified TG1, TG2, tOFF(MIN) parameters
PAGE NUMBER
B
05/17
Clarified Typical Application schematic
4
23, 26
Clarified last paragraph on the Description
1
Clarified FB pin input bias current limits
3
Clarified TG1, TG2, tOFF(MIN) limits
4
37911fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT3791-1
25
LT3791-1
Typical Application
2.5A Buck-Boost 36V SLA Battery Charger
PVIN
9V TO 58V
RIN
0.003Ω
1µF
50Ω
VIN
INTVCC
D1
TG1
IVINP
INTVCC
200k
30.9k
BG1
LT3791-1
RBAT
0.04Ω
SNSN
IVINMON
ISMON
CTRL
BG2
SW2
TG2
FB
ISP
PWM
ISN
C/10
SS
CCM
RT
VREF
SYNC VC
D1, D2: BAT46WJ
22nF
L1: COILCRAFT SER2915L-103K
M1-M4: RENESAS RJK0651DPB
M5: NXP NX7002AK
CIN2: NIPPON CHEMI-CON EMZA630ADA101MJA0G
COUT2: SUNCON 50HVT220M
M3
+
PGND
SGND
100k
L1 10µH
M2
COUT
4.7µF
50V
×2
+
COUT2
220µF
50V
SNSP
CLKOUT
0.1µF
CIN2
100µF
63V
M4
0.1µF
RSENSE
0.004Ω
SHORT
PWMOUT
TEST1
CHARGE CURRENT CONTROL
M1
SWI
OVLO
+
0.1µF
BST1
EN/UVLO
57.6k
4.7µF
D2
BST2
IVINN
470nF
499k
CIN
4.7µF
100V
×2
2.2k
1µF
2.5A
CHARGE
36V
SLA BATTERY
AGM TYPE
41V FLOAT
44V CHARGE
AT 25°C
50Ω
1.00M
INTVCC
20k
84.5k
300kHz
402k
30.1k
M5
22nF
37911 TA03
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT3791
60V, 4-Switch, Synchronous Buck-Boost LED Driver
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VIN: 4.7V to 60V, VOUT Range: 1.2V to 60V, True Color PWM™, Analog,
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LTC®3780
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High Efficiency, Synchronous, 4-Switch Buck-Boost
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VIN: 4V to 38V, VOUT Range: 0.8V to 38V, ISD < 40µA, 4mm × 5mm QFN-28,
SSOP-28 Packages
LT3755/LT3755-1 High Side 60V, 1MHz LED Controller with True Color
LT3755-2
3000:1 PWM Dimming
VIN: 4.5V to 40V, VOUT Range: 5V to 60V, 3000:1 True Color PWM, Analog,
ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
LT3756/LT3756-1 High Side 100V, 1MHz LED Controller with True Color VIN: 6V to 100V, VOUT Range: 5V to 100V, 3000:1 True Color PWM, Analog,
ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
3000:1 PWM Dimming
LT3756-2
LT3596
60V, 300mA Step-Down LED Driver
VIN: 6V to 60V, VOUT Range: 5V to 55V, 10000:1 True Color PWM, Analog,
ISD < 1µA, 5mm × 8mm QFN-52 Package
LT3743
Synchronous Step-Down 20A LED Driver with
Thee-State LED Current Control
VIN: 5.5V to 36V, VOUT Range: 5.5V to 35V, 3000:1 True Color PWM, Analog,
ISD < 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages
37911fb
26
LT 0517 REV B • PRINTED IN USA
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For more information www.linear.com/LT3791-1
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