LTC1562
Very Low Noise, Low Distortion
Active RC Quad Universal Filter
U
FEATURES
■
■
■
■
■
■
■
■
■
■
■
■
■
■
DESCRIPTIO
Continuous Time—No Clock
Four 2nd Order Filter Sections, 10kHz to 150kHz
Center Frequency
±0.5% Typical Center Frequency Accuracy
±0.3% Typical Center Frequency Accuracy (A Grade)
Wide Variety of Response Shapes
Lowpass, Bandpass and Highpass Responses
103dB Typical S/N, ±5V Supply (Q = 1)
97dB Typical S/N, Single 5V Supply (Q = 1)
96dB Typical S/(N + THD) at ±5V Supply, 20kHz Input
Rail-to-Rail Input and Output Voltages
DC Accurate to 3mV (Typ)
“Zero-Power” Shutdown Mode
Single or Dual Supply, 5V to 10V Total
Resistor-Programmable fO, Q, Gain
U
APPLICATIO S
■
■
■
■
■
■
High Resolution Systems (14 Bits to 18 Bits)
Antialiasing/Reconstruction Filters
Data Communications, Equalizers
Dual or I-and-Q Channels (Two Matched 4th Order
Filters in One Package)
Linear Phase Filtering
Replacing LC Filter Modules
The LTC®1562 is a low noise, low distortion continuous-time
filter with rail-to-rail inputs and outputs, optimized for a
center frequency (fO) of 10kHz to 150kHz. Unlike most
monolithic filters, no clock is needed. Four independent 2nd
order filter blocks can be cascaded in any combination, such
as one 8th order or two 4th order filters. Each block’s
response is programmed with three external resistors for
center frequency, Q and gain, using simple design formulas.
Each 2nd order block provides lowpass and bandpass outputs. Highpass response is available if an external capacitor
replaces one of the resistors. Allpass, notch and elliptic
responses can also be realized.
The LTC1562 is designed for applications where dynamic
range is important. For example, by cascading 2nd order
sections in pairs, the user can configure the IC as a dual 4th
order Butterworth lowpass filter with 94dB signal-to-noise
ratio from a single 5V power supply. Low level signals can
exploit the built-in gain capability of the LTC1562. Varying the
gain of a section can achieve a dynamic range as high as
118dB with a ±5V supply.
Other cutoff frequency ranges can be provided upon request.
Please contact LTC Marketing.
, LTC and LT are registered trademarks of Linear Technology Corporation.
U
TYPICAL APPLICATIO
Amplitude Response
Dual 4th Order 100kHz Butterworth Lowpass Filter
R21, 10k
3
5
5V
0.1µF
6
R23, 10k
VIN1
2
8
9
RQ3, 5.62k
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
V–
V + LTC1562
SHDN
AGND
V2 A
V2 D
V1 A
INV A
V1 D
INV D
RIN4, 10k
20
–10
19 RQ2, 13k
–20
18 R22, 10k
VOUT2
–5V
16
15
0.1µF
VOUT1
13
12 R24, 10k
11
RQ4, 13k
1562 TA01
SCHEMATIC INCLUDES PIN
NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN)
ALSO CONNECT TO V –
SEE TYPICAL APPLICATIONS
FOR OTHER CUTOFF FREQUENCIES
DC ACCURATE, NONINVERTING,
UNITY-GAIN, RAIL-TO-RAIL
INPUT AND OUTPUTS. PEAK
SNR ≈ 100dB WITH ±5V SUPPLIES
GAIN (dB)
1
RQ1, 5.62k
RIN3
10k
0
RIN2, 10k
RIN1
10k
VIN2
10
–30
–40
–50
–60
–70
–80
10k
100k
FREQUENCY (Hz)
1M
1562 TA03b
1562fa
1
W
U
U
PACKAGE/ORDER INFORMATION
LTC1562
W W
W
AXI U
U
ABSOLUTE
RATI GS
(Note 1)
Total Supply Voltage (V + to V –) .............................. 11V
Maximum Input Voltage
at Any Pin ....................(V – – 0.3V) ≤ V ≤ (V + + 0.3V)
Storage Temperature Range ................. – 65°C to 150°C
Operating Temperature Range
LTC1562C ............................................... 0°C to 70°C
LTC1562I ............................................ – 40°C to 85°C
Lead Temperature (Soldering, 10 sec).................. 300°C
U
U
W
PACKAGE/ORDER I FOR ATIO
TOP VIEW
INV B
V1 B
V2 B
V –*
V+
SHDN
V –*
V2 A
V1 A
INV A
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
INV C
V1 C
V2 C
V –*
V–
AGND
V –*
V2 D
V1 D
INV D
ORDER PART
NUMBER
LTC1562CG
LTC1562ACG
LTC1562IG
LTC1562AIG
G PACKAGE
20-LEAD PLASTIC SSOP
*G PACKAGE PINS 4, 7, 14, 17 ARE
SUBSTRATE/SHIELD CONNECTIONS
AND MUST BE TIED TO V –
ORDER PART
NUMBER
TOP VIEW
INV B
1
16 INV C
V1 B
2
15 V1 C
V2 B
3
14 V2 C
V+
4
13 V –
SHDN
5
12 AGND
V2 A
6
11 V2 D
V1 A
7
10 V1 D
INV A
8
9
LTC1562CN
INV D
N PACKAGE
16-LEAD PDIP
TJMAX = 150°C, θJA = 90°C/W
TJMAX = 150°C, θJA = 136°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications that apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VS = ±5V, outputs unloaded, SHDN pin to logic “low”,
unless otherwise noted. AC specs are for a single 2nd order section, RIN = R2 = RQ =10k ±0.1%, fO = 100kHz, unless noted.
SYMBOL
PARAMETER
VS
Total Supply Voltage
IS
Supply Current
VOS
CONDITIONS
MIN
TYP
MAX
10.5
V
17.3
19
19.5
21.5
mA
mA
23.5
25.5
mA
mA
4.75
VS = ±2.375V, RL = 5k, CL = 30pF, Outputs at 0V
VS = ±5V, RL = 5k, CL = 30pF, Outputs at 0V
UNITS
VS = ±2.375V, RL = 5k, CL = 30pF, Outputs at 0V
VS = ±5V, RL = 5k, CL = 30pF, Outputs at 0V
●
●
Output Voltage Swing
VS = ±2.375V, RL = 5k, CL = 30pF
VS = ±5V, RL = 5k, CL = 30pF
●
●
DC Offset Magnitude, V2 Outputs
(Lowpass Response)
VS = ±2.375V, Input at AGND Voltage
VS = ±5V, Input at AGND Voltage
●
●
DC AGND Reference Point
VS = Single 5V Supply
2.5
Center Frequency (f O) Error (Note 2)
LTC1562 (SSOP)
LTC1562A (SSOP)
LTC1562 (PDIP)
VS = ±5V, V2 Output Has RL = 5k, CL = 30pF
VS = ±5V, V2 Output Has RL = 5k, CL = 30pF
VS = ±5V, V2 Output Has RL = 5k, CL = 30pF
0.5
0.3
0.6
1.0
0.6
1.5
%
%
%
+ 0.05
+ 0.1
dB
+ 0.2
+ 0.5
dB
HL
LP Passband Gain (V2 Output)
VS = ±2.375V, fIN = 10kHz,
V2 Output Has RL = 5k, CL = 30pF
●
HB
BP Passband Gain (V1 Output)
VS = ±2.375V, fIN = fO,
V2 Output Has RL = 5k, CL = 30pF
●
4.0
9.3
4.6
9.8
3
3
0
VP-P
VP-P
15
15
mV
mV
V
1562fa
2
LTC1562
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications that apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VS = ±5V, outputs unloaded, SHDN pin to logic “low”,
unless otherwise noted. AC specs are for a single 2nd order section, RIN = R2 = RQ =10k ±0.1%, fO = 100kHz, unless noted.
SYMBOL
THD
PARAMETER
CONDITIONS
Q Error
VS = ±2.375V, LP Output Has RL = 5k, CL = 30pF
+3
%
Wideband Output Noise,
Lowpass Response (V2 Output)
VS = ±2.375V, BW = 200kHz, Input AC GND
VS = ±5V, BW = 200kHz, Input AC GND
24
24
µVRMS
µVRMS
Input-Referred Noise, Gain = 100
BW = 200kHz, f O = 100kHz, Q = 1, Input AC GND
4.5
µVRMS
Total Harmonic Distortion,
Lowpass Response (V2 Output)
fIN = 20kHz, 2.8VP-P, V1 and V2 Outputs Have
RL = 5k, CL = 30pF
– 96
dB
fIN = 100kHz, 2.8VP-P, V1 and V2 Outputs Have
RL = 5k, CL = 30pF
– 78
dB
SHDN Pin to V +
SHDN Pin to V +, VS = ±2.375V
1.5
1.0
Shutdown-Input Bias Current
SHDN Pin to 0V
– 10
Shutdown Delay
SHDN Pin Steps from 0V to V +
20
Shutdown Recovery Delay
SHDN Pin Steps from V +
100
µs
5
pA
Shutdown Supply Current
MIN
Shutdown-Input Logic Threshold
TYP
MAX
UNITS
15
µA
µA
– 20
µA
2.5
to 0V
Inverting Input Bias Current, Each Biquad
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
V
µs
Note 2: fO change from ±5V to ±2.375 supplies is – 0.15% typical,
fO temperature coefficient, – 40°C to 85°C, is –25ppm/°C typical.
U W
TYPICAL PERFOR A CE CHARACTERISTICS
fO Error vs Nominal fO (VS = ±5V)
1.25
1.25
1.00
1.00
Q=5
0.25
fO ERROR (%)
fO ERROR (%)
0.50
Q = 2.5
0
– 0.25
– 0.50
– 0.75
0.75
Q=1
0.50
–1.00
–1.25
–1.25
1562 G01
25
Q = 2.5
Q = 10
0
– 0.50
–1.50
TA = 70°C
TA = 25°C
RIN = RQ
Q=5
– 0.25
– 0.75
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
30
0.25
–1.00
–1.50
35
Q ERROR (%)
1.50
0.75
Q Error vs Nominal fO (VS = ±5V)
fO Error vs Nominal fO (VS = ±2.5V)
1.50
Q=1
20
15
10
Q=5
Q = 2.5
5
0
Q=1
–5
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
1562 G02
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
1562 G03
1562fa
3
LTC1562
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Peak BP Gain vs Nominal fO
(VS = ±5V) (Figure 3, V1 Output)
Q Error vs Nominal fO (VS = ±2.5V)
35
3.0
2.5
RIN = RQ
15
Q = 2.5
10
2.0
PEAK BP GAIN (dB)
Q=5
20
1.5
Q=5
1.0
Q = 2.5
0.5
5
Q=1
0
–5
– 0.5
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
1562 G04
55
50
50
45
45
Q = 2.5
30
Q=1
25
35
Q = 2.5
30
25
20
15
15
10
Q=1
10
70
80
90 100 110 120 130 140
NOMINAL fO (kHz)
Q = 2.5
0.5
Q=1
0
– 0.5
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
1562 G6
Distortion vs External Load
Resistance (VS = ±5V, 25°C)
(Figure 8)
Q=5
40
20
60
1.0
THD (AMPLITUDE BELOW FUNDAMENTAL) (dB)
60
55
BP NOISE (µVRMS)
60
35
Q=5
1.5
BP Noise vs Nominal fO
(VS = ±5V, 25°C) (Figure 3,
V1 Output) (RIN = RQ)
Q=5
Q = 10
RIN = RQ
1562 G5
LP Noise vs Nominal fO
(VS = ±5V, 25°C) (Figure 3,
V2 Output) (RIN = R2)
40
TA = 70°C
TA = 25°C
2.0
Q=1
0
50 60 70 80 90 100 110 120 130 140 150
NOMINAL fO (kHz)
2.5
Q = 10
RIN = RQ
Q = 10
25
Q ERROR (%)
3.0
TA = 70°C
TA = 25°C
PEAK BP GAIN (dB)
TA = 70°C
TA = 25°C
30
NOISE (µVRMS)
Peak BP Gain vs Nominal fO
(VS = ±2.5V) (Figure 3, V1 Output)
60
70
80
90 100 110 120 130 140
NOMINAL fO (kHz)
1562 G07
1562 G08
0
–10
–20
–30
2nd ORDER LOWPASS
fO = 100kHz
Q = 0.7
OUTPUT LEVEL 1VRMS (2.83VP-P)
± 5V SUPPLIES
– 40
– 50
– 60
–70
– 80
fIN = 50kHz
– 90
fIN = 20kHz
–100
10k
2k
5k
EXTERNAL LOAD RESISTANCE (Ω)
1k
1562 G09
U
U
U
PI FU CTIO S
Power Supply Pins: The V + and V – pins should be
bypassed with 0.1µF capacitors to an adequate analog
ground or ground plane. These capacitors should be
connected as closely as possible to the supply pins. In the
20-lead SSOP package, the additional pins 4, 7, 14 and 17
are internally connected to V – (Pin 16) and should also be
tied to the same point as Pin 16 for best shielding. Low
noise linear supplies are recommended. Switching supplies are not recommended as they will lower the filter
dynamic range.
Analog Ground (AGND): The AGND pin is the midpoint of
an internal resistive voltage divider, developing a potential
halfway between the V + and V – pins, with an equivalent
series resistance nominally 7kΩ. This serves as an internal ground reference. Filter performance will reflect the
quality of the analog signal ground and an analog ground
plane surrounding the package is recommended. The
analog ground plane should be connected to any digital
ground at a single point. For dual supply operation, the
AGND pin should be connected to the ground plane
1562fa
4
LTC1562
U
U
U
PIN FUNCTIONS
(Figure 1). For single supply operation, the AGND pin
should be bypassed to the ground plane with at least a
0.1µF capacitor (at least 1µF for best AC performance)
(Figure 2). These figures show 20-pin package connections. The same principles apply to the 16-pin package
with allowance for its different pin numbers. The 16-pin
ANALOG
GROUND
PLANE
1
20
2
19
3
18
4
17
V+
5
16
0.1µF
6
LTC1562
20-PIN SSOP
0.1µF
14
8
13
9
12
10
11
Shutdown (SHDN): When the SHDN input goes high or is
open-circuited, the LTC1562 enters a “zero-power” shutdown state and only junction leakage currents flow. The
AGND pin and the amplifier outputs (see Figure 3) assume
a high impedance state and the amplifiers effectively
disappear from the circuit. (If an input signal is applied to
a complete filter circuit while the LTC1562 is in shutdown,
some signal will normally flow to the output through
passive components around the inactive op amps.)
A small pull-up current source at the SHDN input defaults
the LTC1562 to the shutdown state if the SHDN pin is left
floating. Therefore, the user must connect the SHDN pin
to a logic “low” (0V for ±5V supplies, V – for 5V total
supply) for normal operation of the LTC1562. (This convention permits true “zero-power” shutdown since not
even the driving logic must deliver current while the part
is in shutdown.) With a single supply voltage, use V – for
logic “low”— do not connect SHDN to the AGND pin.
15
7
SINGLE-POINT
SYSTEM GROUND
V–
package does not have the four substrate pins (Pins 4, 7,
14, 17 in the 20-pin package).
DIGITAL
GROUND PLANE
(IF ANY)
1562 F01
1/4 LTC1562
*R1 AND C ARE PRECISION
INTERNAL COMPONENTS
Figure 1. Dual Supply Ground Plane Connection
(Including Substrate Pins 4, 7, 14, 17)
ANALOG
GROUND
PLANE
1
20
2
19
3
18
4
17
V+
5
0.1µF
6
LTC1562
20-PIN SSOP
C
–
+
16
V2
15
7
14
8
13
9
12
10
11
SINGLE-POINT
SYSTEM GROUND
1
sR1C*
INV
R2
V1
RQ
1562 F01
1µF
ZIN
+
–
V +/2
REFERENCE
VIN
RESPONSE RESPONSE
ZIN TYPE
AT V1
AT V2
R
BANDPASS LOWPASS
C
HIGHPASS BANDPASS
DIGITAL
GROUND PLANE
(IF ANY)
1562 F01
Figure 2. Single Supply Ground Plane Connection
(Including Substrate Pins 4, 7, 14, 17)
IN EACH CASE,
fO = (100kHz)
(
( )
10kΩ
R2
)
Q = RQ 100kHz
R2
fO
Figure 3. Equivalent Circuit of a Single 2nd Order Section
(Inside Dashed Line) Shown in Typical Connection. Form of ZIN
Determines Response Types at the Two Outputs (See Table)
1562fa
5
LTC1562
U
U
U
PIN FUNCTIONS
INV A, INV B, INV C, INV D: Each of the INV pins is a virtualground summing point for the corresponding 2nd order
section. For each section, external components ZIN, R2,
RQ connect to the INV pin as shown in Figure 3 and
described further in the Applications Information. Note
that the INV pins are sensitive internal nodes of the filter
and will readily receive any unintended signals that are
capacitively coupled into them. Capacitance to the INV
nodes will also affect the frequency response of the filter
sections. For these reasons, printed circuit connections to
the INV pins must be kept as short as possible, less than
one inch (2.5cm) total and surrounded by a ground plane.
V1 A, V1 B, V1 C, V1 D: Output Pins. Provide a bandpass,
highpass or other response depending on external circuitry (see Applications Information section). Each V1 pin
also connects to the RQ resistor of the corresponding 2nd
order filter section (see Figure 3 and Applications Information). Each output is designed to drive a nominal net load
of 5kΩ and 30pF, which includes the loading due to the
external RQ. Distortion performance improves when the
outputs are loaded as lightly as possible. Some earlier
literature refers to these outputs as “BP” rather than V1.
V2 A, V2 B, V2 C, V2 D: Output Pins. Provide a lowpass,
bandpass or other response depending on external circuitry (see Applications Information section). Each V2 pin
also connects to the R2 resistor of the corresponding 2nd
order filter section (see Figure 3 and Applications Information). Each output is designed to drive a nominal net load
of 5kΩ and 30pF, which includes the loading due to the
external R2. Distortion performance improves when the
outputs are loaded as lightly as possible. Some earlier
literature refers to these outputs as “LP” rather than V2.
W
BLOCK DIAGRA
Overall Block Diagram Showing Four 3-Terminal 2nd Order Sections
INV
V1
V2
A
V+
V1
V2
B
–
–
C
∫
+
V+
INV
C
∫
+
SHUTDOWN
SWITCH
V–
R
2ND ORDER SECTIONS
R
SHUTDOWN
SWITCH
SHDN
AGND
D
C
+
+
∫
V–
–
∫
–
C
C
1562 BD
INV
V1
V2
INV
V1
V2
1562fa
6
LTC1562
U
W
U
U
APPLICATIONS INFORMATION
Functional Description
Setting fO and Q
The LTC1562 contains four matched, 2nd order, 3-terminal universal continuous-time filter blocks, each with a
virtual-ground input node (INV) and two rail-to-rail outputs (V1, V2). In the most basic applications, one such
block and three external resistors provide 2nd order
lowpass and bandpass responses simultaneously (Figure
3, with a resistor for ZIN). The three external resistors set
standard 2nd order filter parameters fO, Q and gain. A
combination of internal precision components and external resistor R2 sets the center frequency fO of each 2nd
order block. The LTC1562 is trimmed at manufacture so
that fO will be 100kHz ±0.5% (±0.6% typical for PDIP
package) if the external resistor R2 is exactly 10k.
Each of the four 2nd order sections in the LTC1562 can be
programmed for a standard filter function (lowpass, bandpass or highpass) when configured as in Figure 3 with a
resistor or capacitor for ZIN. These transfer functions all
have the same denominator, a complex pole pair with
center frequency ωO = 2πfO and quality parameter Q. (The
numerators depend on the response type as described
below.) External resistors R2 and RQ set fO and Q as
follows:
However, lowpass/bandpass filtering is only one specific
application for the 2nd order building blocks in the LTC1562.
Highpass response results if the external impedance ZIN in
Figure 3 becomes a capacitor CIN (whose value sets only
gain, not critical frequencies) as described below.
Responses with zeroes are available through other connections (see Notches and Elliptic Responses). Moreover,
the virtual-ground input gives each 2nd order section the
built-in capability for analog operations such as gain
(preamplification), summing and weighting of multiple
inputs, handling input voltages beyond the power supplies
or accepting current or charge signals directly. These
Operational FilterTM frequency-selective building blocks
are nearly as versatile as op amps.
The user who is not copying exactly one of the Typical
Applications schematics shown later in this data sheet is
urged to read carefully the next few sections through at
least Signal Swings, for orientation about the LTC1562,
before attempting to design custom application circuits.
Also available free from LTC, and recommended for designing custom filters, is the general-purpose analog filter
design software FilterCADTM for Windows®. This software
includes tools for finding the necessary f0, Q and gain
parameters to meet target filter specifications such as
frequency response.
fO =
Q =
10kΩ
1
=
(100kHz)
2πC (R1)R2
R2
RQ
=
(R1)R2
RQ
R 100kHz
= Q
R2 fO
(10kΩ)R2
R1 = 10k and C = 159pF are internal to the LTC1562 while
R2 and RQ are external.
A typical design procedure proceeds from the desired fO
and Q as follows, using finite-tolerance fixed resistors.
First find the ideal R2 value for the desired fO:
2
100kHz
R2(Ideal) =
(10kΩ)
fO
Then select a practical R2 value from the available finitetolerance resistors. Use the actual R2 value to find the
desired RQ, which also will be approximated with finite
tolerance:
RQ = Q (10kΩ)R2
The fO range is approximately 10kHz to 150kHz, limited
mainly by the magnitudes of the external resistors
required. As shown above, R2 varies with the inverse
square of fO. This relationship desensitizes fO to R2’s
Operational Filter and FilterCAD are trademarks of Linear Technology Corporation.
Windows is a registered trademark of Microsoft Corporation.
1562fa
7
LTC1562
U
U
W
U
APPLICATIONS INFORMATION
Basic Lowpass
tolerance (by a factor of 2 incrementally), but it also
implies that R2 has a wider range than fO. (RQ and RIN also
tend to scale with R2.) At high fO these resistors fall below
5k, heavily loading the outputs of the LTC1562 and leading
to increased THD and other effects. At the other extreme,
a lower fO limit of 10kHz reflects an arbitrary upper
resistor limit of 1MΩ. The LTC1562’s MOS input circuitry
can accommodate higher resistor values than this, but
junction leakage current from the input protection circuitry may cause DC errors.
When ZIN of Figure 3 is a resistor of value RIN, a standard
2nd order lowpass transfer function results from VIN to V2
(Figure 5):
V2(s)
– HLω O2
= HLP (s) =
VIN(s)
s2 + (ω O / Q)s + ω O2
The DC gain magnitude is HL = R2/RIN. (Note that the
transfer function includes a sign inversion.) Parameters
ωO (= 2πfO) and Q are set by R2 and RQ as above. For a 2nd
order lowpass response the gain magnitude becomes QHL
The 2nd order transfer functions HLP(s), HBP(s) and
HHP(s) (below) are all inverting so that, for example, at DC
the lowpass gain is – HL. If two such sections are cascaded, these phase inversions cancel. Thus, the filter in the
application schematic on the first page of this data sheet
is a dual DC preserving, noninverting, rail-to-rail lowpass
filter, approximating two “straight wires with frequency
selectivity.”
RIN
VIN
RQ
VOUT
INV
fL
fO
Figure 5. Basic Lowpass Configuration
fH
fO
; fO = fL fH
fH – fL
2
1
–1
fL = fO + + 1
2Q
2Q
2
1
1
fH = fO + + 1
2Q
2Q
HIGHPASS RESPONSE
HP
HL
0.707 HL
HP
HH
0.707 HH
fC
fC
fP
f (LOG SCALE)
Q=
1562 F05
GAIN (V/V)
0.707 HB
2
1
1
fC = fO 1 –
+ 1– 2 + 1
2Q2
2Q
1
2
1
1
fC = fO 1 –
+ 1–
+ 1
2
2Q2
2Q
–1
–1
2Q2
1
HP = HL
1
1
1–
Q
4Q2
fP
f (LOG SCALE)
f (LOG SCALE)
fP = fO 1 –
V2
2nd ORDER
LOWPASS RESPONSE
GAIN (V/V)
GAIN (V/V)
BANDPASS RESPONSE
V1
1/4 LTC1562
Figure 4 shows further details of 2nd order lowpass,
bandpass and highpass responses. Configurations to
obtain these responses appear in the next three sections.
HB
R2
1
fP = fO 1 –
2Q2
1
HP = HH
1
1
1–
Q
4Q2
Figure 4. Characteristics of Standard 2nd Order Filter Responses
1562fa
8
LTC1562
U
U
W
U
APPLICATIONS INFORMATION
at frequency fO, and for Q > 0.707, a gain peak occurs at
a frequency below fO, as shown in Figure 4.
Basic Bandpass
There are two different ways to obtain a bandpass function
in Figure 3, both of which give the following transfer
function form:
HBP (s) =
(
)
– HB ω O / Q s
(
)
CIN
VIN
2
s2 + ω O / Q s + ω O
RQ
CIN
RIN
VIN
VIN
R2
RQ
RQ
R2
VOUT
V1
V2
VOUT
INV
V1
V2
2nd ORDER
2nd ORDER
1/4 LTC1562
1/4 LTC1562
1562 F06
(a) Resistive Input
R2
VOUT
ωO = 2πfO and Q are set by R2 and RQ as described previously in Setting fO and Q. When ZIN is a resistor of value
RIN, a bandpass response results at the V1 output (Figure
6a) with a gain parameter HB = RQ/RIN. Alternatively, a
capacitor of value CIN gives a bandpass response at the V2
output (Figure 6b), with the same HBP(s) expression, and
the gain parameter now HB = (RQ/10kΩ)(CIN/159pF). This
transfer function has a gain magnitude of HB (its peak value)
when the frequency equals fO and has a phase shift of 180°
at that frequency. Q measures the sharpness of the peak
(the ratio of fO to – 3dB bandwidth) in a 2nd order bandpass function, as illustrated in Figure 4.
INV
Parameters ωO = 2πfO and Q are set by R2 and RQ as
above. The highpass gain parameter is HH = CIN/159pF.
For a 2nd order highpass response the gain magnitude at
frequency fO is QHH, and approaches HH at high frequencies (f >> fO). For Q > 0.707, a gain peak occurs at a
frequency above fO as shown in Figure 4. The transfer
function includes a sign inversion.
(b) Capacitive Input
Figure 6. Basic Bandpass Configurations
Basic Highpass
When ZIN of Figure 3 is a capacitor of value CIN, a highpass
response appears at the V1 output (Figure 7).
V1(s)
– HHs2
= HHP (s) =
VIN(s)
s2 + (ω O / Q)s + ω O2
INV
V1
V2
2nd ORDER
1/4 LTC1562
1562 F07
Figure 7. Basic Highpass Configuration
Signal Swings
The V1 and V2 outputs are capable of swinging to within
roughly 100mV of each power supply rail. As with any
analog filter, the signal swings in each 2nd order section
must be scaled so that no output overloads (saturates),
even if it is not used as a signal output. (Filter literature
often calls this the “dynamics” issue.) When an unused
output has a larger swing than the output of interest, the
section’s gain or input amplitude must be scaled down to
avoid overdriving the unused output. The LTC1562 can
still be used with high performance in such situations as
long as this constraint is followed.
For an LTC1562 section as in Figure 3, the magnitudes of
the two outputs V2 and V1, at a frequency ω = 2πf, have
the ratio,
| V2( jω ) | (100kHz)
=
| V1( jω ) |
f
regardless of the details of ZIN. Therefore, an input frequency above or below 100kHz produces larger output
amplitude at V1 or V2, respectively. This relationship can
guide the choice of filter design for maximum dynamic
range in situations (such as bandpass responses) where
there is more than one way to achieve the desired frequency response with an LTC1562 section.
1562fa
9
LTC1562
U
W
U
U
APPLICATIONS INFORMATION
Because 2nd order sections with Q ≥ 1 have response
peaks near fO, the gain ratio above implies some rules of
thumb:
fO < 100kHz ⇒ V2 tends to have the larger swing
fO > 100kHz ⇒ V1 tends to have the larger swing.
The following situations are convenient because the
relative swing issue does not arise. The unused output’s
swing is naturally the smaller of the two in these cases:
Lowpass response (resistor input, V2 output, Figure 5)
with fO < 100kHz
Bandpass response (capacitor input, V2 output, Figure
6b) with fO < 100kHz
Bandpass response (resistor input, V1 output, Figure
6a) with fO > 100kHz
Highpass response (capacitor input, V1 output, Figure
7) with fO > 100kHz
The LTC1562-2, a higher frequency derivative of the
LTC1562, has a design center fO of 200kHz compared to
100kHz in the LTC1562. The rules summarized above
apply to the LTC1562-2 but with 200kHz replacing the
100kHz limits. Thus, an LTC1562-2 lowpass filter section
with fO below 200kHz automatically satisfies the desirable
condition of the unused output carrying the smaller signal
swing.
RIN
10k
VIN
RQ
6.98k
INV
R2
10k
V1
V2
2nd ORDER
1/4 LTC1562
CL
30pF
VOUT
RL
(EXTERNAL
LOAD RESISTANCE)
1562 F08
Figure 8. 100kHz, Q = 0.7 Lowpass Circuit for
Distortion vs Loading Test
Low Level or Wide Range Input Signals
The LTC1562 contains a built-in capability for low noise
amplification of low level signals. The ZIN impedance in
each 2nd order section controls the block’s gain. When set
for unity passband gain, a 2nd order section can deliver an
output signal more than 100dB above the noise level. If low
level inputs require further dynamic range, reducing the
value of ZIN boosts the signal gain while reducing the input
referred noise. This feature can increase the SNR for low
level signals. Varying or switching ZIN is also an efficient
way to effect automatic gain control (AGC). From a system
viewpoint, this technique boosts the ratio of maximum
signal to minimum noise, for a typical 2nd order lowpass
response (Q = 1, fO = 100kHz), to 118dB.
Input Voltages Beyond the Power Supplies
Properly used, the LTC1562 can accommodate input
voltage excursions well beyond its supply voltage. This
requires care in design but can be useful, for example,
when large out-of-band interference is to be removed from
a smaller desired signal. The flexibility for different input
voltages arises because the INV inputs are at virtual
ground potential, like the inverting input of an op amp with
negative feedback. The LTC1562 fundamentally responds
to input current and the external voltage VIN appears only
across the external impedance ZIN in Figure 3.
To accept beyond-the-supply input voltages, it is important to keep the LTC1562 powered on, not in shutdown
mode, and to avoid saturating the V1 or V2 output of the
2nd order section that receives the input. If any of these
conditions is violated, the INV input will depart from a
virtual ground, leading to an overload condition whose
recovery timing depends on circuit details. In the event
that this overload drives the INV input beyond the supply
voltages, the LTC1562 could be damaged.
The most subtle part of preventing overload is to consider
the possible input signals or spectra and take care that
none of them can drive either V1 or V2 to the supply limits.
Note that neither output can be allowed to saturate, even
if it is not used as the signal output. If necessary the
passband gain can be reduced (by increasing the impedance of ZIN in Figure 3) to reduce output swings.
The final issue to be addressed with beyond-the-supply
inputs is current and voltage limits. Current entering the
virtual ground INV input flows eventually through the
output circuitry that drives V1 and V2. The input current
magnitude (VIN/ZIN in Figure 3) should be limited by
design to less than 1mA for good distortion performance.
On the other hand, the input voltage VIN appears across the
1562fa
10
LTC1562
U
U
W
U
APPLICATIONS INFORMATION
external component ZIN, usually a resistor or capacitor.
This component must of course be rated to sustain the
magnitude of voltage imposed on it.
Lowpass “T” Input Circuit
The virtual ground INV input in the Operational Filter block
provides a means for adding an “extra” lowpass pole to
any resistor-input application (such as the basic lowpass,
Figure 5, or bandpass, Figure 6a). The resistor that would
otherwise form ZIN is split into two parts and a capacitor
to ground added, forming an R-C-R “T” network (Figure
9). This adds an extra, independent real pole at a frequency:
fP =
1
2πRPCT
where CT is the new external capacitor and RP is the
parallel combination of the two input resistors RINA and
RINB. This pair of resistors must normally have a prescribed series total value RIN to set the filter’s gain as
described above. The parallel value RP can however be set
arbitrarily (to RIN/4 or less) which allows choosing a
convenient standard capacitor value for CT and fine tuning
the new pole with RP.
RINA
A practical limitation of this technique is that the CT capacitor values that tend to be required (hundreds or thousands
of pF) can destabilize the op amp in Figure 3 if RINB is too
small, leading to AC errors such as Q enhancement. For this
reason, when RINA and RINB are unequal, preferably the
larger of the two should be placed in the RINB position.
Highpass “T” Input Circuit
A method similar to the preceding technique adds an
“extra” highpass pole to any capacitor-input application
(such as the bandpass of Figure 6b or the highpass of
Figure 7). This method splits the input capacitance CIN into
two series parts CINA and CINB, with a resistor RT to ground
between them (Figure 10). This adds an extra 1st order
highpass corner with a zero at DC and a pole at the
frequency:
fP =
1
2πRTCP
where CP = CINA + CINB is the parallel combination of the
two capacitors. At the same time, the total series capacitance CIN will control the filter’s gain parameter (HH in
Basic Highpass). For a given series value CIN, the parallel
value CP can still be set arbitrarily (to 4CIN or greater).
CINA
RINB
VIN
CINB
VIN
CT
RQ
INV
V1
R2
V2
RT
RQ
INV
2nd ORDER
V1
R2
V2
2nd ORDER
1/4 LTC1562
1/4 LTC1562
1562 F09
1562 F10
Figure 9. Lowpass “T” Input Circuit
Figure 10. Highpass “T” Input Circuit
The procedure therefore is to begin with the target extra
pole frequency fP. Determine the series value RIN from the
gain requirement. Select a capacitor value CT such that RP
= 1/(2πfPCT) is no greater than RIN/4, and then choose
RINA and RINB that will simultaneously have the parallel
value RP and the series value RIN. Such RINA and RINB can
be found directly from the expression:
The procedure then is to begin with the target corner (pole)
frequency fP. Determine the series value CIN from the gain
requirement (for example, CIN = HH(159pF) for a highpass).
Select a resistor value RT such that CP = 1/(2πRTfP) is at
least 4CIN, and select CINA and CINB that will simultaneously
have the parallel value CP and the series value CIN. Such
CINA and CINB can be found directly from the expression:
1
1
RIN ±
RIN2 – (4RINRP)
2
2
1
1
CP ±
CP2 – (4CINCP)
2
2
1562fa
11
LTC1562
U
W
U
U
APPLICATIONS INFORMATION
This procedure can be iterated, adjusting the value of RT,
to find convenient values for CINA and CINB since resistor
values are generally available in finer increments than
capacitor values.
Different “fO” Measures
Standard 2nd order filter algebra, as in Figure 4 and the
various transfer-function expressions in this data sheet,
uses a center frequency parameter fO (or ωO, which is
2πfO). fO can also be measured in practical ways, including:
• The frequency where a bandpass response has 180°
phase shift
• The frequency where a bandpass response has peak
gain
• The geometric mean of the – 3.01dB gain frequencies in
a bandpass (√ƒLƒH in Figure 4)
An ideal mathematical 2nd order response yields exactly
the same frequency by these three measures. However,
real 2nd order filters with finite-bandwidth circuitry show
small differences between the practical fO measures,
which may be important in critical applications. The issue
is chiefly of concern in high-Q bandpass applications
where, as the data below illustrate, the different f0 measurements tend to converge anyway for the LTC1562. At
low Q the bandpass peak is not sharply defined and the
– 3dB frequencies fL and fH are widely separated from this
peak.
The LTC1562’s fO is trimmed in production to give an
accurate 180° phase shift in the configuration of Figure
6a with resistor values setting f0 = 100kHz and Q = 1.
Table 1 below shows typical differences between fO
values measured via the bandpass 180° criterion and fO
values measured using the two other methods listed
above (Figure 6a, RIN = RQ).
Table 1
fO
(BP 180°)
Q=1
BP-PEAK fO
Q=1
√ƒ
ƒLƒH fO
Q=5
BP-PEAK fO
Q=5
√ƒ
ƒLƒH fO
60kHz
+ 0.3%
+ 0.3%
+ 0.05%
+ 0.05%
100kHz
+ 0.6%
+ 0.6%
+ 0.1%
+ 0.1%
140kHz
+ 0.8%
+ 0.8%
+ 0.15%
+ 0.15%
LTC1562 Demo Board
The LTC1562 demo board is assembled with an LTC1562
or LTC1562A in a 20-pin SSOP package and power supply
decoupling capacitors. Jumpers on the board configure
the LTC1562 for dual or single supply operation and power
shutdown. Pads for surface mount resistors and capacitors are provided to build application-specific filters. Also
provided are terminals for inputs, outputs and power
supplies.
1562fa
12
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
Quad 3rd Order Butterworth Lowpass Filter, Gain = – 1
RIN1B
VIN1
CIN1
VOUT2
1
RQ1
R21
2
3
5
5V
0.1µF
6
R23
8
9
RIN3A
RIN3B
VIN3
RQ3
CIN3
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
V + LTC1562
V–
SHDN
AGND
V2 A
V2 D
V1 A
V1 D
INV A
VOUT3
INV D
RIN2B
20
19
RQ2
18
R22
16
10
RIN2A
VIN2
CIN2
–10
–5V
15
0.1µF
13
R24
–20
–30
–40
12
11
f– 3dB = 100kHz
0
GAIN (dB)
VOUT1
RIN1A
Amplitude Response
RQ4
RIN4B
RIN4A
VIN4
CIN4
VOUT4
1562 TA05a
–50
–60
10k
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
100k
FREQUENCY (Hz)
1M
1562 TA05b
Quad 3rd Order
Butterworth
Lowpass Filters
f – 3dB
20kHz
f – 3dB
40kHz
f – 3dB
60kHz
f – 3dB
80kHz
f – 3dB
100kHz
f – 3dB
120kHz
f – 3dB
140kHz
CIN
RINA
RINB
RQ
R2
220pF
44.2k
205k
249k
249k
1000pF
4.32k
57.6k
61.9k
61.9k
1000pF
3.16k
24.3k
27.4k
27.4k
1000pF
2.43k
13.0k
15.4k
15.4k
1000pF
1.96k
8.06k
10.0k
10.0k
1000pF
1.87k
5.11k
6.98k
6.98k
1000pF
1.69k
3.4k
5.11k
5.11k
All four sections have identical RINA, RINB and CIN values. All resistor values are ±1%
1562fa
13
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
Dual 4th Order Lowpass Filters
Amplitude Response
10
RIN2
VIN2
1
RQ1
R21
2
3
5
5V
0.1µF
6
8
RIN3
VIN1
R23
9
RQ3
10
INV B
INV C
V1 B
V1 C
V2 C
V2 B
V + LTC1562
V–
AGND
SHDN
V2 D
V2 A
V1 D
V1 A
INV D
INV A
0
20
19
RQ2
–10
18
R22
–20
VOUT2
–5V
16
15
GAIN (dB)
RIN1
0.1µF
VOUT1
13
–30
–40
–50
12
R24
–60
11
RQ4
–70
RIN4
BUTTERWORTH
f – 3dB = 100kHz
–80
10k
1562 TA03a
100k
FREQUENCY (Hz)
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
1M
1562 TA03b
Quick Design Formulas for Some Popular Response Types:
Butterworth
(Maximally Flat Passband)
for fC 10kHz to 140kHz
R21, R23, RIN1, RIN3 =
10k
100kHz
ƒC
RQ1, RQ3 =
5.412k
100kHz
ƒC
R22, R24, RIN2, RIN4 =
10k
100kHz
ƒC
RQ2, RQ4 =
13.07k
100kHz
ƒC
Chebyshev
(Equiripple Passband)
for fC 20kHz to 120kHz
2
14.24k
100kHz
ƒC
7.26k
100kHz
ƒC
7.097k
100kHz
ƒC
17.53k
100kHz
ƒC
2
Bessel
(Good Transient Response)
for fC 10kHz to 70kHz
2
3.951k
100kHz
ƒC
5.066k
100kHz
ƒC
4.966k
100kHz
ƒC
3.679k
100kHz
ƒC
2
2
2
Notes: fC is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at fC. For Chebyshev filters with
± 0.1dB passband ripple up to 0.95 fC, use LTC1562 “A” grade.
Example: Butterworth response, fC = 50kHz. from the formulas above, R21 = R23 = RIN1 = RIN3 = 10k(100kHz/50kHz)2
= 40k. RQ1 = RQ3 = 5.412k(100kHz/50kHz) = 10.82k. R22 = R24 = RIN2 = RIN4 = 10k(100kHz/50kHz)2 = 40k.
1562 TA03 TABLE
RQ2 = RQ4 = 13.07k(100kHz/50kHz) = 26.14k. Use nearest 1% values.
1562fa
14
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
8th Order Lowpass Filters
Amplitude Response
10
RIN2
VIN
1
RQ1
R21
2
3
5
5V
0.1µF
6
R23
8
9
RQ3
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
V–
V + LTC1562
SHDN
AGND
V2 A
V2 D
V1 A
V1 D
INV D
INV A
19
RQ2
18
R22
16
15
13
–10
–20
–5V
0.1µF
R24
–30
–40
–50
–60
12
11
CHEBYSHEV
fC = 100kHz
0
20
GAIN (dB)
RIN1
–70
RQ4
–80
–90
10k
RIN4
VOUT
RIN3
100k
FREQUENCY (Hz)
500k
1562 TA04b
1562 TA04a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
Quick Design Formulas for Some Popular Response Types:
Butterworth
(Maximally Flat Passband)
for fC 10kHz to 140kHz
R21 = RIN1 = 10k
100kHz
ƒC
RQ1 = 6.01k
100kHz
ƒC
R22 = RIN2 = 10k
100kHz
ƒC
RQ2 = 9k
100kHz
ƒC
R23 = RIN3 = 10k
100kHz
ƒC
RQ3 = 5.1k
100kHz
ƒC
R24 = RIN4 = 10k
100kHz
ƒC
RQ4 = 25.63k
100kHz
ƒC
Chebyshev
(Equiripple Passband)
for fC 20kHz to 120kHz
2
Bessel
(Good Transient Response)
for fC 10kHz to 70kHz
2
R21 = 7.51k
100kHz
, RIN1 = 2.2R21*
ƒC
RQ1 = 119.3k
2
R22 = RIN2 = 14.99k
RQ2 = 279.9k
2
R23 = RIN3 = 7.15k
100kHz
ƒC
100kHz
ƒC
RQ1 = 3.63k
100kHz
ƒC
R22 = RIN2 = 2.07k
100kHz
ƒC
RQ2 = 5.58k
100kHz
ƒC
R23 = RIN3 = 2.96k
100kHz
ƒC
RQ3 = 3.05k
100kHz
ƒC
R24 = RIN4 = 3.14k
100kHz
ƒC
RQ4 = 2.84k
100kHz
ƒC
2
100kHz
100kHz
ƒC + 2440kHz
ƒC
100kHz
ƒC
RQ3 = 118.1k
2
100kHz
100kHz
ƒC
ƒC + 560kHz
R21 = RIN1 = 2.61k
2
100kHz
100kHz
ƒC
ƒC + 530kHz
2
R24 = 26.7k
100kHz
R24*
, RIN4 =
ƒC
2.2
RQ4 = 8.75k
100kHz
ƒC
2
2
2
2
Notes: fC is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at fC. For Chebyshev filters with
± 0.1dB passband ripple up to 0.95 fC, use LTC1562 “A” grade. *The resistor values marked with an asterisk (*) in the
Chebyshev formulas (R21 and R24) should be rounded to the nearest standard finite-tolerance value before computing
the values dependent on them (RIN1 and RIN4 respectively).
Example: Chebyshev response, fC = 100kHz. The formulas above give R21 = 7.51k, nearest standard 1% value 7.50k.
Using this 1% value gives RIN1 = 16.5k, already a standard 1% value. RQ1 = 18.075k, nearest 1% value 18.2k.
R22 = RIN2 = 14.99k, nearest 1% value 15k. RQ2 = 11.02k, nearest 1% value 11k. R23 = RIN3 = 7.15k, already a
standard 1% value. RQ3 = 18.75k, nearest 1% value 18.7k. R24 = 26.7k, already a standard 1% value. This gives
RIN4 = 12.14k, nearest 1% value 12.1k. RQ4 = 8.75k, nearest 1% value 8.66k.
1562 TA04 TABLE
1562fa
15
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
8th Order Bandpass Filter, Single 5V Supply,
Center Frequency
– 3dB Bandwidth =
10
Amplitude Response
10
RIN2
CIN1
0
VIN
RQ1
R21
2
3
5
5V
0.1µF
6
R23
8
9
RQ3
CIN3
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
V + LTC1562
V–
AGND
SHDN
V2 D
V2 A
V1 D
V1 A
INV D
INV A
20
19
RQ2
18
R22
–10
16
15
fCENTER = 80kHz
– 20
GAIN (dB)
1
1µF
R24
13
– 30
– 40
– 50
– 60
12
–70
RQ4
11
– 80
– 90
VOUT
RIN4
40 48 56 64 72 80 88 96 104 112 120
FREQUENCY (kHz)
1562 TA07a
1562 TA07b
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
Quick Design Formulas for Center Frequency fC (Recommended Range 40kHz to 140kHz):
R21 = R23 = 10.6k
100kHz
ƒC
R22 = R24 = 9.7k
100kHz
ƒC
CIN1 = CIN3 = 159pF
10k
RQ1
2
RQ1 = RQ3 = 164.6k
100kHz
100kHz
ƒC
ƒC + 319kHz
RQ2 = RQ4 = 143.2k
100kHz
100kHz
ƒC
ƒC + 294kHz
2
RIN2 = RIN4 =
R22RQ1CIN1
100kHz
(10k)(10.6pF) ƒC + 286kHz
Notes: RQ1, R22 and CIN1 should be rounded to the nearest standard finite-tolerance value before using these
values in the later formulas.
Example: Center frequency fC of 80kHz. The formulas give R21 = R23 = 16.56k, nearest standard 1% value 16.5k.
RQ1 = RQ3 = 51.56k, nearest 1% value 51.1k. R22 = R24 = 15.15k, nearest 1% value 15k. RQ2 = RQ4 = 47.86k,
nearest 1% value 47.5k. CIN1 = CIN2 = 31.11pF using 51.1k for RQ1, nearest standard 5% capacitor value 33pF.
This and the 1% value R22 = 15k also go into the calculation for RIN2 = RIN4 = 65.20k, nearest 1% value 64.9k.
1562 TA07 TABLE
1562fa
16
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
8th Order Bandpass Filter, Single 5V Supply,
Center Frequency
– 1dB Bandwidth =
10
Amplitude Response
RIN2
VIN
1
RQ1
R21
2
3
5
5V
0.1µF
6
R23
8
9
RQ3
10
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
–
V + LTC1562 V
AGND
SHDN
V2 A
V2 D
V1 A
V1 D
INV D
INV A
0
20
19
RQ2
–10
18
R22
– 20
GAIN (dB)
RIN1
16
15
13
1µF
R24
– 30
– 40
– 50
– 60
12
11
fCENTER = 100kHz
–70
RQ4
– 80
VOUT
RIN4
RIN3
– 90
60 68 76 84 92 100 108 116 124 132 140
FREQUENCY (kHz)
1562 TA06a
1562 TA06b
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
Quick Design Formulas for a Center Frequency fC (Recommended Range 50kHz to 120kHz):
R21 = R23 = 11.7k
100kHz
ƒC
R22 = R24 = 8.66k
100kHz
ƒC
RIN1 = RIN3 =
ƒC + 1736kHz R21
2.56
100kHz
RQ1 = RQ3 = 215.5k
100kHz
100kHz
ƒC + 229kHz
ƒC
RIN2 = RIN4 =
ƒC + 634kHz RQ2
14.36
100kHz
RQ2 = RQ4 = 286.2k
100kHz
100kHz
ƒC + 351kHz
ƒC
2
2
Notes: R21 and RQ2 should be rounded to the nearest standard finite-tolerance value before using these values in
the later formulas. For fC < 100kHz, the maximum peak-to-peak passband input level is (fC /100kHz)5V. Use
LTC1562A for minimum variation of passband gain.
Example: Center frequency fC of 100kHz. The formulas give R21 = R23 = 11.7k, nearest standard 1% value 11.5k.
This value gives RIN1 = RIN3 = 82.46k, nearest 1% value 82.5k. RQ1 = RQ3 = 65.5k, nearest 1% value 64.9k.
R22 = R24 = 8.66k, already a standard 1% value. This gives RIN2 = RIN4 = 32.4k (again already a standard 1% value).
RQ2 = RQ4 = 63.45k, nearest 1% value 63.4k. If LTC1562A is used, resistor tolerances tighter than 1% will further
1562 TA06 TABLE
improve the passband gain accuracy.
1562fa
17
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
8th Order Bandpass (High Frequency) Filter
Center Frequency
– 3dB Bandwidth =
, Gain = 10
10
Amplitude Response
RIN2
VIN
1
RQ1
2
R21
3
5
V+
0.1µF
6
R23
8
9
RQ3
10
30
INV B
INV C
V1 B
V1 C
V2 B
V2 C
–
V + LTC1562 V
AGND
SHDN
V2 A
V2 D
V1 A
V1 D
INV D
INV A
19
RQ2
18
R22
10
0
16
15
fCENTER = 100kHz
20
20
GAIN (dB)
RIN1
V–
0.1µF
R24
13
–10
– 20
– 30
– 40
12
– 50
RQ4
11
– 60
– 70
RIN4
VOUT
40
RIN3
60
80 100 120 140
FREQUENCY (kHz)
1562 TA08a
160
180
1562 TA08b
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
8th Order Bandpass Filter
f
– 3dB BW = CENTER, Gain = 10
10
fCENTER
80kHz
fCENTER
90kHz
fCENTER
100kHz
fCENTER
110kHz
fCENTER
120kHz
fCENTER
130kHz
fCENTER
140kHz
4.64k
46.4k
12.4k
5.23k
52.3k
15.4k
6.34k
42.2k
10.0k
5.11k
38.3k
8.25k
5.11k
34.8k
6.98k
5.49k
32.4k
5.9k
5.62k
30.1k
5.11k
46.4k
46.4k
12.4k
52.3k
52.3k
15.4k
42.2k
42.2k
10.0k
38.3k
38.3k
8.25k
34.8k
34.8k
6.98k
32.4k
32.4k
5.90k
30.1k
30.1k
5.11k
Side B
RIN1
RQ1
R21
Sides A, C, D
RIN2, RIN3, RIN4
RQ2, RQ3, RQ4
R22, R23, R24
All resistor values are ±1%
1562fa
18
LTC1562
U
TYPICAL APPLICATIONS
(Basic)
8th Order Wideband Bandpass Filter
fCENTER = 50kHz, – 3dB BW 40kHz to 60kHz
Amplitude Response
RIN2
69.8k
CIN1
22pF
10
0
RQ1 59k
2
R21 56.2k
3
5
V+
0.1µF
6
8
R23 63.4k
9
RQ3 82.5k
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
20
19
RQ2 48.7k
18
R22 34.8k
– 16
V + LTC1562 V
15
AGND
SHDN
13
V2 D
V2 A
12
V1 D
V1 A
11
INV D
INV A
– 10
GAIN (dB)
1
VIN
V–
1µF
– 20
– 30
– 40
R24 28.7k
– 50
RQ4 100k
– 60
20
CIN3
27pF
100
FREQUENCY (kHz)
VOUT
1562 TA09b
CIN4 47pF
1562 TA09a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
Amplitude Response
8th Order Highpass 0.05dB Ripple Chebyshev Filter fCUTOFF = 30kHz
10
CIN1
150pF
RQ1, 10.2k
R21, 35.7k
2
3
5
5V
0.1µF
6
8
CIN3
150pF
R23, 107k
9
RQ3, 54.9k
10
INV B
INV C
V1 B
V1 C
V2 C
V2 B
V + LTC1562
V–
SHDN
AGND
V2 A
V2 D
V1 A
V1 D
INV D
INV A
0
20
19
RQ2, 22.1k
18
R22, 66.5k
16
15
–10
CIN2
150pF
–20
GAIN (dB)
1
CIN
–5V
0.1µF
13
–30
–40
–50
–60
12
R24, 127k
11
RQ4, 98.9k
CIN4
150pF
–70
–80
–90
1562 TA10a
VOUT
1k
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
10k
100k
FREQUENCY (Hz)
1M
1562 TA10b
TOTAL OUTPUT NOISE = 40µVRMS
Amplitude Response
2nd Order 30kHz Highpass Cascaded with 6th Order 138kHz Lowpass
RIN2, 5.23k
1
VIN
RQ1, 30.1k
R21, 110k
2
3
5
5V
0.1µF
6
8
R23, 5.23k
9
RQ3, 14k
10
RIN3, 8.06k
INV B
INV C
V1 B
V1 C
V2 B
V2 C
20
10
20
19
18
– 16
V + LTC1562 V
15
AGND
SHDN
13
V2 D
V2 A
12
V1 D
V1 A
11
INV D
INV A
RQ2, 5.11k
0
R22, 5.23k
–10
GAIN (dB)
CIN1
150pF
–5V
0.1µF
–40
–60
RQ4, 3.74k
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
–30
–50
R24, 5.23k
RIN4, 3.4k
–20
–70
VOUT
1562 TA11a
–80
10
100
FREQUENCY (kHz)
400
1562 TA11b
ALL RESISTORS = 1% METAL FILM
1562fa
19
LTC1562
U
U
W
U
APPLICATIONS INFORMATION
Notches and Elliptic Responses
The basic (essentially all-pole) LTC1562 circuit techniques described so far will serve many applications.
However, the sharpest-cutoff lowpass, highpass and bandpass filters include notches (imaginary zero pairs) in the
stopbands. A notch, or band-reject, filter has zero gain at
a frequency fN. Notches are also occasionally used by
themselves to reject a narrow band of frequencies. A
number of circuit methods will give notch responses from
an Operational Filter block. Each method exhibits an inputoutput transfer function that is a standard 2nd order bandreject response:
HBR(s) =
(
)
– HN s2 + ω N2
fN. The two signals then cancel out at frequency fN. The
notch depth (the completeness of cancellation) will be
infinite to the extent that the two paths have matching
gains. Three practical circuit methods are presented here,
with different features and advantages.
Examples and design procedures for practical filters using
these techniques appear in a series of articles attached to
this data sheet on the Linear Technology web site
(www.linear-tech.com). Also available free is the analog
filter design software, FilterCAD for Windows, recommended for designing filters not shown in the Typical
Applications schematics in this data sheet.
Elementary Feedforward Notches
s2 + (ω O / Q)s + ω 2O
with parameters ωN = 2πfN and HN set by component
values as described below. (ω0 = 2πf0 and Q are set for the
Operational Filter block by its R2 and RQ resistors as
described earlier in Setting f0 and Q). Characteristically,
the gain magnitude |HBR(j2πf)| has the value HN(fN2/f02) at
DC (f = 0) and HN at high frequencies (f >> fN), so in
addition to the notch, the gain changes by a factor:
High Frequency Gain ƒ 2O
= 2
DC Gain
ƒN
A “textbook” method to get a 180° phase difference at
frequency fN for a notch is to dedicate a bandpass 2nd
order section (described earlier under Basic Bandpass),
which gives 180° phase shift at the section’s center
frequency fO (Figure 11, with CIN1 = 0), so that fN = fO. The
bandpass section of Figure 6a, at its center frequency fO,
has a phase shift of 180° and a gain magnitude of HB =
RQ /RIN. A notch results in Figure 11 if the paths summed
into virtual ground have the same gains at the 180°
frequency (then IO = 0). This requires a constraint on the
resistor values:
RIN2 RQ1
=
RFF2 RIN1
The common principle in the following circuit methods is
to add a signal to a filtered replica of itself having equal gain
and 180° phase difference at the desired notch frequency
CIN1
RIN1
VIN
RQ1
INV
V1
R21
RIN2
V2
IO
VIRTUAL
GROUND
2nd ORDER
1/4 LTC1562
RFF2
RGAIN
–
+
VOUT
1562 F11
Figure 11. Feedforward Notch Configuration for fN ≥ fO
1562fa
20
LTC1562
U
W
U
U
APPLICATIONS INFORMATION
Note that the depth of the notch depends on the accuracy
of this resistor ratioing. The virtual-ground summing
point in Figure 11 may be from an op amp as shown, or in
a practical cascaded filter, the INV input of another Operational Filter block. The transfer function in Figure 11 with
CIN1 = 0 is a “pure” notch (fN = f0) of the HBR(s) form above,
and the parameters are:
ƒN = ƒ O
R
H N = GAIN
RFF2
Feedforward Notches for fN > f0
When CIN1 ≠ 0 in Figure 11, the notch frequency fN is above
the center frequency f0 and the response has a lowpass
shape as well as a notch (Figure 13). CIN1 contributes
phase lead, which increases the notch frequency above
the center frequency of the 2nd order Operational Filter
block. The resistor constraint from the previous section
also applies here and the HBR(s) parameters become:
1
ƒN = ƒ O
Because fN = f0 in this case, the gain magnitude both at DC
and at high frequencies (f >> fN) is the same, HN (assuming
that the op amp in Figure 11 adds no significant frequency
response). Figure 12 shows this. Such a notch is inefficient as a cascaded part of a highpass, lowpass or bandpass filter (the most common uses for notches). Variations of Figure 11 can add a highpass or lowpass shape to
the notch, without using more Operational Filter blocks.
The key to doing so is to decouple the notch frequency fN
from the center frequency f0 of the Operational Filter block
(this is shown in Figures 13 and 15). The next two sections
summarize two variations of Figure 11 with this highpass/
lowpass shaping, and the remaining section shows a
different approach to building notches.
1–
RGAIN ƒ 2O
HN =
RFF2 ƒN2
C is the internal capacitor value in the Operational Filter
block (in the LTC1562, 159pF).
The configuration of Figure 11 is most useful for a
stopband notch in a lowpass filter or as an upper stopband
notch in a bandpass filter, since the two resistors R IN2 and
RFF2 can replace the input resistor RIN of either a lowpass
section (Figure 5) or a resistor-input bandpass section
(Figure 6a) built from a second Operational Filter block.
0
20
DC GAIN = HN
–20
– 40
GAIN (dB)
GAIN (dB)
0
– 60
–100
fN = fO = 100kHz
HN = 1
Q=1
10
100
FREQUENCY (kHz)
1000
AN54 • TA18
Figure 12. Notch Response with fN = fO
( )
fN2
fO2
HIGH FREQ
GAIN = HN
–20
– 40
– 80
RIN1CIN1
RQ1C
– 60
fO = 100kHz
fN = 200kHz
Q=1
DC GAIN = 0dB
10
100
FREQUENCY (kHz)
1000
1562 F13
Figure 13. Notch Response with fN > fO
1562fa
21
LTC1562
U
W
U
U
APPLICATIONS INFORMATION
Feedforward Notches for fN < f0
Just as feedforward around an inverting bandpass section
yields a notch at the section’s f0 (Figure 11 with CIN1 = 0),
feedforward around an inverting lowpass section causes
a notch at zero frequency (which is to say, a highpass
response). Moreover, and this is what makes it useful,
introducing a capacitor for phase lead moves the notch
frequency up from DC, exactly as CIN1 in Figure 11 moves
the notch frequency up from the center frequency f0. In
Figure 14, the inverting lowpass output (V2) of the Operational Filter block is summed, at a virtual ground, with a
fed-forward input signal. Capacitor CIN1 shifts the resulting notch frequency, fN, up from zero, giving a low
frequency notch with a highpass shape (Figure 15). The
HBR(s) response parameters are now:
RIN2 RQ1CIN1
=
RFF2
R1C
R1 and C are the internal precision components (in the
LTC1562, 10k and 159pF respectively) as described above
in Setting f0 and Q.
The configuration of Figure 14 is most useful as a lower
stopband notch in a bandpass filter, because the resistors
RIN2 and RFF2 can replace the input resistor RIN of a
bandpass section made from a second Operational Filter
block, as in Figure 6a. The configuration is robust against
tolerances in the CIN1 value when fN approaches f0 (for f0/
fN ≤ 1.4, as a rule of thumb) which is attractive in narrow
transition-band filters, because of the relative cost of high
accuracy capacitors. Further application details appear in
Part 2 of the series of articles.
20
0
GAIN (dB)
The configuration is robust against tolerances in the CIN1
value when fN approaches f0 (for fN/f0 ≤ 1.4, as a rule of
thumb) which is attractive in narrow transition-band filters, because of the relative cost of high accuracy capacitors. Further application details appear in Part 1 of the
series of articles.
R1 C R21
ƒN = ƒ O 1 –
RQ1 CIN1 RIN1
HN =
DC GAIN = HN
( )
fN
fO2
–20
– 40
RGAIN
RFF2
HIGH FREQ
GAIN = HN
2
fO = 100kHz
fN = 50kHz
Q=1
HIGH FREQ GAIN = 0dB
– 60
10k
100k
FREQUENCY (Hz)
1M
1562 F15
The constraint required for exact cancellation of the two
paths (i.e., for infinite notch depth) becomes:
Figure 15. Notch Response with fN < f0
CIN1
RIN1
VIN
RQ1
INV
V1
R21
RIN2
V2
IO
VIRTUAL
GROUND
2nd ORDER
1/4 LTC1562
RFF2
RGAIN
–
+
VOUT
1562 F14
Figure 14. Feedforward Notch Configuration for fN < fO
1562fa
22
LTC1562
U
U
W
U
APPLICATIONS INFORMATION
R-C Universal Notches
A different way to get 180° phase shift for a notch is to use
the built-in 90° phase difference between the two Operational Filter block outputs along with a further 90° from an
external capacitor. This method achieves deep notches
independent of component matching, unlike the previous
techniques, and it is convenient for cascaded highpass as
well as lowpass and bandpass filters.
The V2 output of an Operational Filter block is a timeintegrated version of V1 (see Figure 3), and therefore lags
V1 by 90° over a wide range of frequencies. In Figure 16,
a notch response occurs when a 2nd order section drives
a virtual-ground input through two paths, one through a
capacitor and one through a resistor. Again, the virtual
ground may come from an op amp as shown, or from
another Operational Filter block’s INV input. Capacitor CN
adds a further 90° to the 90° difference between V1 and
V2, producing a wideband 180° phase difference, but
frequency-dependent amplitude ratio, between currents
IR and IC. At the frequency where IR and IC have equal
magnitude, IO becomes zero and a notch occurs. This
gives a net transfer function from VIN to VOUT in the form
of HBR(s) as above, with parameters:
ƒN =
1
2π RN CNR1C
R
R21
DC Gain = GAIN
RIN1 RN
ƒ 2O
ƒN2
=
High Frequency Gain RN CN
=
DC Gain
R21C
R1 and C are the internal precision components (in the
LTC1562, 10k and 159pF respectively) as described above
in Setting f0 and Q.
Unlike the notch methods of Figures 11 and 14, notch
depth from Figure 16 is inherent, not derived from component matching. Errors in the RN or CN values alter the notch
frequency, fN, rather than the degree of cancellation at fN.
Also, the notch frequency, fN, is independent of the section’s
center frequency f0, so fN can freely be equal to, higher
than or lower than f0 (Figures 12, 13 or 15, respectively)
without changing the configuration. The chief drawback of
Figure 16 compared to the previous methods is a very
practical one—the CN capacitor value directly scales HN
(and therefore the high frequency gain). Capacitor values
are generally not available in increments or tolerances as
fine as those of resistors, and this configuration lacks the
property of the previous two configurations that sensitivity to the capacitor value falls as fN approaches f0. Unlike
the previous notch circuits, this one is also noninverting at
DC.
R
C
HN = – GAIN N
RIN1 C
RIN1
VIN
R21
RQ1
INV
V1
RN
IR
VIRTUAL
GROUND
V2
2nd ORDER
1/4 LTC1562
IO
CN
IC
RGAIN
–
+
VOUT
1562 F16
Figure 16. The R-C Universal Notch Configuration for an Operational Filter Block
1562fa
23
LTC1562
U
TYPICAL APPLICATIONS
(Advanced)
8th Order 50kHz Lowpass Elliptic Filter
with 100dB Stopband Attenuation
CIN2 24pF
Amplitude Response
RIN2 37.4k
1
VIN
RQ1 30.1k
2
R21 31.6k
3
5
5V
0.1µF
6
8
R23 31.6k
9
RQ3 34k
10
INVB
INVC
V1B
V1C
V2B
V2C
V+
V–
LTC1562
SHDN
AGND
V2A
V2D
V1A
V1D
INVA
INVD
20
20
19
RQ2 13k
0
18 R22 57.6k
–20
16
GAIN (dB)
RIN1
48.7k
– 5V
0.1µF
15
13
12
– 60
–80
R24 32.4k
–100
11 RQ4 11.5k
–120
VOUT
RIN4 32.4k
RIN3 31.6k
– 40
10
100
FREQUENCY (kHz)
500
1562 TA12b
CIN3 18pF
CIN4 10pF
USES THREE R-C UNIVERSAL NOTCHES AT fN = 133kHz, 167kHz, 222kHz.
DETAILED DESCRIPTION IN LINEAR TECHNOLOGY DESIGN NOTE 195.
WIDEBAND OUTPUT NOISE 60µVRMS
1562 TA12a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
8th Order 100kHz Elliptic Bandpass Filter
RFF2 301k
RIN2 93.1k
1
VIN
CIN1
5.6pF
RQ1 86.6k
R21 10.7k
2
3
5
5V
0.1µF
R23 10k
RQ3 71.5k
RIN3 294k
6
8
9
10
INVB
INVC
V1B
V1C
V2B
V+
V2C
LTC1562
SHDN
V–
AGND
V2A
V2D
V1A
V1D
INVA
CIN3 18pF
INVD
Amplitude Response
20
19
18
10
RQ2 84.5k
0
–10
R22 10k
–20
16
15
– 5V
0.1µF
13
12
R24 9.53k
11
RQ4 82.5k
GAIN (dB)
RIN1
95.3k
– 30
– 40
– 50
– 60
–70
– 80
– 90
RIN4 95.3k
VOUT
RFF4 332k
25
100
FREQUENCY (kHz)
175
1562 TA13b
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
1562 F13a
1562fa
24
LTC1562
U
TYPICAL APPLICATIONS
(Advanced)
9th Order 22kHz Lowpass Elliptic Filter
RIN2 249k
CIN2 33pF
RIN1A
140k
RIN1B
69.8k
1
VIN
CIN1
390pF
RQ1 95.3k
R21 324k
2
3
5
V+
0.1µF
V–
R23 196k
RQ3 392k
6
8
9
10
CIN3
27pF
INVB
INVC
V1B
V1C
V2B
V2C
V+
SHDN
LTC1562
V–
AGND
V2A
V2D
V1A
V1D
INVA
INVD
TO
PIN 10
20
19
RQ2 182k
18
R22 226k
RIN3
536k
16
V–
0.1µF
15
13
R24 649k
12
RQ4 66.5k
11
RIN4 301k
CIN4 56pF
VOUT
1562 F14a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
Noise + THD vs Frequency
10
– 40
0
– 45
–10
– 50
– 20
– 55
NOISE + THD (dB)
GAIN (dB)
Amplitude Response
– 30
– 40
– 50
– 60
–70
VIN = 1.65VRMS = 4.6VP-P
VS = ± 5V
– 60
– 65
–70
–75
– 80
– 80
– 85
– 90
– 90
5
10
FREQUENCY (kHz)
50
1
10
20
FREQUENCY (kHz)
1562 TA14b
1562 TA14c
1562fa
25
LTC1562
U
TYPICAL APPLICATIONS
(Advanced)
Dual 5th Order Lowpass “Elliptic” Filter
RIN2
CIN2
Amplitude Response
RIN1B
1
VIN1
CIN1
RQ1
2
R21
3
5
5V
0.1µF
R21
8
RQ1
RIN1A
6
9
RIN1B
10
VIN2
INVC
INVB
V1B
V1C
V2B
V2C
V+
V–
LTC1562
SHDN
AGND
V2A
V2D
V1A
V1D
INVA
INVD
CIN1
VOUT1
20
19
RQ2
18
R22
20
fC = 100kHz
0
–20
16
– 5V
GAIN (dB)
RIN1A
0.1µF
15
13
R22
12
RQ2
– 40
– 60
– 80
–100
11
–120
VOUT2
CIN2
RIN2
10
100
FREQUENCY (kHz)
1000
1562 TA15b
1562 TA15a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
fC (Hz)
RIN1A
RIN1B
CIN1
RQ1
R21
RIN2
CIN2
RQ2
R22
100k
5.9k
7.5k
680pF
28k
7.5k
6.34k
68pF
9.31k
11.3k
75k
8.06k
15.4k
560pF
36.5k
13.3k
11.3k
68pF
12.7k
20k
50k
16.9k
35.7k
390pF
56.2k
30.1k
25.5k
68pF
18.7k
44.2k
Construction and Instrumentation Cautions
100dB rejections at hundreds of kilohertz require electrically clean, compact construction, with good grounding
and supply decoupling, and minimal parasitic capacitances in critical paths (such as Operational Filter block
INV inputs). In a circuit with 5k resistances trying for
100dB rejection at 100kHz, a stray coupling of 0.003pF
around the signal path can preclude the 100dB. (By
comparison, the stray capacitance between two adjacent
pins of an IC can be 1pF or more.) Also, high quality supply
bypass capacitors of 0.1µF near the chip provide good
decoupling from a clean, low inductance power source.
But several inches of wire (i.e., a few microhenrys of
inductance) from the power supplies, unless decoupled
by substantial capacitance (≥ 10µF) near the chip, can
cause a high-Q LC resonance in the hundreds of kHz in the
chip’s supplies or ground reference, impairing stopband
rejection and other specifications at those frequencies. In
demanding filter circuits we have often found that a
compact, carefully laid out printed circuit board with good
ground plane makes a difference of 20dB in both stopband
rejection and distortion performance. Highly selective
circuits can even exhibit these issues at frequencies well
below 100kHz. Finally, equipment to measure filter performance can itself introduce distortion or noise floors;
checking for these limits with a wire replacing the filter is
a prudent routine procedure.
1562fa
26
LTC1562
U
PACKAGE DESCRIPTION
G Package
20-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
7.07 – 7.33*
(.278 – .289)
20 19 18 17 16 15 14 13 12 11
7.65 – 7.90
(.301 – .311)
1 2 3 4 5 6 7 8 9 10
5.20 – 5.38**
(.205 – .212)
1.73 – 1.99
(.068 – .078)
0° – 8°
.65
(.0256)
BSC
.55 – .95
(.022 – .037)
.13 – .22
(.005 – .009)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
.05 – .21
(.002 – .008)
.25 – .38
(.010 – .015)
G20 SSOP 0501
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
N Package
16-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
0.770*
(19.558)
MAX
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
0.255 ± 0.015*
(6.477 ± 0.381)
0.130 ± 0.005
(3.302 ± 0.127)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
+0.035
0.325 –0.015
8.255
+0.889
–0.381
)
0.045 – 0.065
(1.143 – 1.651)
0.020
(0.508)
MIN
0.065
(1.651)
TYP
0.125
(3.175)
MIN
0.100
(2.54)
BSC
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
0.018 ± 0.003
(0.457 ± 0.076)
N16 1098
1562fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC1562
U
TYPICAL APPLICATION
Amplitude Response
20
GAIN (dB)
0
Dual 4th Order 12dB Gaussian Lowpass Filter
RIN2
RIN1
1
VIN2
RQ1
R21
2
3
5
5V
0.1µF
6
8
R23
RIN3
RQ3
VIN1
9
10
INV B
INV C
V1 B
V1 C
V2 B
V2 C
V + LTC1562
V–
SHDN
AGND
V2 A
V2 D
V1 A
V1 D
INV D
INV A
– 20
fC = 64kHz
fC = 32kHz
– 40
fC = 16kHz
20
19
RQ2
18
R22
– 60
– 80
VOUT2
16
1
10
1µF
15
100
1562 TA16b
VOUT1
13
12
R24
11
RQ4
RIN4
300
FREQUENCY (kHz)
4-Level Eye Diagram
fC = 16kHz, Data Clock = 32kHz
1562 TA16a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V –
1V/DIV
10µs/DIV
fC (Hz)
RIN1 = RIN3
R21 = R23
RQ1 = RQ3
RIN2 = RIN4
R22 = R24
RQ2 = RQ4
16k
105k
105k
34k
340k
340k
34k
32k
26.1k
26.1k
16.9k
84.5k
84.5k
16.9k
64k
8.45k
6.49k
8.45k
16.2k
21k
8.45k
1562 TA16c
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1068, LTC1068-X
Quad 2-Pole Switched Capacitor Building Block Family
Clock-Tuned
LTC1560-1
5-Pole Elliptic Lowpass, fC = 1MHz/0.5MHz
No External Components, SO8
LTC1562-2
Quad 2-Pole Active RC, 20kHz to 300kHz
Same Pinout as the LTC1562
LTC1563-2/LTC1563-3
4th Order Active RC Lowpass Filters
fCUTOFF(MAX) = 256kHz, Resistor Programmable
LTC1564
10kHz to 150kHz Digitally Controlled Filter and PGA
Continuous Time Low Noise 8th Order with PGA
LTC1565-31
650kHz Continuous Time, Linear Phase Lowpass Filter
7th Order, Differential Inputs and Outputs
LTC1566-1
2.3MHz Continuous Time Lowpass Filter
7th Order, Differential Inputs and Outputs
1562fa
28
Linear Technology Corporation
LT/TP REV A 0102 1.5K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LINEAR TECHNOLOGY CORPORATION 1998
工商网监
湘ICP备2023018690号
