LTC1625
No RSENSETM Current Mode
Synchronous Step-Down
Switching Regulator
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DESCRIPTION
FEATURES
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Highest Efficiency Current Mode Controller
No Sense Resistor Required
Stable High Current Operation
Dual N-Channel MOSFET Synchronous Drive
Wide VIN Range: 3.7V to 36V
Wide VOUT Range: 1.19V to VIN
±1% 1.19V Reference
Programmable Fixed Frequency with Injection Lock
Very Low Drop Out Operation: 99% Duty Cycle
Forced Continuous Mode Control Pin
Optional Programmable Soft Start
Pin Selectable Output Voltage
Foldback Current Limit
Output Overvoltage Protection
Logic Controlled Micropower Shutdown: IQ < 30µA
Available in 16-Lead Narrow SSOP and SO Packages
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APPLICATIONS
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Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
Battery Chargers
Distributed Power
Burst ModeTM operation at low load currents reduces
switching losses and low dropout operation extends operating time in battery-powered systems. A forced continuous mode control pin can assist secondary winding
regulation by disabling Burst Mode operation when the
main output is lightly loaded.
Fault protection is provided by foldback current limiting
and an output overvoltage comparator. An external capacitor attached to the RUN/SS pin provides soft start
capability for supply sequencing. A wide supply range
allows operation from 3.7V (3.9V for LTC1625I) to 36V at
the input and 1.19V to VIN at the output.
, LTC and LT are registered trademarks of Linear Technology Corporation.
No RSENSE and Burst Mode are trademarks of Linear Technology Corporation.
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The LTC®1625 is a synchronous step-down switching
regulator controller that drives external N-Channel power
MOSFETs using few external components. Current mode
control with MOSFET VDS sensing eliminates the need for
a sense resistor and improves efficiency. The frequency of
a nominal 150kHz internal oscillator can be synchronized
to an external clock over a 1.5:1 frequency range.
TYPICAL APPLICATION
Efficiency vs Load Current
VIN
RUN/SS
TK
+
M1
Si4410DY
TG
LTC1625
ITH
RC
10k
CC
2.2nF
SW
CB 0.22µF
BOOST
VPROG INTVCC
SGND
BG
L1
10µH
DB
CMDSH-3
CVCC 4.7µF
+
D1
MBRS140T3
CIN
10µF
30V
×2
+
M2
Si4410DY
VIN
5V TO
28V
VIN = 10V
VOUT = 5V
90
VOUT
3.3V
COUT 4.5A
100µF
10V
×3
EFFICIENCY (%)
CSS
0.1µF
SYNC
100
VOUT = 3.3V
80
70
VOSENSE PGND
1625 F01
Figure 1. High Efficiency Step-Down Converter
60
0.01
0.1
1
LOAD CURRENT (A)
10
1625 TA01
1
LTC1625
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PACKAGE/ORDER I FOR ATIO
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(Note 1)
Input Supply Voltage (VIN, TK) ................. 36V to – 0.3V
Boosted Supply Voltage (BOOST) ............. 42V to – 0.3V
Boosted Driver Voltage (BOOST – SW) ...... 7V to – 0.3V
Switch Voltage (SW).....................................36V to – 5V
EXTVCC Voltage ...........................................7V to – 0.3V
ITH Voltage ................................................2.7V to – 0.3V
FCB, RUN/SS, SYNC Voltages .....................7V to – 0.3V
VOSENSE, VPROG Voltages ........(INTVCC + 0.3V) to – 0.3V
Peak Driver Output Current < 10µs (TG, BG) ............ 2A
INTVCC Output Current ........................................ 50mA
Operating Ambient Temperature Range
LTC1625C............................................... 0°C to 70°C
LTC1625I (Note 5) .............................. – 40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
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ABSOLUTE MAXIMUM RATINGS
ORDER PART
NUMBER
TOP VIEW
EXTVCC 1
16 VIN
SYNC 2
15 TK
RUN/SS 3
14 SW
FCB 4
13 TG
ITH 5
12 BOOST
SGND 6
11 INTVCC
LTC1625CGN
LTC1625CS
LTC1625IGN
LTC1625IS
10 BG
VOSENSE 7
9
VPROG 8
PGND
GN PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/W (GN)
TJMAX = 125°C, θJA = 110°C/W (S)
Consult factory for Military grade parts.
TA = 25°C, VIN = 15V unless otherwise noted.
CONDITIONS
MIN
TYP
MAX
UNITS
10
50
nA
1.190
3.300
5.000
1.202
3.380
5.100
V
V
V
0.001
0.01
%/V
– 0.020
0.035
– 0.2
0.2
%
%
1.19
1.22
V
–1
–2
µA
1.28
1.32
V
– 3.5
3.5
–7
7
µA
µA
500
15
30
µA
µA
Main Control Loop
IINVOSENSE
Feedback Current
VPROG Pin Open, ITH = 1.19V (Note 3)
VOUT
Regulated Output Voltage
1.19V (Adjustable) Selected
3.3V Selected
5V Selected
ITH = 1.19V (Note 3)
VPROG Pin Open
VPROG = 0V
VPROG = INTVCC
VLINEREG
Reference Voltage Line Regulation
VIN = 3.6V to 20V, ITH = 1.19V (Note 3),
VPROG Pin Open
VLOADREG
Output Voltage Load Regulation
ITH = 2V (Note 3)
ITH = 0.5V (Note 3)
●
●
VFCB
Forced Continuous Threshold
VFCB Ramping Negative
●
IFCB
Forced Continuous Current
VFCB = 1.19V
VOVL
Output Overvoltage Lockout
VPROG Pin Open
IPROG
VPROG Input Current
3.3V VOUT
5V VOUT
VPROG = 0V
VPROG = 5V
IQ
Input DC Supply Current
Normal Mode
Shutdown
VRUN/SS
RUN/SS Pin Threshold
IRUN/SS
Soft Start Current Source
∆VSENSE(MAX) Maximum Current Sense Threshold
TG tR
TG tF
2
TG Transition Time
Rise Time
Fall Time
●
●
●
1.178
3.220
4.900
1.16
1.24
EXTVCC = 5V (Note 4)
VRUN/SS = 0V, 3.7V < VIN < 15V
0.8
1.4
2
V
VRUN/SS = 0V
1.2
2.5
4
µA
VOSENSE = 1V, VPROG Pin Open
120
150
170
mV
50
50
150
150
ns
ns
●
CLOAD = 3300pF
CLOAD = 3300pF
LTC1625
ELECTRICAL CHARACTERISTICS
TA = 25°C, VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
BG tR
BG tF
BG Transition Time
Rise Time
Fall Time
CLOAD = 3300pF
CLOAD = 3300pF
MIN
TYP
MAX
UNITS
50
50
150
150
ns
ns
5.2
5.4
V
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
VLDOINT
INTVCC Load Regulation
ICC = 20mA, VEXTVCC = 4V
–1
–2
%
VLDOEXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V
180
300
mV
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, VEXTVCC Ramping Positive
●
●
5.0
4.5
4.7
135
150
V
Oscillator
fOSC
Oscillator Freqency
fH/fOSC
Maximum Synchronized Frequency Ratio
VSYNC
SYNC Pin Threshold (Figure 4)
RSYNC
SYNC Pin Input Resistance
165
kHz
1.5
Ramping Positive
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC1625CGN/LTC1625IGN: TJ = TA + (PD • 130°C/W)
LTC1625CS/LTC1625IS: TJ = TA + (PD • 110°C/W)
0.9
50
1.2
V
kΩ
Note 3: The LTC1625 is tested in a feedback loop that adjusts VOSENSE to
achieve a specified error amplifier output voltage (ITH).
Note 4: Typical in application circuit with EXTVCC tied to VOUT = 5V,
IOUT = 0A and FCB = INTVCC. Dynamic supply current is higher due
to the gate charge being delivered at the switching frequency. See
Applications Information.
Note 5: Minimum input supply voltage is 3.9V at – 40°C for industrial
grade parts.
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LTC1625
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TYPICAL PERFOR A CE CHARACTERISTICS
100
100
ILOAD = 2A
95
80
70
60
VIN = 10V
VOUT = 5V
EXTVCC = VOUT
95
90
ILOAD = 200mA
85
80
75
0.1
0.01
1
LOAD CURRENT (A)
5
10
15
20
INPUT VOLTAGE (V)
25
0
25
VIN – VOUT Dropout Voltage
vs Load Current
3.0
400
FIGURE 1 CIRCUIT
VIN = 20V
2.5 VOUT = 5V
FIGURE 1 CIRCUIT
30
1625 G02
ITH Pin Voltage vs Load Current
– 0.05
FIGURE 1 CIRCUIT
VOUT = 5V – 5% DROP
300
VITH (V)
– 0.10
VIN – VOUT (mV)
2.0
∆VOUT (%)
10
15
20
INPUT VOLTAGE (V)
5
1625 G02
Load Regulation
CONTINUOUS
MODE
1.5
– 0.15
1.0
– 0.20
– 0.25
ILOAD = 200mA
80
30
1625 G01
0
85
70
0
10
90
75
70
50
0.001
FIGURE 1 CIRCUIT
ILOAD = 2A
CONTINUOUS
MODE
EFFICIENCY (%)
EFFICIENCY (%)
90
100
FIGURE 1 CIRCUIT
EFFICIENCY (%)
BURST
MODE
OPERATION
Efficiency vs Input Voltage,
VOUT = 3.3V
Efficiency vs Input Voltage,
VOUT = 5V
Efficiency vs Load Current
Burst Mode
OPERATION
0.5
100
0
1
0
3
2
LOAD CURRENT (A)
4
0
5
1
4
3
5
2
LOAD CURRENT (A)
200
6
0
7
1
3
2
LOAD CURRENT (A)
4
1625 G05
1625 G04
Input and Shutdown Current
vs Input Voltage
5
1625 G06
EXTVCC Switch Drop
vs INTVCC Load Current
INTVCC Load Regulation
1000
0
50
0
500
40
– 0.5
400
20
400
200
10
EXTVCC = 5V
0
0
0
5
20
15
10
25
INPUT VOLTAGE (V)
30
35
1625 G07
4
EXTVCC – INTVCC (mV)
INPUT CURRENT (µA)
SHUTDOWN
SHUTDOWN CURRENT (µA)
30
600
∆INTVCC (%)
EXTVCC OPEN
800
–1.0
–1.5
– 2.0
– 2.5
300
200
100
0
10
30
40
20
INTVCC LOAD CURRENT (mA)
50
1625 G08
0
0
10
30
40
20
INTVCC LOAD CURRENT (mA)
50
1625 G09
LTC1625
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TYPICAL PERFOR A CE CHARACTERISTICS
150
100
50
0
0.2
0.5
0.4
DUTY CYCLE
0.8
300
160
250
150
SYNC = 0V
150
100
50
85
10
35
60
TEMPERATURE (°C)
110
135
0
– 40 –15
60
35
85
10
TEMPERATURE (°C)
1625 G11
1625 G10
110
135
1625 G12
RUN/SS Pin Current
vs Temperature
FCB Pin Current vs Temperature
Soft Start:
Load Current vs Time
0
0
– 0.25
INDUCTOR
CURRENT
2A/DIV
–1
RUN/SS CURRENT (µA)
FCB CURRENT (µA)
200
145
140
– 40 –15
1.0
SYNC = 1.5V
155
FREQUENCY (kHz)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
200
0
Oscillator Frequency
vs Temperature
Maximum Current Sense Voltage
vs Temperature
Maximum Current Sense Voltage
vs Duty Cycle
– 0.50
– 0.75
–1.00
–2
RUN/SS
2V/DIV
–3
20ms/DIV
–1.25
–1.50
– 40 –15
60
35
85
10
TEMPERATURE (°C)
110
135
1625 F06
VIN = 20V
VOUT = 5V
RLOAD = 1Ω
FIGURE 1 CIRCUIT
–4
–5
–40
–15
60
10
85
35
TEMPERATURE (°C)
110
135
1625 G14
1625 G13
Transient Response
(Burst Mode Operation)
Transient Response
Burst Mode Operation
VOUT
50mV/DIV
VOUT
50mV/DIV
VOUT
50mV/DIV
ITH
100mV/DIV
VIN = 20V
VOUT = 5V
ILOAD = 1A TO 4A
FIGURE 1 CIRCUIT
200µs/DIV
1625 F07
VIN = 20V
VOUT = 5V
ILOAD = 50mA TO 1A
FIGURE 1 CIRCUIT
500µs/DIV
1625 F08
VIN = 20V
VOUT = 5V
ILOAD = 50mA
FIGURE 1 CIRCUIT
50µs/DIV
1625 F09
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LTC1625
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PIN FUNCTIONS
EXTVCC (Pin 1): INTVCC Switch Input. When the EXTVCC
voltage is above 4.7V, the switch closes and supplies
INTVCC power from EXTVCC. Do not exceed 7V at this pin.
Leaving VPROG open allows the output voltage to be set by
an external resistive divider between the output and
VOSENSE.
SYNC (Pin 2): Synchronization Input for Internal Oscillator. The oscillator will nominally run at 150kHz when open,
225kHz when tied above 1.2V, and will lock over a 1.5:1
clock frequency range.
PGND (Pin 9): Driver Power Ground. Connects to the
source of the bottom N-channel MOSFET, the (–) terminal
of CVCC and the (–) terminal of CIN.
RUN/SS (Pin 3): Run Control and Soft Start Input. A
capacitor to ground at this pin sets the ramp time to full
current output (approximately 1s/µF). Forcing this pin
below 1.4V shuts down the device.
FCB (Pin 4): Forced Continuous Input. Tie this pin to
ground to force synchronous operation at low load, to a
resistive divider from the secondary output when using
a secondary winding, or to INTVCC to enable Burst Mode
operation at low load.
ITH (Pin 5): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.4V.
SGND (Pin 6): Signal Ground. Connect to the (–) terminal
of COUT.
BG (Pin 10): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
INTVCC (Pin 11): Internal 5.2V Regulator Output. The
driver and control circuits are powered from this voltage.
Decouple this pin to power ground with a minimum of
4.7µF tantalum capacitance.
BOOST (Pin 12): Topside Floating Driver Supply. The (+)
terminal of the bootstrap capacitor connects here. This pin
swings from a diode drop below INTVCC to VIN + INTVCC.
TG (Pin 13): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage swing equal to INTVCC minus a
diode drop, superimposed on the switch node voltage.
SW (Pin 14): Switch Node. The (–) terminal of the bootstrap capacitor connects here. This pin swings from a
diode drop below ground up to VIN.
VOSENSE (Pin 7): Output Voltage Sense. Feedback input
from the remotely sensed output voltage or from an
external resistive divider across the output.
TK (Pin 15): Top MOSFET Kelvin Sense. MOSFET VDS
sensing requires this pin to be routed to the drain of the top
MOSFET separately from VIN.
VPROG (Pin 8): Output Voltage Programming. When
VOSENSE is connected to the output, VPROG < 0.8V selects
a 3.3V output and VPROG > 3.5V selects a 5V output.
VIN (Pin 16): Main Supply Input. Decouple this pin to
ground with an RC filter (4.7Ω, 0.1µF) for applications
above 3A.
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LTC1625
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FUNCTIONAL DIAGRA
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TK
+
2 SYNC
VIN
15
TA
× 11
+
–
CIN
–
BA
× 11
0.95V
+
ITH
+–
5
OSC
0.6V
RC
–
–
+
gm = 1m
BOOST
12
B
SLEEP
CB
TG
SWITCH
LOGIC/
DROPOUT
COUNTER
Ω
14
INTVCC
11
OVERVOLTAGE
+–
0.6V
DB
+
CVCC
BG
FCNT
10
3µA
M2
PGND
+
3
M1
13
SW
SHUTDOWN
EA
–
9
6V
1.28V
1.19V
REF
–
+
6
–
TOP
+
–
CL
1.19V
SGND
R
0.5V
VFB
CSS
I1
–
+
RUN/SS
S
Q
ITHB
0.6V
I2
+
CC1
+
REV
VIN
16
OV
5.2V
LDO REG
1.19V
+
+
4.7V
F
–
–
1µA
L1
8 VPROG
7 VOSENSE
4 FCB
1 EXTVCC
+
COUT
1625 BD
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LTC1625
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OPERATIO
Main Control Loop
The LTC1625 is a constant frequency, current mode
controller for DC/DC step-down converters. In normal
operation, the top MOSFET is turned on when the RS latch
is set by the on-chip oscillator and is turned off when the
current comparator I1 resets the latch. While the top
MOSFET is turned off, the bottom MOSFET is turned on
until either the inductor current reverses, as determined
by the current reversal comparator I2, or the next cycle
begins. Inductor current is measured by sensing the VDS
potential across the conducting MOSFET. The output of
the appropriate sense amplifier (TA or BA) is selected by
the switch logic and applied to the current comparator.
The voltage on the ITH pin sets the comparator threshold
corresponding to peak inductor current. The error amplifier EA adjusts this voltage by comparing the feedback
signal VFB from the output voltage with the internal 1.19V
reference. The VPROG pin selects whether the feedback
voltage is taken directly from the VOSENSE pin or is derived
from an on-chip resistive divider. When the load current
increases, it causes a drop in the feedback voltage relative
to the reference. The ITH voltage then rises until the
average inductor current again matches the load current.
The internal oscillator can be synchronized to an external
clock applied to the SYNC pin and can lock to a frequency
between 100% and 150% of its nominal 150kHz rate.
When the SYNC pin is left open, it is pulled low internally
and the oscillator runs at its normal rate. If this pin is taken
above 1.2V, the oscillator will run at its maximum 225kHz
rate.
Pulling the RUN/SS pin low forces the controller into its
shutdown state and turns off both MOSFETs. Releasing
the RUN/SS pin allows an internal 3µA current source to
charge up an external soft start capacitor CSS. When this
voltage reaches 1.4V, the controller begins switching, but
with the ITH voltage clamped at approximately 0.8V. As
CSS continues to charge, the clamp is raised until full range
operation is restored.
The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is normally recharged
from INTVCC through a diode DB when the top MOSFET is
turned off. As VIN decreases towards VOUT, the converter
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will attempt to turn on the top MOSFET continuously
(‘’dropout’’). A dropout counter detects this condition and
forces the top MOSFET to turn off for about 500ns every
tenth cycle to recharge the bootstrap capacitor.
An overvoltage comparator OV guards against transient
overshoots and other conditions that may overvoltage the
output. In this case, the top MOSFET is turned off and the
bottom MOSFET is turned on until the overvoltage condition is cleared.
Foldback current limiting for an output shorted to ground
is provided by a transconductance amplifer CL. As VFB
drops below 0.6V, the buffered ITH input to the current
comparator is gradually pulled down to a 0.95V clamp.
This reduces peak inductor current to about one fifth of its
maximum value.
Low Current Operation
The LTC1625 is capable of Burst Mode operation at low
load currents. If the error amplifier drives the ITH voltage
below 0.95V, the buffered ITH input to the current comparator will remain clamped at 0.95V. The inductor current
peak is then held at approximately 30mV/RDS(ON)(TOP). If
ITH then drops below 0.5V, the Burst Mode comparator B
will turn off both MOSFETs. The load current will be
supplied solely by the output capacitor until ITH rises
above the 50mV hysteresis of the comparator and switching is resumed. Burst Mode operation is disabled by
comparator F when the FCB pin is brought below 1.19V.
This forces continuous operation and can assist secondary winding regulation.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the internal circuitry of the LTC1625 is derived from the
INTVCC pin. When the EXTVCC pin is left open, an internal
5.2V low dropout regulator supplies the INTVCC power
from VIN. If EXTVCC is raised above 4.7V, the internal
regulator is turned off and an internal switch connects
EXTVCC to INTVCC. This allows a high efficiency source,
such as the primary or a secondary output of the converter
itself, to provide the INTVCC power.
LTC1625
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APPLICATIONS INFORMATION
Power MOSFET Selection
The LTC1625 requires two external N-channel power
MOSFETs, one for the top (main) switch and one for the
bottom (synchronous) switch. Important parameters for
the power MOSFETs are the breakdown voltage V(BR)DSS,
threshold voltage VGS(TH), on-resistance RDS(ON), reverse
transfer capacitance CRSS and maximum current ID(MAX).
The ρT is a normalized term accounting for the significant
variation in RDS(ON) with temperature, typically about
0.4%/°C as shown in Figure 2. Junction to case temperature TJC is around 10°C in most applications. For a
maximum ambient temperature of 70°C, using ρ80°C ≅ 1.3
in the above equation is a reasonable choice. This equation
is plotted in Figure 3 to illustrate the dependence of
maximum output current on RDS(ON). Some popular
MOSFETs from Siliconix are shown as data points.
2.0
ρT NORMALIZED ON RESISTANCE
The basic LTC1625 application circuit is shown in Figure 1.
External component selection is primarily determined by
the maximum load current and begins with the selection of
the sense resistance and power MOSFETs. Because the
LTC1625 uses MOSFET VDS sensing, the sense resistance
is the RDS(ON) of the MOSFETs. The operating frequency
and the inductor are chosen based largely on the desired
amount of ripple current. Finally, CIN is selected for its
ability to handle the large RMS current into the converter
and COUT is chosen with low enough ESR to meet the
output voltage ripple specification.
RDS(ON)(MAX) ≅
120mV
(IO(MAX) )(ρT )
1.0
0.5
0
– 50
The gate drive voltage is set by the 5.2V INTVCC supply.
Consequently, logic level threshold MOSFETs must be
used in LTC1625 applications. If low input voltage operation is expected (VIN < 5V), then sub-logic level threshold
MOSFETs should be used. Pay close attention to the
V(BR)DSS specification for the MOSFETs as well; many of
the logic level MOSFETs are limited to 30V or less.
50
100
0
JUNCTION TEMPERATURE (°C)
150
1625 F02
Figure 2. RDS(ON) vs Temperature
10
MAXIMUM OUTPUT CURRENT (A)
The MOSFET on-resistance is chosen based on the
required load current. The maximum average output current IO(MAX) is equal to the peak inductor current less half
the peak-to-peak ripple current ∆IL. The peak inductor
current is inherently limited in a current mode controller
by the current threshold ITH range. The corresponding
maximum VDS sense voltage is about 150mV under normal conditions. The LTC1625 will not allow peak inductor
current to exceed 150mV/RDS(ON)(TOP). The following
equation is a good guide for determining the required
RDS(ON)(MAX) at 25°C (manufacturer’s specification), allowing some margin for ripple current, current limit and
variations in the LTC1625 and external component values:
1.5
8
Si4420
6
Si4410
4
Si4412
2
0
Si9936
0
0.02
0.06
0.04
RDS(ON) (Ω)
0.08
0.10
1625 F03
Figure 3. Maximum Output Current vs RDS(ON) at VGS = 4.5V
The power dissipated by the top and bottom MOSFETs
strongly depends upon their respective duty cycles and
the load current. When the LTC1625 is operating in continuous mode, the duty cycles for the MOSFETs are:
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LTC1625
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APPLICATIONS INFORMATION
7V
V
Top Duty Cycle = OUT
VIN
V –V
Bottom Duty Cycle = IN OUT
VIN
The MOSFET power dissipations at maximum output
current are:
1.2V
1µs
4µs
±
1625 F04
0
V
2
PTOP = OUT (IO(MAX) )(ρT(TOP) )(RDS(ON) )
VIN
2
+ (k)(VIN )(IO(MAX) )(CRSS)( f)
V –V
2
PBOT = IN OUT (IO(MAX) )(ρT(BOT ) )(RDS(ON) )
VIN
Both MOSFETs have I2R losses and the PTOP equation
includes an additional term for transition losses, which are
largest at high input voltages. The constant k = 1.7 can be
used to estimate the amount of transition loss. The bottom
MOSFET losses are greatest at high input voltage or during
a short circuit when the duty cycle is nearly 100%.
Operating Frequency and Synchronization
The choice of operating frequency and inductor value is a
trade-off between efficiency and component size. Low
frequency operation improves efficiency by reducing
MOSFET switching losses, both gate charge loss and
transition loss. However, lower frequency operation
requires more inductance for a given amount of ripple
current.
The internal oscillator runs at a nominal 150kHz frequency
when the SYNC pin is left open or connected to ground.
Pulling the SYNC pin above 1.2V will increase the frequency by 50%. The oscillator will injection lock to a clock
signal applied to the SYNC pin with a frequency between
165kHz and 200kHz. The clock high level must exceed
1.2V for at least 1µs and no longer than 4µs as shown in
Figure 4. The top MOSFET turn-on will synchronize with
the rising edge of the clock.
10
Figure 4. SYNC Clock Waveform
Inductor Value Selection
Given the desired input and output voltages, the inductor
value and operating frequency directly determine the
ripple current:
V
V
∆IL = OUT 1 – OUT
VIN
( f)(L)
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Thus, highest efficiency operation is obtained at
low frequency with small ripple current. To achieve this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of IO(MAX). Note that the largest ripple
current occurs at the highest VIN. To guarantee that ripple
current does not exceed a specified maximum, the inductor should be chosen according to:
VOUT
VOUT
–
1
L≥
VIN(MAX)
( f)( ∆IL(MAX) )
Burst Mode Operation Considerations
The choice of RDS(ON) and inductor value also determines
the load current at which the LTC1625 enters Burst Mode
operation. When bursting, the controller clamps the peak
inductor current to approximately:
IBURST(PEAK) =
30mV
RDS(ON)
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The corresponding average current depends on the amount
of ripple current. Lower inductor values (higher ∆IL) will
reduce the load current at which Burst Mode operation
begins.
The output voltage ripple can increase during Burst Mode
operation if ∆IL is substantially less than IBURST. This will
primarily occur when the duty cycle is very close to unity
(VIN is close to VOUT) or if very large value inductors are
chosen. This is generally only a concern in applications
with VOUT ≥ 5V. At high duty cycles, a skipped cycle
causes the inductor current to quickly descend to zero.
However, it takes multiple cycles to ramp the current back
up to IBURST(PEAK). During this interval, the output capacitor must supply the load current and enough charge may
be lost to cause significant droop in the output voltage. It
is a good idea to keep ∆IL comparable to IBURST(PEAK).
Otherwise, one might need to increase the output capacitance in order to reduce the voltage ripple or else disable
Burst Mode operation by forcing continuous operation
with the FCB pin.
maximum values for RDS(ON), but not a minimum. A
reasonable, but perhaps overly conservative, assumption
is that the minimum RDS(ON) lies the same amount below
the typical value as the maximum RDS(ON) lies above it.
Consult the MOSFET manufacturer for further guidelines.
The LTC1625 includes current foldback to help further
limit load current when the output is shorted to ground. If
the output falls by more than half, then the maximum
sense voltage is progressively lowered from 150mV to
30mV. Under short-circuit conditions with very low duty
cycle, the LTC1625 will begin skipping cycles in order to
limit the short-circuit current. In this situation the bottom
MOSFET RDS(ON) will control the inductor current trough
rather than the top MOSFET controlling the inductor
current peak. The short-circuit ripple current is determined by the minimum on-time tON(MIN) of the LTC1625
(approximately 0.5µs), the input voltage, and inductor
value:
∆IL(SC) = tON(MIN) VIN /L.
The resulting short-circuit current is:
Fault Conditions: Current Limit and Output Shorts
The LTC1625 current comparator can accommodate a
maximum sense voltage of 150mV. This voltage and the
sense resistance determine the maximum allowed peak
inductor current. The corresponding output current limit
is:
ILIMIT =
(R
150mV
)( ρ )
DS(ON)
T
1
– ∆IL
2
The current limit value should be checked to ensure that
ILIMIT(MIN) > IO(MAX). The minimum value of current limit
generally occurs with the largest VIN at the highest ambient temperature, conditions which cause the highest power
dissipation in the top MOSFET. Note that it is important to
check for self-consistency between the assumed junction
temperature of the top MOSFET and the resulting value of
ILIMIT which heats the junction.
Caution should be used when setting the current limit
based upon RDS(ON) of the MOSFETs. The maximum
current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and
ISC =
1
+ ∆ IL(SC)
RDS(ON)(BOT ) ρT 2
(
30mV
)( )
Normally, the top and bottom MOSFETs will be of the same
type. A bottom MOSFET with lower RDS(ON) than the top
may be chosen if the resulting increase in short-circuit
current is tolerable. However, the bottom MOSFET should
never be chosen to have a higher nominal RDS(ON) than the
top MOSFET.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
the inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Kool Mµ is a registered trademark of Magnetics, Inc.
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Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses rapidly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
which do not increase the height significantly.
Schottky Diode Selection
The Schottky diode D1 shown in Figure 1 conducts during
the dead time between the conduction of the power
MOSFETs. This prevents the body diode of the bottom
MOSFET from turning on and storing charge during the
dead time, which could cost as much as 1% in efficiency.
A 1A Schottky diode is generally a good size for 3A to 5A
regulators. The diode may be omitted if the efficiency loss
can be tolerated.
CIN and COUT Selection
In continuous mode, the drain current of the top MOSFET
is approximately a square wave of duty cycle VOUT / VIN. To
prevent large input voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS current is given by:
V
V
IRMS ≅ IO(MAX) OUT IN − 1
VIN VOUT
1/ 2
This formula has a maximum at VIN = 2VOUT, where IRMS
= IO(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that ripple current ratings
from capacitor manufacturers are often based on only
12
2000 hours of life. This makes it advisable to further derate
the capacitor or to choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
placed in parallel to meet size or height requirements in the
design.
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple. The output ripple
∆VOUT is approximately bounded by:
1
∆VOUT ≤ ∆IL ESR +
(8 )(f )(COUT )
Since ∆IL increases with input voltage, the output ripple is
highest at maximum input voltage. Typically, once the ESR
requirement is satisfied the capacitance is adequate for
filtering and has the required RMS current rating.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be placed in parallel to meet the ESR requirement.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
tested for use in switching power supplies. An excellent
choice is the AVX TPS series of surface mount tantalum,
available in case heights ranging from 2mm to 4mm. Other
capacitor types include Sanyo OS-CON, Nichicon PL series, and Sprague 593D and 595D series. Consult the
manufacturer for other specific recommendations.
INTVCC Regulator
An internal P-channel low dropout regulator produces the
5.2V supply which powers the drivers and internal circuitry within the LTC1625. The INTVCC pin can supply up
to 50mA and must be bypassed to ground with a minimum
of 4.7µF tantalum or low ESR electrolytic capacitance.
Good bypassing is necessary to supply the high transient
currents required by the MOSFET gate drivers.
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High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the LTC1625
to exceed its maximum junction temperature rating. Most
of the supply current drives the MOSFET gates unless an
external EXTVCC source is used. The junction temperature
can be estimated from the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC1625CGN
is limited to less than 14mA from a 30V supply:
TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in continuous mode at high VIN.
3. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
greater than 4.7V. This can be done with either an
inductive boost winding as shown in Figure 5a or a
capacitive charge pump as shown in Figure 5b.
4. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range (EXTVCC < VIN),
it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements.
The LTC1625 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
Whenever the EXTVCC pin is above 4.7V the internal 5.2V
regulator shuts off, the switch closes and INTVCC power is
supplied via EXTVCC until EXTVCC drops below 4.5V. This
allows the MOSFET gate drive and control power to be
derived from the output or other external source during
normal operation. When the output is out of regulation
(start-up, short circuit) power is supplied from the internal
regulator. Do not apply greater than 7V to the EXTVCC pin
and ensure that EXTVCC ≤ VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current supplying
the driver and control currents will be scaled by a factor of
Duty Cycle/Efficiency. For 5V regulators this simply means
connecting the EXTVCC pin directly to VOUT. However, for
3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5.2V regulator resulting
in an efficiency penalty of up to 10% at high input
voltages.
VIN
+
EXTVCC Connection
CIN
VIN
TK
OPTIONAL
EXTVCC
CONNECTION
5V < VSEC < 7V
VSEC
1N4148
•
TG
LTC1625
EXTVCC
SW
R4
+
T1
1:N
FCB
R3
CSEC
1µF
VOUT
• +
COUT
BG
SGND
PGND
1625 F05a
Figure 5a: Secondary Output Loop and EXTVCC Connection
VPUMP ≈ 2(VOUT – VD)
+
+
VIN
1µF
VIN
CIN
BAT85
BAT85
0.22µF
TK
TG
LTC1625
BAT85
VN2222LL
SW
EXTVCC
L1
VOUT
+
COUT
BG
PGND
1625 F05b
Figure 5b: Capacitive Charge Pump for EXTVCC
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
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Note that RDS(ON) also varies with the gate drive level. If
gate drives other than the 5.2V INTVCC are used, this must
be accounted for when selecting the MOSFET RDS(ON).
Particular care should be taken with applications where
EXTVCC is connected to the output. When the output
voltage is between 4.7V and 5.2V, INTVCC will be connected to the output and the gate drive is reduced. The
resulting increase in RDS(ON) will also lower the current
limit. Even applications with VOUT > 5.2V will traverse this
region during start-up and must take into account the
reduced current limit.
VPROG pin is left open and the VOSENSE pin is connected to
feedback resistors as shown in Figure 6b. The output
voltage is set by the divider as:
R2
VOUT = 1.19V 1 +
R1
LTC1625
VOUT = 5V: INTVCC
GND
VOUT = 3.3V:
VOUT
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor (CB in the functional
diagram) connected to the BOOST pin supplies the gate
drive voltage for the topside MOSFET. This capacitor is
charged through diode DB from INTVCC when the SW node
is low. Note that the voltage across CB is about a diode
drop below INTVCC. When the top MOSFET turns on, the
switch node voltage rises to VIN and the BOOST pin rises
to approximately VIN + INTVCC. During dropout operation,
CB supplies the top driver for as long as ten cycles between
refreshes. Thus, the boost capacitance needs to store
about 100 times the gate charge required by the top
MOSFET. In many applications 0.22µF is adequate.
When adjusting the gate drive level , the final arbiter is the
total input current for the regulator. If you make a change
and the input current decreases, then you improved the
efficiency. If there is no change in input current, then there
is no change in efficiency.
Output Voltage Programming
The LTC1625 has a pin selectable output voltage determined by the VPROG pin as follows:
VPROG = 0V
VPROG = INTVCC
VPROG = Open
VOUT = 3.3V
VOUT = 5V
VOUT = Adjustable
Remote sensing of the output voltage is provided by the
VOSENSE pin. For fixed 3.3V and 5V output applications an
internal resistive divider is used and the VOSENSE pin is
connected directly to the output voltage as shown in
Figure 6a. When using an external resistive divider, the
14
VPROG
VOSENSE
+
COUT
SGND
1625 F06a
Figure 6a. Fixed 3.3V or 5V VOUT
LTC1625
OPEN
VPROG
R2
+
COUT
VOSENSE
R1
SGND
1625 F06b
Figure 6b. Adjustable VOUT
Run/Soft Start Function
The RUN/SS pin is a dual purpose pin that provides a soft
start function and a means to shut down the LTC1625. Soft
start reduces surge currents from VIN by gradually increasing the controller’s current limit ITH(MAX). This pin
can also be used for power supply sequencing.
Pulling the RUN/SS pin below 1.4V puts the LTC1625 into
a low quiescent current shutdown (IQ < 30µA). This pin can
be driven directly from logic as shown in Figure 7. Releasing the RUN/SS pin allows an internal 3µA current source
to charge up the external capacitor CSS. If RUN/SS has
been pulled all the way to ground there is a delay before
starting of approximately:
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then VSEC will droop. An external resistor divider from
VSEC to the FCB pin sets a minimum voltage VSEC(MIN):
1.4V
tDELAY =
CSS = 0.5s / µF CSS
3µA
(
)
When the voltage on RUN/SS reaches 1.4V the LTC1625
begins operating with a clamp on ITH at 0.8V. As the
voltage on RUN/SS increases to approximately 3.1V, the
clamp on ITH is raised until its full 2.4V range is restored.
This takes an additional 0.5s/µF. During this time the load
current will be folded back to approximately 30mV/RDS(ON)
until the output reaches half of its final value.
Diode D1 in Figure 7 reduces the start delay while allowing
CSS to charge up slowly for the soft start function. This
diode and CSS can be deleted if soft start is not needed. The
RUN/SS pin has an internal 6V zener clamp (See Functional Diagram).
3.3V
OR 5V
RUN/SS
RUN/SS
R4
VSEC(MIN) ≅ 1.19 V 1 +
R3
If VSEC drops below this level, the FCB voltage forces
continuous operation until VSEC is again above its
minimum.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC1625 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays and
the amount of gate charge required to turn on the top
MOSFET. Low duty cycle applications may approach this
minimum on-time limit and care should be taken to ensure
that:
D1
CSS
CSS
1625 F07
Figure 7. RUN/SS Pin Interfacing
FCB Pin Operation
When the FCB pin drops below its 1.19V threshold,
continuous synchronous operation is forced. In this case,
the top and bottom MOSFETs continue to be driven
regardless of the load on the main output. Burst Mode
operation is disabled and current reversal is allowed in the
inductor.
In addition to providing a logic input to force continuous
operation, the FCB pin provides a means to regulate a
flyback winding output. It can force continuous synchronous operation when needed by the flyback winding,
regardless of the primary output load.
The secondary output voltage VSEC is normally set as
shown in Figure 5a by the turns ratio N of the transformer:
VSEC ≅ (N + 1)VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
tON(MIN) <
VOUT
(VIN)( f)
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC1625 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple current and ripple voltage will increase.
The minimum on-time for the LTC1625 is generally about
0.5µs. However, as the peak sense voltage (IL(PEAK) •
RDS(ON)) decreases, the minimum on-time gradually
increases up to about 0.7µs. This is of particular concern
in forced continuous applications with low ripple current
at light loads. If the duty cycle drops below the minimum
on-time limit in this situation, a significant amount of
cycle skipping can occur with correspondingly larger
current and voltage ripple.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power (×100%). Percent efficiency can be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
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where L1, L2, etc. are the individual losses as a percentage
of input power. It is often useful to analyze individual
losses to determine what is limiting the efficiency and
which change would produce the most improvement.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1625 circuits:
4. LTC1625 VIN supply current. The VIN current is the DC
supply current to the controller excluding MOSFET gate
drive current. Total supply current is typically about
850µA. If EXTVCC is connected to 5V, the LTC1625 will
draw only 330µA from VIN and the remaining 520µA will
come from EXTVCC. VIN current results in a small
(< 1%) loss which increases with VIN.
1. INTVCC current. This is the sum of the MOSFET driver
and control currents. The driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched on and then off,
a packet of gate charge Qg moves from INTVCC to
ground. The resulting current out of INTVCC is typically
much larger than the control circuit current. In continuous mode, IGATECHG = f(Qg(TOP) + Qg(BOT)).
Other losses including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead time
and inductor core losses, generally account for less
than 2% total additional loss.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of Duty Cycle/
Efficiency. For example, in a 20V to 5V application at
400mA load, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the loss
from 10% (if the driver was powered directly from VIN)
to about 3%.
2. DC I2R Losses. Since there is no separate sense resistor, DC I2R losses arise only from the resistances of the
MOSFETs and inductor. In continuous mode the average output current flows through L, but is “chopped”
between the top MOSFET and the bottom MOSFET. If
the two MOSFETs have approximately the same RDS(ON),
then the resistance of one MOSFET can simply be
summed with the resistance of L to obtain the DC I2R
loss. For example, if each RDS(ON) = 0.05Ω and RL =
0.15Ω, then the total resistance is 0.2Ω. This results in
losses ranging from 2% to 8% as the output current
increases from 0.5A to 2A for a 5V output. I2R losses
cause the efficiency to drop at high output currents.
3. Transition losses apply only to the topside MOSFET,
and only when operating at high input voltages (typically 20V or greater). Transition losses can be estimated from:
Transition Loss = (1.7)(VIN2)(IO(MAX))(CRSS)(f)
16
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT immediately shifts
by an amount equal to (∆ILOAD)(ESR), where ESR is the
effective series resistance of COUT, and COUT begins to
charge or discharge. The regulator loop acts on the
resulting feedback error signal to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a
stability problem. The ITH pin external components shown
in Figure 1 will provide adequate compensation for most
applications.
A second, more severe transient is caused by connecting
loads with large (> 1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive in order
to limit the inrush current to the load.
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main battery line in an
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automobile is the source of a number of nasty potential
transients, including load dump, reverse and double
battery.
Load dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 8 is the most straightforward
approach to protect a DC/DC converter from the ravages
of an automotive battery line. The series diode prevents
current from flowing during reverse battery, while the
transient suppressor clamps the input voltage during load
dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamp the input voltage below breakdown of the converter.
Although the LTC1625 has a maximum input voltage of
36V, most applications will be limited to 30V by the
MOSFET V(BR)DSS.
50A IPK
RATING
12V
For 40% ripple current at maximum VIN the inductor
should be:
L≥
3.3V
3.3V
1–
= 16µH
(225kHz)(0.4)(2A)
22V
Choosing a standard value of 15µH results in a maximum
ripple current of:
∆IL(MAX) =
Next, check that the minimum value of the current limit is
acceptable. Assume a junction temperature close to a
70°C ambient with ρ80°C = 1.3.
ILIMIT ≥
1
150mV
– 0.83A = 2.3A
(0.042Ω)(1.3) 2
This is comfortably above IO(MAX) = 2A. Now double-check
the assumed TJ:
3.3V
(2.3A)2 (1.3)(0.042Ω) +
22V
(1.7)(22)2(2.3A)(180pF )(225kHz)
= 43mW + 77mW = 120mW
PTOP =
VIN
LTC1625
3.3V
3.3V
1–
= 0.83A
22V
(225kHz)(15µH)
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
TJ = 70°C + (120mW)(50°C/W) = 76°C
PGND
1625 F08
Since ρ(76°C) ≅ ρ(80°C), the solution is self-consistent.
Figure 8. Automotive Application Protection
A short circuit to ground will result in a folded back
current of:
Design Example
As a design example, take a supply with the following
specifications: VIN = 12V to 22V (15V nominal), VOUT =
3.3V, IO(MAX) = 2A, and f = 225kHz. The required RDS(ON)
can immediately be estimated:
RDS(ON) =
120mV
= 0.046Ω
(2A)(1.3)
A 0.042Ω Siliconix Si4412DY MOSFET (θJA = 50°C/W) is
close to this value.
ISC =
1 (15V)(0.5µs)
30mV
+
= 1.2A
(0.03Ω)(1.1) 2
15µH
with a typical value of RDS(ON) and ρ(50°C) = 1.1. The
resulting power dissipated in the bottom MOSFET is:
PBOT =
15V – 3.3V
(1.2A)2 (1.1)(0.03Ω) = 37mW
15V
which is less than under full load conditions.
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1
CSS
0.1µF
CC1
470pF
INTVCC
RC
10k
OPEN
CC2
220pF
2
EXTVCC
VIN
SYNC
TK
3
+
16
15
M1
Si4412DY
14
SW
RUN/SS
LTC1625
13
TG
FCB
5
12
BOOST
ITH
L1
15µH
4
6
7
8
SGND
INTVCC
BG
VOSENSE
VPROG
PGND
11
CIN
22µF
35V
×2
CVCC
DB
CB
4.7µF
CMDSH-3 0.1µF
10
9
VOUT
3.3V
2A
+
+
VIN
12V TO 22V
M2
Si4412DY
COUT
100µF
10V
0.065Ω
×2
D1
MBRS140T3
1625 F09
CIN: AVX TPSE226M035R0300
COUT: AVX TPSD107M010R0065
L1: SUMIDA CDRH125-150MC
Figure 9. 3.3V/2A Fixed Output at 225kHz
CIN is chosen for an RMS current rating of at least 1A at
temperature. COUT is chosen with an ESR of 0.033Ω for
low output ripple. The output ripple in continuous mode
will be highest at the maximum input voltage and is
approximately:
∆VO = (∆IL(MAX))(ESR) = (0.83A)(0.033Ω) = 27mV
The complete circuit is shown in Figure 9.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1625. These items are also illustrated graphically in
the layout diagram of Figure 10. Check the following in
your layout:
1) Connect the TK lead directly to the drain of the topside
MOSFET. Then connect the drain to the (+) plate of CIN.
This capacitor provides the AC current to the top
MOSFET.
2) The power ground pin connects directly to the source of
the bottom N-channel MOSFET. Then connect the source
to the anode of the Schottky diode and (–) plate of CIN,
which should have as short lead lengths as possible.
18
3) The LTC1625 signal ground pin must return to the (–)
plate of COUT. Connect the (–) plate of COUT to power
ground at the source of the bottom MOSFET
4) Keep the switch node SW away from sensitive smallsignal nodes. Ideally the switch node should be placed
on the opposite side of the power MOSFETs from the
LTC1625.
5) Connect the INTVCC decoupling capacitor CVCC closely
to the INTVCC pin and the power ground pin. This
capacitor carries the MOSFET gate drive current.
6) Does the VOSENSE pin connect directly to the (+) plate of
COUT? In adjustable applications, the resistive divider
(R1, R2) must be connected between the (+) plate of
COUT and signal ground. Place the divider near the
LTC1625 in order to keep the high impedance VOSENSE
node short.
7) For applications with multiple switching power converters connected to the same VIN, ensure that the input
filter capacitance for the LTC1625 is not shared with the
other converters. AC input current from another converter will cause substantial input voltage ripple that
may interfere with proper operation of the LTC1625. A
few inches of PC trace or wire (≈100nH) between CIN
and VIN is sufficient to prevent sharing.
LTC1625
U
U
W
U
APPLICATIONS INFORMATION
+
OPTIONAL 5V EXTVCC
CONNECTION
2
EXT
CLK
CSS
CC1
1
EXTVCC
VIN
SYNC
TK
16
15
M1
3
14
RUN/SS
SW
LTC1625
4
13
FCB
TG
OPEN
RC
5
ITH
L1
CB
DB
6
R2
R1
SGND
7
VOSENSE
8
OPEN
VPROG
VIN
12
BOOST
CVCC
11
INTVCC
+
+
10
BG
D1
M2
9
PGND
CIN
–
–
OUTPUT DIVIDER
REQUIRED
WITH VPROG OPEN
+
VOUT
COUT
+
1625 F10
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 10. LTC1625 Layout Diagram
U
TYPICAL APPLICATIONS
5V/1.2A Fixed Output at 225kHz
1
CSS
0.1µF
CC
330pF RC
10k
INTVCC
OPEN
2
EXTVCC
VIN
SYNC
TK
3
+
16
15
M1
1/2 Si9936DY
L1
39µH
14
SW
RUN/SS
LTC1625
13
TG
FCB
5
12
BOOST
ITH
4
6
7
8
CIN: AVX TPSD156M035R0300
COUT: AVX TPSD107M010R0100
L1: SUMIDA CD104-390MC
SGND
VOSENSE
VPROG
INTVCC
BG
PGND
11
DB
CMDSH-3
9
VIN
5V TO 28V
VOUT
5V
1.2A
CVCC
CB 4.7µF
0.1µF
+
+
10
CIN
15µF
35V
M2
1/2 Si9936DY
COUT
100µF
10V
0.100Ω
1625 TA02
19
LTC1625
U
TYPICAL APPLICATIONS
2.5V/2.8A Adjustable Output
RF
4.7Ω
1
CSS
0.1µF
2
3
CC1
1nF
EXTVCC
VIN
SYNC
TK
RUN/SS
SW
16
15
OPEN
8
VOSENSE
VPROG
BG
PGND
M1
1/2 Si4920DY
14
LTC1625
13
RC OPEN 4 FCB
TG
10k
5
12
BOOST
ITH
CC2
330pF
6
11
INTVCC
SGND
7
+
CF
0.1µF
CIN
22µF
35V
×2
L1
15µH
DB
CMDSH-3
VOUT
2.5V
2.8A
R2
11k
1%
CVCC
CB 4.7µF
0.22µF
+
10
+
R1
10k
1%
M2
1/2 Si4920DY
9
VIN
5V TO 28V
COUT
100µF
10V
0.065Ω
×2
D1
MBRS140T3
1625 TA03
CIN: AVX TPSE226M020R0300
COUT: AVX TPSD107M010R0065
L1: SUMIDA CDRH125-150MC
3.3V/7A Fixed Output
RF
4.7Ω
1
CSS
0.1µF
CC1
2.2nF
RC
10k
CC2
220pF
EXTVCC
VIN
EXT 2
TK
SYNC
CLK 3
SW
RUN/SS
LTC1625
4
TG
FCB
OPEN
5
BOOST
ITH
6
7
8
SGND
VOSENSE
VPROG
INTVCC
BG
PGND
16
+
CF
0.1µF
15
M1
FDS6680A
14
L1
7µH
13
12
11
CVCC
DB
CB 4.7µF
CMDSH-3 0.22µF
+
+
10
9
M2
FDS6680A
1625 TA05
20
VIN
5V TO 28V
VOUT
3.3V
7A
D1
MBRS140T3
CIN: SANYO 30SC10M
COUT: SANYO 6SA150M
CIN
10µF
30V
×3
COUT
150µF
6.3V
0.03Ω
×2
LTC1625
U
TYPICAL APPLICATIONS
3.3V/4A Fixed Output with 12V/120mA Auxiliary Output
RF
4.7Ω
CF
0.1µF
VIN
6V TO 20V
CIN
10µF
30V
×2
+
M1
IRLR3103
• T1
1
CSS
0.1µF
2
EXT
CLK
VIN
SYNC
TK
•
15
14
SW
RUN/SS
LTC1625
4
13
TG
FCB
5
12
BOOST
ITH
CC2
220pF
6
7
8
SGND
INTVCC
BG
VOSENSE
VPROG
PGND
CS
0.1µF
8µH
1:2.53
DS
SM4003TR*
3
RC
10k
CC1
470pF
EXTVCC
16
CB
0.22µF
DB
CMDSH-3
11
VSEC
12V
120mA
RS
100k
M3
NDT410EL
+
CSEC
3.3µF
35V
R1
4.7k
CVCC
4.7µF
D2
CDMSH-3
+
10
9
+
C1
0.01µF
M2
IRLR3103
R4
95.3k
1%
R3
11k
1%
VOUT
3.3V
COUT
4A
100µF
10V
0.065Ω
×3
D1
MBRS140T3
1625 TA04
CIN: SANYO 30SC10M
COUT: AVX TPSD107M010R0065
CSEC: AVX TAJB335M035R
T1: BH ELECTRONICS 510-1079
*YES! USE A STANDARD RECOVERY DIODE
12V/2.2A Adjustable Output
RF
4.7Ω
1
CSS
0.1µF
2
EXTVCC
VIN
SYNC
TK
3
CC
470pF
RC
22k
15
M1
Si4412DY
14
L1
27µH
RUN/SS
SW
LTC1625
13
FCB
TG
5
12
ITH
BOOST
7
8
SGND
VOSENSE
VPROG
INTVCC
BG
PGND
11
DB
CMDSH-3
CVCC
CB 4.7µF
0.1µF
9
M2
Si4412DY
VIN
12.5V TO 28V
VOUT
12V
2A
R2
35.7k
1% +
+
10
CIN
22µF
35V
×2
+
CF
0.1µF
4
6
OPEN
16
R1
3.92k
1%
COUT
68µF
20V
0.15Ω
×2
1625TA06
CIN: AVX TPSE226M020R0300
COUT: AVX TPSE686M020R0150
L1: SUMIDA CDRH127-270MC
21
LTC1625
U
TYPICAL APPLICATIONS
– 5V/4.5A Positive to Negative Converter
RF
4.7Ω
VIN
5V TO 10V
CF
0.1µF
1
2
EXTVCC
VIN
SYNC
TK
16
3
CC1
2.2nF
RC
10k
14
SW
RUN/SS
LTC1625
4
13
TG
FCB
5
12
BOOST
ITH
CSS
0.1µF
CC2
220pF
6
7
8
SGND
INTVCC
VOSENSE
VPROG
BG
DB
CMDSH-3
11
10
CIN
220µF
16V
L1
6µH
CB
0.22µF
+
PGND
+
M1
FDS6670A
15
D1
MBR140T3
+
M2
FDS6670A
CVCC
4.7µF
COUT
470µF
6.3V
VOUT
–5V
4.5A
9
1625TA08
CIN: SANYO 16SV220M
COUT: SANYO 6SV470M
L1: MAGNETICS Kool-Mµ 77120-A7, 9 TURNS, 17 GAUGE
Single Inductor, Positive Output Buck Boost
RF
4.7Ω
CF
0.1µF
1
CSS
0.1µF
2
3
4
RC
10k
5
CC2
220pF
CC1
2.2nF
6
7
R1
3.92k
8
EXTVCC
VIN
SYNC
TK
SW
RUN/SS
FCB
LTC1625
ITH
SGND
VOSENSE
VPROG
TG
BOOST
INTVCC
BG
PGND
VIN
6V TO 18V
+
16
M1
Si4420DY
15
CIN
68µF
20V
x2
VIN
IOUT
18
12
6
4.0
3.3
2.0
D2
MBRS340T3
14
L1
18µH
13
VOUT
12V
CB
0.33µF
12
R1
100k
DB
CMDSH-3
11
10
+
CVCC
4.7µF
9
M2
Si4420DY
D1
MBRS
340T3
R2
35.7k
Z1
MMBZ
5240
10V
1
D3
BAT85
8
+
M3
Si4420DY
7
2 1/2
LTC1693-2
3
D4
BAT85
M4
Si4425DY
C1
470pF
4
5
6 1/2
LTC1693-2
C2
0.1µF
D5
BAT85
1625TA09
CIN: SANYO 20S68M
COUT: SANYO 16SA100M
L1: 7A, 18µH Kool-Mµ 77120-A7, 15 TURNS, 17 GAUGE
22
COUT
100µF
16V
30mΩ
x2
LTC1625
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
GN Package
16-Lead Plastic SSOP (Narrow 0.150)
(LTC DWG # 05-08-1641)
0.189 – 0.196*
(4.801 – 4.978)
0.009
(0.229)
REF
16 15 14 13 12 11 10 9
0.229 – 0.244
(5.817 – 6.198)
0.150 – 0.157**
(3.810 – 3.988)
1
0.015 ± 0.004
× 45°
(0.38 ± 0.10)
0.007 – 0.0098
(0.178 – 0.249)
2 3
4
5 6
0.053 – 0.068
(1.351 – 1.727)
8
7
0.004 – 0.0098
(0.102 – 0.249)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.025
(0.635)
BSC
0.008 – 0.012
(0.203 – 0.305)
* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
GN16 (SSOP) 0398
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
2
3
4
5
6
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
8
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
0.406 – 1.270
7
0.050
(1.270)
TYP
S16 0695
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LTC1625
U
TYPICAL APPLICATION
3.3V/1.8A Fixed Output
1
CSS
0.1µF
2
EXTVCC
VIN
16
+
15
TK
14
SW
RUN/SS
LTC1625
4
13
TG
FCB
5
12
BOOST
ITH
SYNC
M1
1/2 Si4936DY
3
CC1
1nF
RC
10k
OPEN
CC2
100pF
6
7
8
SGND
VOSENSE
VPROG
INTVCC
BG
PGND
11
CIN
15µF
35V
×2
L1
27µH
CVCC
DB
CB 4.7µF
CMDSH-3 0.1µF
9
VOUT
3.3V
1.8A
COUT
100µF
10V
0.1Ω
×2
+
+
10
VIN
5V TO 28V
M2
1/2 Si4936DY
D1
MBRS140T3
1625 TA07
CIN: AVX TPSD156M035R0300
COUT: AVX TPSD107M010R0100
L1: SUMIDA CDRH125-270MC
RELATED PARTS
PART NUMBER
LTC1435A
LTC1436A-PLL
LTC1438
LTC1530
DESCRIPTION
High Efficiency Synchronous Step-Down Controller
High Efficiency Low Noise Synchronous Step-Down Controller
Dual High Efficiency Step-Down Controller
High Power Synchronous Step-Down Controller
LTC1538-AUX
LTC1649
Dual High Efficiency Step-Down Controller
3.3V Input High Power Step-Down Controller
24
Linear Technology Corporation
COMMENTS
Optimized for Low Duty Cycle Battery to CPU Power Applications
PLL Synchronization and Auxiliary Linear Regulator
Power-On Reset and Low-Battery Comparator
SO-8 with Current Limit, No RSENSE Saves Space, Fixed
Frequency Ideal for 5V to 3.3V
5V Standby Output and Auxiliary Linear Regulator
2.7V to 5V Input, 90% Efficiency, Ideal for 3.3V to 1.xV – 2.xV
Up to 20A
1625f LT/TP 1298 4K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1998
工商网监
湘ICP备2023018690号
