LTC1872B
Constant Frequency
Current Mode Step-Up
DC/DC Controller in ThinSOT
Features
Description
Burst Mode™ Operation Disabled for Lower Output
Ripple at Light Loads
nn High Efficiency: Over 90%
nn High Output Currents Easily Achieved
nn Wide V Range: 2.5V to 9.8V
IN
nn V
OUT Limited Only by External Components
nn Constant Frequency 550kHz Operation
nn Current Mode Operation for Excellent Line and Load
Transient Response
nn Shutdown Mode Draws Only 8µA Supply Current
nn Low Profile (1mm) ThinSOT™ Package
The LTC®1872B is a constant frequency current mode
step-up DC/DC controller providing excellent AC and DC
load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the
LTC1872B when the input voltage falls below 2.0V.
nn
Applications
Optical Communications
Lithium-Ion-Powered Applications
nn Cellular Telephones
nn Wireless Devices
nn Portable Computers
nn Scanners
nn
The LTC1872B provides a ± 2.5% output voltage accuracy
and consumes only 270µA of quiescent current. In shutdown, the device draws a mere 8µA.
High constant operating frequency of 550kHz allows the
use of a small external inductor. The constant frequency
operation is maintained down to very light loads, resulting
in less low frequency noise generation over a wide load
current range.
The LTC1872B is available in a 6-lead low profile (1mm)
ThinSOT package. For a Burst Mode operation enabled
version of the LTC1872B, please refer to the LTC1872
data sheet.
nn
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Typical Application
C1
10µF
10V
147k
220pF
80.6k
1
2
3
ITH/RUN
VIN
LTC1872B
GND
SENSE –
VFB
NGATE
5
M1
D1
90
C2
4× 10µF
10V
VOUT
5V
1A
422k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R106K010AL
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: Si2302DS
R1: DALE 0.25W
VIN = 3.3V
VOUT = 5V
95
L1
4.7µH
4
6
100
EFFICIENCY (%)
R1
0.03Ω
Typical Efficiency vs Load Current*
VIN
3.3V
85
80
75
70
65
1872B TA01
1
10
100
LOAD CURRENT (mA)
1000
1872B TA01b
Figure 1. LTC1872B High Output Current 3.3V to 5V Boost Converter
*Output ripple waveforms for the circuit of Figure 1 appear in Figure 2.
1872bfa
For more information www.linear.com/LTC1872B
1
LTC1872B
Absolute Maximum Ratings
Pin Configuration
(Note 1)
Input Supply Voltage (VIN).......................... – 0.3V to 10V
SENSE–, NGATE Voltages.............. –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages............................... –0.3V to 2.4V
NGATE Peak Output Current (< 10µs) ......................... 1A
Storage Ambient Temperature Range.....– 65°C to 150°C
Operating Temperature Range (Note 2)....– 40°C to 85°C
Junction Temperature (Note 3).............................. 150°C
Lead Temperature (Soldering, 10 sec).................... 300°C
TOP VIEW
ITH/RUN 1
6 NGATE
GND 2
5 VIN
4 SENSE –
VFB 3
S6 PACKAGE
6-LEAD PLASTIC SOT-23
TJMAX = 150°C, θJA = 230°C/W
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC1872BES6#PBF
LTC1872BES6#TRPBF
LTXY
16-Lead Plastic SOT-23
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on nonstandard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Undervoltage Lockout Threshold
Typicals at VIN = 4.2V (Note 4)
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V
VIN < UVLO Threshold
VIN Falling
VIN Rising
Shutdown Threshold (at ITH/RUN)
Start-Up Current Source
Regulated Feedback Voltage
VFB Input Current
Oscillator Frequency
Gate Drive Rise Time
Gate Drive Fall Time
Peak Current Sense Voltage
VITH/RUN = 0V
0°C to 70°C(Note 5)
– 40°C to 85°C(Note 5)
(Note 5)
VFB = 0.8V
CLOAD = 3000pF
CLOAD = 3000pF
(Note 6)
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1872BE is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the – 40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA°C/W)
2
MIN
TYP
MAX
UNITS
l
1.55
1.85
270
230
8
6
2.00
2.10
420
370
22
10
2.35
2.40
µA
µA
µA
µA
V
V
l
0.15
0.25
0.780
0.770
0.35
0.5
0.800
0.800
10
550
40
40
120
0.55
0.85
0.820
0.830
50
650
V
µA
V
V
nA
kHz
ns
ns
mV
l
l
l
500
114
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1872B is tested in a feedback loop that servos VFB to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense
voltage is VREF/6.67 at duty cycle 7.5% by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Low Load Current Operation
Under very light load current conditions, the ITH/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will assure that the current
comparator remains tripped (even at zero load current) and
the regulator will start to skip cycles, as it must, in order
to maintain regulation. This behavior allows the regulator
to maintain constant frequency down to very light loads,
resulting in less low frequency noise generation over a
wide load current range.
For more information www.linear.com/LTC1872B
1872bfa
LTC1872B
Operation
(Refer to Functional Diagram)
Figure 2 illustrates this result for the circuit of Figure 1
using both an LTC1872 in Burst Mode operation and an
LTC1872B (non-Burst Mode operation). At an output
current of 50mA, the Burst Mode operation part exhibits
an output ripple of approximately 80mVP-P, whereas the
non-Burst Mode operation part has an output ripple of
≈45mVP-P. At lower output current levels, the improvement
is even greater. This comes at a trade off of slightly lower
efficiency for the non-Burst Mode operation part. Also
notice the constant frequency operation of the LTC1872B,
even at 5% of maximum output current.
Slope Compensation and Inductor’s Peak Current
The inductor’s peak current is determined by:
IPK =
VITH −0.7
10 (RSENSE )
when the LTC1872B is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor
current. The amount of reduction is given by the curves
in Figure 3.
110
Undervoltage Lockout
100
To prevent operation of the N-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorporated into the LTC1872B. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
and all circuitry is turned off except the undervoltage block,
which draws only several microamperes.
SF = IOUT/IOUT(MAX) (%)
90
80
70
60
50
IRIPPLE = 0.4IPK
AT 5% DUTY CYCLE
IRIPPLE = 0.2IPK
AT 5% DUTY CYCLE
40
30
Overvoltage Protection
20
The overvoltage comparator in the LTC1872B will turn the
external MOSFET off when the feedback voltage has risen
7.5% above the reference voltage of 0.8V. This comparator
has a typical hysteresis of 20mV.
20mV AC/DIV
10
VIN = 4.2V
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1872B F03
Figure 3. Maximum Output Current vs Duty Cycle
20mV AC/DIV
VIN = 3.3V
VOUT = 5V
IOUT = 50mA
5µs/DIV
1872B F02a
(2a) VOUT Ripple for Figure 1 Circuit
Using LTC1872 Burst Mode Operation
VIN = 3.3V
VOUT = 5V
IOUT = 50mA
5µs/DIV
1872B F02b
(2b) VOUT Ripple for Figure 1 Circuit Using
LTC1872B Non-Burst Mode Operation
Figure 2. Output Ripple Waveforms for the Circuit of Figure 1
1872bfa
For more information www.linear.com/LTC1872B
5
LTC1872B
Operation
(Refer to Functional Diagram)
Short-Circuit Protection
Since the power switch in a boost converter is not in
series with the power path from input to load, turning off
the switch provides no protection from a short-circuit at
the output. External means such as a fuse in series with
the boost inductor must be employed to handle this fault
condition.
Applications Information
The basic LTC1872B application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1 and
RSENSE (= R1). Next, the power MOSFET and the output
diode D1 is selected followed by CIN(= C1) and COUT(= C2).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator
determines the inductor’s peak current. The output current
the LTC1872B can provide is given by:
⎛ 0.12 I
⎞ V
IN
IOUT = ⎜
− RIPPLE ⎟
2 ⎠ VOUT + VD
⎝ R SENSE
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section) and VD is the
forward drop of the output diode at the full rated output
current.
A reasonable starting point for setting ripple current is:
IRIPPLE = (O.4) (IOUT )
VOUT + VD
VIN
⎛ V
⎞
IN
⎜
⎟
(10) (IOUT ) (100) ⎝ VOUT + VD ⎠
SF
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor ripple
current. However, this is at the expense of efficiency due
to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple
current. The ripple current, IRIPPLE, decreases with higher
inductance or frequency and increases with higher VOUT.
The inductor’s peak-to-peak ripple current is given by:
V ⎛ V + V − V ⎞
IRIPPLE = IN ⎜ OUT D IN ⎟
f (L ) ⎝ VOUT + VD ⎠
where f is the operating frequency. Accepting larger values
of IRIPPLE allows the use of low inductances, but results
in higher output voltage ripple and greater core losses.
A reasonable starting point for setting ripple current is:
⎛ V + V ⎞
IRIPPLE = 0.4 IOUT(MAX ) ⎜ OUT D ⎟
⎝ VIN ⎠
(
Rearranging the above equation, it becomes:
⎛ V
⎞
1
IN
RSENSE =
⎜
⎟
(10) ( IOUT) ⎝ VOUT + VD ⎠
for Duty Cycle
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