LTC1877
High Efficiency
Monolithic Synchronous
Step-Down Regulator
DESCRIPTION
FEATURES
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High Efficiency: Up to 95%
Very Low Quiescent Current: Only 10μA
During Operation
600mA Output Current at VIN = 5V
2.65V to 10V Input Voltage Range
550kHz Constant-Frequency Operation
No Schottky Diode Required
Low Dropout Operation: 100% Duty Cycle
Synchronizable from 400kHz to 700kHz
Selectable Burst Mode® Operation or
Pulse-Skipping Mode
0.8V Reference Allows Low Output Voltages
Shutdown Mode Draws < 1μA Supply Current
±2% Output Voltage Accuracy
Current Mode Control for Excellent Line and Load
Transient Response
Overcurrent and Overtemperature Protected
Available in 8-Lead MSOP Package
The LTC®1877 s a high efficiency monolithic synchronous
buck regulator using a constant-frequency, current mode
architecture. Supply current during operation is only 10μA
and drops to < 1μA in shutdown. The 2.65V to 10V input
voltage range makes the LTC1877 ideally suited for both
single and dual Li-Ion battery-powered applications. 100%
duty cycle provides low dropout operation, extending battery life in portable systems.
Switching frequency is internally set at 550kHz, allowing
the use of small surface mount inductors and capacitors.
For noise sensitive applications the LTC1877 can be externally synchronized from 400kHz to 700kHz. Burst Mode
operation is inhibited during synchronization or when the
SYNC/MODE pin is pulled low, preventing low frequency
ripple from interfering with audio circuitry.
The internal synchronous switch increases efficiency
and eliminates the need for an external Schottky diode.
Low output voltages are easily supported with the 0.8V
feedback reference voltage. The LTC1877 is available in a
space saving 8-lead MSOP package. Lower input voltage
applications (less than 7V abs max) should refer to the
LTC1878 data sheet.
APPLICATIONS
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Cellular Telephones
Wireless Modems
Personal Information Appliances
Portable Instruments
Distributed Power Systems
Battery-Powered Equipment
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
TYPICAL APPLICATION
Efficiency vs Output Current
100
High Efficiency Step-Down Converter
5
SW
SYNC
6
LTC1877
VIN
1
RUN
3
2
ITH GND VFB
7
10μF**
CER
10μH*
20pF
+
47μF***
887k
280k
220pF
VIN = 10V
85
80
75
VIN = 7.2V
VIN = 5V
70
65
4
*TOKO D62CB A920CY-100M
**TAIYO-YUDEN CERAMIC LMK325BJ106MN
***SANYO POSCAP 6TPA47M
†
VOUT CONNECTED TO VIN FOR 2.65V < VIN < 3.3V
VIN = 3.6V
90
†
VOUT
3.3V
EFFICIENCY (%)
VIN
2.65V
TO 10V
95
60
55
50
0.1
1877 TA01
VOUT = 3.3V
L = 10μH
Burst Mode OPERATION
1.0
100
10
OUTPUT CURRENT (mA)
1000
1877 TA02
1877fb
1
LTC1877
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
Input Supply Voltage (VIN) ..........................–0.3V to 11V
ITH, PLL LPF Voltage ................................. –0.3V to 2.7V
RUN, VFB Voltages ...................................... –0.3V to VIN
SYNC/MODE Voltage .................................. –0.3V to VIN
SW Voltage ................................... –0.3V to (VIN + 0.3V)
P-Channel MOSFET Source Current (DC)............ 800mA
N-Channel MOSFET Sink Current (DC)................ 800mA
Peak SW Sink and Source Current .......................... 1.5A
Operating Temperature Range (Note 2) .....–40°C to 85°C
Junction Temperature (Note 3) ............................ 125°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................... 300°C
TOP VIEW
RUN
ITH
VFB
GND
1
2
3
4
8
7
6
5
PLL LPF
SYNC/MODE
VIN
SW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 150°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC1877EMS8#PBF
LTC1877EMS8#TRPBF
LTLU
8-Lead Plastic MSOP
–40°C to 85°C
LTC1877IMS8#PBF
LTC1877IMS8#TRPBF
LTLV
8-Lead Plastic MSOP
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
IVFB
Feedback Current
(Note 4)
l
MIN
VFB
Regulated Output Voltage
(Note 4) 0°C ≤ TA ≤ 85°C
(Note 4) –40°C ≤ TA ≤ 85°C
l
ΔVOVL
Output Overvoltage Lockout
ΔVOVL = VOVL – VFB
l
ΔVFB
Reference Voltage Line Regulation
VIN = 2.65V to 10V (Note 4)
VLOADREG
Output Voltage Load Regulation
Measured in Servo Loop; VITH = 0.9V to 1.2V
Measured in Servo Loop; VITH = 1.6V to 1.2V
VIN
Input Voltage Range
IQ
Input DC Bias Current
Pulse-Skipping Mode
Burst Mode Operation
Shutdown
(Note 5)
2.65V < VIN < 10V, VSYNC/MODE = 0V, IOUT = 0A
VSYNC/MODE = VIN, IOUT = 0A
VRUN = 0V, VIN = 10V
fOSC
Oscillator Frequency
VFB = 0.8V
VFB = 0V
fSYNC
SYNC Capture Range
TYP
MAX
4
30
nA
0.784
0.74
0.8
0.8
0.816
0.84
V
V
20
50
110
mV
0.05
0.15
%/V
0.1
–0.1
0.5
–0.5
%
%
10
V
230
10
0
350
15
1
μA
μA
μA
550
80
605
kHz
kHz
700
kHz
l
l
l
2.65
495
400
UNITS
1877fb
2
LTC1877
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
IPLL LPF
Phase Detector Output Current
Sinking Capability
Sourcing Capability
fPLLIN < fOSC
fPLLIN > fOSC
l
l
MIN
TYP
MAX
UNITS
3
–3
10
–10
20
–20
μA
μA
RPFET
RDS(ON) of P-Channel MOSFET
ISW = 100mA
0.65
0.85
Ω
RNFET
RDS(ON) of N-Channel MOSFET
ISW = –100mA
0.75
0.95
Ω
IPK
Peak Inductor Current
VFB = 0.7V, Duty Cycle < 35%
1.0
1.25
A
ILSW
SW Leakage
VRUN = 0V, VSW = 0V or 8.5V, VIN = 8.5V
±0.01
±1
μA
VSYNC/MODE
SYNC/MODE Threshold
1.0
1.5
V
±0.01
±1
μA
0.7
1.5
V
±0.01
±1
μA
ISYNC/MODE
SYNC/MODE Leakage Current
VRUN
RUN Threshold
IRUN
RUN Input Current
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC1877E is guaranteed to meet specifications from 0°C to
85°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls. The LTC1877I is guaranteed over the full –40°C to 85°C
operating temperature range.
0.8
l
0.3
l
0.3
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1877EMS8: TJ = TA + (PD)(150°C/W)
Note 4: The LTC1877 is tested in a feedback loop which servos VFB to the
balance point for the error amplifier (VITH = 1.2V).
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
ILOAD = 100mA I
LOAD = 10mA
95
90
80
85
70
ILOAD = 300mA
80
ILOAD = 1mA
75
70
ILOAD = 0.1mA
65
60
2
6
8
4
INPUT VOLTAGE (V)
VIN = 3.6V
90
VIN = 7.2V
85
VIN = 7.2V
VIN = 3.6V
50
40
PULSE-SKIPPING MODE
Burst Mode OPERATION
20
10
50
0
60
30
VOUT = 2.5V
L = 10μH
Burst Mode OPERATION
55
Efficiency vs Output Current
95
90
EFFICIENCY (%)
EFFICIENCY (%)
Efficiency vs Output Current
100
10
12
1877 G01
0
0.1
VOUT = 2.5V
L = 10μH
1.0
100
10
OUTPUT CURRENT (mA)
1000
1877 G02
EFFICIENCY (%)
100
L = 15μH
L = 10μH
80
75
70
65
60
55
0.1
VIN = 10V
VOUT = 3.3V
Burst Mode OPERATION
1.0
100
10
OUTPUT CURRENT (mA)
1000
1877 G03
1877fb
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LTC1877
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Output Current
Oscillator Frequency
vs Temperature
Reference Voltage vs Temperature
605
0.814
95
VIN = 5V
VIN = 3.6V
90
EFFICIENCY (%)
VIN = 10V
VIN = 5V
75
70
65
60
50
0.1
0.799
0.794
545
535
50
25
75
0
TEMPERATURE (°C)
100
495
–50
125
2.53
595
2.52
585
2.51
555
545
535
525
100
RDS(ON) vs Input Voltage
PULSE-SKIPPING MODE
VIN = 4.2V
L = 10μH
1.0
2.50
SYNCHRONOUS
SWITCH
0.8
2.49
2.48
2.47
0.6
MAIN SWITCH
0.4
2.46
2.45
515
125
1.2
RDS(ON) (Ω)
565
25
50
75
0
TEMPERATURE (°C)
1877 G06
Output Voltage vs Load Current
605
575
–25
1877 G05
OUTPUT VOLTAGE (V)
OSCILLATOR FREQUENCY (kHz)
555
515
0.784
–50 –25
Oscillator Frequency
vs Supply Voltage
0.2
2.44
505
2.43
0
2
6
8
4
SUPPLY VOLTAGE (V)
10
200
600
400
LOAD CURRENT (mA)
0
12
0.8
VIN = 5V
0.6
SYNCHRONOUS SWITCH
MAIN SWITCH
50
25
75
0
TEMPERATURE (°C)
VOUT = 1.8V
150
100
50
125
1877 G10
8
9
10
PULSE-SKIPPING MODE
200
150
100
50
Burst Mode OPERATION
100
3 4 5 6 7
INPUT VOLTAGE (V)
2
VIN = 5V
250
200
SUPPLY CURRENT (μA)
DC SUPPLY CURRENT (μA)
1.2
0.2
–50 –25
300
PULSE-SKIPPING MODE
VIN = 3V
1
DC Supply Current vs Temperature
250
1.0
0
1877 G09
DC Supply Current vs Input Voltage
RDS(ON) vs Temperature
1.4
0.4
0
800
1877 G08
1877 G07
RDS(ON) (Ω)
565
525
1877 G04
495
575
505
1000
1.0
100
10
OUTPUT CURRENT (mA)
0.804
0.789
L = 10μH
VOUT = 2.5V
55
585
FREQUENCY (kHz)
VIN = 7.2V
80
REFERENCE VOLTAGE (V)
0.809
85
VIN = 5V
595
Burst Mode OPERATION
0
0
2
6
4
INPUT VOLTAGE (V)
8
10
1877 G11
0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1877 G11b
1877fb
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LTC1877
TYPICAL PERFORMANCE CHARACTERISTICS
Switch Leakage vs Temperature
Switch Leakage vs Input Voltage
1400
VIN = 10V
RUN = 0V
1000
MAIN SWITCH
800
600
400
SYNCHRONOUS
SWITCH
200
RUN = 0V
SW
5V/DIV
12
10
50
25
75
0
TEMPERATURE (°C)
100
6
4
0
125
VOUT
20mV/DIV
AC COUPLED
SYNCHRONOUS
SWITCH
8
IL
200mA/DIV
2
0
–50 –25
Burst Mode Operation
14
SWITCH LEAKAGE (nA)
SWITCH LEAKAGE (nA)
1200
16
MAIN SWITCH
0
2
4
6
INPUT VOLTAGE (V)
8
1877 G12
10
1877 G13
Start-Up from Shutdown
VIN = 5V
VOUT = 1.5V
CIN = 10μF
10μs/DIV
L = 10μH
COUT = 47μF
ILOAD = 50mA
1877 G14
Load Step Response
RUN
5V/DIV
VOUT
50mV/DIV
AC COUPLED
VOUT
1V/DIV
IL
500mA/DIV
IL
500mA/DIV
ITH
1V/DIV
50μs/DIV
1877 G16
1877 G15
CIN = 10μF
COUT = 47μF
ILOAD = 500mA
VIN = 5V
VOUT = 1.5V
L = 10μH
VIN = 5V
VOUT = 1.5V
L = 10μH
Load Step Response
40μs/DIV
CIN = 10μF
COUT = 47μF
ILOAD = 50mA TO 500mA
PULSE-SKIPPING MODE
Load Step Response
VOUT
50mV/DIV
AC COUPLED
VOUT
50mV/DIV
AC COUPLED
IL
500mA/DIV
10Ms/DIV
IL
500mA/DIV
ITH
1V/DIV
ITH
1V/DIV
VIN = 5V
VOUT = 1.5V
L = 10μH
1877 G17
40μs/DIV
CIN = 10μF
COUT = 47μF
ILOAD = 50mA TO 500mA
Burst Mode OPERATION
1877 G18
VIN = 5V
VOUT = 1.5V
L = 10μH
20μs/DIV
CIN = 10μF
COUT = 47μF
ILOAD = 200mA TO 500mA
PULSE-SKIPPING MODE
1877fb
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LTC1877
PIN FUNCTIONS
RUN (Pin 1): Run Control Input. Forcing this pin below
0.4V shuts down the LTC1877. In shutdown, all functions
are disabled drawing 6.25% by turning the main switch off and keeping it off
until the fault is removed.
Burst Mode Operation
The LTC1877 is capable of Burst Mode operation in which
the internal power MOSFETs operate intermittently based
on load demand. To enable Burst Mode operation, simply
tie the SYNC/MODE pin to VIN or connect it to a logic
high (VSYNC/MODE > 1.5V). To disable Burst Mode operation and enable PWM pulse-skipping mode, connect the
SYNC/MODE pin to GND. In this mode, the efficiency is
lower at light loads, but becomes comparable to Burst
Mode operation when the output load exceeds 50mA. The
advantage of pulse-skipping mode is lower output ripple
and less interference to audio circuitry.
When the converter is in Burst Mode operation, the peak
current of the inductor is set to approximately 250mA,
even though the voltage at the ITH pin indicates a lower
value. The voltage at the ITH pin drops when the inductor’s
average current is greater than the load requirement. As the
ITH voltage drops below approximately 0.55V, the BURST
comparator trips, causing the internal sleep line to go
high and forces off both power MOSFETs. The ITH pin is
then disconnected from the output of the EA amplifier and
parked a diode voltage above ground.
In sleep mode, both power MOSFETs are held off and a
majority of the internal circuitry is partially turned off,
reducing the quiescent current to 10μA. The load current
is now being supplied solely from the output capacitor.
When the output voltage drops, the ITH pin reconnects
to the output of the EA amplifier and the top MOSFET is
again turned on and this process repeats.
Short-Circuit Protection
When the output is shorted to ground, the frequency of
the oscillator is reduced to about 80kHz, one-seventh the
nominal frequency. This frequency foldback ensures that
the inductor current has ample time to decay, thereby
preventing runaway. The oscillator’s frequency will progressively increase to 550kHz (or the synchronized frequency)
when VFB rises above 0.3V.
Frequency Synchronization
A phase-locked loop (PLL) is available on the LTC1877 to
allow the internal oscillator to be synchronized to an external
source connected to the SYNC/MODE pin. The output of
the phase detector at the PLL LPF pin operates over a 0V
to 2.4V range corresponding to 400kHz to 700kHz. When
locked, the PLL aligns the turn-on of the top MOSFET to
the rising edge of the synchronizing signal.
When the LTC1877 is clocked by an external source, Burst
Mode operation is disabled; the LTC1877 then operates in
PWM pulse-skipping mode. In this mode, when the output
load is very low, current comparator ICOMP may remain
tripped for several cycles and force the main switch to stay
off for the same number of cycles. Increasing the output
load slightly allows constant-frequency PWM operation
to resume. This mode exhibits low output ripple as well
as low audio noise and reduced RF interference while
providing reasonable low current efficiency.
Frequency synchronization is inhibited when the feedback
voltage VFB is below 0.6V. This prevents the external clock
from interfering with the frequency foldback for shortcircuit protection.
1877fb
8
LTC1877
OPERATION
Dropout Operation
When the input supply voltage decreases toward the output
voltage, the duty cycle increases toward the maximum
on-time. Further reduction of the supply voltage forces the
main switch to remain on for more than one cycle until it
reaches 100% duty cycle. The output voltage will then be
determined by the input voltage minus the voltage drop
across the internal P-channel MOSFET and the inductor.
Low Supply Operation
The LTC1877 is designed to operate down to an input supply
voltage of 2.65V although the maximum allowable output
current is reduced at this low voltage. Figure 1 shows the
reduction in the maximum output current as a function of
input voltage for various output voltages.
1200
Slope compensation provides stability in constant-frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current signal
at duty cycles in excess of 40%. As a result, the maximum
inductor peak current is reduced for duty cycles >40%.
This is shown in the decrease of the inductor peak current
as a function of duty cycle graph in Figure 2.
MAXIMUM INDUCTOR PEAK CURRENT (mA)
1000
MAX OUTPUT CURRENT (mA)
Slope Compensation and Inductor Peak Current
1100
VOUT = 2.5V
VIN = 5V
1000
VOUT = 1.5V
800
VOUT = 5V
600
VOUT = 3.3V
400
200
L = 10μH
0
Another important detail to remember is that at low input
supply voltages, the RDS(ON) of the P-channel switch increases. Therefore, the user should calculate the power
dissipation when the LTC1877 is used at 100% duty cycle
with a low input voltage (see Thermal Considerations in
the Applications Information section).
0
2
4
6
VIN (V)
8
10
12
1877 F01
Figure 1. Maximum Output Current vs Input Voltage
900
800
700
600
0
20
60
40
DUTY CYCLE (%)
80
100
1877 F02
Figure 2. Maximum Inductor Peak Current vs Duty Cycle
APPLICATIONS INFORMATION
The basic LTC1877 application circuit is shown on the first
page. External component selection is driven by the load
requirement and begins with the selection of L followed
by CIN and COUT.
Inductor Value Calculation
The inductor selection will depend on the operating frequency of the LTC1877. The internal nominal frequency is
550kHz, but can be externally synchronized from 400kHz
to 700kHz.
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of
smaller inductor and capacitor values. However, operating
at a higher frequency generally results in lower efficiency
because of increased internal gate charge losses.
The inductor value has a direct effect on ripple current.
The ripple current ΔIL decreases with higher inductance
or frequency and increases with higher VIN or VOUT.
⎛ V
⎞
1
ΔIL =
VOUT ⎜1− OUT ⎟
VIN ⎠
⎝
( f) (L)
(1)
1877fb
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LTC1877
APPLICATIONS INFORMATION
Accepting larger values of ΔIL allows the use of low inductance, but results in higher output voltage ripple and
greater core losses. A reasonable starting point for setting
ripple current is ΔIL = 0.4(IMAX).
The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when
the inductor current peaks fall to approximately 250mA.
Lower inductor values (higher ΔIL) will cause this to occur
at lower load currents, which can cause a dip in efficiency
in the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to increase.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy,
or Kool Mμ cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates hard, which means that
inductance collapses abruptly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Kool Mμ (from Magnetics, Inc.) is a very good, low loss core
material for toroids with a soft saturation characteristic.
Molypermalloy is slightly more efficient at high (>200kHz)
switching frequencies but quite a bit more expensive. Toroids are very space efficient, especially when you can use
several layers of wire, while inductors wound on bobbins
are generally easier to surface mount. New designs for
surface mount inductors are available from Coiltronics,
Coilcraft, Dale and Sumida.
CIN and COUT Selection
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum RMS
capacitor current is given by:
1/2
⎡⎣ VOUT ( VIN − VOUT )⎤⎦
CIN required IRMS ≅ IOMAX
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do
not offer much relief. Note the capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of
life. This makes it advisable to further derate the capacitor,
or choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the manufacturer if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering.
The output ripple ΔVOUT is determined by:
⎛
1 ⎞
ΔVOUT ≅ ΔIL ⎜ESR +
⎟
8fCOUT ⎠
⎝
where f = operating frequency, COUT = output capacitance
and ΔIL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ΔIL increases
with input voltage. For the LTC1877, the general rule for
proper operation is:
COUT required ESR < 0.25Ω
The choice of using a smaller output capacitance increases
the output ripple voltage due to the frequency dependent
term but can be compensated for by using capacitor(s) of
very low ESR to maintain low ripple voltage. The ITH pin
compensation components can be optimized to provide
stable high performance transient response regardless of
the output capacitor selected.
ESR is a direct function of the volume of the capacitor.
Manufacturers such as Taiyo Yuden, AVX, Sprague, Kemet
and Sanyo should be considered for high performance ca1877fb
10
LTC1877
APPLICATIONS INFORMATION
pacitors. The POSCAP solid electrolytic capacitor available
from Sanyo is an excellent choice for output bulk capacitors
due to its low ESR/size ratio. Once the ESR requirement
for COUT has been met, the RMS current rating generally
far exceeds the IRIPPLE(P-P) requirement.
When using tantalum capacitors, it is critical that they are
surge tested for use in switching power supplies. A good
choice is the AVX TPS series of surface mount tantalum,
available in case heights ranging from 2mm to 4mm. Other
capacitor types include KEMET T510 and T495 series and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
The output of the phase detector is a pair of complementary
current sources charging or discharging the external filter
network on the PLL LPF pin. The relationship between the
voltage on the PLL LPF pin and operating frequency is
shown in Figure 4. A simplified block diagram is shown
in Figure 5.
If the external frequency (VSYNC/MODE) is greater than
550kHz, the center frequency, current is sourced continuously, pulling up the PLL LPF pin. When the external
frequency is less than 550kHz, current is sunk continuously,
pulling down the PLL LPF pin. If the external and internal
frequencies are the same but exhibit a phase difference,
the current sources turn on for an amount of time corresponding to the phase difference. Thus, the voltage on
the PLL LPF pin is adjusted until the phase and frequency
of the external and internal oscillators are identical. At
800
⎛ R2 ⎞
VOUT = 0.8V ⎜1+ ⎟
⎝ R1 ⎠
OSCILLATOR FREQUENCY (kHz)
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing, as shown in Figure 3.
0.8V b VOUT b 10V
R2
700
600
500
400
VFB
LTC1877
300
R1
0
0.4
GND
0.8
1.2
VPLL LPF (V)
1.6
2.0
1877 F04
1877 F03
Figure 3. Setting the LTC1877 Output Voltage
Figure 4. Relationship Between Oscillator
Frequency and Voltage at PLL LPF Pin
Phase-Locked Loop and Frequency Synchronization
The LTC1877 has an internal voltage-controlled oscillator
and phase detector comprising a phase-locked loop. This
allows the top MOSFET turn-on to be locked to the rising
edge of an external frequency source. The frequency range
of the voltage-controlled oscillator is 400kHz to 700kHz.
The phase detector used is an edge sensitive digital type
that provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector will
not lock up on input frequencies close to the harmonics
of the VCO center frequency. The PLL hold-in range ΔfH is
equal to the capture range, ΔfH = ΔfC = ±150kHz.
RLP
PHASE
DETECTOR
2.4V
CLP
PLL LPF
SYNC/
MODE
DIGITAL
PHASE/
FREQUENCY
DETECTOR
VCO
1877 F05
Figure 5. Phase-Locked Loop Block Diagram
1877fb
11
LTC1877
APPLICATIONS INFORMATION
The loop filter components CLP and RLP smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
component’s CLP and RLP determine how fast the loop
acquires lock. Typically RLP = 10k and CLP is 2200pF
to 0.01μF. When not synchronized to an external clock,
the internal connection to the VCO is disconnected. This
disallows setting the internal oscillator frequency by a DC
voltage on the VPLL LPF pin.
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of
the losses in LTC1877 circuits: VIN quiescent current and
I2R losses. The VIN quiescent current loss dominates the
efficiency loss at very low load currents, whereas the
I2R loss dominates the efficiency loss at medium to high
load currents. In a typical efficiency plot, the efficiency
curve at very low load currents can be misleading since
the actual power lost is of no consequence, as illustrated
in Figure 6.
1. The VIN quiescent current is due to two components:
the DC bias current as given in the Electrical Characteristics section and the internal main switch and synchronous switch gate charge currents. The gate charge
current results from switching the gate capacitance
of the internal power MOSFET switches. Each time
the gate is switched from high to low to high again, a
packet of charge dQ moves from VIN to ground. The
resulting dQ/dt is the current out of VIN that is typically
larger than the DC bias current. In continuous mode,
IGATECHG = f(QT + QB) where QT and QB are the gate
charges of the internal top and bottom switches. Both
the DC bias and gate charge losses are proportional
to VIN and thus their effects will be more pronounced
at higher supply voltages.
2. I2R losses are calculated from the resistances of the
internal switches, RSW , and external inductor RL. In
continuous mode, the average output current flowing through inductor L is chopped between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add
RSW to RL and multiply the result by the square of the
average output current.
Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less
than 2% total additional loss.
1
0.1
POWER LOST (W)
this stable operating point the phase comparator output
is high impedance and the filter capacitor CLP holds the
voltage.
0.01
VIN = 4.2V
L = 10μH
VOUT = 1.5V
VOUT = 2.5V
VOUT = 3.3V
Burst Mode OPERATION
0.001
0.0001
0.00001
0.1
1
10
100
LOAD CURRENT (mA)
1000
1877 F06
Figure 6. Power Lost vs Load Current
Thermal Considerations
In most applications the LTC1877 does not dissipate much
heat due to its high efficiency. But, in applications where the
LTC1877 is running at high ambient temperature with low
supply voltage and high duty cycles, such as in dropout,
the heat dissipated may exceed the maximum junction
1877fb
12
LTC1877
APPLICATIONS INFORMATION
temperature of the part. If the junction temperature reaches
approximately 150°C, both power switches will be turned
off and the SW node will become high impedance.
To avoid the LTC1877 from exceeding the maximum junction temperature, the user will need to do some thermal
analysis. The goal of the thermal analysis is to determine
whether the power dissipated exceeds the maximum
junction temperature of the part. The temperature rise is
given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to
the ambient temperature.
The junction temperature, TJ, is given by:
TJ = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC1877 in dropout at an
input voltage of 3V, a load current of 500mA, and an ambient temperature of 70°C. From the typical performance
graph of switch resistance, the RDS(ON) of the P-channel
switch at 70°C is approximately 0.9Ω. Therefore, power
dissipated by the part is:
PD = ILOAD2 • RDS(ON) = 0.225W
For the MSOP package, the θJA is 150°C/W. Thus, the
junction temperature of the regulator is:
TJ = 70°C + (0.225)(150) = 104°C
which is below the maximum junction temperature of
125°C.
Note that at higher supply voltages, the junction temperature
is lower due to reduced switch resistance (RDS(ON)).
Checking Transient Response
The regulator loop response can be checked by looking at the load transient response. Switching regulators
take several cycles to respond to a step in load current.
When a load step occurs, VOUT immediately shifts by an
amount equal to (ΔILOAD • ESR), where ESR is the effective
series resistance of COUT. ΔILOAD also begins to charge
or discharge COUT, which generates a feedback error
signal. The regulator loop then acts to return VOUT to its
steady-state value. During this recovery time VOUT can be
monitored for overshoot or ringing that would indicate a
stability problem. The internal compensation provides
adequate compensation for most applications. But if additional compensation is required, the ITH pin can be used
for external compensation using RC, CC1, as shown in
Figure 7. The 220pF capacitor, CC2, is typically needed for
noise decoupling.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator
can deliver enough current to prevent this problem if the
load switch resistance is low and it is driven quickly. The
only solution is to limit the rise time of the switch drive
so that the load rise time is limited to approximately
(25 • CLOAD). Thus, a 10μF capacitor charging to 3.3V
would require a 250μs rise time, limiting the charging
current to about 130mA.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1877. These items are also illustrated graphically
in the layout diagram of Figure 7. Check the following in
your layout:
1. Are the signal and power grounds segregated? The
LTC1877 signal ground consists of the resistive divider,
the optional compensation network (RC and CC1) and
CC2. The power ground consists of the (–) plate of CIN,
the (–) plate of COUT and Pin 4 of the LTC1877. The power
ground traces should be kept short, direct and wide. The
signal ground and power ground should converge to a
common node in a star-ground configuration.
2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected
between the (+) plate of COUT and signal ground.
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
4. Keep the switching node SW away from sensitive small
signal nodes.
1877fb
13
LTC1877
APPLICATIONS INFORMATION
Design Example
A 15μH inductor works well for this application. For best
efficiency choose a 1A inductor with less than 0.25Ω
series resistance.
As a design example, assume the LTC1877 is used in a
single lithium-ion battery-powered cellular phone application. The input voltage will be operating from a maximum
of 4.2V down to about 2.7V. The load current requirement
is a maximum of 0.3A but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both
low and high load currents is important. Output voltage
is 2.5V. With this information we can calculate L using
Equation (1),
⎛ V
⎞
1
L=
VOUT ⎜1− OUT ⎟
VIN ⎠
⎝
( f) (ΔIL )
(3)
CIN will require an RMS current rating of at least 0.15A
at temperature and COUT will require an ESR of less than
0.25Ω. In most applications, the requirements for these
capacitors are fairly similar.
For the feedback resistors, choose R1 = 412k. R2 can then
be calculated from Equation (2) to be:
⎛V
⎞
R2 = ⎜ OUT – 1⎟ R1= 875.5k; use 887k
⎝ 0.8
⎠
Figure 8 shows the complete circuit along with its efficiency curve.
Substituting VOUT = 2.5V, VIN = 4.2V, ΔIL=120mA and
f = 550kHz in Equation (3) gives:
L=
⎛ 2.5V ⎞
2.5V
⎜1−
⎟ = 15.3μH
550kHz(120mA) ⎝ 4.2V ⎠
CC2
LTC1877
OPTIONAL
1
CC1
RC
2
3
4
R1
R2
RUN
ITH
PLL LPF
SYNC/MODE
VFB
VIN
GND
SW
8
7
BOLD LINES INDICATE
HIGH CURRENT PATHS
6
5
+
L1
+
+
+
CIN
COUT
VOUT
VIN
–
–
1877 F07
Figure 7. LTC1877 Layout Diagram
2
3
4
RUN
ITH
PLL LPF
SYNC/MODE
LTC1877
VFB
VIN
GND
SW
8
85
7
6
5
15μH*
+
887k
412k
VIN = 3.0V
90
20pF
VOUT
2.5V
47μF***
EFFICIENCY (%)
220pF
1
95
VIN
2.7V TO 4.2V
10μF**
CER
80
VIN = 4.2V
VIN = 3.6V
75
70
65
60
1877 F08a
*SUMIDA CD54-150
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***SANYO POSCAP 6TPA47M
VOUT = 2.5V
L = 15μH
55
50
0.1
100
1.0
10
OUTPUT CURRENT (mA)
1000
1877 F08b
Figure 8. Single Lithium-Ion to 2.5V/0.3A Regulator from Design Example
1877fb
14
LTC1877
TYPICAL APPLICATIONS
Dual Lithium-Ion to 2.5V/0.6A Regulator
Using All Ceramic Capacitors
1
2
RUN
ITH
220pF
PLL LPF
SYNC/MODE
1
8
CIN**
10μF
CER
4
VFB
VIN
GND
SW
6
2
VIN
b 8.4V
7
LTC1877
3
4-to 6-Cell NiCd/NiMH to 1.8V/0.6A Regulator
Using All Ceramic Capacitors
4
VOUT
2.5V/0.6A
887k
VFB
VIN
GND
SW
1877 TA03
2
ITH
220pF
PLL LPF
SYNC/MODE
8
7
LTC1877
3
4
VFB
GND
VIN
SW
EXT
CLOCK
700kHz
5
VOUT
1.8V/0.6A
10μH*
887k
698k
*TOKO D62CB A920CY-100M
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***TAIYO YUDEN CERAMIC JMK325BJ226MM
1877 TA04
VIN
2.85V
TO 10V
1
CIN**
10μF
CER
10k
2
RUN
ITH
220pF
VOUT
2.5V/
0.6A
10μH*
20pF
PLL LPF
SYNC/MODE
8
CIN**
10μF
CER
7
LTC1877
3
5
4
VFB
VIN
GND
SW
6
5
20pF
COUT***
22μF
CER
1877 TA05
VOUT
2.5V/0.3A
15μH*
887k
412k
COUT***
22μF
CER
Low Noise 2.5V/0.3A Regulator
6
*TOKO D62CB A920CY-100M
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***TAIYO YUDEN CERAMIC JMK325BJ226MM
6
VIN
4V
TO 10V
0.01μF
RUN
VIN
b 9V
CIN**
10μF
CER
20pF
Externally Synchronized 2.5V/0.6A Regulator
Using All Ceramic Capacitors
1
7
COUT***
22μF
CER
412k
*SUMIDA CD54-150
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***TAIYO YUDEN CERAMIC JMK325BJ226MM
SYNC/MODE
8
LTC1877
3
15μH*
PLL LPF
ITH
220pF
5
20pF
RUN
*SUMIDA CD54-150
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***SANYO POSCAP 6TPA47M
887k
COUT***
47μF
6.3V
412k
1877 TA06
1877fb
15
LTC1877
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MS8 Package
8-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1660 Rev F)
0.889 ± 0.127
(.035 ± .005)
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.65
(.0256)
BSC
0.42 ± 0.038
(.0165 ± .0015)
TYP
8
7 6 5
0.52
(.0205)
REF
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0° – 6° TYP
GAUGE PLANE
1
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
2 3
4
1.10
(.043)
MAX
0.86
(.034)
REF
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.65
(.0256)
NOTE:
BSC
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MS8) 0307 REV F
1877fb
16
LTC1877
REVISION HISTORY
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
PAGE NUMBER
B
11/11
Part marking for LTC1877IMS8 corrected from LTLU to LTLV
2
1877fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
17
LTC1877
TYPICAL APPLICATION
Single Lithium-Ion to 3.3V/0.3A Regulator
1
2
RUN
ITH
220pF
PLL LPF
SYNC/MODE
10μF**
8
+
–
7
VIN
Li-Ion BATTERY
3V TO 4.2V
LTC1877
3
4
VFB
VIN
GND
SW
6
5
VOUT
3.3V/0.25A
10μH*
20pF
887k
47μF***
280k
*TOKO D62CB A920CY-100M
**TAIYO YUDEN CERAMIC LMK325BJ106MN
***SANYO POSCAP 6TPA47M
1877 TA07
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1174/LTC1174-3.3/
LTC1174-5
High Efficiency Step-Down and Inverting DC/DC Converters
Monolithic Switching Regulators, IOUT to 450mA,
Burst Mode Operation
LTC1265
1.2A, High Efficiency Step-Down DC/DC Converter
Constant Off-Time, Monolithic, Burst Mode Operation
LT®1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
High Frequency, Small Inductor, High Efficiency
LTC1436/LTC1436-PLL
High Efficiency, Low Noise, Synchronous Step-Down Converters 24-Pin Narrow SSOP
LTC1474/LTC1475
Low Quiescent Current Step-Down DC/DC Converters
Monolithic, IOUT to 250mA, IQ = 10μA, 8-Pin MSOP
LTC1504A
Monolithic Synchronous Step-Down Switching Regulator
Low Cost, Voltage Mode IOUT to 500mA, VIN: 4V to 10V
LTC1622
Low Input Voltage Current Mode Step-Down DC/DC Controller
High Frequency, High Efficiency, 8-Pin MSOP
LTC1626
Low Voltage, High Efficiency Step-Down DC/DC Converter
Monolithic, Constant Off-Time, IOUT to 600mA, Low Supply
Voltage Range: 2.5V to 6V
LTC1627
Monolithic Synchronous Step-Down Switching Regulator
Constant Frequency, IOUT to 500mA, Secondary Winding
Regulation, VIN: 2.65V to 8.5V
LTC1701
Monolithic Current Mode Step-Down Switching Regulator
Constant Off-Time, IOUT to 500mA, 1MHz Operation,
VIN: 2.5V to 5.5V
LTC1707
Monolithic Synchronous Step-Down Switching Regulator
1.19V VREF Pin, Constant Frequency, IOUT to 600mA,
VIN: 2.65V to 8.5V
LTC1735
High Efficiency, Synchronous Step-Down Converter
16-Pin SO and SSOP, VIN Up to 36V, Fault Protection
LTC1772
Low Input Voltage Current Mode Step-Down DC/DC Controller
550kHz, 6-Pin SOT-23, IOUT Up to 5A, VIN: 2.2V to 10V
LTC1878
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, VIN Up to 6V, IQ = 10μA, IOUT to 600mA
1877fb
18 Linear Technology Corporation
LT 1111 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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