LTC3119
18V, 5A Synchronous
Buck-Boost DC/DC Converter
Features
Description
Input Voltage Range: 2.5V to 18V
nn Runs Down to V = 250mV After Start-Up
IN
nn Output Voltage Range: 0.8V to 18V
nn 5A Output Current in Buck Mode, V > 6V
IN
nn 3A Output Current for V = 3.6V, V
IN
OUT = 5V
nn Programmable Switching Frequency: 400kHz to 2MHz
nn Synchronizable with an External Clock Up to 2MHz
nn Accurate Run Comparator Threshold
nn Burst Mode® Operation, No-Load I = 35µA
Q
nn Ultralow Noise Buck-Boost PWM
nn Current Mode Control
nn Maximum Power Point Control
nn Power Good Indicator
nn Internal Soft-Start
nn 28-Lead 4mm × 5mm QFN and TSSOP Packages
The LTC®3119 is a high efficiency 18V monolithic buckboost converter that can deliver up to 5A of continuous
output current. Extensive feature integration and very
low resistance internal power switches minimize the total
solution footprint for even the most demanding applications. A proprietary 4-switch PWM architecture provides
seamless low noise operation from input voltages above,
equal to, or below the output voltage.
Applications
Other features include: output short-circuit protection,
thermal overload protection, less than 3µA shutdown
current, power good indicator, Burst Mode operation, and
maximum power point control.
nn
Wide Input Range Power Supplies
1- to 4-Cell Lithium Battery Powered Products
nn RF Power Supplies
nn Solar Battery Chargers
nn System Backup Power Supplies
nn Lead Acid to 12V Regulator
nn
nn
External frequency programming as well as synchronization using an internal PLL enable operation over a wide
switching frequency range of 400kHz to 2MHz. The wide
2.5V to 18V input range is well suited for operation from
unregulated power sources including battery stacks and
backup capacitors. After start-up, operation is possible
with input voltages as low as 250mV.
The LTC3119 is offered in thermally enhanced 28-lead
4mm × 5mm QFN and TSSOP packages.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and PowerPath and No RSENSE are trademarks of Analog Devices, Inc. All other trademarks are
the property of their respective owners.
Typical Application
Wide Input Range 5V Regulator
Efficiency Efficiency
100
3.3µH
0.1µF
VIN
2.5V TO 18V
10µF
100k
SW1
BST1
PVIN
VIN
RUN
BURST PWM
4.7µF
SW2
BST2
PVOUT
150µF
LTC3119
536k
PWM/SYNC
FB
MPPC
VCC
PGOOD
SVCC
VC
RT
PGND GND
105k
VOUT
5V AT 5A, VIN > 6V
5V AT 3A, VIN = 3.6V
3119 TA01a
80
EFFICIENCY (%)
0.1µF
Burst Mode OPERATION
90
70
60
PWM
50
40
30
102k
78.7k
680pF
VIN = 3V
VIN = 5V
VIN = 9V
20
10
100μ
1m
10m
100m
LOAD CURRENT (A)
1
10
3119 TA01b
3119fb
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1
LTC3119
Absolute Maximum Ratings
(Note 1)
VIN, PVIN, PVOUT, RUN, PGOOD.................. –0.3V to 19V
FB, VC, RT, SYNC, MPPC, VCC, SVCC............ –0.3V to 6V
BST1 Voltage...................... (SW1 – 0.3V) to (SW1 + 6V)
BST2 Voltage......................(SW2 – 0.3V) to (SW2 + 6V)
Operating Junction Temperature (Notes 2, 3)
LTC3119E/LTC3119I............................ –40°C to 125°C
LTC3119H........................................... –40°C to 150°C
LTC3119MP........................................ –55°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
FE...................................................................... 300°C
Pin Configuration
TOP VIEW
N/C
1
28 PWM/SYNC
26 BST1
28 27 26 25 24 23
PGND
4
25 PGND
SW2
5
24 SW1
N/C
27 N/C
3
N/C
2
N/C
N/C
BST2
BST2
BST1
PWM/SYNC
TOP VIEW
PGND 1
22 PGND
SW2 2
21 SW1
PVOUT 3
20 PVIN
PVOUT 4
19 PVIN
29
PGND
SW2 5
18 SW1
17 PGND
PGND 6
PGOOD 7
16 VIN
SVCC 8
RT
VCC
MPPC
SGND
FB
7
SW2
8
PGND
9
23 PVIN
29
PGND
SVCC 11
9 10 11 12 13 14
VC
6
PGOOD 10
15 RUN
22 PVIN
21 SW1
20 PGND
19 VIN
18 RUN
FB 12
17 RT
VC 13
16 VCC
SGND 14
UFD PACKAGE
28-LEAD (4mm × 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 22°C/W
EXPOSED PAD (PIN 29) IS PGND, MUST BE SOLDERED TO PCB
Order Information
PVOUT
PVOUT
15 MPPC
FE PACKAGE
28-LEAD PLASTIC TSSOP
TJMAX = 150°C, θJA = 21°C/W
EXPOSED PAD (PIN 29) IS PGND, MUST BE SOLDERED TO PCB
http://www.linear.com/product/LTC3119#orderinfo
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3119EUFD#PBF
LTC3119EUFD#TRPBF
3119
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3119IUFD#PBF
LTC3119IUFD#TRPBF
3119
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3119EFE#PBF
LTC3119EFE#TRPBF
3119
28-Lead Plastic Enhanced TSSOP
–40°C to 125°C
LTC3119IFE#PBF
LTC3119IFE#TRPBF
3119
28-Lead Plastic Enhanced TSSOP
–40°C to 125°C
LTC3119HFE#PBF
LTC3119HFE#TRPBF
3119
28-Lead Plastic Enhanced TSSOP
–40°C to 150°C
LTC3119MPFE#PBF
LTC3119MPFE#TRPBF
3119
28-Lead Plastic Enhanced TSSOP
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
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LTC3119
Electrical Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = PVIN = 12V, PVOUT = 5V, RT = 76.8k unless otherwise stated.
PARAMETER
Input Operating Voltage
CONDITIONS
MIN
After Start-Up (Note 4)
Output Operating Voltage
VCC Undervoltage Lockout Threshold
VCC Rising
VCC Falling
TYP
MAX
2.5
0.25
18
18
V
V
l
0.8
18
V
l
l
2.18
2.4
V
V
VCC Undervoltage Lockout Hysteresis
2.35
2.25
60
Input Current in Shutdown
RUN = 0V
Input Current in Sleep
FB = 0.9V
UNITS
l
l
mV
3
31
µA
Oscillator Frequency
l
900
1100
kHz
Oscillator Operating Frequency
l
400
2000
kHz
PWM/SYNC Frequency Range
l
400
2000
kHz
PWM/SYNC Logic Threshold
PWM/SYNC Pulse Width
l
Minimum Low or High Duration
0.3
1000
µA
0.7
1
Soft-Start Duration
6
Feedback Voltage
l
787
779
Feedback Pin Current
Error Amplifier Transconductance
VRUN Rising
795
795
803
811
1
50
0.3
0.8
1
l
1.17
1.205
1.24
RUN Pin Hysteresis Voltage
PGOOD Hysteresis
Percentage of FB Voltage
l
–9.5
–8
700
VPGOOD = 18V
MPPC Pin Threshold
l
774
l
7
MPPC Pin Current
Inductor Current Limit
(Note 3)
Burst Mode Operation Inductor
Current Limit
VIN > VOUT (Note 3)
Maximum Duty Cycle
Percentage of Period SW2 is Low in Boost Mode
l
90
Minimum Duty Cycle
Percentage of Period SW1 is High in Buck Mode
l
0
SW1, SW2 Minimum Low Time
V
V
mV
–6.5
1.2
PGOOD Pull Down Resistance
nA
µA
90
Percentage of FB Voltage Falling
mV
mV
µS
0.25
PGOOD Threshold
nA
ms
l
RUN Pin Hysteresis Current
PGOOD Leakage
50
120
RUN Pin Logic Threshold
V
ns
PWM/SYNC Pin Current
RUN Pin Comparator Threshold
1.1
100
%
%
2000
Ω
1
40
nA
798
822
mV
1
50
nA
8
A
0.6
A
92
%
90
ns
%
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3
LTC3119
Electrical Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = PVIN = 12V, PVOUT = 5V, RT = 76.8k unless otherwise stated.
PARAMETER
CONDITIONS
N-Channel Switch Resistance
Switch A (PVIN to SW1)
Switch B (SW1 to PGND)
Switch C (SW2 to PGND)
Switch D (SW2 to PVOUT)
MIN
30
30
30
30
N-Channel Switch Leakage
PVIN = PVOUT = 18V, SW1 = SW2 = 0V, 18V
1
10
3.70
3.85
VCC Regulation Voltage
VCC Dropout Voltage
l
90
VCC Current Limit
VCC Reverse Current
3.55
VCC Current = 50mA, VIN = 3V
TYP
180
VCC = 5V, VIN = 3V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3119 is tested under pulsed load conditions such
that TJ ≈ TA. The LTC3119E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specification over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3119I specifications are guaranteed over the –40°C to 125°C operating
junction temperature range. The LTC3119H specifications are guaranteed
over the –40°C to 150°C operating junction temperature range. The
LTC3119MP specifications are guaranteed and tested over the –55°C to
150°C operating junction temperature range. High temperatures degrade
operating lifetimes; operating lifetime is derated for junction temperatures
greater than 125°C.
MAX
UNITS
mΩ
mΩ
mΩ
mΩ
µA
V
mV
mA
5
µA
Note 3: Current measurements are performed when the LTC3119 is
not switching. The current limit values measured in operation will be
somewhat higher due to the propagation delay of the comparators.
Note 4: Minimum input voltage is governed by the VCC UVLO threshold. If
VCC is maintained though external bootstrapping, the part will continue to
operate until power transfer to the output is no longer possible.
Note 5: Switch timing measurements are made in an open-loop test
configuration. Timing in the application may vary somewhat from these
values due to differences in the switch pin voltage during the non-overlap
durations when switch pin voltage is influenced by the magnitude and
direction of the inductor current.
Note 6: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
junction temperature may impair device reliability or permanently damage
the device.
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LTC3119
Typical Performance Characteristics
Output
Voltage Line Regulation
Line Regulation
0.4
CHANGE IN VOUT (%)
0.6
0.2
0
–0.2
–0.4
0.0
–0.2
–0.4
–0.6
–0.8
–0.8
6
8
10 12 14
INPUT VOLTAGE (V)
16
–1.0
0.001
18
0.01
0.1
LOAD CURRENT (A)
1
35
38
32
26
20
–60
6
–30
0
30
60
90
TEMPERATURE (° C)
120
150
3119 G03
Run Pin Hysteresis Current vs
Temperature
Power Switch Resistance vs VCC
260
34
HYSTERESIS CURRENT (nA)
255
33
32
31
30
29
28
27
44
3119 G02
3119 G01
SWITCH RESISTANCE (mΩ)
250
245
240
235
230
225
2
2.5
3
3.5
VCC VOLTAGE (V)
220
–60
4
3119 G04
Run Pin Comparator Threshold vs
Temperature
vs Temperature
1.35
1.25
1.20
1.15
1.10
1.05
–60
–30
0
30
60
90
TEMPERATURE (° C)
120
150
0
30
60
90
120
150
3119 G05
Run Pin Logic Threshold vs
Temperature
vs Temperature
850
RISING
FALLING
1.30
–30
TEMPERATURE ( ° C)
RISING
FALLING
THRESHOLD VOLTAGE (mV)
4
50
0.2
–0.6
2
Temperature
vs
Temperature
PWM/SYNC = HIGH
0.8
0.4
THRESHOLD VOLTAGE (V)
CHANGE IN VOUT (%)
1.0
LOAD = 1A
PWM/SYNC = HIGH
0.6
–1
Power Switch Resistance vs
Output Voltage Load Regulation
SWITCH RESISTANCE (mΩ)
1
0.8
TA = 25°C, unless otherwise noted.
770
690
610
530
450
–60
3119 G06
–30
0
30
60
90
TEMPERATURE (° C)
120
150
3119 G07
3119fb
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5
LTC3119
Typical Performance Characteristics
Run Pin Current vs Run Pin
Oscillator Frequency vs RT
2.0
25
1.8
20
15
10
5
0
–5
0
2
4
6
8 10 12
VOLTAGE (V)
14
16
1.6
1.4
1.2
1.0
0.8
0.6
0.4
18
1.0
CHANGE FROM VIN = 12V (%)
30
OSCILLATOR FREQUENCY (MHz)
RUN PIN CURRENT (μA)
Voltage
TA = 25°C, unless otherwise noted.
20
50
80
110
140
170
RT PIN RESISTOR (kΩ)
Oscillator Frequency vs
Temperature
–0.5
–1.0
–1.5
2
4
6
8
10 12 14
INPUT VOLTAGE (V)
16
18
3119 G10
FB
FB Voltage
Voltage vs
vs Temperature
Temperature
1.0
0.1
CHANGE FROM 25 ° C (%)
0.7
CHANGE FROM 25 ° C (%)
0
3119 G09
3119 G08
0.4
0.1
–0.2
–0.5
–0.8
–60
–30
0
30
60
90
TEMPERATURE (° C)
120
0.0
–0.1
–0.2
–0.3
–0.4
–60
150
3119 G11
0
30
60
90
120
150
3119 G12
SW1, SW2 Minimum Low Time vs
Temperature
5
0.2
4
0.1
CHANGE FROM 25 ° C (%)
0.0
–0.1
–0.2
–0.3
–0.4
–0.5
–0.6
–60
–30
TEMPERATURE (° C)
MPPC Pin Voltage vs
Temperature
MPPC Voltage vs Temperature
CHANGE FROM 25 ° C (%)
0.5
–2.0
200
Oscillator Frequency vs VIN
IN
3
2
1
0
–1
–2
–3
–4
–30
0
30
60
90
TEMPERATURE ( ° C)
120
150
–5
–60
3119 G13
–30
0
30
60
90
TEMPERATURE (° C)
120
150
3119 G14
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LTC3119
Typical Performance Characteristics
SW2 Maximum Duty Cycle vs
Temperature
94.0
70
93.5
60
93.0
40
30
20
10
2
92.0
91.5
2.5
3
3.5
4
VCC VOLTAGE (V)
4.5
0
30
60
90
TEMPERATURE ( ° C)
35
fSW = 1MHz
120
16.0
–60
150
17
16
15
14
13
12
3119 G16
VCC Current vs Switching
Frequency
VCC Current vs Switching Frequency
3
3.5
4
VCC VOLTAGE (V)
4.5
25
20
15
80
–1.0
0
30
60
90
120
150
3119 G20
Inductor Current Limit vs
Temperature
9.0
0.4
8.5
0.2
0.1
0.0
–0.1
–0.2
–0.3
–1.2
–30
TEMPERATURE (° C)
CURRENT LIMIT (A)
–0.8
3119 G17
90
60
–60
2.0
0.5
VCC CHANGE (%)
VCC CHANGE (%)
–0.6
–1.4
0.6 0.8 1 1.2 1.4 1.6 1.8
SWITCHING FREQUENCY (MHz)
0.3
–0.4
150
100
VCC Line Regulation
–0.2
120
110
3119 G19
fSW = 1MHz
90
70
3119 G18
VCC
CC Load Regulation
60
120
5
0.4
5
30
130
10
2.5
0
VCC Dropout Voltage vs
Temperature
11
2
–30
TEMPERATURE (° C)
30
VCC CURRENT (mA)
VCC CURRENT (mA)
–30
3119 G15
18
0.0
17.0
16.5
90.0
–60
5
VCC
CC Current vs VCC
CC
19
10
fSW = 1MHz
90.5
–10
20
92.5
91.0
0
VVCC
Current vs
vs Temperature
Temperature
CC Current
17.5
VCC CURRENT (mA)
DUTY CYCLE (%)
CHANGE (%)
50
–20
18.0
DROPOUT VOLTAGE (mV)
80
SW1, SW2 Minimum Low Time
vs VCC
TA = 25°C, unless otherwise noted.
8.0
7.5
7.0
6.5
–0.4
0
10 20 30 40 50 60 70 80 90 100
VCC LOAD CURRENT (mA)
3119 G21
–0.5
4
6
8
10
12
14
INPUT VOLTAGE (V)
16
18
3119 G22
6.0
–60
–30
0
30
60
90
TEMPERATURE (° C)
120
150
3119 G23
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7
LTC3119
Typical Performance Characteristics
TA = 25°C, unless otherwise noted.
VCC UVLO Threshold vs
Temperature
Efficiency vs Switching Frequency
2.40
95
RISING
FALLING
2.38
94
2.34
93
EFFICIENCY (%)
THRESHOLD VOLTAGE (V)
2.36
2.32
2.30
2.28
2.26
92
91
2.24
90
2.22
2.20
–60
–30
0
30
60
90
120
TEMPERATURE ( ° C)
89
0.4
150
200
175
0.8
150
0.7
0.6
0.5
0.4
4
6
8
10 12 14
INPUT VOLTAGE (V)
16
100
75
50
25
0
18
2
4
6
8
10 12 14
INPUT VOLTAGE (V)
16
18
3119 G27
Maximum Reverse Current During
0.83)
MPPC Control (V MPPC 6V
PVOUT
3119 F07
Figure 7. Diode-OR of Input Supply and VOUT
Powers VCC Regulator
Programming Custom VIN UVLO Thresholds
With the addition of an external resistive divider connected
to VIN as shown in Figure 8, the RUN pin can be used to
program the input voltage at which the LTC3119 is enabled
and disabled.
For a rising input voltage, the LTC3119 is enabled when VIN
reaches a threshold given by the following equation, where
R1 and R2 are the values of the resistor divider resistors:
Therefore, the rising UVLO threshold and the amount of
hysteresis can be independently programmed via appropriate selection of resistors R1 and R2. For high levels
of hysteresis the value of R1 can become larger than is
desirable in a practical implementation. In such cases, the
amount of hysteresis can be increased further through the
addition of an additional resistor RH, as shown in Figure 9.
When using the additional RH resistor, the rising RUN pin
threshold remains as given by the original equation and
the hysteresis is given by the following expression:
R R2+RHR1+R1R2
0.25µA + R1+R2 0.09V
VHYST = H
R2
R2
VIN
R1+R2
VTH(RISING) = 1.205V
R2
R1
RH
LTC3119
RUN
R2
To ensure robust operation in the presence of noise, the RUN
pin has two forms of hysteresis. A fixed 90mV hysteresis
within the RUN pin comparator provides hysteresis equal
to 7.5% of the input turn-on voltage independent of the
resistor divider values. In addition, an internal hysteresis
current that is sourced from the RUN pin during operation generates an additive level of hysteresis which can
be programmed by the value of R1 to increase the overall
hysteresis to suit the requirements of specific applications.
Once the IC is enabled, it will remain enabled until the
input voltage drops below the comparator threshold by
the hysteresis voltage, VHYST, as given by the following
equation where R1 and R2 are the values of the voltage
divider:
R1+R2
• 0.09V
VHYST =R1• 0.25µA +
R2
GND
3119 F09
Figure 9. Increasing Input UVLO hysteresis
To improve the noise robustness and accuracy of the UVLO
threshold, the RUN pin input can be filtered by adding a
470pF capacitor from RUN to GND. Larger valued capacitors should not be utilized because they could interfere
with operation of the hysteresis.
Switching Frequency Selection
The switching frequency is set by the value of a resistor
connected between the RT pin and ground. The switching
frequency is related to the resistor value by the following
equation where RT is the resistance:
fSW =
100MHz
8 + (1.2• R T / kΩ)
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LTC3119
Applications Information
Higher switching frequencies facilitate the use of smaller
inductors as well as smaller input and output filter capacitors which results in a smaller solution size and reduced
component height. However, higher switching frequencies
also generally reduce conversion efficiency due to the
increased switching losses.
Output Capacitor Selection
A low ESR output capacitor should be utilized at the buckboost converter output in order to minimize output voltage
ripple. Multilayer ceramic capacitors are an excellent option
as they have low ESR and are available in small footprints.
The capacitor value should be chosen large enough to
reduce the output voltage ripple to acceptable levels.
Neglecting the capacitor ESR and ESL, the peak-to-peak
output voltage ripple can be calculated by the following
formulas, where fSW is the switching frequency, COUT is
the output capacitance, tLOW is the switch pin minimum
low time, and ILOAD is the output current. Curves for the
value of tLOW as a function of VCC voltage and temperature
can be found in Typical Performance Characteristics section of this data sheet.
∆VP-P(BUCK ) @
ILOAD tLOW
COUT
∆VP-P(BOOST ) @
ILOAD VOUT − VIN + tLOW fSW VIN
fSW COUT
VOUT
The output voltage ripple increases with load current and
is generally higher in boost mode than in buck mode.
These expressions only take into account the output
voltage ripple that results from the output current being
discontinuous. They provide a good approximation to the
ripple at any significant load current but underestimate
the output voltage ripple at very light loads where output
voltage ripple is dominated by the inductor current ripple.
In addition to output voltage ripple generated across the
output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportional to the series resistance of the output capacitor
and is given by the following expressions where RESR is
the series resistance of the output capacitor and all other
terms are as previously defined.
∆VP-P(BUCK ) @
ILOADRESR
@I
R
1− tLOW fSW LOAD ESR
∆VP-P(BOOST ) @
V
ILOADRESR VOUT
@ILOADRESR OUT
VIN (1− tLOW fSW )
VIN
Input Capacitor Selection
The PVIN pin carries the full inductor current and provides
power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 10μF
should be located as close to this pin as possible. The
traces connecting this capacitor to PVIN and the ground
plane should be made as short as possible. The VIN pin
provides power to the VCC regulator and other internal
circuitry. If the PCB trace connecting VIN to PVIN is long, it
may be necessary to add an additional small value bypass
capacitor near the VIN pin.
When powered through long leads or from a high ESR
power source, a larger value bulk input capacitor may be
required. In such applications, a 47μF to 100μF electrolytic
capacitor in parallel with a 1μF ceramic capacitor generally
yields a high performance, low cost solution.
When powered through an inductive connection such as
a long cable, the inductance of the power source and the
input bypass capacitor form a high-Q resonant LC filter. In
such applications, hot-plugging into a powered source can
lead to a significant voltage overshoot, even up to twice the
nominal input source voltage. Care must be taken in such
situations to ensure that the absolute maximum input voltage
rating of the LTC3119 is not violated. See Linear Technology
Application Note 88 for solutions to increase damping in
the input filter and minimize this voltage overshoot.
Inductor Selection
The choice of inductor used in LTC3119 application circuits
influences the maximum deliverable output current, the
converter bandwidth, the magnitude of the inductor current
ripple and the overall converter efficiency. The inductor must
have a low DC series resistance, when compared to the
3119fb
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LTC3119
Applications Information
internal switch resistance (30mΩ), or output current capability and efficiency will be compromised. Larger inductor
values reduce inductor current ripple but may not increase
output current capability as is the case with peak current
mode control as described in the Maximum Output Current
section. Larger value inductors also tend to have a higher
DC series resistance for a given case size, which will have a
negative impact on efficiency. Larger values of inductance
will also lower the right half plane zero (RHPZ) frequency
when operating in boost mode, which can compromise loop
stability. Nearly all LTC3119 application circuits deliver the
best performance with an inductor value between 1.5μH
and 15μH. Buck mode only applications can use the larger
inductor values as they are unaffected by the RHPZ, while
mostly boost applications generally require inductance on the
lower end of this range depending on how large the step-up
ratio is. Regardless of inductor value, the saturation current
rating should be selected such that it is greater than the worst
case average inductor current plus half of the ripple current.
The peak-to-peak inductor current ripple for each operational mode can be calculated from the following formula,
where fSW is the programmed switching frequency, L is
the inductance and tLOW is the switch pin minimum low
time, typically 90ns.
V −V 1
V
∆IL(P-P )(BUCK ) @ OUT IN OUT
− tLOW
L
VIN
fSW
∆IL(P-P )(BOOST ) @
VIN VOUT − VIN 1
−
t
LOW
L VOUT fSW
It should be noted that the worst-case peak-to-peak inductor ripple current occurs when the duty cycle in buck
mode is minimum (highest VIN) and in boost mode when
the duty cycle is 50% (VOUT @ 2 • VIN).
As an example, if VIN (minimum) = 2.5V and VIN (maximum) = 15V, VOUT = 5V, fSW = 1MHz and L = 4.7μH, the
peak-to-peak inductor ripples at the voltage extremes (15V
VIN for buck and 2.5V VIN for boost) are:
5 15− 5
∆IL (P-P )(BUCK ) @
• 910ns = 645mA
4.7µH 15
2.5 5− 2.5
∆IL (P-P )(BOOST ) @
• 910ns = 242mA
4.7µH 5
One half of this inductor ripple current must be added to
the highest expected average inductor current in order to
select the proper saturation current rating for the inductor.
Programming the Output Voltage
The output voltage is set via the external resistive divider
comprised of resistors RTOP and RBOT as shown in Figure 4.
The resistor divider values determine the output regulation
voltage according to:
R
VOUT = 0.795V • 1+ TOP
RBOT
Programming the MPPC Voltage
The LTC3119 includes an MPPC function to optimize
performance when operating from current limited input
sources. Using an external voltage divider from VIN (refer to
Figure 5), the MPPC function takes control of the average
inductor current when necessary to maintain a minimum
input voltage VMPPC, as programmed by the user.
R5
VMPPC = 0.798V • 1+
R6
This is useful for such applications as photovoltaic powered
converters, since the maximum power transfer point occurs
when the photovoltaic panel is operated at approximately
75% of its open-circuit voltage. For example, when operating from a photovoltaic panel with an open-circuit voltage
of 10V, the maximum power transfer point will be when the
panel is loaded such that its output voltage is about 7.5V.
When using the MPPC function, the input capacitor should
be sized between 100µF and 470µF. Resistor R6 should
be chosen between 50k and 250k. Lower values will result
in smaller undershoot of the MPPC tracking point during
line and load transient conditions, but will draw more
current from the input supply. For this example, a value
of 100k will be used.
V
7.5
R5 = MPPC −1 • R6 =
−1 • 100kΩ
0.798
0.798V
= 838kΩ @ 845kΩ
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21
LTC3119
Applications Information
Compensation of the Buck-Boost Converter
The LTC3119 incorporates an average current mode control
architecture which consists of two control loops. Both the
inner average current mode control loop and outer control
loop require compensation to maintain stability. The inner
current mode control loop is internally compensated to
maintain wide bandwidth and good transient response.
For many applications, the inner current loop can be
treated like a voltage controlled current source (VCCS).
This current source is commanded by the voltage error
amplifier to regulate the output load formed primarily by
the load resistance (RLOAD) and output capacitor (COUT).
This simplified version is illustrated in Figure 10, showing the key components that need to be considered when
compensating the converter.
VOLTAGE
CONTROLLED
CURRENT
SOURCE
+
–
The MPPC loop requires compensation to maintain
stability of the input voltage regulation loop. This can be
accomplished by means of a pole-zero pair on the MPPC
pin created with a series RC network in parallel with the
lower MPPC resistor R6.
The pole and zero locations should be selected to create
a low frequency pole at or below approximately 360Hz
and a zero at a frequency that is scaled based on the size
of the input capacitor. The equations for determining the
values for the compensation capacitor CC2, and zero
resistor RC2 are:
CC2 =
1
2π • R6 • 360Hz
CIN
RC2 =
2π • CC2
1
= 4.42nF @ 4.7nF
2π•100kΩ•360Hz
220µF
RC2 =
= 7.45kΩ @ 7.50kΩ
2π• 4.7nF
CC2 =
22
VOUT
FB
0.795V
COUT
RTOP
RBOT
RCOSER
RLOAD
VC
gm = 10.8A/V
0.96V
GND
RZ
CP2
CP1
3119 F10
Figure 10. Simplified Representation
of Control Loop Components
The bandwidth of the output voltage control loop should
be set low enough to avoid the small signal effects of
the inner current loop. The maximum loop bandwidth is
determined using the inductor value to be approximately:
Using the divider values determined previously, and a
220µF input capacitor, the following compensation values
are obtained:
VOLTAGE LOOP
ERROR
AMP
gm
+
–
Using these resistor values, the MPPC function is programmed to control the maximum input current so as
to maintain VIN at a minimum of 7.56V. Note that if the
photovoltaic panel can provide more power than the
LTC3119 can draw or the load requires, the input voltage
will rise above the programmed MPPC point. Higher input
voltages do not present a problem so long as the input
voltage does not exceed the maximum operating input
voltage. For photovoltaic panel applications, it may be also
desirable to use the programmable RUN feature to disable
the part when VIN drops too low due to lack of sufficient
light. Using the RUN pin provides a well controlled behavior
when the input power source is dropping out. Preventing
switching while under these conditions is important to
minimize discharging of the output storage element due
to switching while the input source is dropping out. This
custom input UVLO voltage should be programmed to be
below the MPPC tracking voltage with sufficient margin to
ensure the part does not disable under transient conditions.
FVLOOP ≤
(4.7µH •100kHz)
10 •L
With a full-scale command on VC, the LTC3119 buckboost converter will generate an average inductor current
of 8A. With a VC voltage range of 220mV to 960mV, the
resulting current gain for the inner average current loop
is 10.8A/V. Similar to peak current mode control, the inner
average current mode control loop effectively turns the
inductor into a current source over the frequency range
of interest, resulting in a frequency response from the
For more information www.linear.com/LTC3119
3119fb
LTC3119
Applications Information
power stage that exhibits a single pole (–20dB/decade)
roll off. The output capacitor (COUT) and load resistance
(RLOAD) form the normally dominant low frequency pole
and the effective series resistance of the output capacitor
and its capacitance form a zero, usually at a high enough
frequency to be ignored. A potentially troublesome right
half plane zero (RHPZ) is also encountered if the LTC3119
is operated in boost mode. The RHPZ causes an increase
in gain, like a zero, but a decrease in phase, like a pole.
This will ultimately limit the maximum converter bandwidth
that can be achieved with the LTC3119. The RHPZ is not
present when operating in buck mode. The overall open
loop gain at DC is the product of the following terms:
Voltage Error Amplifier Gain:
GEA = gm • REA = 120µS • 5MΩ = 600
Voltage Divider Gain:
VFB 0.795V
=
VOUT
VOUT
Current Loop Transconductance:
GCS =
8A
= 10.8A / V
0.74V
It is important to note that GCS is the transconductance
gain from the control voltage VC to the inductor current
level, which equals the output current level in buck mode.
In boost mode, the output current level will be reduced
by the efficiency divided by the boost ratio. Refer to the
typical curves for efficiency information.
GCS(OUT) = 10.8A / V
GCS(OUT) = 10.8A / V •
(Buck Mode)
VIN
•Eff
VOUT
(Boost Mode)
Frequency dependent terms that affect the loop gain include:
Output Load Pole (P1)
fP1 =
1
2π•RLOAD •COUT
Error Amplifier Compensation (P2, Z1)
fP2 =
1
Hz (close to DC)
2π • REA C P1
fZ1 =
1
Hz
2π•R ZCP1
Right Half Plane Zero (RHPZ)
fRHPZ =
VIN 2 •RLOAD
VOUT 2 •2π•L
Hz
In some cases it may not be possible to achieve sufficient
loop bandwidth and phase margin using a simple RC network connected to the VC pin. In these cases, additional
compensation may be required. This is accomplished by the
addition of a feed forward RC network in parallel with the
top resistor of the feedback divider. A small feed forward
capacitor alone may be sufficient in some applications.
A common situation that may require a feed forward network is when the converter is operating in boost mode
and the closed loop crossover frequency (fCC) is close the
Right Half Plane Zero (RHPZ). This may be done in order
to reduce output capacitance requirements by increasing
the loop bandwidth. Due to the phase additions of the
RHPZ, a simple compensator on the VC pin may not be
able to provide sufficient phase boost to stabilize the loop.
Compensation Example
This section will demonstrate how to derive and select the
compensation components for a 5V output supplying 2A
from an input voltage as low as 3V. Designing compensation for most other applications is simply a matter of
substituting in different values to the equations given in
the example and reviewing the resulting Bode Plot, adjusting as needed. Since the compensation design procedure
uses a simplified model of the LTC3119, results should
be checked using time domain step response tests to
validate the effectiveness of the compensation chosen. It
is assumed that values and types for capacitors and the
inductor will be selected based on the guidance given
elsewhere in this data sheet. Particular attention should
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23
LTC3119
Applications Information
be paid to voltage biasing effects on capacitors used for
input and output bypassing. Similarly, it is assumed that
inductor values and current ratings are selected based on
application requirements.
Example Operating Conditions:
VIN = 3V to 15V
Since the converter will be operating in boost mode, the
GCS term must be scaled to represent the commanded
output current.
Looking in the Typical Curves section, we find the efficiency
to be roughly 77%. Using this information, the effective
output current gain can be calculated.
V
GCS(OUT) = GCS • IN • Eff = 4.99 A V
VOUT
VOUT = 5V
ILOAD(MAX) = 2A
COUT = 150µF
Using this information, the gain and phase contributions
from the output filter are calculated.
L = 3.3µH
fSW = 1MHz
First it is necessary to determine the lowest frequency for
fRHPZ. This will determine the maximum bandwidth that
can safely be configured for the converter while operating
in boost mode.
fRHPZ =
VIN 2 • RLOAD
VOUT 2 • 2π • L
= 43.4kHz
In order to ensure sufficient safety margin, a closed loop
crossover frequency (fCC) should be sufficiently below the
RHPZ frequency to account for variability of the internal
components of the IC as well as variability of external influences on the converter response at the cost of possibly
higher loop bandwidth. If sufficient phase margin exists
at the crossover frequency, a higher loop bandwidth may
be realizable while still maintaining stability and good
transient response. In this example, we will use cross
over frequency equal to one sixth of the RHPZ frequency.
fCC =
fRHPZ
@7.24kHz
6
The RHPZ will have a negligible effect on the gain at the
loop crossover, however it will have a phase contribution
that must be considered.
fcc
= tan−1 1 = 9.5o
ϕRHPZ = tan−1
6
fRHPZ
GOUT = GCS(OUT) •
RLOAD 2
= 0.729
fCC 2
fP1 +1
fcc
= tan−1 7240 = 86.5°
ϕP1 = tan−1
424
fP1
Choosing a phase margin of 50 degrees, the required
phase boost from the compensation network is determined
by summing together the phase contributions that were
calculated above. A phase contribution of 90° is assumed
for P2.
ϕZ1 = 50 +ϕP2 +ϕP1 +ϕRHPZ −180 = 56°
The compensation network gain is used to adjust the loop
gain to crossover at the desired frequency. Using the
feedback divider gain and output gain, the compensation
network gain is calculated.
−1
V
REF
GCOMP =
• GOUT = 8.57
VOUT
The compensation network resistor is then found using
the error amplifier transconductance and the required
compensation gain found above.
RZ =
GCOMP 8.57
=
= 71.4kΩ
gm
120µs
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LTC3119
Applications Information
With the value of RZ now known, the compensation
capacitor can be chosen to place the zero Z1 in the correct
location.
CP1 =
tan ( ϕZ1)
= 455pF
2π • fCC • R Z
Selecting standard value components, values of
RZ = 71.5kΩ and CP1 = 470pF are used.
PCB Layout Considerations
The LTC3119 buck-boost converter switches large currents at high frequencies. Special attention should be
paid to the PC board layout to ensure a stable, noise-free
and efficient application circuit. Figures 11 and 12 show
a representative PCB layout for each package option to
outline some of the primary considerations. A few key
guidelines are provided below:
1. The parasitic inductance and resistance of all circulating high current paths should be minimized. This
can be accomplished by keeping the routes to all bold
components in Figures 11 and 12 as short and as wide
as possible. Capacitor ground connections should via
down to the ground plane by way of the shortest route
possible. The bypass capacitors on PVIN, PVOUT and
VCC should be placed as close to the IC as possible
and should have the shortest possible paths to ground.
2. The exposed pad is the electrical ground connection
for the LTC3119. Multiple vias should connect the
back pad directly to the ground plane. In addition,
maximization of the metallization connected to the
back pad will improve the thermal environment and
improve the power handling capabilities of the IC in
both the FE and UFD packages.
3. There should be an uninterrupted ground plane under
the entire converter in order to minimize the crosssectional area of the high frequency current loops.
This minimizes EMI and reduces the inductive drops
in these loops thereby minimizing SW pin overshoot
and ringing.
4. Connections to all of the components shown in bold
should be made as wide as possible to reduce the
series resistance. This will improve efficiency and
maximize the output current capability of the buckboost converter.
5. To prevent large circulating currents in the ground
plane from disrupting operation of the LTC3119, all
small-signal grounds should return directly to GND
by way of a dedicated Kelvin route. This includes the
ground connection for the RT pin resistor, and the
ground connection for the feedback network as shown
in Figures 11 and 12.
6. Keep the routes connecting to the high impedance,
noise sensitive inputs FB and RT as short as possible
to reduce noise pick-up.
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25
LTC3119
Applications Information
N/C
1
28 PWM/SYNC
N/C
2
27 N/C
BST2
3
26 BST1
PGND
4
25 PGND
SW2
5
24 SW1
PVOUT
6
23 PVIN
PVOUT
7
22 PVIN
SW2
8
21 SW1
PGND
9
20 PGND
PGOOD 10
KELVIN
TO VOUT
19 VIN
SVCC 11
FB
12
RTOP
RBOT
18 RUN
RT
17
VC 13
SGND
RT
16 VCC
14
15 MPPC
UNINTERRUPTED GROUND PLANE SHOULD EXIST UNDER ALL COMPONENTS
SHOWN IN BOLD AND UNDER TRACES CONNECTING TO THOSE COMPONENTS
VIA TO GROUND PLANE
(AND TO INNER LAYER WHERE SHOWN)
3119 F11
23 BST1
24 N/C
25 PWM/SYNC
26 N/C
27 N/C
28 BST2
Figure 11. PCB Layout Recommended for the FE Package
PGND
1
22 PGND
PVOUT
4
19 PVIN
SW2
5
18 SW1
PGND
6
17 PGND
PGOOD
7
16 VIN
SVCC
8
15 RUN
SGND
MPPC 12
FB
KELVIN
TO VOUT
VCC 13
RT
14
20 PVIN
11
21 SW1
3
VC 10
2
9
SW2
PVOUT
RT
RBOT
UNINTERRUPTED GROUND PLANE SHOULD EXIST UNDER ALL COMPONENTS
SHOWN IN BOLD AND UNDER TRACES CONNECTING TO THOSE COMPONENTS
VIA TO GROUND PLANE
(AND TO INNER LAYER WHERE SHOWN)
3119 F12
Figure 12. PCB Layout Recommended for the UFD Package
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LTC3119
Typical Applications
3.3V, 400kHz Wide Input Regulator
3.3µH
0.1µF
0.1µF
SW1
BST1
VIN
2.5V TO 18V
10µF
SW2
BST2
PVIN
VIN
100k
220µF
LTC3119
RUN
PWM/SYNC
MPPC
VCC
SVCC
RT
22pF
316k
FB
PGOOD
PGND GND
100k
VC
39.2k
3119 TA03
4.7µF
VOUT
3.3V
PVOUT
200k
560pF
VIN = 2.5; IOUT = 2.4A; EFFICIENCY = 78%
VIN = 5.0; IOUT = 5A; EFFICIENCY = 84%
3.3V, 750kHz Wide Input Regulator
3.3µH
0.1µF
0.1µF
SW1
BST1
VIN
2.5V TO 18V
10µF
100k
SW2
BST2
PVIN
VIN
PVOUT
330µF
LTC3119
RUN
PWM/SYNC
MPPC
VCC
SVCC
RT
316k
FB
PGOOD
PGND GND
4.7µF
100k
VC
3119 TA04
105k
VOUT
3.3V
80.4k
560pF
VIN = 2.5; IOUT = 2A; EFFICIENCY = 78%
VIN = 5.0; IOUT = 5A; EFFICIENCY = 82%
3119fb
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27
LTC3119
Typical Applications
3.3V, 1.5MHz Wide Input Regulator
2.2µH
0.1µF
0.1µF
SW1
BST1
VIN
2.5V TO 18V
10µF
100k
SW2
BST2
PVIN
VIN
PVOUT
150µF
LTC3119
RUN
316k
PWM/SYNC
MPPC
VCC
SVCC
RT
FB
PGOOD
PGND GND
100k
VC
3119 TA05
4.7µF
VOUT
3.3V
52.3k
48.7k
470pF
VIN = 2.5; IOUT = 1.75A; EFFICIENCY = 73%
VIN = 5.0; IOUT = 5A; EFFICIENCY = 80%
3.3V, 500kHz Wide Input Regulator
4.7µH
0.1µF
VIN
1.3V TO 18V
STARTS AT 2.5V
0.1µF
SW1
BST1
10µF
100k
SW2
BST2
PVIN
VIN
PVOUT
220µF
LTC3119
RUN
PWM/SYNC
MPPC
VCC
SVCC
RT
4.7µF
FB
PGOOD
PGND GND
VC
3119 TA06
162k
316k
VOUT
3.3V AT 5A, VIN > 4V
3.3V AT 1A, VIN = 1.6V
100k
78.7k
820pF
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LTC3119
Typical Applications
5V, 500kHz Wide Input Regulator
4.7µH
0.1µF
0.1µF
SW1
BST1
VIN
2.5V TO 18V
10µF
SW2
BST2
PVIN
VIN
100k
220µF
LTC3119
RUN
536k
PWM/SYNC
MPPC
VCC
SVCC
RT
FB
PGOOD
PGND GND
102k
VC
3119 TA07
4.7µF
VOUT
5.0V
PVOUT
100k
162k
560pF
VIN = 2.5; IOUT = 1.5A; EFFICIENCY = 78%
VIN = 6.0; IOUT = 5A; EFFICIENCY = 88%
5V, 1MHz Wide Input Regulator
2.2µH
0.1µF
0.1µF
SW1
BST1
VIN
2.5V TO 18V
10µF
100k
SW2
BST2
PVIN
VIN
RUN
PVOUT
150µF
LTC3119
VOUT
5.0V
536k
PWM/SYNC
MPPC
VCC
SVCC
RT
FB
PGOOD
PGND GND
3119 TA08
4.7µF
76.8k
102k
VC
127k
330pF
VIN =2.5; IOUT = 1.2A; EFFICIENCY = 73%
VIN =6.0; IOUT = 5A; EFFICIENCY = 83%
3119fb
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29
LTC3119
Typical Applications
5V, 2MHz Wide Input Regulator
12V to 12V, 1MHz Line Conditioner with
8.5V Undervoltage Lockout Threshold
1.5µH
2.2µH
0.1µF
VIN
2.5V TO 18V
10µF
100k
SW1
BST1
SW2
BST2
PVIN
VIN
PVOUT
100µF
LTC3119
RUN
PWM/SYNC
MPPC
VCC
SVCC
RT
0.1µF
VOUT
5.0V
VIN
9V TO 15V
RUN
10µF
MPPC
VCC
SVCC
RT
PGOOD
PGND GND
102k
34.8k
124k
4.7µF
270pF
SW1
BST1
8.0
SW2
BST2
PVIN
VIN
47µF
VOUT
12V
1370k
FB
PGOOD
GND
VC
3119 TA11
34.8k
FB
PGOOD
PGND GND
97.6k
VC
100k
180pF
97.6k
155k
350pF
PULSED LOAD
CONTINUOUS LOAD
7.0
0.1µF
PVOUT
LTC3119
RUN
PWM/SYNC
MPPC
VCC
SVCC
RT
PGND
4.7µF
1370k
Maximum Output Current
VOUT = 12V
OUTPUT CURRENT (A)
100k
VOUT
12.0V
VIN = 9; IOUT =3A; EFFICIENCY = 92.5%
VIN = 13; IOUT =5A; EFFICIENCY = 93.5%
ENABLE AT VIN > 8.5V
1.5µH
10µF
47µF
LTC3119
76.8k
12V, 2MHz Wide Input Regulator
VIN
2.5V TO 18V
PVOUT
3119 TA10
VIN = 2.5; IOUT = 1A; EFFICIENCY = 70%
VIN = 6.0; IOUT = 4.5A; EFFICIENCY = 80%
0.1µF
0.1µF
SW2
BST2
PWM/SYNC
200k
FB
VC
SW1
BST1
PVIN
VIN
1210k
536k
3119 TA09
4.7µF
0.1µF
6.0
5.0
4.0
3.0
2.0
DC2129A DEMO BOARD
fSW = 2MHz
100Hz PULSE LOAD
20% DUTY CYCLE
1.0
0
2
4
6
8
10 12 14
INPUT VOLTAGE (V)
16
18
3119 TA11b
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LTC3119
Typical Applications
12V, 750kHz Regulator with Input Supply Rundown
3.3µH
4.7µF
0.1µF
0.1µF
SW1
BST1
PVIN
VIN
SW2
BST2
VIN
4.22M
1.87M
4700μF
FB
MPPC
VCC
PGOOD
SVCC
VC
RT
PGND GND
VIN (ENABLE) = 5V
VIN (DISABLE) = 0V
80k
3119 TA12a
4.7µF
105k
1370k
97.6k
2200pF
Rundown Behavior After Input
Disconnect
Output Holdup Time vs
Input Bulk Capacitor Size
150
ILOAD = 300mA
ILOAD = 600mA
125
HOLDUP TIME (ms)
INDUCTOR
CURRENT
1.0A/DIV
47µF
LTC3119
VOUT
12V
PWM/SYNC
604k
4.7µF
RUN
PVOUT
VOUT
10V/DIV
VIN
10V/DIV
INPUT SUPPLY REMOVED
20ms/DIV
3119 TA12b
100
75
50
25
2
4
6
8
10
INPUT BULK CAPACITANCE (mF)
12
3119 TA12c
3119fb
For more information www.linear.com/LTC3119
31
LTC3119
Typical Applications
Selectable 12V or 3.3V Output, 1MHz Regulator
3.3µH
0.1µF
VIN
5V TO 18V
10µF
100k
SW1
BST1
PVIN
VIN
RUN
BURST PWM
0.1µF
VOUT SELECTABLE
VOUT = 3.3V/12V, IOUT = 2A
PVOUT
VIN
100µF
LTC3119
1400k
PWM/SYNC
MPPC
VCC
SVCC
RT
4.7µF
SW2
BST2
FB
PGOOD
PGND GND
442k
130k
VC
60k
GPIO
76.8k
1200pF
VOUT SELECT
12V
3.3V
741G05
500k
GPIO
3119 TA13a
Output Voltage Transition
3.3V to 12V
VOUT = 3.3V/12V
fSW = 1MHz
VOUT SELECT
5V/DIV
VOUT
10V/DIV
PGOOD
10V/DIV
200µs/DIV
3119 TA13b
3119fb
32
For more information www.linear.com/LTC3119
LTC3119
Package Description
Please refer to http://www.linear.com/product/LTC3119#packaging for the most recent package drawings.
UFD Package
28-Lead Plastic QFN (4mm × 5mm)
(Reference LTC DWG # 05-08-1712 Rev C)
0.70 ±0.05
4.50 ±0.05
3.10 ±0.05
2.50 REF
2.65 ±0.05
3.65 ±0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
3.50 REF
4.10 ±0.05
5.50 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ±0.10
(2 SIDES)
0.75 ±0.05
R = 0.05
TYP
PIN 1 NOTCH
R = 0.20 OR 0.35
× 45° CHAMFER
2.50 REF
R = 0.115
TYP
27
28
0.40 ±0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
5.00 ±0.10
(2 SIDES)
3.50 REF
3.65 ±0.10
2.65 ±0.10
(UFD28) QFN 0816 REV C
0.200 REF
0.00 – 0.05
0.25 ±0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGHD-3).
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
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33
LTC3119
Package Description
Please refer to http://www.linear.com/product/LTC3119#packaging for the most recent package drawings.
FE Package
28-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663 Rev K)
Exposed Pad Variation EA
9.60 – 9.80*
(.378 – .386)
7.56
(.298)
7.56
(.298)
28 2726 25 24 23 22 21 20 19 18 1716 15
6.60 ±0.10
4.50 ±0.10
3.05
(.120)
SEE NOTE 4
0.45 ±0.05
EXPOSED
PAD HEAT SINK
ON BOTTOM OF
PACKAGE
6.40
3.05 (.252)
(.120) BSC
1.05 ±0.10
0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.25
REF
1.20
(.047)
MAX
0° – 8°
0.65
(.0256)
BSC
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1 2 3 4 5 6 7 8 9 10 11 12 13 14
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE28 (EA) TSSOP REV K 0913
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3119fb
34
For more information www.linear.com/LTC3119
LTC3119
Revision History
REV
DATE
DESCRIPTION
A
02/17
Added Maximum Output Current Curves
PAGE NUMBER
B
03/17
Modified Schematic
10, 11, 30
32, 36
Modified BST2 pin description
12
Output Capacitor Selection, changed switching frequency to VCC voltage
20
Modified Voltage Error Amplifier Gain equation
23
3119fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LTC3119
35
LTC3119
Typical Application
Photovoltaic Panel Input Lead-Acid Charger with Temperature Correction (1MHz)
4.7µH
0.1µF
SW1
BST1
VIN
220µF
1020k
SW2
BST2
PVIN
VIN
100k
RUN
PHOTOVOLTAIC
PANEL
V(MPP) = 9V
PVOUT
10µF
LTC3119
PWM/SYNC
4.7nF
MPPC
VCC
SVCC
RT
100k
7.5k
PGOOD
PGND GND
6-CELL
LEAD-CELL
BATTERY
2000k
NTC
68k
FB
220k
TEMPERATURE
COMPENSATION
VC
3119 TA02
4.7µF
0.1µF
52.3k
76.8k
127k
5.6nF
Related Parts
PART NUMBER DESCRIPTION
COMMENTS
LTC3113
5V, 3A Synchronous Buck-Boost
VIN = 1.8V to 5.5V, VOUT = 1.8V to 5.25V, IQ = 30μA, ISD <
1μA, DFN and TSSOP Packages
LTC3129/
LTC3129-1
15V, 200mA Synchronous Buck-Boost with 1.3µA IQ
VIN = 2.42V to 15V, VOUT = 2.5V to 14V, IQ = 1.3μA, ISD =
10nA, QFN and MSOP Packages
LTC3112
15V, 2.5A Synchronous Buck-Boost
VIN = 2.7V to 15V, VOUT = 2.5V to 14V, IQ = 40μA, ISD < 1μA,
DFN and TSSOP Packages
LTC3118
18V, 2A Dual Input PowerPath™ Buck-Boost Converter
VIN = 2.2V to 18V, VOUT = 2V to 14V, IQ = 50μA, ISD < 2μA,
QFN and TSSOP Packages
LTC3130/
LTC3130-1
25V, 600mA Synchronous Buck-Boost Converter
VIN = 2.4V to 25V, VOUT = 1V to 25V, IQ = 1.2µA, ISD = 500nA
LTC3114-1
40V, 1A Synchronous Buck-Boost
VIN = 2.2V to 40V, VOUT = 2.7V to 15V, IQ = 30µA, ISD < 3µA,
DFN and TSSOP Packages
LTC3115-1/
LTC3115-2
40V, 2A Synchronous Buck-Boost
VIN = 2.7V to 40V, VOUT = 2.7V to 40V, IQ = 30μA, ISD < 3μA,
DFN and TSSOP Packages
LTC3785
10V, High Efficiency, Synchronous, No RSENSE™ Buck-Boost Controller
VIN = 2.7V to 10V, VOUT = 2.7V to 10V, IQ = 86μA, ISD < 15μA,
QFN Package
LTC3789
38V, High Efficiency, Synchronous, 4-Switch Buck-Boost Controller
VIN = 4V to 38V, VOUT = 0.8V to 38V, IQ = 3mA, ISD < 60µA,
SSOP-28, QFN-28 Packages
LT3790
60V, Synchronous, 4-Switch Buck-Boost Controller
VIN = 4.7V to 60V, VOUT = 1.2V to 60V, IQ = 3mA, ISD < 1µA,
TSSOP Package
QFN and MSOP Packages
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36
LT 0317 REV B • PRINTED IN USA
For more information www.linear.com/LTC3119
www.linear.com/LTC3119
LINEAR TECHNOLOGY CORPORATION 2016
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