LTC3546IFE#PBF

LTC3546 Dual Synchronous, 3A/1A or 2A/2A Configurable Step-Down DC/DC Regulator FEATURES DESCRIPTION VIN Range: 2.3V to 5.5V VOUT Range: 0.6V to 5V n Programmable Frequency Operation; 2.25MHz, or Adjustable Between 0.75MHz to 4MHz n Low R DS(ON) Internal Switches n High Efficiency: Up to 96% n No Schottky Diodes Required n Short-Circuit Protected n Current Mode Operation for Excellent Line and Load Transient Response n Low Ripple Burst Mode Operation (30mV P-P), IQ = 160µA n Ultralow Shutdown Current: I < 1µA Q n Low Dropout Operation: 100% Duty Cycle n Power Good Output For Each Channel n Externally or Internally Programmable Burst Level n External or Internal Soft-Start or Supply Tracking n Available in Thermally Enhanced 28-Lead (4mm × 5mm) QFN and TSSOP Packages The LTC®3546 is a dual, constant-frequency, synchronous step down DC/DC converter for medium power applications. The design consists of 2A and 1A primary output switches. In addition to the 2A/1A capability, a 1A dependant output switch can be externally connected to either of the primary outputs to produce 3A/1A dual regulator or 2A/2A dual regulator configurations. n n APPLICATIONS n n n n Supply operation is from 2.3V to 5.5V. The switching frequency can be set to 2.25MHz, adjustable from 0.75MHz to 4MHz, or synchronized to an external clock. Each output is adjustable from 0.6V to 5V and has output tracking on power-up. Internal synchronous low RDS(ON) power switches provide high efficiency without external Schottky diodes. User-selectable modes (Burst Mode® operation, pulse skipping and forced continuous) allow a trade-off between ripple noise and power efficiency. Burst Mode operation provides high efficiency at light loads. Pulse-skipping mode provides low ripple noise at light loads. The device is capable of low dropout configurations and both channels can operate at 100% duty cycle. In shutdown, the device draws (VIN – 0.5V), Burst Mode operation is selected for both regulators. When the voltage on the SYNC/MODE pin is 0.63V). The Burst Mode peak inductor current can be set externally via the BMC pin. When this pin is set somewhere between 0V to 0.6V, the voltage on this pin controls the Burst Mode clamp level. When the BMC pin is pulled to VIN, an internal Burst Mode clamp level is used. The SYNC/MODE pin selects what mode the LTC3546 is in. The SYNC/MODE pin sets the mode for both regulators. For lower output voltage ripple at low currents, pulseskipping mode can be used. In this mode, the LTC3546 continues to switch at constant frequency down to very low currents, where it will eventually begin skipping pulses. Finally, in forced continuous mode, the inductor current is constantly cycled which creates a fixed output voltage ripple at all output current levels. This feature is desirable in telecommunications since the noise is a constant frequency and is thus easy to filter out. Another advantage of this mode is that the regulator is capable of both sourcing current into a load and sinking some current from the output. In forced 14 Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases to 100% which is the dropout condition. In the dropout condition, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFETs and inductors. Low Supply Operation The LTC3546 incorporates an undervoltage lockout circuit which shuts down the part when the input voltage drops below about 2.14V to prevent unstable operation. 3546fd For more information www.linear.com/3546 LTC3546 APPLICATIONS INFORMATION A general LTC3546 application circuit is shown in Figure  7. External component selection is driven by the load requirement, and begins with the selection of the inductors L1, and L2. Once L1 and L2 are chosen, CIN, COUT1, and COUT2 can be selected. Assuming a worst-case minimum on-time of 150ns, this can be calculated as: Operating Frequency The minimum frequency is limited by leakage and noise coupling due to the large resistance of RT. Selection of the operating frequency is a trade-off between efficiency and component size. High frequency operation allows the use of smaller inductor and capacitor values. Operation at lower frequencies improves efficiency by reducing internal gate charge losses but requires larger inductance values and/or capacitance to maintain low output ripple voltage. The operating frequency, fO, of the LTC3546 is determined by pulling the FREQ pin to VIN, for 2.25MHz operation, by connecting an external resistor from FREQ to ground, or by driving an external clock signal into SYNC/MODE. When using an external resistor to set the oscillator frequency use the following equation: 2.51•1011 RT = (Ω) – 20kΩ fO for 0.75MHz ≤ fO ≤ 4MHz. Or use Figure 1 to select the value for RT. The maximum operating frequency is also constrained by the minimum on-time (typically 70ns) and duty cycle, especially when forced continuous mode is selected. ⎛ V ⎞ fO(MAX) ≈ 6.67 ⎜ OUT ⎟ (MHz ) ⎝ VIN(MAX) ⎠ Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current ΔIL decreases with higher inductance and increases with higher VIN or VOUT. ΔIL = VOUT ⎛ VOUT ⎞ 1− fO •L ⎜⎝ VIN ⎟⎠ Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is ΔIL = 0.35ILOAD(MAX), where ILOAD(MAX) is the maximum output current. The largest ripple ΔIL occurs at the maximum input voltage. To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation: L≥ ⎞ VOUT ⎛ V 1− OUT ⎟ ⎜ fO • ΔIL ⎝ VIN(MAX) ⎠ 500 450 400 RT (kΩ) 350 300 250 200 150 100 50 0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 FREQUENCY (MHz) 3546 F01 Figure 1. Frequency vs RT 3546fd For more information www.linear.com/3546 15 LTC3546 APPLICATIONS INFORMATION Burst Mode Operation Considerations Inductor Core Selection There are two factors that determine the load current at which the LTC3546 enters Burst Mode operation: the inductor value and the BMC pin voltage. The transition from low current operation begins when the peak inductor current falls below a level set by the burst clamp. Lower inductor values result in higher ripple current which causes Burst Mode operation to occur at lower load currents. Lower inductor values will also cause a dip in efficiency in the upper range of low current operation. Lower inductor values will also cause the burst frequency to increase in Burst Mode operation. Different core materials and shapes will change the size/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements of any radiated field/EMI requirements than on what the LTC3546 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3546 applications. The burst clamp level can be set by the voltage on the BMC pin. If BMC is tied to VIN, an internally set level is used. A BMC pin voltage between 0V and 0.6V will set the burst clamp level (see charts OUT1 Minimum Peak Current vs VBMC1 and OUT2 Minimum Peak Current vs VBMC2 in the Typical Performance Characteristics section). Generally, a higher clamp level results in improved light load efficiency and higher output voltage ripple, while a lower clamp level results in small output voltage ripple at the expense of efficiency. The BMC pin should be connected to ground when Burst Mode operation is not selected. Input Capacitor (CIN) Selection In continuous mode, the input current of the converter can be approximated by the sum of two square waves with duty cycles of approximately VOUT1/VIN and VOUT2/VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. Some capacitors have a derating spec for maximum RMS current. If the capacitor being used has this requirement it is necessary to calculate Table 1. MANUFACTURER PART NUMBER VALUE (µH) MAX DC CURRENT (A) DCR DIMENSIONS L × W × H (mm) Würth Elektronik WE-PD2 MS 7447745012 1.2 4.6 0.017 5.2 × 5.8 × 2 Würth Elektronik WE-PD2 MS 74477450056 0.56 6.5 0.0078 5.2 × 5.8 × 2 Vishay IHLP-1616AB-11 1.2 3.75 0.068 4.06 × 4.45 × 1.20 Vishay IHLP-1616AB-11 0.47 6 0.019 4.06 × 4.45 × 1.20 Coilcraft LPS6225-122 1.2 5.4 0.04 6.2 × 6.2 × 2.5 Coilcraft DO1813H-561 0.56 7.7 0.01 6.10 × 8.89 × 5.00 SD20-1R2 1.2 2.55 0.0275 5.2 × 5.2 × 2 SD20-R47 0.47 4 0.02 5.2 × 5.2 × 2 CDRH3D23NP-1R5NC 1 2.8 0.025 3.8 × 3.8 × 2.3 Coiltronics Coiltronics Sumida 16 3546fd For more information www.linear.com/3546 LTC3546 APPLICATIONS INFORMATION the maximum RMS current. The RMS current calculation is different if the part is used in-phase or out-of-phase. For in-phase, there are two different equations: VOUT1 > VOUT2: IRMS = ( ) ( ) ( ) ( ) 2 •I1•I2 •D2 (1–D1) +I22 D2 –D22 +I12 D1–D12 VOUT2 > VOUT1: IRMS = 2 •I1•I2 •D1(1–D2) +I22 D2 –D22 +I12 D1–D12 Where: Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1µF to 1µF ceramic capacitor is also recommended on VIN for high frequency decoupling, when not using an all ceramic capacitor solution. Output Capacitor (COUT1 and COUT2) Selection V V D1= OUT1 and D2 = OUT2 VIN VIN When D1 = D2, then the equation simplifies to: when VOUT1 – VIN/2 = VOUT2 and when VOUT2 – VIN/2 = VOUT1. As a good rule of thumb, the amount of worst-case ripple is about 75% of the worst-case ripple in the in-phase mode. Note, that when VOUT1 = VOUT2 = VIN/2 and I1 = I2, the ripple is at its minimum. IRMS = (I1+I2) D(1–D) The selection of COUT1 and COUT2 is driven by the required ESR to minimize voltage ripple and load step transients. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (ΔVOUT) is determined by: or IRMS = (I1+I2) VOUT ( VIN – VOUT ) VIN where the maximum average output currents I1 and I2 equals the peak current minus half the peak-to-peak ripple current: ΔI I1=ILIM1 – L1 2 ΔI I2 =ILIM2 – L2 2 These formula have a maximum at VIN = 2VOUT, where IRMS = (I1 + I2)/2. This simple worst-case is commonly used to determine the worst-case IRMS. For out-of-phase (PHASE pin is at ground), the ripple current can be lower than the in-phase. In the out-of-phase case, the maximum IRMS does not occur when VOUT1 = VOUT2. The maximum typically occurs ⎛ ⎞ 1 ΔVOUT ≈ ΔIL ⎜ ESR+ 8• fO •COUT ⎟⎠ ⎝ where fO = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Once the ESR requirements for COUT have been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement, except for an all ceramic solution. In surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolytic, special polymer, ceramic and dry tantalum capacitors are all available in surface mount packages. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Special polymer capacitors, such as Sanyo POSCAP, offer very low ESR, but have a lower capacitance density than other types. Tantalum capacitors have the highest capacitance 3546fd For more information www.linear.com/3546 17 LTC3546 APPLICATIONS INFORMATION density, but it has a larger ESR and it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors have a significantly larger ESR, and are often used in extremely cost-sensitive applications provided that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have the lowest ESR and cost but also have the lowest capacitance density, high voltage and temperature coefficient and exhibit audible piezoelectric effects. In addition, the high Q of ceramic capacitors along with trace inductance can lead to significant ringing. Other capacitor types include the Panasonic specialty polymer (SP) capacitors. Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Because the LTC3546 control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Great care must be taken when using only ceramic input and output capacitors. When a ceramic capacitor is used at the input and the power is being supplied through long wires, such as from a wall adapter, a load step at the output can induce ringing at the VIN pin. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, the ringing at the input can be large enough to disrupt circuit operation or damage the part. to support the load. The time required for the feedback loop to respond is dependent on the compensation components and the output capacitor size. Typically, 3 to 4 cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP, is usually about 2 to 3 times the linear droop of the first cycle. Thus, a good place to start is with the output capacitor size of approximately: COUT ≈ 2.5 More capacitance may be required depending on the duty cycle and load step requirements. In most applications, the input capacitor is merely required to supply high frequency bypassing, since impedance to the supply is very low. A 10µF ceramic capacitor is usually enough for these conditions. Setting the Output Voltage The LTC3546 generates a 0.6V reference voltage between the feedback pin, VFB1 and VFB2, and the signal ground. The output voltage is set by a resistive divider according to the following formula: ⎛ R1⎞ VOUT1 ≈ 0.6V ⎜ 1+ ⎟ ⎝ R2 ⎠ ⎛ R3 ⎞ VOUT2 ≈ 0.6V ⎜ 1+ ⎟ ⎝ R4 ⎠ Resistor locations are shown in Figure 2. VOUT1 CFF1 Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough 18 ΔIOUT fOVDROOP VOUT2 R1 R3 CFF2 LTC3546 VFB1 R2 VFB2 3546 F02 R4 Figure 2. Setting Output Voltages 3546fd For more information www.linear.com/3546 LTC3546 APPLICATIONS INFORMATION Keeping the current small (