LTC3606B
800mA Synchronous
Step-Down DC/DC with
Average Input Current Limit
DESCRIPTION
FEATURES
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Programmable Average Input Current Limit:
±5% Accuracy
Step-Down Output: Up to 96% Efficiency
Low Noise Pulse-Skipping Operation at Light Loads
Input Voltage Range: 2.5V to 5.5V
Output Voltage Range: 0.6V to 5V
2.25MHz Constant-Frequency Operation
Power Good Output Voltage Monitor
Low Dropout Operation: 100% Duty Cycle
Internal Soft-Start
Current Mode Operation for Excellent Line and Load
Transient Response
±2% Output Voltage Accuracy
Short-Circuit Protected
Shutdown Current ≤ 1μA
Available in Small Thermally Enhanced 8-Lead
3mm × 3mm DFN Package
APPLICATIONS
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The LTC®3606B is an 800mA monolithic synchronous
buck regulator using a constant frequency current mode
architecture.
The input supply voltage range is 2.5V to 5.5V, making it
ideal for Li-Ion and USB powered applications. 100% duty
cycle capability provides low dropout operation, extending
the run time in battery-operated systems. Low output
voltages are supported with the 0.6V feedback reference
voltage. The LTC3606B can supply 800mA output current.
The LTC3606B’s programmable average input current limit
is ideal for USB applications and for point-of-load power
supplies because the LTC3606B’s limited input current
will still allow its output to deliver high peak load currents without collapsing the input supply. The operating
frequency is internally set at 2.25MHz allowing the use of
small surface mount inductors. Internal soft-start reduces
in-rush current during start-up. The LTC3606B is available
in an 8-Lead 3mm × 3mm DFN package.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815,
6304066, 6498466, 6580258, 6611131.
High Peak Load Current Applications
USB Powered Devices
Supercapacitor Charging
Radio Transmitters and Other Handheld Devices
TYPICAL APPLICATION
Monolithic Buck Regulator with Input Current Limit
1.5μH
VIN 3.4V
TO 5.5V
VIN
CIN
10μF
RUN
+
PGOOD
VFB
RLIM
PGOOD
GND
1000pF
VOUT
3.4V AT
800mA
SW
LTC3606B
499k
116k
1210k
GSM Pulse Load
2.2mF
s2
SuperCap
VIN
AC-COUPLED
1V/DIV
IOUT
500mA/DIV
255k
3606B TA01
ILIM = 475mA
VOUT
200mV/DIV
IIN
500mA/DIV
1ms/DIV
3606B TA01b
VIN = 5V, 500mA COMPLIANT
ILOAD = 0A to 2.2A
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LTC3606B
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Input Supply Voltage (VIN) ........................... –0.3V to 6V
VFB ................................................... –0.3V to VIN + 0.3V
RUN, RLIM ....................................... –0.3V to VIN + 0.3V
SW ................................................... –0.3V to VIN + 0.3V
PGOOD............................................. –0.3V to VIN + 0.3V
P-Channel SW Source Current (DC) (Note 2) ..............1A
N-Channel SW Source Current (DC) (Note 2)..............1A
Peak SW Source and Sink Current (Note 2) ............. 2.7A
Operating Junction Temperature Range
(Notes 3, 6, 8) ........................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 125°C
Reflow Peak Body Temperature ............................ 260°C
GND
1
RLIM
2
GND
3
SW
4
8 VFB
9
GND
7 RUN
6 PGOOD
5 VIN
DD PACKAGE
8-LEAD (3mm s 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 40°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3606BEDD#PBF
LTC3606BEDD#TRPBF
LFMB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3606BIDD#PBF
LTC3606BIDD#TRPBF
LFMB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3606B
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, unless otherwise noted.
SYMBOL
PARAMETER
VIN
VIN Operating Voltage Range
CONDITIONS
MIN
VUV
VIN Undervoltage Lockout
IFB
Feedback Pin Input Current
VFBREG
Feedback Voltage
LTC3606BE, –40°C < TJ < 85°C (Note 7)
LTC3606BI, –40°C < TJ < 125°C (Note 7)
VLINEREG
VFB Line Regulation
VLOADREG
l
TYP
MAX
UNITS
5.5
V
2.5
V
±30
nA
0.600
0.600
0.612
0.618
V
V
VIN = 2.5V to 5.5V (Note 7)
0.01
0.25
%/V
VFB Load Regulation
ILOAD = 0mA to 800mA (Note 7)
0.5
Supply Current
Active Mode (Note 4)
Shutdown
VFB = 0.95 × VFBREG
VRUN = 0V, VIN = 5.5V
420
650
1
μA
μA
fOSC
Oscillator Frequency
VFB = VFBREG
1.8
2.25
2.7
MHz
ILIM(PEAK)
Peak Switch Current Limit
VIN = 5V, VFB < VFBREG , Duty Cycle 100pF at the RLIM pin.
Each application input current limit will call for different
CLIM value to optimize its response time. Using a large CLIM
capacitor requires longer time for the RLIM pin voltage to
charge. For example, consider the application 500mA input
current limit, 5V input and 1A, 2.5V output with a 50% duty
cycle. When an instantaneous 1A output pulse is applied,
the current out of the RLIM pin becomes 1A/55k = 18.2μA
during the 50% on-time or 9.1μA full duty cycle. With a
CLIM capacitor of 1μF, RLIM of 116k, and using I = CdV/dt,
it will take 110ms for CLIM to charge from 0V to 1V. This is
the time after which the LTC3606B will start input current
limiting. Any current within this time must be considered
in each application to determine if it is tolerable.
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LTC3606B
OPERATION
Figure 1a shows VIN (IIN) current below input current
limit with a CLIM capacitor of 0.1μF. When the load pulse
is applied, under the specified condition, ILIM current is
1.1A/55k • 0.66 = 13.2μA, where 0.66 is the duty cycle.
It will take a little more than 7.5ms to charge the CLIM
capacitor from 0V to 1V, after which the LTC3606B begins
to limit input current. The IIN current is not limited during
this 7.5ms time and is more than 725mA. This current
transient may cause the input supply to temporarily
droop if the supply current compliance is exceeded, but
recovers after the input current limit engages. The output
will continue to deliver the required current load while the
output voltage droops to allow the input voltage to remain
regulated during input current limit.
and the output must deliver the required current load.
This may cause the input voltage to droop if the current
compliance is exceeded. Depending on how long this time
is, the VIN supply decoupling capacitor can provide some
of this current before VIN droops too much. In applications
with a bigger VIN supply decoupling capacitor and where
VIN supply is allow to droop closer to dropout, the CLIM
capacitor can be increased slightly. This will delay the
start of input current limit and artificially regulated VOUT
before input current limit is engaged. In this case, within
the 577μs load pulse, the VOUT voltage will stay artificially
regulated for 92μs out of the total 577μs before the input
current limit activates. This approach may be used if a
faster recovery on the output is desired.
For applications with short load pulse duration, a smaller
CLIM capacitor may be the better choice as in the example
shown in Figure 1b. In this example, a 577μs, 0A to 2A
output pulse is applied once every 4.7ms. A CLIM capacitor
of 2.2nF requires 92μs for VRLIM to charge from 0V to 1V.
During this 92μs, the input current limit is not yet engaged
Selecting a very small CLIM will speed up response time
but it can put the device within threshold of interfering
with normal operation and input current limit in every
few switching cycles. This may be undesirable in terms
of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as
a starting point, R being RLIM, C being CLIM.
VOUT
2V/DIV
VOUT
200mV/DIV
IIN
500mA/DIV
VIN
AC-COUPLED
1V/DIV
VRLIM
1V/DIV
IOUT
500mA/DIV
IIN
500mA/DIV
IL
1A/DIV
50ms/DIV
3606B F01a
VIN = 5V, 500mA COMPLIANT
RLIM = 116k, CLIM = 0.1μF
ILOAD = 0A to 1.1A, COUT = 2.2mF, VOUT = 3.3V
ILIM = 475mA
Figure 1a. Input Current Limit Within 100ms Load Pulses
1ms/DIV
3606B F01b
VIN = 5V, 500mA COMPLIANT
RLIM = 116k, CLIM = 2200pF
ILOAD = 0A to 2A, COUT = 2.2mF, VOUT = 3.3V
ILIM = 475mA
Figure 1b. Input Current Limit Within
577μs, 2A Repeating Load Pulses
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LTC3606B
APPLICATIONS INFORMATION
A general LTC3606B application circuit is shown in Figure 2.
External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once
the inductor is chosen, CIN and COUT can be selected.
Inductor Selection
Although the inductor does not influence the operating frequency, the inductor value has a direct effect on
ripple current. The inductor ripple current IL decreases
with higher inductance and increases with higher VIN or
VOUT :
V
V
IL = OUT • 1 OUT
(1)
fO • L
VIN
Accepting larger values of IL allows the use of low
inductances, but results in higher output voltage ripple,
greater core losses, and lower output current capability.
A reasonable starting point for setting ripple current is
40% of the maximum output load current. So, for a 800mA
regulator, IL = 320mA (40% of 800mA).
The inductor value will also have an effect on Burst Mode
operation. The transition to low current operation begins
when the peak inductor current falls below a level set by
the internal burst clamp. Lower inductor values result in
higher ripple current which causes the transition to occur
at lower load currents. This causes a dip in efficiency in
the upper range of low current operation. Furthermore,
lower inductance values will cause the bursts to occur
with increased frequency.
L1
VIN
2.5V TO 5.5V
VIN
RPGD
CIN
VOUT
SW
LTC3606B
CF
RUN
COUT
PGOOD
VFB
RLIM
PGOOD
GND
RLIM
R2
R1
CLIM
3606B F02
Figure 2. LTC3606B General Schematic
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and do not radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. The choice of which style
inductor to use often depends more on the price versus
size requirements, and any radiated field/EMI requirements,
than on what the LTC3606B requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3606B applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER
PART NUMBER
MAX DC
VALUE CURRENT
DCR
HEIGHT
Coilcraft
LPS4012-152ML
LPS4012-222ML
LPS4012-332ML
LPS4012-472ML
LPS4018-222ML
LPS4018-332ML
LPS4018-472ML
1.5μH
2.2μH
3.3μH
4.7μH
2.2μH
3.3μH
4.7μH
2200mA
1750mA
1450mA
1450mA
2300mA
2000mA
1800mA
0.070Ω
0.100Ω
0.100Ω
0.170Ω
0.070Ω
0.080Ω
0.125Ω
1.2mm
1.2mm
1.2mm
1.2mm
1.8mm
1.8mm
1.8mm
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7μH
3.3μH
2.2μH
1100mA
1200mA
1300mA
0.11Ω
0.1Ω
0.08Ω
1mm
1mm
1mm
LQH32CN4R7M23 4.7μH
450mA
0.2Ω
2mm
ELT5KT4R7M
4.7μH
950mA
0.2Ω
1.2mm
CDRH2D18/LD
CDH38D11SNP3R3M
CDH38D11SNP2R2M
4.7μH
3.3μH
630mA
1560mA
0.086Ω
0.115Ω
2mm
1.2mm
2.2μH
1900mA
0.082Ω
1.2mm
2.2μH
2.2μH
3.3μH
2.2μH
4.7μH
510mA
530mA
410mA
1100mA
750mA
0.13Ω
0.33Ω
0.27Ω
0.1Ω
0.19Ω
1.6mm
1.25mm
1.6mm
1mm
1mm
4.7μH
700mA
0.28Ω
1mm
3.3μH
870mA
0.17Ω
1mm
2.2μH
1000mA
0.12Ω
1mm
2.2μH
1500mA
0.076Ω
1.2mm
3.3μH
1700mA
0.095Ω
1.2mm
2.2μH
2300mA
0.059Ω
1.4mm
Murata
Panasonic
Sumida
Taiyo Yuden CB2016T2R2M
CB2012T2R2M
CB2016T3R3M
NR30102R2M
NR30104R7M
TDK
VLF3010AT4R7MR70
VLF3010AT3R3MR87
VLF3010AT2R2M1R0
VLF4012AT-2R2
M1R5
VLF5012ST-3R3
M1R7
VLF5014ST-2R2
M2R3
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LTC3606B
APPLICATIONS INFORMATION
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is a
square wave with a duty cycle of approximately VOUT / VIN .
To prevent large voltage transients, a low equivalent series
resistance (ESR) input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor
current is given by:
IRMS IMAX
VOUT (VIN VOUT )
VIN
Where the maximum average output current IMAX equals
the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – IL /2. This formula has a maximum at
VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case
is commonly used to design because even significant
deviations do not offer much relief. Note that capacitor
manufacturer’s ripple current ratings are often based on
only 2000 hours lifetime. This makes it advisable to further
derate the capacitor, or choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet the size or height requirements of the
design. An additional 0.1μF to 1μF ceramic capacitor is
also recommended on VIN for high frequency decoupling
when not using an all-ceramic capacitor solution.
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. The
output ripple VOUT is determined by:
1
VOUT IL ESR+
8fOCOUT
where fO = operating frequency, COUT = output capacitance
and IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since IL increases with input voltage.
If tantalum capacitors are used, it is critical that the
capacitors are surge tested for use in switching power
supplies. An excellent choice is the AVX TPS series of
surface mount tantalum. These are specially constructed
and tested for low ESR so they give the lowest ESR for a
given volume. Other capacitor types include Sanyo POSCAP,
Kemet T510 and T495 series, and Sprague 593D and
595D series. Consult the manufacturer for other specific
recommendations.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them
ideal for switching regulator applications. Because the
LTC3606B control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input. When a ceramic capacitor is used at the
input and the power is supplied by a wall adapter through
long wires, a load step at the output can induce ringing at
the input, VIN. At best, this ringing can couple to the output
and be mistaken as loop instability. At worst, a sudden
inrush of current through the long wires can potentially
cause a voltage spike at VIN, large enough to damage the
part. For more information, see Application Note 88.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage
characteristics of all the ceramics for a given value and
size.
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LTC3606B
APPLICATIONS INFORMATION
Setting the Output Voltage
The LTC3606B regulates the VFB pin to 0.6V during
regulation. Thus, the output voltage is set by a resistive
divider, Figure 2, according to the following formula:
VOUT = 0.6V 1+
R2
R1
(2)
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to
Application Note 76.
To improve the frequency response of the main control
loop, a feedback capacitor (CF) may also be used. Great
care should be taken to route the VFB line away from noise
sources, such as the inductor or the SW line.
In some applications, a more severe transient can be caused
by switching in loads with large (>1μF) input capacitors. The
discharged input capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the switch
connecting the load has low resistance and is driven quickly.
The solution is to limit the turn-on speed of the load switch
driver. A Hot Swap™ controller is designed specifically for
this purpose and usually incorporates current limiting,
short-circuit protection, and soft-starting.
Checking Transient Response
Efficiency Considerations
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ILOAD • ESR, where ESR is the effective series
resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the
regulator to return VOUT to its steady-state value. During
this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem.
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard second
order overshoot/DC ratio cannot be used to determine
the phase margin. In addition, feedback capacitors (CF)
can be added to improve the high frequency response, as
shown in Figure 2. Capacitor CF provides phase lead by
creating a high frequency zero with R2 which improves
the phase margin.
Although all dissipative elements in the circuit produce
losses, four sources usually account for the losses in
LTC3606B circuits: 1) VIN quiescent current, 2) switching
losses, 3) I2R losses, 4) other system losses.
Keeping the current small (< 10μA) in these resistors
maximizes efficiency, but making it too small may allow
stray capacitance to cause noise problems or reduce the
phase margin of the error amp loop.
% Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc., are the individual losses as a percentage
of input power.
1. The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET
driver and control currents. VIN current results in a
small (