LTC3703IGN#TRPBF

LTC3703 100V Synchronous Switching Regulator Controller Description Features High Voltage Operation: Up to 100V Large 1Ω Gate Drivers No Current Sense Resistor Required Step-Up or Step-Down DC/DC Converter Dual N-Channel MOSFET Synchronous Drive Excellent Line and Load Transient Response Programmable Constant Frequency: 100kHz to 600kHz n ±1% Reference Accuracy n Synchronizable up to 600kHz n Selectable Pulse-Skip Mode Operation n Low Shutdown Current: 50µA Typ n Programmable Current Limit n Undervoltage Lockout n Programmable Soft-Start n 16-Pin Narrow SSOP and 28-Pin SSOP Packages n n n n n n The LTC®3703 is a synchronous step-down switching regulator controller that can directly step down voltages from up to 100V, making it ideal for telecom and automotive applications. The LTC3703 drives external N-channel MOSFETs using a constant frequency (up to 600kHz), voltage mode architecture. n A precise internal reference provides 1% DC accuracy. A high bandwidth error amplifier and patented line feedforward compensation provide very fast line and load transient response. Strong 1Ω gate drivers allow the LTC3703 to drive multiple MOSFETs for higher current applications. The operating frequency is user programmable from 100kHz to 600kHz and can also be synchronized to an external clock for noise-sensitive applications. Current limit is programmable with an external resistor and utilizes the voltage drop across the synchronous MOSFET to eliminate the need for a current sense resistor. For applications requiring up to 60V operation with logic-level MOSFETS, refer to the LTC3703-5 data sheet. Applications 48V Telecom and Base Station Power Supplies Networking Equipment, Servers n Automotive and Industrial Control n n PARAMETER Maximum VIN MOSFET Gate Drive VCC UV+ VCC UV– L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and ThinSOT and No RSENSE are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5408150, 5055767, 6677210, 5847554, 5481178, 6304066, 6580258. LTC3703-5 60V 4.5V to 15V 3.7V 3.1V LTC3703 100V 9.3V to 15V 8.7V 6.2V Typical Application VCC 9.3V TO 15V + 22µF 25V BAS19 MODE/SYNC VIN 10k 1000pF 470pF 15k 8.06k 1% 0.1µF 113k 1% 100Ω 2200pF + fSET BOOST LTC3703 TG COMP FB SW IMAX VCC INV RUN/SS GND DRVCC BG BGRTN 100 68µF 0.1µF 8µH 270µF 16V 10Ω VIN = 25V VIN = 50V 95 Si7456DP VOUT 12V 5A + EFFICIENCY (%) 30k Efficiency vs Load Current VIN 15V TO 100V VIN = 75V 90 85 Si7456DP MBR1100 10µF 80 1µF 3703 F01 Figure 1. High Efficiency High Voltage Step-Down Converter 0 1 3 2 LOAD (A) 4 5 3703 F01b 3703fc 1 LTC3703 Absolute Maximum Ratings (Note 1) Supply Voltages VCC, DRVCC ........................................... –0.3V to 15V (DRVCC – BGRTN), (BOOST – SW)........ –0.3V to 15V BOOST.................................................. –0.3V to 115V BGRTN........................................................ –5V to 0V VIN Voltage............................................... –0.3V to 100V SW Voltage (Note 10).................................. –1V to 100V RUN/SS Voltage........................................... –0.3V to 5V MODE/SYNC, INV Voltages........................ –0.3V to 15V fSET, FB, IMAX Voltages................................ –0.3V to 3V Peak Output Current 125°C RUN/SS = 0V MIN l TYP 9.3 15 l l UNITS V 100 V 1.7 50 2.5 mA µA 0 0 5 5 µA µA 360 360 0 500 800 5 µA µA µA 0.800 0.808 0.812 V V 0.007 0.05 %/V 0.01 0.1 % 0.81 0.87 V l l MAX Main Control Loop VFB Feedback Voltage (Note 4) l ∆VFB(LINE) Feedback Voltage Line Regulation 9V < VCC < 15V (Note 4) l ∆VFB(LOAD) VMODE/SYNC Feedback Voltage Load Regulation 1V < VCOMP < 2V (Note 4) l MODE/SYNC Threshold MODE/SYNC Rising ∆VMODE/SYNC MODE/SYNC Hysteresis 0.75 20 IMODE/SYNC MODE/SYNC Current VINV Invert Threshold IINV Invert Current 0 ≤ VINV ≤ 15V IVIN VIN Sense Input Current VIN = 100V RUN/SS = 0V, VIN = 10V IMAX IMAX Source Current VIMAX = 0V VOS(IMAX) VIMAX Offset Voltage VRUN/SS Shutdown Threshold IRUN/SS VUV 0.792 0.788 0 ≤ VMODE/SYNC ≤ 15V 1 mV 0 1 µA 1.5 2 V 0 1 µA 100 0 140 1 µA µA 10.5 12 13.5 µA |VSW| – VIMAX at IRUN/SS = 0µA H Grade –25 –25 10 10 55 65 mV mV 0.7 0.9 1.2 V RUN/SS Source Current RUN/SS = 0V 2.5 4 5.5 µA Maximum RUN/SS Sink Current |VSW| – VIMAX ≥ 200mV, VRUN/SS = 3V Undervoltage Lockout VCC Rising VCC Falling RSET = 25k l l 9 17 25 µA 8.0 5.7 8.7 6.2 9.3 6.8 V V 270 300 330 kHz 600 kHz Oscillator fOSC Oscillator Frequency fSYNC External Sync Frequency Range tON(MIN) Minimum On-Time DCMAX Maximum Duty Cycle 100 200 f < 200kHz 89 93 1.5 2 1.5 2 ns 96 % Driver IBG(PEAK) BG Driver Peak Source Current RBG(SINK) BG Driver Pull-Down RDS(ON) ITG(PEAK) TG Driver Peak Source Current RTG(SINK) TG Driver Pull-Down RDS(ON) (Note 8) 1 (Note 8) 1 A 1.5 Ω A 1.5 Ω Feedback Amplifier AVOL Op Amp DC Open Loop Gain (Note 4) fU Op Amp Unity Gain Crossover Frequency (Note 6) IFB FB Input Current 0 ≤ VFB ≤ 3V ICOMP COMP Sink/Source Current 74 85 dB 25 MHz 0 ±5 ±10 1 µA mA 3703fc 3 LTC3703 Electrical Characteristics Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3703E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3703I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LTC3703H is guaranteed over the full –40°C to 150°C operating junction temperature range. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3703: TJ = TA + (PD • 100 °C/W) G Package Note 4: The LTC3703 is tested in a feedback loop that servos VFB to the reference voltage with the COMP pin forced to a voltage between 1V and 2V. Note 5: The dynamic input supply current is higher due to the power MOSFET gate charge being delivered at the switching frequency (QG • fOSC). Note 6: Guaranteed by design. Not subject to test. Note 7: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 8: RDS(ON) guaranteed by correlation to wafer level measurement. Note 9: High junction temperatures degrade operating lifetimes. Operating lifetime at junction temperatures greater than 125°C is derated to 1000 hours. Note 10: Transient voltages (such as due to inductive ringing) are allowed beyond this range provided that the voltage does not exceed 10V below ground and duration does not exceed 20ns per switching cycle. Typical Performance Characteristics TA = 25°C unless otherwise noted. Efficiency vs Input Voltage Efficiency vs Load Current IOUT = 5A 95 95 EFFICIENCY (%) EFFICIENCY (%) 90 IOUT = 0.5A 85 80 0 10 20 70 30 40 50 60 INPUT VOLTAGE (V) VIN = 45V 90 VIN = 75V 85 IOUT 2A/DIV 80 70 80 VOUT 50mV/DIV VIN = 15V VOUT = 5V f = 250kHz PULSE SKIP ENABLED 75 VOUT = 12V f = 300kHz PULSE SKIP DISABLED 75 70 Load Transient Response 100 100 VIN = 50V 50µs/DIV VOUT = 12V 1A TO 5A LOAD STEP 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 LOAD CURRENT (A) 3703 G02 3703 G01 VCC Current vs VCC Voltage 4 3.5 VCC Shutdown Current vs VCC Voltage VCC Current vs Temperature 100 90 2.0 VFB = 0V 1.5 1.0 80 COMP = 1.5V 3 VCC CURRENT (µA) COMP = 1.5V 2.5 VCC CURRENT (mA) VCC CURRENT (mA) 3.0 2 VFB = 0V 1 70 60 50 40 30 20 0.5 0 3703 G03 10 6 8 10 12 VCC VOLTAGE (V) 14 16 3703 G04 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G05 0 6 8 12 10 VCC VOLTAGE (V) 14 16 3703 G06 3703fc 4 LTC3703 Typical Performance Characteristics VCC Shutdown Current vs Temperature Reference Voltage vs Temperature 70 Normalized Frequency vs Temperature 0.803 1.20 1.15 55 50 45 40 0.802 NORMALIZED FREQUENCY 60 REFERENCE VOLTAGE (V) VCC CURRENT (µA) 65 0.801 0.800 0.799 30 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 0.798 –50 –25 0 0.95 0.90 VCC = 10V PEAK SOURCE CURRENT (A) 1.4 2.6 Driver Peak Source Current vs Supply Voltage 3.0 1.6 VCC = 10V 1.2 2.4 RDS(ON) (Ω) 2.2 2.0 1.8 1.6 1.0 0.8 0.6 0.4 1.4 0.2 1.2 1.0 –50 –25 0 25 50 75 0 –50 –25 100 125 150 TEMPERATURE (°C) 0 70 1.0 0.5 0.7 15 3703 G13 8 9 10 11 12 13 14 15 7 DRVCC/BOOST VOLTAGE (V) 8 DRVCC, BOOST = 10V 7 RISE 50 40 30 FALL 20 10 14 6 RUN/SS Pull-Up Current vs Temperature RUN/SS CURRENT (µA) RISE/FALL TIME (ns) 0.8 5 3703 G12 60 1.0 8 9 10 11 12 13 DRVCC/BOOST VOLTAGE (V) 1.5 Rise/Fall Time vs Gate Capacitance 1.1 7 2.0 3703 G11 Driver Pull-Down RDS(ON) vs Supply Voltage 6 2.5 0 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G10 0.9 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G09 Driver Pull-Down RDS(ON) vs Temperature 3.0 2.8 0 3703 G08 Driver Peak Source Current vs Temperature PEAK SOURCE CURRENT (A) 1.00 0.80 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G07 RDS(ON) (Ω) 1.05 0.85 35 0.6 1.10 0 6 5 4 3 2 1 0 6000 2000 4000 8000 GATE CAPACITANCE (pF) 10000 3703 G14 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1573 G15 3703fc 5 LTC3703 Typical Performance Characteristics RUN/SS Sink Current vs SW Voltage 6 25 5 20 4 3 2 1 0 Max % DC vs RUN/SS Voltage 100 IMAX = 0.3V 90 80 15 MAX DUTY CYCLE (%) RUN/SS SINK CURRENT (µA) RUN/SS PULLUP CURRENT (µA) RUN/SS Pull-Up Current vs VCC Voltage 10 5 0 6 8 10 12 VCC VOLTAGE (V) 14 16 0 0.1 0.2 0.3 0.4 0.5 |SW| VOLTAGE (V) 0.6 0.7 12 1.0 2.0 1.5 RUN VOLTAGE (V) 2.5 Max % DC vs Frequency and Temperature 100 VIN = 10V 95 60 VIN = 75V 40 VIN = 50V VIN = 25V 20 25 50 75 100 125 150 TEMPERATURE (°C) 0 3.0 3703 G18 MAX DUTY CYCLE (%) 80 –45°C 90 25°C 85 90°C 80 150°C 75 0.5 0.75 1.00 1.25 1.50 COMP (V) 1.75 70 2.00 0 100 200 300 400 500 FREQUENCY (kHz) 3703 G20 3703 G19 Shutdown Threshold vs Temperature 125°C 600 700 3703 G21 tON(MIN) vs Temperature 1.4 200 1.2 180 160 1.0 tON(MIN) (ns) SHUTDOWN THRESHOLD (V) 20 –10 0.5 100 DUTY CYCLE (%) IMAX SOURCE CURRENT (µA) 30 % Duty Cycle vs COMP Voltage 13 0.8 0.6 0.4 140 120 100 80 0.2 0 –50 –25 40 3703 G17 IMAX Current vs Temperature 0 50 0 3703 G16 11 –50 –25 60 10 –5 –10 70 60 0 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G22 40 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3703 G23 3703fc 6 LTC3703 Pin Functions (GN16/G28) MODE/SYNC (Pin 1/Pin 6): Pulse-Skip Mode Enable/Sync Pin. This multifunction pin provides pulse-skip mode enable/disable control and an external clock input for synchronization of the internal oscillator. Pulling this pin below 0.8V or to an external logic-level synchronization signal disables pulse-skip mode operation and forces continuous operation. Pulling the pin above 0.8V enables pulse-skip mode operation. This pin can also be connected to a feedback resistor divider from a secondary winding on the inductor to regulate a second output voltage. fSET (Pin 2/Pin 7): Frequency Set. A resistor connected to this pin sets the free running frequency of the internal oscillator. See Applications Information section for resistor value selection details. COMP (Pin 3/Pin 8): Loop Compensation. This pin is connected directly to the output of the internal error amplifier. An RC network is used at the COMP pin to compensate the feedback loop for optimal transient response. FB (Pin 4/Pin 9): Feedback Input. Connect FB through a resistor divider network to VOUT to set the output voltage. Also connect the loop compensation network from COMP to FB. IMAX (Pin 5/Pin 10): Current Limit Set. The IMAX pin sets the current limit comparator threshold. If the voltage drop across the bottom MOSFET exceeds the magnitude of the voltage at IMAX, the controller goes into current limit. The IMAX pin has an internal 12µA current source, allowing the current threshold to be set with a single external resistor to ground. See the Current Limit Programming section for more information on choosing RIMAX. INV (Pin 6/Pin 11): Top/Bottom Gate Invert. Pulling this pin above 2V sets the controller to operate in step-up (boost) mode with the TG output driving the synchronous MOSFET and the BG output driving the main switch. Below 1V, the controller will operate in step-down (buck) mode. RUN/SS (Pin 7/Pin 13): Run/Soft-Start. Pulling RUN/SS below 0.9V will shut down the LTC3703, turn off both of the external MOSFET switches and reduce the quiescent supply current to 50µA. A capacitor from RUN/SS to ground will control the turn-on time and rate of rise of the output voltage at power-up. An internal 4µA current source pull-up at the RUN/SS pin sets the turn-on time at approximately 750ms/µF. GND (Pin 8/Pin 14): Ground Pin. BGRTN (Pin 9/Pin 15): Bottom Gate Return. This pin connects to the source of the pull-down MOSFET in the BG driver and is normally connected to ground. Connecting a negative supply to this pin allows the synchronous MOSFET’s gate to be pulled below ground to help prevent false turn-on during high dV/dt transitions on the SW node. See the Applications Information section for more details. BG (Pin 10/Pin 19): Bottom Gate Drive. The BG pin drives the gate of the bottom N-channel synchronous switch MOSFET. This pin swings from BGRTN to DRVCC. DRVCC (Pin 11/Pin 20): Driver Power Supply Pin. DRVCC provides power to the BG output driver. This pin should be connected to a voltage high enough to fully turn on the external MOSFETs, normally 10V to 15V for standard threshold MOSFETs. DRVCC should be bypassed to BGRTN with a 10µF, low ESR (X5R or better) ceramic capacitor. VCC (Pin 12/Pin 21): Main Supply Pin. All internal circuits except the output drivers are powered from this pin. VCC should be connected to a low noise power supply voltage between 9V and 15V and should be bypassed to GND (Pin 8) with at least a 0.1µF capacitor in close proximity to the LTC3703. SW (Pin 13/Pin 26): Switch Node Connection to Inductor and Bootstrap Capacitor. Voltage swing at this pin is from a Schottky diode (external) voltage drop below ground to VIN. TG (Pin 14/Pin 27): Top Gate Drive. The TG pin drives the gate of the top N-channel synchronous switch MOSFET. The TG driver draws power from the BOOST pin and returns to the SW pin, providing true floating drive to the top MOSFET. BOOST (Pin 15/Pin 28): Top Gate Driver Supply. The BOOST pin supplies power to the floating TG driver. The BOOST pin should be bypassed to SW with a low ESR (X5R or better) 0.1µF ceramic capacitor. An additional fast recovery Schottky diode from DRVCC to BOOST will create a complete floating charge-pumped supply at BOOST. VIN (Pin 16/Pin 1): Input Voltage Sense Pin. This pin is connected to the high voltage input of the regulator and is used by the internal feedforward compensation circuitry to improve line regulation. This is not a supply pin. 3703fc 7 LTC3703 Functional Diagram RSET 2 fSET GN16 OVERCURRENT 12µA 4µA – 5 IMAX RMAX + 50mV – ± + RUN/SS 5 – CSS 0.9V 3.2V 1 ± CHIP SD + UVSD OTSD MODE/SYNC SYNC DETECT EXT SYNC OSC 4 R2 R1 16 12 DB – INV REVERSE CURRENT BOOST TG COMP 0.8V FB VCC + FORCED CONTINUOUS 3 INV + + FB – ÷ % DC LIMIT SW – PWM + DRIVE LOGIC VIN DRVCC BG +MIN– +MAX– BGRTN INV VCC (> VOUT, the top MOSFETs’ “on” resistance is normally less important for overall efficiency than its input capacitance at operating frequencies above 300kHz. MOSFET manufacturers have designed special purpose devices that provide reasonably low “on” resistance with significantly reduced input capacitance for the main switch application in switching regulators. Selection criteria for the power MOSFETs include the “on” resistance RDS(ON), input capacitance, breakdown voltage and maximum output current. The most important parameter in high voltage applications is breakdown voltage BVDSS. Both the top and bottom MOSFETs will see full input voltage plus any additional ringing on the switch node across its drain-to-source during its off-time and must be chosen with the appropriate 3703fc 13 LTC3703 Applications Information breakdown specification. Since many high voltage MOSFETs have higher threshold voltages (typically, VGS(MIN) ≥ 6V), the LTC3703 is designed to be used with a 9V to 15V gate drive supply (DRVCC pin). For maximum efficiency, on-resistance RDS(ON) and input capacitance should be minimized. Low RDS(ON) minimizes conduction losses and low input capacitance minimizes transition losses. MOSFET input capacitance is a combination of several components but can be taken from the typical “gate charge” curve included on most data sheets (Figure 9). VIN VGS MILLER EFFECT a V b QIN CMILLER = (QB – QA)/VDS + VGS – +V DS – 3703 F09 Figure 9. Gate Charge Characteristic The curve is generated by forcing a constant input current into the gate of a common source, current source loaded stage and then plotting the gate voltage versus time. The initial slope is the effect of the gate-to-source and the gate-to-drain capacitance. The flat portion of the curve is the result of the Miller multiplication effect of the drain-to-gate capacitance as the drain drops the voltage across the current source load. The upper sloping line is due to the drain-to-gate accumulation capacitance and the gate-to-source capacitance. The Miller charge (the increase in coulombs on the horizontal axis from a to b while the curve is flat) is specified for a given VDS drain voltage, but can be adjusted for different VDS voltages by multiplying by the ratio of the application VDS to the curve specified VDS values. A way to estimate the CMILLER term is to take the change in gate charge from points a and b on a manufacturers data sheet and divide by the stated VDS voltage specified. CMILLER is the most important selection criteria for determining the transition loss term in the top MOSFET but is not directly specified on MOSFET data sheets. CRSS and COS are specified sometimes but definitions of these parameters are not included. When the controller is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT VIN Synchronous Switch Duty Cycle = VIN – VOUT VIN The power dissipation for the main and synchronous MOSFETs at maximum output current are given by: PMAIN = VOUT 2 IMAX ) (1+ δ)RDS(ON) + ( VIN 2 IMAX VIN (RDR )(CMILLER )• 2  1  1 +   (f)  VCC – VTH(IL) VTH(IL)  V −V PSYNC = IN OUT (IMAX )2 (1+ δ)RDS(0N) VIN where δ is the temperature dependency of RDS(ON), RDR is the effective top driver resistance (approximately 2Ω at VGS = VMILLER), VIN is the drain potential and the change in drain potential in the particular application. VTH(IL) is the data sheet specified typical gate threshold voltage specified in the power MOSFET data sheet at the specified drain current. CMILLER is the calculated capacitance using the gate charge curve from the MOSFET data sheet and the technique described above. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which peak at the highest input voltage. For VIN < 25V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 25V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short circuit when the synchronous switch is on close to 100% of the period. 3703fc 14 LTC3703 Applications Information The term (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, and typically varies from 0.005/°C to 0.01/°C depending on the particular MOSFET used. Multiple MOSFETs can be used in parallel to lower RDS(ON) and meet the current and thermal requirements if desired. The LTC3703 contains large low impedance drivers capable of driving large gate capacitances without significantly slowing transition times. In fact, when driving MOSFETs with very low gate charge, it is sometimes helpful to slow down the drivers by adding small gate resistors (5Ω or less) to reduce noise and EMI caused by the fast transitions. Schottky Diode Selection The Schottky diode D1 shown in Figure 1 conducts during the dead time between the conduction of the power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on and storing charge during the dead time and requiring a reverse recovery period that could cost as much as 1% to 2% in efficiency. A 1A Schottky diode is generally a good size for 3A to 5A regulators. Larger diodes result in additional losses due to their larger junction capacitance. The diode can be omitted if the efficiency loss can be tolerated. Input Capacitor Selection In continuous mode, the drain current of the top MOSFET is approximately a square wave of duty cycle VOUT/VIN which must be supplied by the input capacitor. To prevent large input transients, a low ESR input capacitor sized for the maximum RMS current is given by:  V  V ICIN(RMS) ≅IO(MAX) OUT  IN – 1 VIN  VOUT  1/2 This formula has a maximum at VIN = 2VOUT, where IRMS = IO(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher tempera- ture than required. Several capacitors may also be placed in parallel to meet size or height requirements in the design. Because tantalum and OS-CON capacitors are not available in voltages above 30V, for regulators with input supplies above 30V, choice of input capacitor type is limited to ceramics or aluminum electrolytics. Ceramic capacitors have the advantage of very low ESR and can handle high RMS current, however ceramics with high voltage ratings (>50V) are not available with more than a few microfarads of capacitance. Furthermore, ceramics have high voltage coefficients which means that the capacitance values decrease even more when used at the rated voltage. X5R and X7R type ceramics are recommended for their lower voltage and temperature coefficients. Another consideration when using ceramics is their high Q which if not properly damped, may result in excessive voltage stress on the power MOSFETs. Aluminum electrolytics have much higher bulk capacitance, however, they have higher ESR and lower RMS current ratings. A good approach is to use a combination of aluminum electrolytics for bulk capacitance and ceramics for low ESR and RMS current. If the RMS current cannot be handled by the aluminum capacitors alone, when used together, the percentage of RMS current that will be supplied by the aluminum capacitor is reduced to approximately: % IRMS,ALUM ≈ 1 1+(8fCRESR )2 •100% where RESR is the ESR of the aluminum capacitor and C is the overall capacitance of the ceramic capacitors. Using an aluminum electrolytic with a ceramic also helps damp the high Q of the ceramic, minimizing ringing. Output Capacitor Selection The selection of COUT is primarily determined by the ESR required to minimize voltage ripple. The output ripple (∆VOUT) is approximately equal to:  1  ∆VOUT ≤ ∆IL  ESR + 8fCOUT   3703fc 15 LTC3703 Applications Information Since ∆IL increases with input voltage, the output ripple is highest at maximum input voltage. ESR also has a significant effect on the load transient response. Fast load transitions at the output will appear as voltage across the ESR of COUT until the feedback loop in the LTC3703 can change the inductor current to match the new load current value. Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering and has the required RMS current rating. Manufacturers such as Nichicon, Nippon Chemi-Con and Sanyo should be considered for high performance throughhole capacitors. The OS-CON (organic semiconductor dielectric) capacitor available from Sanyo has the lowest product of ESR and size of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the effect of their lead inductance. In surface mount applications, multiple capacitors placed in parallel may be required to meet the ESR, RMS current handling and load step requirements. Dry tantalum, special polymer and aluminum electrolytic capacitors are available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Several excellent surge-tested choices are the AVX TPS and TPSV or the KEMET T510 series. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-driven applications providing that consideration is given to ripple current ratings and long term reliability. Other capacitor types include Panasonic SP and Sanyo POSCAPs. Output Voltage The LTC3703 output voltage is set by a resistor divider according to the following formula:  R1 VOUT = 0.8V  1+   R2  The external resistor divider is connected to the output as shown in the Functional Diagram, allowing remote voltage sensing. The resultant feedback signal is compared with the internal precision 800mV voltage reference by the error amplifier. The internal reference has a guaranteed tolerance of ±1%. Tolerance of the feedback resistors will add additional error to the output voltage. 0.1% to 1% resistors are recommended. MOSFET Driver Supplies (DRVCC and BOOST) The LTC3703 drivers are supplied from the DRVCC and BOOST pins (see Figure 3), which have an absolute maximum voltage of 15V. If the main supply voltage, VIN, is higher than 15V a separate supply with a voltage between 9V and 15V must be used to power the drivers. If a separate supply is not available, one can easily be generated from the main supply using one of the circuits shown in Figure 10. If the output voltage is between 10V and 15V, the output can be used to directly power the drivers as shown in Figure 10a. If the output is below 10V, Figure 10b shows an easy way to boost the supply voltage to a sufficient level. This boost circuit uses the LT1613 in a ThinSOT™ package and a chip inductor for minimal extra area ( VIN. For hard shorts, the inductor current is limited only by the input supply capability. Refer to Current Limit Programming for buck mode for further considerations for current limit programming. Boost Converter: Feedback Loop/Compensation Compensating a voltage mode boost converter is unfortunately more difficult than for a buck converter. This is due to an additional right-half plane (RHP) zero that is present in the boost converter but not in a buck. The additional phase lag resulting from the RHP zero is difficult if not impossible to compensate even with a Type 3 loop, so the best approach is usually to roll off the loop gain at a lower frequency than what could be achievable in buck converter. A typical gain/phase plot of a voltage mode boost converter is shown in Figure 16. The modulator gain and phase can be measured as described for a buck converter or can be estimated as follows: GAIN (COMP-to-VOUT DC gain) = 20Log(VOUT2/VIN) Dominant Pole: fP = VIN 1 • VOUT 2π LC Since significant phase shift begins at frequencies above the dominant LC pole, choose a crossover frequency no greater than about half this pole frequency. The gain of the compensation network should equal –GAIN at this frequency so that the overall loop gain is 0dB here. The 3703 F16 Figure 16. Transfer Function of Boost Modulator compensation component to achieve this, using a Type 1 amplifier (see Figure 12), is: G = 10–GAIN/20 C1 = 1/(2π • f • G • R1) Run/Soft-Start Function The RUN/SS pin is a multipurpose pin that provide a softstart function and a means to shut down the LTC3703. Soft-start reduces the input supply’s surge current by gradually increasing the duty cycle and can also be used for power supply sequencing. Pulling RUN/SS below 0.9V puts the LTC3703 into a low quiescent current shutdown (IQ ≅ 50µA). This pin can be driven directly from logic as shown in Figure 17. Releasing the RUN/SS pin allows an internal 4µA current source to RUN/SS 2V/DIV VOUT 5V/DIV IL 2A/DIV VIN = 50V ILOAD = 2A CSS = 0.01µF 2ms/DIV 3703 F17 Figure 17. LTC3703 Start-Up Operation 3703fc 23 LTC3703 Applications Information charge up the soft-start capacitor CSS. When the voltage on RUN/SS reaches 0.9V, the LTC3703 begins operating at its minimum on-time. As the RUN/SS voltage increases from 1.4V to 3V, the duty cycle is allowed to increase from 0% to 100%. The duty cycle control minimizes input supply inrush current and eliminates output voltage overshoot at start-up and ensures current limit protection even with a hard short. The RUN/SS voltage is internally clamped at 4V. If RUN/SS starts at 0V, the delay before starting is approximately: tDELAY,START = 1V C = (0.25s/µF)CSS 4µA SS plus an additional delay, before the output will reach its regulated value, of: tDELAY,REG ≥ 3V – 1V C = (0.5s/µF)CSS 4µA SS The start delay can be reduced by using diode D1 in Figure 18. 3.3V OR 5V RUN/SS RUN/SS D1 CSS CSS 3703 F18 Figure 18. RUN/SS Pin Interfacing MODE/SYNC Pin (Operating Mode and Secondary Winding Control) The MODE/SYNC pin is a dual function pin that can be used for enabling or disabling pulse-skip mode operation and also as an external clock input for synchronizing the internal oscillator (see next section). Pulse-skip mode is enabled when the MODE/SYNC pin is above 0.8V and is disabled, i.e., forced continuous, when the pin is below 0.8V. In addition to providing a logic input to force continuous operation and external synchronization, the MODE/SYNC pin provides a means to regulate a flyback winding output as shown in Figure 10c. The auxiliary output is taken from a second winding on the core of the inductor, converting it to a transformer. The auxiliary output voltage is set by the main output voltage and the turns ratio of the extra winding to the primary winding as follows: VSEC ≈ (N + 1)VOUT Since the secondary winding only draws current when the synchronous switch is on, load regulation at the auxiliary output will be relatively good as long as the main output is running in continuous mode. As the load on the primary output drops and the LTC3703 switches to pulse-skip mode operation, the auxiliary output may not be able to maintain regulation, especially if the load on the auxiliary output remains heavy. To avoid this, the auxiliary output voltage can be divided down with a conventional feedback resistor string with the divided auxiliary output voltage fed back to the MODE/SYNC pin. The MODE/SYNC threshold is trimmed to 800mV with 20mV of hysteresis, allowing precise control of the auxiliary voltage and is set as follows:  R1 VSEC(MIN) ≈ 0.8V  1+   R2  where R1 and R2 are shown in Figure 10c. If the LTC3703 is operating in pulse-skip mode and the auxiliary output voltage drops below VSEC(MIN), the MODE/ SYNC pin will trip and the LTC3703 will resume continuous operation regardless of the load on the main output. Thus, the MODE/SYNC pin removes the requirement that power must be drawn from the inductor primary in order to extract power from the auxiliary winding. With the loop in continuous mode (MODE/SYNC < 0.8V), the auxiliary outputs may nominally be loaded without regard to the primary output load. The following table summarizes the possible states available on the MODE/SYNC pin: Table 1 MODE/SYNC PIN CONDITION DC Voltage: 0V to 0.75V Forced Continuous Current Reversal Enabled DC Voltage: ≥ 0.87V Pulse-Skip Mode Operation No Current Reversal Feedback Resistors Regulating a Secondary Winding Ext. Clock: 0V to ≥ 2V Forced Continuous Current Reversal Enabled 3703fc 24 LTC3703 Applications Information MODE/SYNC Pin (External Synchronization) The internal LTC3703 oscillator can be synchronized to an external oscillator by applying and clocking the MODE/ SYNC pin with a signal above 2VP-P . The internal oscillator locks to the external clock after the second clock transition is received. When external synchronization is detected, LTC3703 will operate in forced continuous mode. If an external clock transition is not detected for three successive periods, the internal oscillator will revert to the frequency programmed by the RSET resistor. The internal oscillator can synchronize to frequencies between 100kHz and 600kHz, independent of the frequency programmed by the RSET resistor. However, it is recommended that an RSET resistor be chosen such that the frequency programmed by the RSET resistor is close to the expected frequency of the external clock. In this way, the best converter operation (ripple, component stress, etc) is achieved if the external clock signal is lost. Fault Conditions: Output Overvoltage Protection (Crowbar) The output overvoltage crowbar is designed to blow a system fuse in the input lead when the output of the regulator rises much higher than nominal levels. This condition causes huge currents to flow, much greater than in normal operation. This feature is designed to protect against a shorted top MOSFET; it does not protect against a failure of the controller itself. The comparator (MAX in the Functional Diagram) detects overvoltage faults greater than 5% above the nominal output voltage. When this condition is sensed the top MOSFET is turned off and the bottom MOSFET is forced on. The bottom MOSFET remains on continuously for as long as the 0V condition persists; if VOUT returns to a safe level, normal operation automatically resumes. Minimum On-Time Considerations (Buck Mode) Minimum on-time tON(MIN) is the smallest amount of time that the LTC3703 is capable of turning the top MOSFET on and off again. It is determined by internal timing delays and the amount of gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: tON = VOUT >t VIN • f ON(MIN) where tON(MIN) is typically 200ns. If the duty cycle falls below what can be accommodated by the minimum on-time, the LTC3703 will begin to skip cycles. The output will be regulated, but the ripple current and ripple voltage will increase. If lower frequency operation is acceptable, the on-time can be increased above tON(MIN) for the same step-down ratio. Pin Clearance/Creepage Considerations The LTC3703 is available in two packages (GN16 and G28) both with identical functionality. The GN16 package gives the smallest size solution, however the 0.013" (minimum) space between pins may not provide sufficient PC board trace clearance between high and low voltage pins in higher voltage applications. Where clearance is an issue, the G28 package should be used. The G28 package has four unconnected pins between the all adjacent high voltage and low voltage pins, providing 5(0.0106") = 0.053" clearance which will be sufficient for most applications up to 100V. For more information, refer to the printed circuit board design standards described in IPC-2221 (www.ipc.org). Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power (x100%). Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. It is often useful to analyze the individual losses to determine what is limiting the efficiency and what change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3703 circuits: 1) LTC3703 VCC current, 2) MOSFET gate current, 3) I2R losses, 4) Topside MOSFET transition losses. 3703fc 25 LTC3703 Applications Information 1. VCC supply current. The VCC current is the DC supply current given in the Electrical Characteristics table which powers the internal control circuitry of the LTC3703. Total supply current is typically about 2.5mA and usually results in a small (