LTC3803
Constant Frequency
Current Mode Flyback
DC/DC Controller in ThinSOT
FEATURES
DESCRIPTION
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The LTC®3803 is a constant frequency current mode flyback
controller optimized for driving N-channel MOSFETs in high
input voltage applications. Constant frequency operation
is maintained down to very light loads, resulting in less
low frequency noise generation over a wide range of load
currents. Slope compensation can be programmed with
an external resistor.
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VIN and VOUT Limited Only by External Components
Adjustable Slope Compensation
Internal Soft-Start
Constant Frequency 200kHz Operation
±1.5% Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
No Minimum Load Requirement
Low Quiescent Current: 240μA
Low Profile (1mm) SOT-23 Package
The LTC3803 provides ±1.5% output voltage accuracy and
consumes only 240μA of quiescent current. Ground-referenced current sensing allows LTC3803-based converters
to accept input supplies beyond the LTC3803’s absolute
maximum VCC. A micropower hysteretic start-up feature
allows efficient operation at high input voltages. For simplicity, the LTC3803 can also be powered from a high VIN
through a resistor, due to its internal shunt regulator. An
internal undervoltage lockout shuts down the LTC3803
when the input voltage is too low to provide sufficient
gate drive to the external MOSFET.
APPLICATIONS
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Telecom Power Supplies
42V and 12V Automotive Power Supplies
Auxiliary/Housekeeping Power Supplies
Power Over Ethernet
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT and No RSENSE are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners.
The LTC3803 is available in a low profile (1mm) 6-lead
SOT-23 (ThinSOT™) package.
TYPICAL APPLICATION
5V Output Nonisolated Telecom Housekeeping Power Supply
90
UPS840
T1
10μF
10V
X5R
0.0022μF
•
•
300μF*
6.3V
X5R
VCC
ITH/RUN NGATE
56k
4.7μF
100V
X5R
80
VOUT
5V
2A MAX
FDC2512
VIN = 36V
70
EFFICIENCY (%)
VIN
36V TO 72V
10k
Efficiency vs Load Current
VIN = 48V
60
50
40
VIN = 60V
VIN = 72V
30
LTC3803
GND
VFB
20k
20
SENSE
10
68mΩ
0
0.1
105k
3803 TA01
T1: COOPER CTX02-15242
*THREE 100μF UNITS IN PARALLEL
1
10
IOUT (A)
3803 TA02
3803fc
1
LTC3803
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
VCC to GND
Low Impedance Source ........................... –0.3V to 8V
Current Fed ........................................25mA into VCC*
NGATE Voltage ............................................–0.3V to VCC
VFB, ITH/RUN Voltages............................... –0.3V to 3.5V
SENSE Voltage ............................................. –0.3V to 1V
NGATE Peak Output Current ( VTURNON, VITH/RUN Falling
LTC3803E
LTC3803H, LTC3803I, LTC3803MP
l
l
ICC(UV)
VITHSHDN
Shutdown Threshold (at ITH /RUN)
IITHSTART
Start-Up Current Source
VITH/RUN = 0V
VFB
Regulated Feedback Voltage
(Note 5)
LTC3803E:
0°C ≤ TJ ≤ 85°C
–40°C ≤ TJ ≤ 85°C
LTC3803I:
0°C ≤ TJ ≤ 85°C
–40°C ≤ TJ ≤ 125°C
LTC3803H:
0°C ≤ TJ ≤ 85°C
–40°C ≤ TJ ≤ 150°C
LTC3803MP:
0°C ≤ TJ ≤ 85°C
–55°C ≤ TJ ≤ 150°C
gm
Error Amplifier Transconductance
ITH/RUN Pin Load = ±5μA (Note 5)
ΔVO(LINE)
Output Voltage Line Regulation
(Note 5)
ΔVO(LOAD)
Output Voltage Load Regulation
IFB
V
V
240
350
μA
40
40
90
110
μA
μA
0.15
0.09
0.28
0.28
0.45
0.46
V
V
0.2
0.3
0.4
μA
l
0.788
0.780
0.800
0.800
0.812
0.816
V
V
l
0.788
0.780
0.800
0.800
0.812
0.820
V
V
l
0.788
0.780
0.800
0.800
0.812
0.820
V
V
l
0.788
0.780
0.800
0.800
0.812
0.820
V
V
333
500
200
μA/V
0.05
mV/V
ITH/RUN Sinking 5μA (Note 5)
ITH/RUN Sourcing 5μA (Note 5)
3
3
mV/μA
mV/μA
VFB Input Current
(Note 5)
10
50
nA
fOSC
Oscillator Frequency
VITH/RUN = 1.3V
200
240
kHz
DCON(MIN)
Minimum Switch On Duty Cycle
VITH/RUN = 1.3V, VFB = 0.8V
6
8
%
DCON(MAX)
Maximum Switch On Duty Cycle
VITH/RUN = 1.3V, VFB = 0.8V
80
90
%
tRISE
Gate Drive Rise Time
CLOAD = 3000pF
40
ns
tFALL
Gate Drive Fall Time
CLOAD = 3000pF (Note 7)
40
ns
VIMAX
Peak Current Sense Voltage
RSL = 0 (Note 6)
LTC3803E
LTC3803H, LTC3803I
LTC3803MP
ISLMAX
Peak Slope Compensation Output Current
tSFST
Soft-Start Time
(Note 7)
180
70
l
l
l
90
90
85
100
100
100
115
120
120
mV
mV
mV
5
μA
1.4
ms
3803fc
3
LTC3803
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3803 is tested under pulsed load conditions such that TJ ≈ TA.
The LTC3803E is guaranteed to meet specifications from 0°C to 85°C
junction temperature. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3803I is guaranteed to
meet performance specifications over the –40°C to 125°C operating junction
temperature range, the LTC3803H is guaranteed to meet performance
specifications over the –40°C to 150°C operating junction temperature range
and the LTC3803MP is tested and guaranteed over the full –55°C to 150°C
operating junction temperature range. High junction temperatures degrade
operating lifetimes; operating lifetime is derated for junction temperatures
greater than 125°C. Note that the maximum ambient temperature consistent
with these specifications is determined by specific operating conditions in
conjunction with board layout, the rated package thermal impedance and
other environmental factors.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • 230°C/W).
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC3803 is tested in a feedback loop that servos VFB to the
output of the error amplifier while maintaining ITH/RUN at the midpoint of
the current limit range.
Note 6: Peak current sense voltage is reduced dependent on duty cycle
and an optional external resistor in series with the SENSE pin (RSL). For
details, refer to the programmable slope compensation feature in the
Applications Information section.
Note 7: Guaranteed by design.
TYPICAL PERFORMANCE CHARACTERISTICS
Reference Voltage
vs Supply Voltage
Reference Voltage vs Temperature
820
Reference Voltage
vs VCC Shunt Regulator Current
801.0
VCC = 8V
815
804
TA = 25°C
VCC ≤ VCLAMP1mA
800.8
TA = 25°C
803
800.6
805
800
795
790
802
800.4
VFB VOLTAGE (mV)
VFB VOLTAGE (mV)
VFB VOLTAGE (mV)
810
800.2
800.0
799.8
799.6
–30
0
30
60
90
TEMPERATURE (°C)
120
150
799.0
6
6.5
8
7.5
8.5
VCC SUPPLY VOLTAGE (V)
7
9
3803 G01
210
TA = 25°C
210
200
190
206
204
202
200
198
196
194
192
0
30
60
90
TEMPERATURE (°C)
120
150
3803 G04
15
ICC (mA)
20
25
TA = 25°C
206
204
202
200
198
196
194
192
190
–30
10
208
OSCILLATOR FREQUENCY (kHz)
220
5
Oscillator Frequency
vs VCC Shunt Regulator Current
208
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
210
230
0
3803 G03
Oscillator Frequency
vs Supply Voltage
VCC = 8V
180
–60
796
9.5
3803 F02
Oscillator Frequency
vs Temperature
240
799
797
799.2
780
–60
800
798
799.4
785
801
190
6
6.5
7.5
8
8.5
7
VCC SUPPLY VOLTAGE (V)
9
3803 G05
0
5
15
10
ICC (mA)
20
25
3803 G06
3803fc
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LTC3803
TYPICAL PERFORMANCE CHARACTERISTICS
VCC Undervoltage Lockout
Thresholds vs Temperature
VCC Shunt Regulator Voltage
vs Temperature
350
10.0
8.5
9.9
VTURNON
8.0
9.7
7.0
9.6
6.5
VTURNOFF
6.0
ICC = 1mA
9.4
9.3
5.0
9.2
4.5
9.1
–30
ICC = 25mA
9.5
5.5
4.0
–60
0
30
60
90
TEMPERATURE (°C)
120
–30
0
30
60
90
TEMPERATURE (°C)
120
3803 G07
60
50
40
30
20
10
120
800
400
350
300
250
200
100
–60
150
–30
0
30
60
90
120
120
150
VCC = VTURNON + 0.1V
VITH/RUN = 0V
600
500
400
300
200
100
0
–60
150
–30
0
30
60
90
TEMPERATURE (°C)
3803 G11
Peak Current Sense Voltage
vs Temperature
120
150
3803 G12
Soft-Start Time vs Temperature
4.0
VCC = 8V
3.5
110
3.0
SOFT-START TIME (ms)
115
105
100
95
90
2.5
2.0
1.5
1.0
0.5
85
80
–60
700
TEMPERATURE (°C)
3803 G10
120
0
30
60
90
TEMPERATURE (°C)
ITH /RUN Start-Up Current Source
vs Temperature
150
0
30
60
90
TEMPERATURE (°C)
–30
3803 G09
ITH/RUN PIN CURRENT SOURCE (nA)
SHUTDOWN THRESHOLD (mV)
70
SENSE PIN VOLTAGE (mV)
START-UP SUPPLY CURRENT (μA)
200
–60
150
450
80
–30
250
ITH /RUN Shutdown Threshold
vs Temperature
VCC = VTURNON – 0.1V
0
–60
275
3803 G08
Start-Up ICC Supply Current
vs Temperature
90
300
225
9.0
–60
150
SUPPLY CURRENT (μA)
7.5
VCC = 8V
VITH/RUN = 1.3V
325
9.8
VCC (V)
VCC UNDERVOLTAGE LOCKOUT (V)
9.0
ICC Supply Current
vs Temperature
–30
0
30
60
90
120
150
0
–60
–30
0
30
60
90
120
150
TEMPERATURE (°C)
TEMPERATURE (°C)
3803 G13
3803 G14
3803fc
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LTC3803
PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It serves
as the error amplifier compensation point as well as the
run/shutdown control input. Nominal voltage range is 0.7V
to 1.9V. Forcing this pin below the shutdown threshold
(VITHSHDN) causes the LTC3803 to shut down. In shutdown
mode, the NGATE pin is held low.
SENSE (Pin 4): This pin performs two functions. It monitors
switch current by reading the voltage across an external
current sense resistor to ground. It also injects a current
ramp that develops slope compensation voltage across
an optional external programming resistor.
GND (Pin 2): Ground Pin.
VCC (Pin 5): Supply Pin. Must be closely decoupled to
GND (Pin 2).
VFB (Pin 3): Receives the feedback voltage from an external
resistive divider across the output.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VCC.
BLOCK DIAGRAM
5
VCC
0.3μA 0.28V
800mV
REFERENCE
VCC
SHUNT
REGULATOR
+
SHUTDOWN
COMPARATOR
VCC < VTURNON
–
SHUTDOWN
SOFTSTART
CLAMP
+
3
VFB
GND
2
UNDERVOLTAGE
LOCKOUT
–
ERROR
AMPLIFIER
CURRENT
COMPARATOR
VCC
R
+
Q
S
–
20mV
1.2V
200kHz
OSCILLATOR
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
GATE
DRIVER NGATE
SLOPE
COMP
CURRENT
RAMP
SENSE
1
6
4
ITH/RUN
3803 BD
3803fc
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LTC3803
OPERATION
The LTC3803 is a constant frequency current mode controller for flyback and DC/DC boost converter applications in
a tiny ThinSOT package. The LTC3803 is designed so that
none of its pins need to come in contact with the input or
output voltages of the power supply circuit of which it is
a part, allowing the conversion of voltages well beyond
the LTC3803’s absolute maximum ratings.
Main Control Loop
Due to space limitations, the basics of current mode
DC/DC conversion will not be discussed here; instead, the
reader is referred to the detailed treatment in Application
Note 19, or in texts such as Abraham Pressman’s Switching Power Supply Design.
Please refer to the Block Diagram and the Typical Application on the front page of this data sheet. An external
resistive voltage divider presents a fraction of the output
voltage to the VFB pin. The divider must be designed so
that when the output is at the desired voltage, the VFB pin
voltage will equal the 800mV from the internal reference.
If the load current increases, the output voltage will decrease slightly, causing the VFB pin voltage to fall below
800mV. The error amplifier responds by feeding current
into the ITH/RUN pin. If the load current decreases, the
VFB voltage will rise above 800mV and the error amplifier
will sink current away from the ITH/RUN pin.
The voltage at the ITH/RUN pin commands the pulse-width
modulator formed by the oscillator, current comparator
and RS latch. Specifically, the voltage at the ITH/RUN pin
sets the current comparator’s trip threshold. The current
comparator monitors the voltage across a current sense
resistor in series with the source terminal of the external
MOSFET. The LTC3803 turns on the external power MOSFET
when the internal free-running 200kHz oscillator sets
the RS latch. It turns off the MOSFET when the current
comparator resets the latch or when 80% duty cycle is
reached, whichever happens first. In this way, the peak
current levels through the flyback transformer’s primary
and secondary are controlled by the ITH/RUN voltage.
Since the ITH/RUN voltage is increased by the error amplifier whenever the output voltage is below nominal, and
decreased whenever output voltage exceeds nominal, the
voltage regulation loop is closed. For example, whenever
the load current increases, output voltage will decrease
slightly, and sensing this, the error amplifier raises the
ITH/RUN voltage by sourcing current into the ITH/RUN pin,
raising the current comparator threshold, thus increasing
the peak currents through the transformer primary and
secondary. This delivers more current to the load, bringing
the output voltage back up.
The ITH/RUN pin serves as the compensation point for
the control loop. Typically, an external series RC network
is connected from ITH/RUN to ground and is chosen for
optimal response to load and line transients. The impedance
of this RC network converts the output current of the error
amplifier to the ITH/RUN voltage which sets the current
comparator threshold and commands considerable influence over the dynamics of the voltage regulation loop.
Start-Up/Shutdown
The LTC3803 has two shutdown mechanisms to disable
and enable operation: an undervoltage lockout on the
VCC supply pin voltage, and a forced shutdown whenever
external circuitry drives the ITH/RUN pin low. The LTC3803
transitions into and out of shutdown according to the state
diagram (Figure 1).
LTC3803
SHUT DOWN
VCC < VTURNOFF
(NOMINALLY 5.7V)
> VITHSHDN
V
VITH/RUN < VITHSHDN ITH/RUN
AND VCC > VTURNON
(NOMINALLY 0.28V)
(NOMINALLY 8.7V)
LTC3803
ENABLED
3803 F01
Figure 1. Start-Up/Shutdown State Diagram
3803fc
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LTC3803
OPERATION
The undervoltage lockout (UVLO) mechanism prevents the
LTC3803 from trying to drive a MOSFET with insufficient
VGS. The voltage at the VCC pin must exceed VTURNON
(nominally 8.7V) at least momentarily to enable LTC3803
operation. The VCC voltage is then allowed to fall to VTURNOFF
(nominally 5.7V) before undervoltage lockout disables the
LTC3803. This wide UVLO hysteresis range supports the
use of a bias winding on the flyback transformer to power
the LTC3803—see the section Powering the LTC3803.
The ITH/RUN pin can be driven below VITHSHDN (nominally
0.28V) to force the LTC3803 into shutdown. An internal
0.3μA current source always tries to pull this pin towards
VCC. When the ITH/RUN pin voltage is allowed to exceed
VITHSHDN, and VCC exceeds VTURNON, the LTC3803 begins
to operate and an internal clamp immediately pulls the
ITH/RUN pin up to about 0.7V. In operation, the ITH/RUN
pin voltage will vary from roughly 0.7V to 1.9V to represent
current comparator thresholds from zero to maximum.
Internal Soft-Start
An internal soft-start feature is enabled whenever the
LTC3803 comes out of shutdown. Specifically, the ITH/
RUN voltage is clamped and is prevented from reaching
maximum until roughly 1.4ms has passed. This allows
the input and output currents of LTC3803-based power
supplies to rise in a smooth and controlled manner on
start-up.
Powering the LTC3803
In the simplest case, the LTC3803 can be powered from
a high voltage supply through a resistor. A built-in shunt
regulator from the VCC pin to GND will draw as much
current as needed through this resistor to regulate the
VCC voltage to around 9.5V as long as the VCC pin is not
forced to sink more than 25mA. This shunt regulator is
always active, even when the LTC3803 is in shutdown,
since it serves the vital function of protecting the VCC pin
from seeing too much voltage.
For higher efficiency or for wide VIN range applications,
flyback controllers are typically powered through a separate
bias winding on the flyback transformer. The LTC3803 has
the wide UVLO hysteresis and small VCC supply current
draw that is needed to support such bootstrapped hysteretic
start-up schemes.
The VCC pin must be bypassed to ground immediately
adjacent to the IC pins with a minimum of a 10μF ceramic
or tantalum capacitor. Proper supply bypassing is necessary to supply the high transient currents required by the
MOSFET gate driver.
Adjustable Slope Compensation
The LTC3803 injects a 5μA peak current ramp out through
its SENSE pin which can be used for slope compensation in
designs that require it. This current ramp is approximately
linear and begins at zero current at 6% duty cycle, reaching peak current at 80% duty cycle. Additional details are
provided in the Applications Information section.
3803fc
8
LTC3803
APPLICATIONS INFORMATION
Many LTC3803 application circuits can be derived from
the topology shown in Figure 2.
The LTC3803 itself imposes no limits on allowed power
output, input voltage VIN or desired regulated output voltage
VOUT; these are all determined by the ratings on the external
power components. The key factors are: Q1’s maximum
drain-source voltage (BVDSS), on-resistance (RDS(ON))
and maximum drain current, T1’s saturation flux level and
winding insulation breakdown voltages, CIN and COUT’s
maximum working voltage, ESR, and maximum ripple
current ratings, and D1 and RSENSE’s power ratings.
T1
LBIAS
D2
R3
•
VIN
D1
VOUT
•
RSTART
CIN LPRI
LSEC
COUT
•
5
CVCC
1
CC
2
VCC
ITH/RUN NGATE
LTC3803
GND
SENSE
6
4
VFB
R1
3
Q1
RSL
RSENSE
R2
3803 F02
Figure 2. Typical LTC3803 Application Circuit
TRANSFORMER DESIGN CONSIDERATIONS
Transformer specification and design is perhaps the
most critical part of applying the LTC3803 successfully.
In addition to the usual list of caveats dealing with high
frequency power transformer design, the following should
prove useful.
Turns Ratios
Due to the use of the external feedback resistor divider
ratio to set output voltage, the user has relative freedom
in selecting transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2,
etc. can be employed which yield more freedom in setting
total turns and mutual inductance. Simple integer turns
ratios also facilitate the use of “off-the-shelf” configurable transformers such as the Coiltronics VERSA-PAC™
series in applications with high input to output voltage
ratios. For example, if a 6-winding VERSA-PAC is used
with three windings in series on the primary and three
windings in parallel on the secondary, a 3:1 turns ratio
will be achieved.
Turns ratio can be chosen on the basis of desired duty
cycle. However, remember that the input supply voltage
plus the secondary-to-primary referred version of the
flyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating.
SELECTING FEEDBACK RESISTOR DIVIDER VALUES
Leakage Inductance
The regulated output voltage is determined by the resistor
divider across VOUT (R1 and R2 in Figure 2). The ratio
of R2 to R1 needed to produce a desired VOUT can be
calculated:
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to occur after the
output switch (Q1) turn-off. This is increasingly prominent
at higher load currents, where more stored energy must
be dissipated. In some cases a “snubber” circuit will be
required to avoid overvoltage breakdown at the MOSFET’s
drain node. Application Note 19 is a good reference on
snubber design.
R2 =
VOUT – 0.8V
• R1
0.8V
Choose resistance values for R1 and R2 to be as large as
possible in order to minimize any efficiency loss due to
the static current drawn from VOUT, but just small enough
so that when VOUT is in regulation, the error caused by
the nonzero input current to the VFB pin is less than 1%.
A good rule of thumb is to choose R1 to be 80k or less.
A bifilar or similar winding technique is a good way to
minimize troublesome leakage inductances. However,
remember that this will limit the primary-to-secondary
breakdown voltage, so bifilar winding is not always
practical.
3803fc
9
LTC3803
APPLICATIONS INFORMATION
CURRENT SENSE RESISTOR CONSIDERATIONS
The external current sense resistor (RSENSE in Figure 2)
allows the user to optimize the current limit behavior for
the particular application. As the current sense resistor
is varied from several ohms down to tens of milliohms,
peak switch current goes from a fraction of an ampere to
several amperes. Care must be taken to ensure proper
circuit operation, especially with small current sense
resistor values.
For example, a peak switch current of 5A requires a
sense resistor of 0.020Ω. Note that the instantaneous
peak power in the sense resistor is 0.5W and it must be
rated accordingly. The LTC3803 has only a single sense
line to this resistor. Therefore, any parasitic resistance
in the ground side connection of the sense resistor will
increase its apparent value. In the case of a 0.020Ω sense
resistor, one milliohm of parasitic resistance will cause a
5% reduction in peak switch current. So the resistance of
printed circuit copper traces and vias cannot necessarily
be ignored.
PROGRAMMABLE SLOPE COMPENSATION
The LTC3803 injects a ramping current through its SENSE
pin into an external slope compensation resistor (RSL in
Figure 2). This current ramp starts at zero right after the
NGATE pin has been high for the LTC3803’s minimum
duty cycle of 6%. The current rises linearly towards a
peak of 5μA at the maximum duty cycle of 80%, shutting
off once the NGATE pin goes low. A series resistor (RSL)
connecting the SENSE pin to the current sense resistor
(RSENSE) thus develops a ramping voltage drop. From
the perspective of the SENSE pin, this ramping voltage
adds to the voltage across the sense resistor, effectively
reducing the current comparator threshold in proportion
to duty cycle. This stabilizes the control loop against
subharmonic oscillation. The amount of reduction in the
current comparator threshold (ΔVSENSE) can be calculated
using the following equation:
ΔVSENSE =
Duty Cycle – 6%
• 5μA • RSL
74%
Note: LTC3803 enforces 6% < Duty Cycle < 80%.
A good starting value for RSL is 5.9k, which gives a 30mV
drop in current comparator threshold at 80% duty cycle.
Designs not needing slope compensation may replace RSL
with a short circuit.
INTERNAL WIDE HYSTERESIS UNDERVOLTAGE
LOCKOUT
The LTC3803 is designed to implement DC/DC converters
operating from input voltages of typically 48V or more.
The standard operating topology employs a third transformer winding (LBIAS in Figure 2) on the primary side that
provides power for the LTC3803 via its VCC pin. However,
this arrangement is not inherently self-starting. Start-up is
affected by the use of an external “trickle-charge” resistor
(RSTART in Figure 2) and the presence of an internal wide
hysteresis undervoltage lockout circuit that monitors VCC
pin voltage. Operation is as follows:
“Trickle charge” resistor RSTART is connected to VIN and
supplies a small current, typically on the order of 100μA
to 120μA, to charge CVCC. After some time, the voltage
on CVCC reaches the VCC turn-on threshold. The LTC3803
then turns on abruptly and draws its normal supply current. The NGATE pin begins switching and the external
MOSFET (Q1) begins to deliver power. The voltage on
CVCC begins to decline as the LTC3803 draws its normal
supply current, which exceeds that delivered by RSTART.
After some time, typically tens of milliseconds, the output
voltage approaches its desired value. By this time, the third
transformer winding is providing virtually all the supply
current required by the LTC3803.
One potential design pitfall is undersizing the value of
capacitor CVCC. In this case, the normal supply current
drawn by the LTC3803 will discharge CVCC too rapidly;
before the third winding drive becomes effective, the VCC
turn-off threshold will be reached. The LTC3803 turns off,
3803fc
10
LTC3803
APPLICATIONS INFORMATION
and the VCC node begins to charge via RSTART back up to
the VCC turn-on threshold. Depending on the particular
situation, this may result in either several on-off cycles
before proper operation is reached or permanent relaxation
oscillation at the VCC node.
VIN
RVCC
LTC3803
VCC
GND
CVCC
3803 F03
Component selection is as follows:
Resistor RSTART should be made small enough to yield a
worst-case minimum charging current greater than the
maximum rated LTC3803 start-up current, to ensure there
is enough current to charge CVCC to the VCC turn-on threshold. It should be made large enough to yield a worst-case
maximum charging current less than the minimum rated
LTC3803 supply current, so that in operation, most of the
LTC3803’s supply current is delivered through the third
winding. This results in the highest possible efficiency.
Capacitor CVCC should then be made large enough to avoid
the relaxation oscillation behavior described above. This
is complicated to determine theoretically as it depends on
the particulars of the secondary circuit and load behavior.
Empirical testing is recommended.
The third transformer winding should be designed so that
its output voltage, after accounting for the D2’s forward
voltage drop, exceeds the maximum VCC turn-off threshold.
Also, the third winding’s nominal output voltage should
be at least 0.5V below the minimum rated VCC clamp voltage to avoid running up against the LTC3803’s VCC shunt
regulator, needlessly wasting power.
Figure 3. Powering the LTC3803 Via the Internal Shunt Regulator
The shunt regulator can draw up to 25mA through the
VCC pin to GND to drop enough voltage across RVCC to
regulate VCC to around 9.5V. For applications where VIN
is low enough such that the static power dissipation in
RVCC is acceptable, using the VCC shunt regulator is the
simplest way to power the LTC3803.
EXTERNAL PREREGULATOR
The circuit in Figure 4 shows a third way to power the
LTC3803. An external series preregulator consisting of
series pass transistor Q1, Zener diode D1, and bias resistor RB brings VCC to at least 7.6V nominal, well above the
maximum rated VCC turn-off threshold. Resistor RSTART
momentarily charges the VCC node up to the VCC turn-on
threshold, enabling the LTC3803.
VIN
RB
Q1
RSTART
LTC3803
VCC
VCC SHUNT REGULATOR
In applications including a third transformer winding,
the internal VCC shunt regulator serves to protect the
LTC3803 from overvoltage transients as the third winding is powering up.
D1
8.2V
CVCC
GND
3803 F04
Figure 4. Powering the LTC3803 with an External Preregulator
In applications where a third transformer winding is
undesirable or unavailable, the shunt regulator allows
the LTC3803 to be powered through a single dropping
resistor from VIN to VCC, in conjunction with a bypass
capacitor, CVCC, that closely decouples VCC to GND (see
Figure 3). This simplicity comes at the expense of reduced
efficiency due to the static power dissipation in the RVCC
dropping resistor.
3803fc
11
LTC3803
TYPICAL APPLICATIONS
2W Isolated Housekeeping Telecom Converter
BAS516
PRIMARY SIDE
10V, 100mA
OUTPUT
T1
•
2.2μF
1μF
VIN
36V TO 75V
•
22k
806Ω
2.2μF
BAS516
9.2k
1nF
BAS516
1k
1
LTC3803
6
ITH/RUN NGATE
2
5
3
GND
VFB
VCC
SENSE
4
220k
•
SECONDARY SIDE
10V, 100mA
OUTPUT
SECONDARY
SIDE GROUND
FDC2512
T1: PULSE ENGINEERING PA0648
OR TYCO TTI8698
5.6k
1μF
PRIMARY GROUND
0.1Ω
3803 TA03
3803fc
12
LTC3803
TYPICAL APPLICATIONS
4:1 Input Range 3.3V Output Isolated Flyback DC/DC Converter
T1
PA1277NL
VIN+
18V TO 72V
•
2.2μF
220k
MMBTA42
100k
PDZ6.8B
100μF
6.3V
×3
PDS1040
•
GND
BAS516
68Ω
150pF
VCC
10Ω
22Ω BAS516
680Ω
•
1
ITH/RUN
2
0.1μF
6
GND
FDC2512
GATE
VCC
5
LTC3803
3
4
SENSE
VFB
VOUT+
4.7k
BAT760
0.1μF
0.040Ω
270Ω
VCC
1
6.8k
BAS516
PS2801-1
0.1μF
1
2
0.33μF
BAS516
2
3
VIN
OPTO
LT4430
GND
OC
COMP
FB
VOUT+
6
2.2nF
5
56k
47pF
100k
4
22.1k
3803 TA05
Efficiency vs Load Current
84
82
80
EFFICIENCY (%)
VIN–
VOUT+
3.3V
3A
78
76
74
72
70
VIN = 48V
VIN = 24V
0
1
2
IOUT (A)
3
4
3803 TA05a
3803fc
13
LTC3803
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3803fc
14
LTC3803
REVISION HISTORY
(Revision history begins at Rev C)
REV
DATE
DESCRIPTION
PAGE NUMBER
C
6/10
MP-grade part added. Reflected throughout the data sheet.
1 to 16
3803fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3803
TYPICAL APPLICATIONS
Synchronous Flyback 3.3VOUT
EFFICIENCY (%)
91
90% Efficient Synchronous Flyback Converter
VIN
36V TO 72V
VOUT*
3.3V
1.5A
T1
•
Q2
CIN
270k
90
89
CO
•
88
0.5
D1
33k 1
2
8.06k
3
ITH/RUN
GATE
LTC3803
GND
VCC
VFB = 0.8V SENSE
6
Q1
5
560Ω
4
5k
•
1μF
10V
VOUT
2.0
3803 TA04b
Synchronous Flyback 5VOUT
92
3803 TA04a
25.5k*
RFB
1.5
1.0
OUTPUT CURRENT (A)
0.1μF
RCS
T1: PULSE ENGINEERING PA1006
Q1: FAIRCHILD FDC2512
Q2: VISHAY Si9803
D1: PHILIPS BAS516
CIN: TDK 1μF, 100V, X5R
CO: TDK 100μF, 6.3V, X5R
RCS: VISHAY OR IRC, 80mΩ
*FOR 5V OUTPUT CHANGE RFB TO 42.2k
91
90
EFFICIENCY (%)
1n
89
88
87
86
85
0.5
1.0
1.5
2.0
OUTPUT CURRENT (A)
2.5
3803 TA04c
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3573
Isolated Flyback Switching Regulator with 60V
Integrated Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or Third Winding Required, Up to 7W
Output Power, MSOP-16E
LTC3805/
LTC3805-5
Adjustable Constant Frequency Flyback, Boost, SEPIC
DC/DC Controller
VIN and VOUT Limited Only by External Components, 3mm × 3mm DFN-10,
MSOP-10E Packages
LTC3873/
LTC3873-5
No RSENSE™ Constant Frequency Flyback, Boost, SEPIC
Controller
VIN and VOUT Limited Only by External Components, 8-pin ThinSOT or
2mm × 3mm DFN-8 Packages
LT3757
Boost, Flyback, SEPIC and Inverting Controller
2.9V ≤ VIN ≤ 40V, 100kHz to 1MHz Programmable Operating Frequency,
3mm × 3mm DFN-10 and MSOP-10E Package
LT3758
Boost, Flyback, SEPIC and Inverting Controller
5.5V ≤ VIN ≤ 100V, 100kHz to 1MHz Programmable Operating Frequency,
3mm × 3mm DFN-10 and MSOP-10E
LTC1871/LTC1871-1/ Wide Input Range, No RSENSE Low Quiescent Current
LTC1871-7
Flyback, Boost and SEPIC Controller
Programmable Operating Frequency, 2.5V ≤ VIN ≤ 36V, Burst Mode®
Operation at Light Load, MSOP-10
3803fc
16 Linear Technology Corporation
LT 0610 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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