LTC3873
No RSENSETM Constant
Frequency Current Mode
Boost/Flyback/SEPIC
DC/DC Controller
FEATURES
DESCRIPTION
n
n
n
n
n
n
n
VIN and VOUT Limited Only by External Components
Internal or Programmable External Soft-Start
Constant Frequency 200kHz Operation
Adjustable Current Limit
Current Sense Resistor Optional
Maximum 60V on SW Node with RDS(ON) Sensing
±1.5% Voltage Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
n Low Quiescent Current: 300µA
n Low Profile (1mm) ThinSOTTM and (0.75mm)
2mm × 3mm DFN Package
The LTC®3873 is a constant frequency current mode,
boost, flyback or SEPIC DC/DC controller that drives an
N-channel power MOSFET in high input and output voltage converter applications. Soft-start can be programmed
using an external capacitor.
APPLICATIONS
The LTC3873 is available in 8-lead ThinSOT and 2mm ×
3mm DFN packages.
n
The LTC3873 provides ±1.5% output voltage accuracy and
consumes only 300μA quiescent current during normal
operation and only 55μA during micropower start-up.
Using a 9.3V internal shunt regulator, the LTC3873 can
be powered from a high input voltage through a resistor
or it can be powered directly from a low impedance DC
voltage of 9V or less.
Telecom Power Supplies
n 42V and 12V Automotive Power Supplies
n Portable Electronic Equipment
PARAMETER
VCC UV+
VCC UV–
n
LTC3873
8.4V
4V
LTC3873-5
4.1V
2.9V
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
TYPICAL APPLICATION
Efficiency and Power Loss vs Load Current
5V Output Nonisolated Telecom Power Supply
221k
10µF
10V
X5R
7.5k
2.2nF
0.1µF
D1
•
•
VCC
ITH
NGATE
LTC3873
RUN/SS
GND
SW
IPRG
VFB
12.1k
100µF
6.3V
X5R
×3
100
VOUT
5V
2A MAX
90
M1
3000
EFFICIENCY
80
2500
70
2000
60
50
1000
30
VIN = 72V
VIN = 60V
VIN = 48V
VIN = 36V
20
10
68mΩ
0
38.3k
1500
POWER LOSS
40
500
10
1000
10
3873 TA01a
POWER LOSS (mW)
4.7µF
100V
X5R
D2
T1
EFFICIENCY (%)
VIN
36V TO 72V
LOAD CURRENT (mA)
3873 TA01b
3873fb
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1
LTC3873
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VCC to GND
Low Impedance Source............................ –0.3V to 9V
Current Fed...........................................25mA Into VCC
RUN/SS......................................................... –0.3V to 9V
IPRG Voltage..................................–0.3V to (VCC + 0.3V)
VFB, ITH Voltages........................................ –0.3V to 2.4V
SW Voltage................................................. –0.3V to 60V
Operating Temperature Range (Note 2).... –40°C to 85°C
Junction Temperature (Note 3).............................. 125°C
Storage Temperature Range.................... –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
TS8 Package...................................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
IPRG 1
ITH 2
VFB 3
GND 4
8 SW
7 RUN/SS
6 VCC
5 NGATE
8
VFB 2
7
VCC
6
RUN/SS
5
SW
ITH 3
9
IPRG 4
TS8 PACKAGE
8-LEAD PLASTIC TSOT-23
NGATE
GND 1
DDB PACKAGE
8-LEAD (3mm × 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 230°C/W
TJMAX = 125°C, θJA = 76°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
(http://www.linear.com/product/LTC3873#orderinfo)
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3873ETS8#PBF
LTC3873ETS8\#TRPBF
LTCSN
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC3873EDDB#PBF
LTC3873EDDB#TRPBF
LCSK
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3873ETS8
LTC3873ETS8#TR
LTCSN
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC3873EDDB
LTC3873EDDB#TR
LCSK
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
2
3873fb
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LTC3873
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 9V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Shutdown
UVLO
Typicals
VITH = 1.9V
VRUN/SS = 0V
VCC = UVLO Threshold – 100mV, VRUN/SS = VCC
300
55
45
400
100
60
µA
µA
µA
Undervoltage Lockout Threshold
VCC Rising
VCC Falling
VCC Hysteresis
l
l
l
7.9
3.5
4.0
8.4
4.0
4.4
8.8
4.4
4.8
V
V
V
Shutdown Threshold (at RUN/SS)
VRUN/SS Falling
VRUN/SS Rising
l
0.5
0.6
0.7
0.8
0.9
1.0
V
V
Regulated Feedback Voltage
(Note 5)
l
1.182
1.2
1.218
V
Feedback Voltage Line Regulation
5.6V < VCC < 9V (Note 5)
0.12
mV/V
Feedback Voltage Load Regulation
VITH = 1.6V (Note 5)
VITH = 1V (Note 5)
0.05
–0.05
%
%
VFB Input Current
(Note 5)
Maximum Duty Cycle
RUN/SS Pull-Up Current
VRUN/SS = 0V
VRUN/SS = 1.3V
25
50
nA
70
78
84
%
1.5
5
3
15
4.5
25
µA
µA
ISLMAX, Peak Slope Compensation Current
20
Oscillator Frequency
160
200
µA
240
40
kHz
Gate Drive Rise Time
CLOAD = 3000pF (Note 6)
Gate Drive Fall Time
CLOAD = 3000pF (Note 6)
Peak Current Sense Voltage
IPRG = GND
IPRG = Float
IPRG = VIN
l
l
l
95
165
265
110
185
295
125
210
325
mV
mV
mV
VIN Shunt Regulator Voltage
IIN = 1mA, IIN = 25mA, VRUN/SS = 0V
l
9
9.3
9.6
V
40
Default Internal Soft-Start
3.3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3873E is guaranteed to meet performance specifications
from 0°C to 85°C junction temperature. Specifications over the –40°C
to 85°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA)
ns
ns
ms
Note 4: The dynamic input supply current is higher due to power MOSFET
gate charging (QG • fOSC). See Applications Information.
Note 5: The LTC3873 is tested in a feedback loop which servos VFB to
the reference voltage with the ITH pin forced to the midpoint of its voltage
range (0.7V ≤ VITH ≤ 1.9V, midpoint = 1.3V).
Note 6: Rise and fall times are measured at 10% and 90% levels.
VCC = 5.6V.
3873fb
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3
LTC3873
TYPICAL PERFORMANCE CHARACTERISTICS
Feedback Voltage Line Regulation
1.24
1.2020
1.23
1.2015
1.22
1.21
1.20
1.19
0
1.2010
1.2005
40 60 80 100 120
TEMPERATURE (°C)
1.0
0.5
5
6
7
0
10
9
8
3873 G03
Gate Drive Rise and Fall Time
vs CLOAD
Shutdown IQ vs Temperature
70
100
80
65
90
70
50
45
40
35
30
80
60
70
50
TIME (ns)
SHUTDOWN MODE IQ (µA)
55
40
30
4
5
7
6
VIN (V)
8
9
10
RISE TIME
60
FALL TIME
50
40
30
20
20
10
25
3
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
RUN/SS VOLTAGE (V)
3873 G02
Shutdown Mode IQ vs VIN
60
0
VIN (V)
3873 G01
SHUTDOWN MODE IQ (µA)
1.5
1.2000
1.1990
20
VIN = 5V
2.0
1.1995
1.18
–60 –40 –20
20
ITH Voltage vs RUN/SS Voltage
2.5
ITH VOLTAGE (V)
1.2025
VFB VOLTAGE (V)
VFB VOLTAGE (V)
Feedback Voltage vs Temperature
1.25
10
0
–60 –40 –20
0
0 20 40 60 80 100 120
TEMPERATURE (°C)
3873 G04
0
2000
6000
4000
CLOAD (pF)
8000
10000
3873 G06
3873 G05
Shunt Regulation Voltage
vs ISHUNT
RUN Threshold vs Temperature
1.0
10.2
10.1
0.8
0.7
REGULATION VOLTAGE (V)
RUN THRESHOLDS (V)
0.9
RISING
FALLING
0.6
10.0
9.9
9.8
9.7
9.6
9.5
9.4
9.3
0.5
–60 –40 –20
0 20 40 60 80 100 120
TEMPERATURE (°C)
9.2
0
3873 G07
4
5
10
15
20 25 30
ISHUNT (mA)
35
40
45
3873 G08
3873fb
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LTC3873
TYPICAL PERFORMANCE CHARACTERISTICS
Maximum Sense Threshold
vs Temperature
Frequency vs Temperature
300
MAXIMUM SENSE THRESHOLD (mV)
250
FREQUECY (kHz)
230
210
190
170
150
–60 –40 –20
0 20 40 60 80 100 120
TEMPERATURE (°C)
IPRG = VIN
250
200
150
IPRG = GND
100
50
0
–60 –40 –20
0 20 40 60 80 100 120
TEMPERATURE (°C)
3873 G10
3873 G09
PIN FUNCTIONS
IPRG = FLOAT
(TS8/DDB)
IPRG (Pin 1/Pin 4): Current Sense Limit Select Pin.
ITH (Pin 2/Pin 3): This pin serves as the error amplifier
compensation point. Nominal voltage range for this pin
is 0.7V to 1.9V.
VFB (Pin 3/Pin 2): This pin receives the feedback voltage
from an external resistor divider across the output.
GND (Pin 4/Pin 1): Ground Pin.
NGATE (Pin 5/Pin 8): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VIN.
RUN/SS (Pin 7/Pin 6): Shutdown and External Soft-Start
Pin. In shutdown, all functions are disabled and the NGATE
pin is held low.
SW (Pin 8/Pin 5): Switch node connection to inductor and
current sense input pin through external slope compensation resistor. Normally, the external N-channel MOSFET’s
drain is connected to this pin.
Exposed Pad (NA/Pin 9): Ground. Must be soldered to PCB
for electrical contact and rated thermal performance.
VCC (Pin 6/Pin 7): Supply Pin. This pin must be closely
decoupled to GND (Pin 4).
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5
LTC3873
FUNCTIONAL DIAGRAM
GND
VCC
UV
UNDERVOLTAGE
LOCKOUT
VOLTAGE
REFERENCE
1.2V
SW
VCC 9.5V
SHUNT
REGULATOR
SLOPE
COMPENSATION
–
SHUTDOWN
COMPARATOR
+
IPRG
CURRENT
COMPARATOR
3µA
ILIM
+
SHDN
RUN/SS
–
ITH
BUFFER
RS
LATCH
R
S
Q
CURRENT LIMIT
CLAMP
VIN
SWITCHING
LOGIC CIRCUIT
+
INTERNAL
SOFT-START
RAMP
NGATE
VFB
–
ERROR
AMPLIFIER
200kHz
OSCILLATOR AND
MAX DUTY CYCLE
1.2V
ITH
3873 FD
6
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LTC3873
OPERATION
Main Control Loop
The LTC3873 is a general purpose N-channel switching
DC/DC converter for boost, flyback and SEPIC applications.
Its No RSENSE sensing technique improves efficiency,
increases power density and reduces the cost of the
overall solution.
For circuit operation, please refer to the Functional Diagram
of the IC and the Typical Application on the front page.
During normal operation, the power MOSFET is turned
on when the oscillator sets the PWM latch and is turned
off when the current comparator resets the latch. The
divided-down output voltage is compared to an internal
1.2V reference by the error amplifier, which outputs an
error signal at the ITH pin. The voltage on the ITH pin
sets the current comparator input threshold. When the
load current increases, a fall in the VFB voltage relative to
the reference voltage causes the ITH pin to rise, causing
the current comparator to trip at a higher peak inductor
current value. The average inductor current will therefore
rise until it equals the load current, thereby maintaining
output regulation.
L
VIN
D
VOUT
VCC
+
SW
VSW
LTC3873
COUT
The LTC3873 can be used either by sensing the voltage
drop across the power MOSFET or by connecting the SW
pin to a conventional sensing resistor in the source of the
power MOSFET. Sensing the voltage across the power
MOSFET maximizes converter efficiency and minimizes the
component count; the maximum rating for this pin, 60V,
allows MOSFET sensing in a wide output voltage range.
Shunt Regulator
A built-in shunt regulator from the VCC pin to GND limits
the voltage on the VCC pin to approximately 9.3V as long as
the shunt regulator is not forced to sink more than 25mA.
The shunt regulator permits the use of a wide variety of
powering schemes that exceed the LTC3873’s absolute
maximum ratings. Further details on powering schemes
are described in the Application Information section.
Start-Up/Shutdown
The LTC3873 has two shutdown mechanisms to disable
and enable operation: an undervoltage lockout on the
VCC supply pin voltage and a threshold RUN/SS pin. The
LTC3873 transitions into and out of shutdown according
to the state diagram shown in Figure 3.
The undervoltage lockout (UVLO) mechanism prevents the
LTC3873 from trying to drive a MOSFET with insufficient
voltage. The voltage at the VCC pin must exceed VTURNON
NGATE
GND
GND
3873 F01
LTC3873
SHUT DOWN
Figure 1. SW Pin (Internal Sense Pin)
Connection for Maximum Efficiency
L
VIN
D
VOUT
VSW
VCC
NGATE
+
VIN < VTURNOFF
(NOMINALLY 4V)
VRUN/SS < VSHDN
(NOMINALLY 0.8V)
VRUN/SS > VSHDN
AND VIN > VTURNON
(NOMINALLY 8.4V)
COUT
LTC3873
SW
GND
LTC3873
ENABLED
RSENSE
GND
3873 F02
3873 F03
Figure 2. SW Pin (Internal Sense Pin)
Connection for Sensing Resistor
Figure 3. Start-Up/Shutdown State Diagram
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7
LTC3873
OPERATION
(nominally 8.4V) at least momentarily to enable LTC3873
operation. The VCC voltage is then allowed to fall to VTURNOFF
(nominally 4V) before undervoltage lockout disables the
LTC3873. This wide UVLO hysteresis range supports the
use of trickle charger on the flyback transformer to power
the LTC3873—see the section, VCC Bias Power. The RUN/SS
pin can be driven below VSHDN (nominally 0.7V) to force
the LTC3873 into shutdwn. When the chip is off, the input
supply current is typically only 55μA.
Soft-Start
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Leave the RUN/SS pin open to use the internal 3.3ms
soft-start. During the internal soft-start, a voltage ramp
limits the VITH. 3.3ms is required for ITH to ramp from
zero current level to full current level. The soft-start can
be lengthened by placing an external capacitor from the
RUN/SS pin to the GND. A 3μA current will charge the
capacitor, pulling the RUN/SS pin above the shutdown
threshold and a 15µA pull-up current will continue to ramp
RUN/SS to limit VITH during the start-up. When RUN/SS
is driven by an external logic, a minimum of 2.75V logic
is recommended to allow the maximum ITH range.
Light Load Operation
Under very light load current conditions, the ITH pin voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will assure that the current
comparator remains tripped (even at zero load current) and
the regulator will start to skip cycles in order to maintain
regulation. This behavior allows the regulator to maintain
constant frequency down to very light loads, resulting in low
output ripple as well as low audible noise and reduced RF
interference while providing high light load efficiency.
Current Sense
During the switch on-time, the control circuit limits the
maximum voltage drop across the current sense component to about 295mV, 110mV and 185mV at low duty
cycle with IPRG tied to VIN, GND or left floating respectively. It is reduced with increasing duty cycle as shown
in Figure 4.
300
250
IPRG = HIGH
200
IPRG = FLOAT
150
IPRG = LOW
100
50
0
1
20
40
60
DUTY CYCLE (%)
80
100
3873 F04
Figure 4. Maximum SENSE Threshold Voltage vs Duty Cycle
8
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LTC3873
APPLICATIONS INFORMATION
VCC Bias Power
The VCC pin must be bypassed to the GND pin with a
minimum 10μF ceramic or tantalum capacitor located
immediately adjacent to the two pins. Proper supply bypassing is necessary to supply the high transient currents
required by the MOSFET gate driver.
For maximum flexibility, the LTC3873 is designed so
that it can be operated from voltages well beyond the
LTC3873’s absolute maximum ratings. In the simplest case,
the LTC3873 can be powered with a resistor connected
between the input voltage and VCC. The built-in shunt
regulator limits the voltage on the VCC pin to around 9.3V
as long as the shunt regulator is not forced to sink more
than 25mA. This powering scheme has the drawback that
the power loss in the resistor reduces converter efficiency
and the 25mA shunt regulator maximum may limit the
maximum-minimum range of input voltage.
In some cases, the input or the output voltage is within
the operational range of VCC for the LTC3873. In this case,
the LTC3873 is operated directly from either the input or
output voltage. The typical application circuit on the first
page of this data sheet shows a 5V output converter in
which RSTART and CVCC form a start-up trickle charger while
D1 powers VCC from the output once the converter is in
normal operation. Note that RSTART need only supply the
very small 55µA micropower start-up current while CVCC
is charged to VTURNON. At this point, VRUN/SS > VSHDN,
the converter begins switching the external MOSFET and
ramps up the converter output voltage at a rate set by the
capacitor CRUN/SS on the RUN/SS pin. Since RSTART cannot
supply enough current to operate the external MOSFET, CVCC
begins discharging and VCC drops. The soft-start must be
fast enough so that the output voltage reaches its target
value of 5V before VCC drops to VTURNOFF or the converter
will fail to start. Otherwise more CVCC capacitor is needed
to hold the input voltage when soft-start is too long.
Figure 5 shows a different flyback converter bias power
strategy for a case in which neither the input or the output
is suitable for providing the bias power to the LTC3873.
The trickle charger is identical to that described in the
prior paragraph. However, the flyback transformer has an
additional bias winding to provide bias power. Note that this
topology is very powerful because, by appropriate choice
of the transformer turn ratio, the output voltage can be
chosen without regard to the value of the input voltage or
the VCC bias power for the LTC3873. The number of the
turns in the bias winding is chosen according to:
NBIAS = NSEC
VCC + VD2
VOUT + VD1
where NBIAS is the number of turns in the bias winding,
NSEC is the number of turns in the secondary winding,
VCC is the desired voltage to power the LTC3873, VOUT
is the converter output voltage, VD1 is the forward drop
voltage of D1 and VD2 is the forward drop voltage of D2.
Note that since VOUT is regulated by the converter control
loop, VCC is also regulated although not precisely. The
value of VCC is often constrained since NBIAS and NSEC are
often a limited range of small integer numbers. For proper
operation, the value of VCC must be between VTURNON and
VTURNOFF . Since the ratio of VTURNON to VTURNOFF is over
two to one, the requirement is relative easy to satisfy.
Finally, as with all trickle charger start-up schemes, the
soft-start must be fast enough so that the power supplied
by the bias winding is available before the discharge of
CVCC down to VTURNOFF .
T1
NBIAS
D2
•
VIN
CIN
RSTART
R3
CVCC
NPRI
VCC
RUN/SS NGATE
CVIN
D1
VOUT
•
•
NSEC
COUT
Q1
LTC3873
CC
ITH
SW
GND
RSENSE
VFB
R1
RSL
R2
3873 F05
Figure 5. Typical LTC3873 Application Circuit
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9
LTC3873
APPLICATIONS INFORMATION
The circuit in Figure 6 shows a third way to power the
LTC3873. An external series pre-regulator consisting of
series pass transistor Q1, zener diode D1 and bias resistor RB brings VCC to at least 7.6V nominal, well above
the maximum rated VCC turn-off threshold of 4V. Resistor
RSTART momentarily charges the VCC node up to the VCC
turn-on threshold, enabling the LTC3873.
VIN
RB
D1
8.2V
Q1
RSTART
CVCC
0.1µF
LTC3873
VCC
GND
Figure 6
Slope Compensation
The LTC3873 has built-in internal slope compensation to
stabilize the control loop against sub-harmonic oscillation.
It also provides the ability to externally increase slope
compensation by injecting a ramping current out of its SW
pin into an external slope compensation resistor (RSL in
Figure 5). This current ramp starts at zero right after the
NGATE pin has been set high. The current rises linearly
towards a peak of 20µA at the maximum duty cycle of
80%, shutting off once the NGATE pin goes low. A series
resistor (RSL) connecting the SW pin to the current sense
resistor (RSENSE) thus develops a ramping voltage drop.
From the perspective of the SW pin, this ramping voltage
adds to the voltage across the sense resistor, effectively
reducing the current comparator threshold in proportion
to duty cycle. The amount of reduction in the current
comparator threshold (ΔVSENSE) can be calculated using
the following equation:
Duty Cycle – 6%
20µA • RSLOPE
80%
Note the external programmable slope compensation is
only needed when the internal slope compensation is not
sufficient. In most applications RSL can be shorted. For the
LTC3873, when the RDS(ON) sensing technique is used, the
10
Output Voltage Programming
The output voltage is set by a resistor divider according
to the following formula:
⎛ R2⎞
VO = 1.2V • ⎜1+ ⎟
⎝ R1⎠
3873 F06
ΔVSENSE =
ringing on the SW pin disrupts the tiny slope compensation current out of the pin. It is not recommended to add
external slope compensation in this case.
The external resistor divider is connected to the output
as shown in Figure 5, allowing remote voltage sensing.
Choose resistance values for R1 and R2 to be as large as
possible in order to minimize any efficiency loss due to
the static current drawn from VOUT, but just small enough
so that when VOUT is in regulation, the error caused by
the nonzero input current to the VFB pin is less than 1%.
A good rule of thumb is to choose R1 to be 24k or less.
Transformer Design Considerations
Transformer specification and design is perhaps the
most critical part of applying the LTC3873 successfully.
In addition to the usual list of caveats dealing with high
frequency power transformer design, the following should
prove useful.
Turns Ratios
Due to the use of the external feedback resistor divider
ratio to set output voltage, the user has relative freedom
in selecting a transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2,
etc. can be employed which yield more freedom in setting
total turns and mutual inductance. Simple integer turns
ratios also facilitate the use of “off-the-shelf” configurable
transformers such as the Coiltronics VERSA-PAC series
in applications with high input-to-output voltage ratios.
For example, if a 6-winding VERSA-PAC is used with three
windings in series on the primary and three windings in
parallel on the secondary, a 3:1 turns ratio will be achieved.
Turns ratio can be chosen on the basis of desired duty
cycle. However, remember that the input supply voltage
3873fb
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LTC3873
APPLICATIONS INFORMATION
plus the secondary-to-primary referred voltage of the
flyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating.
Leakage Inductance
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to occur after the
output switch (Q1) turn-off. This is increasingly prominent
at higher load currents where more stored energy must
be dissipated. In some cases a “snubber” circuit will be
required to avoid overvoltage breakdown at the MOSFET’s
drain node. Application Note 19 is a good reference on
snubber design. A bifilar or similar winding technique is a
good way to minimize troublesome leakage inductances.
However, remember that this will limit the primary-tosecondary breakdown voltage, so bifilar winding is not
always practical.
Power MOSFET Selection
The power MOSFET serves two purposes in the LTC3873:
it represents the main switching element in the power path
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFET’s
thermal resistances (RTH(JC) and RTH(JA)).
Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), which determines output ripple
voltage and affects efficiency. Low ESR ceramic capacitors are often used to minimize the output ripple. Boost
regulators have large RMS ripple current in the output
capacitor that must be rated to handle the current. The
output ripple current (RMS) is:
IRMS(COUT) ≈ IOUT(MAX) •
VOUT – VIN(MIN)
VIN(MIN)
Output ripple is then simply:
VOUT = RESR(ΔIL(RMS))
The output capacitor for flyback converter should have a
ripple current rating greater than:
IRMS = IOUT •
DMAX
1– DMAX
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular, and
does not contain large square wave currents as found in
the output capacitor. The input voltage source impedance
determines the size of the capacitor that is typically 10μF
to 100μF. A low ESR is recommended although not as
critical as the output capacitor can be on the order of 0.3Ω.
For boost applications with RDS(ON) sensing, refer to
the LTC3872 data sheet for the selection of MOSFET
RDS(ON).
The RMS input ripple current for a boost converter is:
MOSFETs have conduction losses (I2R) and switching
losses. For VDS < 20V, high current efficiency generally
improves with large MOSFETs with low RDS(ON), while
for VDS > 20V the transition losses rapidly increase to the
point that the use of a higher RDS(ON) device with lower
reverse transfer capacitance, CRSS, actually provides
higher efficiency.
IRMS(CIN) = 0.3 •
VIN(MIN)
L•f
• DMAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to the
input of the converter and solid tantalum capacitors can
fail catastrophically under these conditions.
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11
LTC3873
APPLICATIONS INFORMATION
In a flyback converter, the input flows in pulses placing
severe demands on the input capacitors. Select an input
capacitor with a ripple current rating greater than:
IRMS =
PIN
VIN(MIN)
1– DMAX
DMAX
impacted by duty factor. Unfortunately duty factor cannot be adjusted to simultaneously optimize all of these
requirements. In general, avoid extreme duty factors since
this severely impacts the current stress on most of the
components. A reasonable target for duty factor is 50% at
nominal input voltage. Using this rule of thumb, the ideal
transformer turns ratio is:
Duty Cycle Considerations
The LTC3873 imposes a maximum duty cycle limit of
80% typical. For a flyback converter, the maximum duty
cycle prevents the transformer core from saturation. In
a boost converter application, however, it sets a limit on
the maximum step-up ratio or maximum output voltage
with the given input voltage of:
VOUT(MAX) =
VIN(MIN)
1– 0.8
– VD
Current and voltage stress on the power switch and
synchronous rectifiers, input and output capacitor RMS
currents and transformer utilization (size vs power) are
VIN
36V TO 72V
15k
0.1µF
ITH
NGATE
LTC3873
GND
VCC
RUN/SS
12.06k
RFB*
21.5k
VOUT
VOUT 1– D VOUT
•
=
VIN
D
VIN
Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
T1
4.7µF
100V
221k
2.2nF
NIDEAL =
•
•
D2
UPS840
100µF
6.3V
×3
VOUT*
3.3V
3A
Q1
FAN2512
•
D1
BAS516
51Ω
0.1µF
IPRG
VFB = 1.2V SW
3873 F07
4.7µF
10V
68mΩ
*FOR 5V OUTPUT CHANGE RFB TO 42.2k
Figure 7. 3.3V Output Nonisolated Telecom DC/DC Converter
12
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For more information www.linear.com/LTC3873
LTC3873
TYPICAL APPLICATIONS
9V to 15V VIN, 12V VOUT SEPIC Converter
VIN
9V TO 15V
10µF
×3
+
1
2
3
10nF
33.2k
11k
4
LTC3873
IPRG
ITH
SW
RUN/SS
VFB = 1.2V VCC
GND
100µF
20V 2
•
•
3
10µF
25V
301Ω
100k
T1
4.56µH
BH510-1009
BH ELECTRONICS
1
4
NGATE
UPS840
+
Si4840
8
47µF
16V
×3
10µF
16V
VOUT
12V
2A
7
6
5
4.7µF
0.1µF
3873 TA05
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For more information www.linear.com/LTC3873
13
LTC3873
TYPICAL APPLICATIONS
10W Isolated Telecom Converter
TR1
ISOLATION BARRIER
VIN
36V TO 72V
4.7µF
100V
221k
OPT
MMBTA42
OPT
PDZ6.8B
OPT
221k
BAS516
2
3
4
IRPG ISENSE
ITH RUN/SS
FB
GND
VCC
GATE
•
100µF
6.3V
×3
51Ω
5•
2
FDC2512
LTC3873
1
4•
1
VOUT
3.3V
3A
7
8
9
10
UPS840
2.2Ω
2
1
8
BAT54CWT1G
7
0.068Ω
6
3
5
4.7µF
1210
AND
0805
1µF
OPT
0.1µF
6.8k
BAT760
274Ω
4
NEC
PS2801-1
1
3
2
BAS516
LT4430
1
2
3
VIN
OPTO
GND COMP
OC 0.6V FB
1µF
6
5
4
22nF
100k
330pF
3.01k
22.1k
3873 TA04
2200pF
250V AC
14
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For more information www.linear.com/LTC3873
LTC3873
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/product/LTC3873#packaging for the most recent package drawings.
TS8 Package
8-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1637 Rev A)
0.40
MAX
2.90 BSC
(NOTE 4)
0.65
REF
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.22 – 0.36
8 PLCS (NOTE 3)
0.65 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.95 BSC
TS8 TSOT-23 0710 REV A
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
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For more information www.linear.com/LTC3873
15
LTC3873
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/product/LTC3873#packaging for the most recent package drawings.
DDB Package
8-Lead Plastic DFN (3mm × 2mm)
(Reference LTC DWG # 05-08-1702 Rev B)
0.61 ±0.05
(2 SIDES)
0.70 ±0.05
2.55 ±0.05
1.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
2.20 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
5
R = 0.05
TYP
0.40 ± 0.10
8
2.00 ±0.10
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.56 ± 0.05
(2 SIDES)
0.200 REF
0.75 ±0.05
0 – 0.05
4
0.25 ± 0.05
1
PIN 1
R = 0.20 OR
0.25 × 45°
CHAMFER
(DDB8) DFN 0905 REV B
0.50 BSC
2.15 ±0.05
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
16
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For more information www.linear.com/LTC3873
LTC3873
REVISION HISTORY
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
B
03/16
Changed LTC3873-5 VCC UV+ to 4.1V
PAGE NUMBER
1
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LTC3873
17
LTC3873
TYPICAL APPLICATIONS
Non-Insulated Boost Converter for Industrial Applications
VIN
4.5V-5.5V
+
C1
OPT
22µF
4.7pF
IPRG
VCC
GND
NGATE
0.1µF
LTC3873-5
RUN/SS
ITH
15k
47pF
VFB
2.2nF
L1
8.2µH
XAL1010-822ME
Q1
D1
BSC0906NS UPS840
SW
0.005Ω
2.37k
21.5k
C2
22µF
+
VOUT
12V AT 2A
C3
150µF
3873 TA06
C1, C2: MURATA GRM32ER71E226
C3: PANASONIC 16TQC150
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3748
100V Isolated Flyback Controller
5V ≤ VIN ≤ 100V, No-Opto Flyback , MSOP-16 with High Voltage Spacing
LT8309
Secondary-Side Synchronous Rectifier Driver
Works with DCM, BCM and Critical Conduction Mode Flyback
Topologies, Supports up to 150V MOSFETs, SOT-23 Package
LT3511/LT3512
100V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,
MSOP-16(12)
LT8301/LT8302
42VIN Micropower No-Opto Isolated Flyback Converters 2.7V to 42V Input Voltage Range, SOT-23 and S8 Packages
with 65V, 1.2A/3.6A Internal DMOS Power Switch
LTC3786
38V Low IQ Synchronous Boost Controller
4.5V (Down to 2.5V after Start-Up) ≤ VIN ≤ 38V, VOUT up to 60V, 50kHz to
900kHz Fixed Operating Frequency, 3mm × 3mm QFN-32, MSOP-16E
LTC3769
60V Low IQ Synchronous Boost Controller
4.5V (Down to 2.3V after Start-up) ≤ VIN ≤ 60V, VOUT up to 60V, 50kHz to
900kHz Fixed Operating Frequency, 4mm × 4mm QFN-24, TSSOP-20
LTC3803/LTC3803-3/ 200kHz/300kHz Flyback Controller
LTC3803-5
VIN and VOUT Limited by External Components, SOT-23
LTC3805/LTC3805-5
Adjustable Frequency Flyback Controller
VIN and VOUT Limited by External Components
LT3757A
Boost, Flyback, SEPIC and Inverting Controller
2.9V ≤ VIN ≤ 40V, Current Mode Control, 100kHz to 1MHz Programmable
Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Packages
LT3798
Off-Line Isolated No Opto-Coupler Flyback Controller
with Active PFC
VIN and VOUT Limited Only by External Components
LT3825
Isolated No-Opto Synchronous Flyback Controller
25V to 75V Input Voltage Range, Up to 80W
LT3837
Isolated No-Opto Synchronous Flyback Controller
4.5V to 20V Input Voltage Range, Up to 60W
LTC3872/LTC3872-1
No RSENSE Current Mode Boost DC/DC Controller
2.75V ≤ VIN ≤ 9.8V, VOUT Up to 60V, 550kHz Operation Frequency,
SOT-23 and 2mm × 3mm DFN Packages
18 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LTC3873
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LTC3873
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LT 0316 REV B • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 2007
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