LTC7852EUFD-1#TRPBF

LTC7852/LTC7852-1 Dual Output, 6-Phase, Multiphase Current Mode Synchronous Controller with Current Monitoring DESCRIPTION FEATURES Sub-Milliohm DCR Sensing or DrMOS with Current Sense Improves Efficiency n Operates with Power Blocks, DrMOS or External Gate Drivers and MOSFETs n ±0.5% Total Output Voltage Accuracy n Flexible Phase Configuration n Dual Output Current Monitoring n t ON(MIN) = 40ns, Capable of Very Low Duty Cycles at High Frequency n Dual Differential Remote Sensing Amplifiers n Programmable Frequency Range of 250kHz to 1.2MHz n V Range Is Not Limited by IC IN n V CC Range: 4.5V to 5.5V n V OUT Range: 0.5V to 2.0V n 48 Lead (5mm × 6mm) GQFN for LTC7852 n 36 Lead (4mm × 5mm) QFN for LTC7852-1 n APPLICATIONS Computer Systems Telecom and Datacom Systems n DC Power Distribution Systems n n All registered trademarks and trademarks are the property of their respective owners. Protected by U.S. patents, including 9525351, 9793800, 10848051. The LTC®7852/LTC7852-1 is a six-phase, dual output current mode synchronous step-down switching regulator controller that works in conjunction with external power train devices such as DrMOS, power blocks or discrete N-channel MOSFETs and associated gate drivers. Its flexible design enables 1-, 2-, 3-, 4-, 5-, and 6-phase configurations. The LTC7852 offers a unique feature that enhances the signal-to-noise ratio of the current sense signal, allowing the use of inductors with very low DC winding resistances for maximum efficiency. The controller achieves a minimum on-time of just 40ns, permitting the use of high switching frequency at high step-down ratios. 8-, 10- or 12 phases with two ICs can be paralleled for very high current requirements up to 400A. The remote sense differential amplifiers and a precise reference provide accurate output voltages between 0.5V and 2.0V. The input voltage is not limited by the controller. Hiccup mode protection from output shorts or overcurrent minimizes the thermal dissipation. The LTC7852-1 is designed specifically for DrMOS with an internal current sense signal. TYPICAL APPLICATION VIN 5V TO 13V + VCC 4.5V TO 5.5V 2.2µF 4.7µF VDD VCC VOUT1 0.9V 120A L1~4 0.25µH + DrMOS PGOOD1 PGOOD2 PWM1-4 PWM5-6 330µF ×12 + LTC7852 715Ω SNSP1-4 220nF SNSP5-6 220nF 220nF SNSAVG1-4 220nF 715Ω 2.32k SNSN1-4 SNSN5-6 VOSNS1+ VOSNS2+ VOSNS1– VOSNS2– 3.01k 3.3nF 28k 20k 3.01k ITH1 SS1 150pF 0.22µF 330µF X6 VOUT2 1.2V 60A SNSAVG5-6 2.32k 16k 20k PINS NOT SHOWN IN THIS CIRCUIT: IMON1 IMON2 V1P5 PLLIN CLKOUT RUN1 RUN2 L4~5 0.25µH DrMOS ITH2 SS2 ILIM1 FREQ GND PHCFG ILIM2 150pF 3.3nF 0.22µF 78521 TA01a 37.4k Rev. B Document Feedback For more information www.analog.com 1 LTC7852/LTC7852-1 TABLE OF CONTENTS Features............................................................................................................................. 1 Applications........................................................................................................................ 1 Typical Application ................................................................................................................ 1 Description......................................................................................................................... 1 Absolute Maximum Ratings...................................................................................................... 3 Order Information.................................................................................................................. 3 Electrical Characteristics......................................................................................................... 4 Typical Performance Characteristics........................................................................................... 6 Pin Functions....................................................................................................................... 8 Functional Block Diagrams....................................................................................................... 9 Operation..........................................................................................................................11 Applications Information........................................................................................................15 Typical Applications..............................................................................................................27 Package Description.............................................................................................................29 Revision History..................................................................................................................31 Typical Application...............................................................................................................32 Related Parts......................................................................................................................32 Rev. B 2 For more information www.analog.com LTC7852/LTC7852-1 ABSOLUTE MAXIMUM RATINGS (Note 1) RUN1,2, PGOOD1,2, VCC Voltage.................. –0.3V to 6V SNSN, SNSAVG (LTC7852 Only), SNSP.............................................–0.3V to (VCC + 0.3V) All Other Pin Voltages....................–0.3V to (VCC + 0.3V) Operating Junction Temperature Range.... –40° to 125°C Storage Temperature Range................... –65°C to 150°C PIN CONFIGURATION LTC7852 LTC7852-1 SNSP4 2 49 49 SNSAVG5 4 PWM1 PWM2 PWM3 36 35 34 33 32 31 30 29 37 SNSP3 SNSN4 3 PWM4 38 SNSAVG3 PWM5 GND SNSAVG4 1 PWM6 TOP VIEW 48 47 46 45 44 43 42 41 40 39 CLKOUT VDD CLKOUT PWM1 PWM2 PWM3 PWM4 PWM5 PWM6 PLLIN GND TOP VIEW 36 SNSN3 PLLIN 1 28 VDD 35 SNSAVG2 SNSN 2 27 SNSP3 26 SNSP2 SNSP5 5 34 SNSP2 SNSP4 3 SNSN5 6 33 SNSN2 SNSP5 4 32 SNSAVG1 SNSP6 5 31 SNSP1 PGOOD2 6 30 SNSN1 SS2 7 22 SS1 29 PGOOD1 ITH2 8 21 ITH1 28 SS1 VOSNS2– 9 20 VOSNS1– 27 ITH1 VOSNS2+ 10 19 VOSNS1+ VOSNS2– 13 26 VOSNS1– VOSNS2+ 14 25 VOSNS1+ 11 12 13 14 15 16 17 18 UFD PACKAGE 36-LEAD (4mm × 5mm) PLASTIC QFN RUN1 ILIM1 FREQ PHCFG VCC IMON1 V1P5 IMON2 RUN2 ILIM2 15 16 17 18 19 20 21 22 23 24 GND 49 23 PGOOD1 RUN1 49 ITH2 12 24 SNSP1 ILIM1 SS2 11 FREQ PGOOD2 10 PHCFG SNSN6 9 VCC SNSP6 8 25 SNSN 37 GND RUN2 49 GND ILIM2 SNSAVG6 7 TJMAX = 125°C, θJA = 43°C/W RHE PACKAGE 48-LEAD (5mm × 6mm) PLASTIC GQFN TJMAX = 125°C, θJA = 30°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC7852ERHE#PBF LTC7852ERHE#TRPBF 7852 48-LEAD (5mm x 6mm) Plastic GQFN –40°C to 125°C LTC7852IRHE#PBF LTC7852IRHE#TRPBF 7852 48-LEAD (5mm x 6mm) Plastic GQFN –40°C to 125°C LTC7852EUFD-1#PBF LTC7852EUFD-1#TRPBF 78521 36-LEAD (4mm x 5mm) Plastic QFN –40°C to 125°C LTC7852IUFD-1#PBF LTC7852IUFD-1#TRPBF 78521 36-LEAD (4mm x 5mm) Plastic QFN –40°C to 125°C Contact the factory for parts specified with wider operating temperature ranges. Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. Rev. B For more information www.analog.com 3 LTC7852/LTC7852-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating temperature range, otherwise specifications are at TA = 25°C (Note 2). VCC = VRUN = 5V unless otherwise specified. SYMBOL PARAMETER VCC Minimum VOUT VOSNS+ Bias Supply Input Output Voltage Range Regulated Feedback Voltage IOSNS+ VREFLNREG VLOADREG UVLO UVLOHYS ISNSAVG ISNSP Feedback Current Reference Voltage Line Reg. Output Voltage Load Regulation Transconductance Amplifier gm Differential Amplifier UnityGain Crossover Frequency Feedback Overvoltage Lockout Input DC Supply Current Normal Mode Shutdown Undervoltage Lockout UVLO Hysteresis Sense Pin Bias Currents Sense Pin Bias Currents ISS VSNSN Soft-Start Charge Current Sense Pin Bias Voltage AVT_SNS Total Sense Signal Gain to Current Comparator RUN Pin ON Threshold RUN Pin ON Hysteresis RUN Pin Pull-Up Current RUN < ON Threshold RUN > ON Threshold Maximum Current Sense Threshold gm1,2 f0dB VOVL IQ VRUN VRUN_HYS IRUN CONDITIONS LTC7852 Only (Note 2) ITH = 1.2V (Note 4) –40°C to 125°C 0°C to 85°C VCC = 4.5V to 5.5V ΔITH = 0.7V to 1.2V ΔITH = 1.2V to 1.6V ITH =1.7V, Sink/Source 10µA Measured at VOSNS+ l l l 496 498 VRUN Rising VØMIS Standard Deviation of Phase to Phase Current Sensed Voltage Mismatching tON(MIN) Minimum On-Time (Note 6) 500 500 0.01 0.01 2.75 l l MAX 504 502 –100 0.1 0.1 0.1 4 l VRUN = 0V VVCC Falling VSNSAVG = 1.0V LTC7852 Only LTC7852 SNSP = 1.0V LTC7852-1 SNSP = 1.5V VSS = 0V –3mA < ISNSN < 3mA LTC7852-1 Only LTC7852 Only TYP 4.5 2.0 (Note 5) RUN < 1.1V RUN > 1.34V LTC7852 ITH = 2.2V, VSNSN =1.0V VSNSAVG = VSNSN ILIM = 0V ILIM = 1/4VCC ILIM = Float ILIM = 3/4VCC ILIM = VCC LTC7852-1 ITH = 2.2V ILIM = 0V ILIM = 1/4Vcc ILIM = Float ILIM = 3/4VCC ILIM = VCC ILIM = Float, PHCFG = Float ITH = 2.2V VSENSE(MAX) MIN 5 3.6 7.5 15 1.2 4.0 200 l –4.5 –5 1.5 10 4.3 –5.5 5 l 1.1 1.22 140 V V mV mV nA % % % mmho MHz ±30 ±50 l UNITS % mA mA V mV nA nA µA V V/V 1.34 –1.3 –7.7 V mV µA µA l l l l l 9 14 19 24 28.5 10 15 20 25 30 11 16 21 26 31.5 mV mV mV mV mV l l l l l 45 70 95 120 142.5 50 75 100 125 150 0.5 55 80 105 130 157.5 mV mV mV mV mV mV 40 ns Rev. B 4 For more information www.analog.com LTC7852/LTC7852-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating temperature range, otherwise specifications are at TA = 25°C (Note 2). VCC = VRUN = 5V unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Power Good VPGOOD(ON) PGOOD Pull Down Resistance IPGOOD(OFF) PGOOD Leakage Current tPGOOD VPGOOD High to Low Delay VPGD PGOOD Trip Level VPG1(HYST) PGOOD Trip Level Hysteresis VPGOOD = 5V 200 Ω 2 µA 45 VFB with Respect to Set Output Voltage VFB Ramping Up VFB Ramping Down 5 –5 7.5 –7.5 µs 10 –10 15 % % mV Oscillator and Phase-Locked Loop fOSC Oscillator Frequency RFREQ = 30.1kΩ RFREQ = 47.5kΩ RFREQ = 54.9kΩ RFREQ = 75.0kΩ 215 550 675 0.875 Sync. Freq. Range IFREQ FREQ Pin Output Current l VFREQ = 0.8V 250 600 750 1.05 0.25 –18.5 20 285 650 825 1.225 kHz kHz kHz MHz 1.2 MHz –21.5 µA RPLLIN PLLIN Input Resistance 200 kΩ VPLLIN PLLIN Input Threshold VPLLIN Rising VPLLIN Falling 2 1.2 V V VCLKOUT Low Output Voltage High Output Voltage ILOAD = 500µA ILOAD = –500µA 0.2 5 V V 3.3 V VDD Output VDD Internal VDD Voltage PWM Outputs PWM ILOAD = –500µA l PWM Output Low Voltage ILOAD = 500µA l PWM Output Current in Hi-Z State PWM = 0V PWM = 3.3V PWM Output High Voltage 3.1 3.3 3.5 V 0.5 V –1 1 µA µA V IMON Outputs (LTC7852 Only) V1P5 IMON 1.5V Regulator Output Voltage VSNS = 0, –3mA < IV1P5 < 3mA IMON Output Voltage VSNS = VSNSMAX, ILIM = Float Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC7852/LTC7852-1 is tested under pulsed load conditions such that TJ ≈ TA. The LTC7852E/LTC7852-1E is guaranteed to meet performance specifications from 0°C to 85°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC7852I/LTC7852-1I is guaranteed to meet performance specifications over the full –40°C to 125°C operating junction temperature range. The maximum ambient temperature consistent with these specifications is determined by specific l 1.4 1.5 1.6 l V1P5 +142.5 V1P5 +150 V1P5 +157.5 mV operating conditions in conjunction with board layout, the package thermal impedance and other environmental factors. TJ is calculated from the ambient temperature, TA, and power dissipation, PD, according to the following formula: LTC7852RHE: TJ = TA + (PD • 30°C/W) LTC7852UFD-1: TJ = TA + (PD • 43°C/W) Note 3: Output voltage range of LTC7852-1 is determined by the DrMOS. Note 4: The LTC7852/LTC7852-1 is tested in a feedback loop that servos VITH to a specified voltage and measures the resultant VFB. Note 5: Guaranteed by design. Note 6: The minimum on-time condition is specified for an inductor peak-to-peak ripple current >40% of IMAX (See Minimum On-Time Considerations in the Applications Information section). Rev. B For more information www.analog.com 5 LTC7852/LTC7852-1 TYPICAL PERFORMANCE CHARACTERISTICS 160 30.0 CURRENT SENSE THRESHOLD (mV) ILIM = VCC ILIM = 3/4 VCC 25.0 ILIM = 1/2 VCC 20.0 ILIM = 1/4 VCC 15.0 ILIM = 0 10.0 5.0 0 0.5 1 1.5 VSENSE COMMON MODE VOLTAGE (V) 120 100 20.4 20.3 80 60 40 20 0 –20 –40 2 20.5 ILIM = 0 ILIM = 1/4 VCC ILIM = 1/2 VCC ILIM = 3/4 VCC ILIM = VCC 140 FREQ PIN CURRENT (µA) 35.0 CURRENT SENSE THRESHOLD (mV) FREQ Pin Source Current vs Temperature Current Sense Threshold vs ITH Voltage Maximum Current Sense Threshold vs Common Mode Voltage 0 TA = 25°C, VCC = 5V unless otherwise noted. 7852 G01 20.1 20.0 19.9 19.8 19.7 19.6 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2.0 2.2 VITH (V) 19.5 –50 –25 1800 Prebias Startup at 0.5V VFREQ = VCC RUN PIN 2V/DIV ILOAD 30A TO 45A 20µs/DIV 7852 G04 OSCILLATOR FREQUENCY (kHz) 1600 IL1, IL2 5A/DIV 25 50 75 100 125 150 TEMPERATURE (°C) 7852 G03 Oscillator Frequency vs Temperature VOUT AC–COUPLED 50mV/DIV 0 7852 G02 Load Step VIN = 12V VOUT = 1V fSW = 400kHz L = 0.25µH 20.2 1400 OUTPUT 200mV/DIV 1200 1000 800 VFREQ = 0.95V SS VOLTAGE 200mV/DIV PGOOD 5V/DIV 600 400 VFREQ = GND 200 0 –50 –25 0 500µs/DIV 7852 G06 25 50 75 100 125 150 TEMPERATURE (°C) 7852 G05 Rev. B 6 For more information www.analog.com LTC7852/LTC7852-1 TYPICAL PERFORMANCE CHARACTERISTICS Regulated Feedback Voltage vs Temperature Quiescent Current vs Temperature 15.40 15.35 15.30 15.25 15.20 15.15 15.10 15.05 15.00 –50 –25 0 1.4 503 1.3 502 RUN THRESHOLD (V) REGULATED FEEDBACK VOLTAGE (mV) 15.45 501 500 499 498 OFF 1.1 1.0 496 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) ON 1.2 497 0 0.9 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) Undervoltage Lockout Threshold (VCC) vs Temperature 5.20 4.40 1.28 5.15 4.30 5.10 4.20 1.26 1.24 5.05 ISS (µA) 1.18 5.00 4.95 1.16 4.90 1.14 0 25 50 75 100 125 150 TEMPERATURE (°C) 4.80 –50 –25 RISING 4.10 4.00 FALLING 3.90 3.80 3.70 4.85 1.12 1.10 –50 –25 UVLO THRESHOLD (V) 1.30 1.20 25 50 75 100 125 150 TEMPERATURE (°C) 7852 G09 SS Pull-Up Current vs Temperature Shutdown Current vs Temperature 1.22 0 7852 G08 7852 G07 SHUTDOWN CURRENT (mA) RUN Threshold vs Temperature 504 15.50 QUIESCENT CURRENT (mA) TA = 25°C, VCC = 5V unless otherwise noted. 3.60 0 25 50 75 100 125 150 TEMPERATURE (°C) 7852 G11 7852 G10 3.50 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 7852 G12 Rev. B For more information www.analog.com 7 LTC7852/LTC7852-1 PIN FUNCTIONS (GQFN/QFN, LTC7852/LTC7852-1) V1P5 (Pin 18, LTC7852 Only): Internally Generated 1.5V Voltage Regulator Output Pin. Bypass this pin to SGND with a low ESR 2.2µF capacitor. IMON1, IMON2 (Pins 19, 17, LTC7852 Only): Output Current Monitors. The differential voltage between each IMON pin and the V1P5 pin provides a linear indication of output current from the corresponding channel. VCC (Pin 20/Pin 13): External 5V Input. The control circuits are powered from this voltage. Bypass this pin to GND with a capacitor (0.1μF to 1μF ceramic) in close proximity to the chip. PHCFG (Pin 21/Pin 14): Phase Configuration Pin. This pin selects the phases powering output 1 and output 2. FREQ (Pin 22/Pin 15): Frequency Set/Select Pin. A resistor between this pin and SGND sets the switching frequency. This pin sources 20µA. ILIM1, ILIM2 (Pins 23, 16/Pins 16, 12): Current Comparator Sense Voltage Limit Selection Pin. RUN1, RUN2 (Pins 24, 15/Pins 17, 11): Enable Control Inputs. A voltage above 1.22V turns on the IC. There is a 1µA pull-up current on this pin. Once the RUN pin rises above the 1.22V threshold, the pull-up increases to 7.7µA. VOSNS1+, VOSNS2+ (Pins 25, 14/Pins 19, 10): Remote Sense Differential Amplifier Non-Inverting inputs. Connect to feedback divider center tap with the divider across the output load. The remote sense differential amplifier’s output is internally connected to the error amplifier’s inverting input. VOSNS1–, VOSNS2– (Pins 26, 13/Pins 20, 9): Remote Sense Differential Amplifier Inverting Inputs. Connect to sense ground at the output load. ITH1, ITH2 (Pins 27, 12/Pins 21, 8): Current Control Thresholds and Error Amplifier Compensation Points. The current comparator’s threshold increases with the ITH control voltage. SS1, SS2 (Pins 28, 11/Pins 22, 7): Soft-Start Inputs. The voltage ramp rate at this pin sets the voltage ramp rate of the output. A capacitor to ground programs soft-start. This pin has a 5µA pull-up current. The minimum required soft-start capacitor is 22nF. PGOOD1, PGOOD2 (Pins 29, 10/Pins 23, 6): Power Good Indicator Outputs. Open drain output that pulls to ground when output voltage is not in regulation. 8 SNSN1, SNSN2, SNSN3, SNSN4, SNSN5, SNSN6 (Pins 30, 33, 36, 3, 6, 9, LTC7852 Only): Second Negative Current Sense Comparator Inputs. This input senses the signal from the output inductor’s DCR with a filter bandwidth of five times the inductor’s L/DCR value when low DCR current sensing is enabled. SNSP1, SNSP2, SNSP3, SNSP4, SNSP5, SNSP6 (Pins 31, 34, 37, 2, 5, 8 /Pins 24, 26, 27, 3, 4, 5): Positive Current Sense Comparator Inputs. SNSAVG1, SNSAVG2, SNSAVG3, SNSAVG4, SNSAVG5, SNSAVG6 (Pins 32, 35, 38, 1, 4, 7, LTC7852 Only): First Negative Current Sense Comparator Inputs. This input senses the signal from the output inductor’s DCR with a filter which has a bandwidth at 3/5 of the inductor’s L/DCR value. Tie to VCC for DCR sensing with DCR >1mΩ or DrMOS current sensing. SNSN (Pins 2, 25, LTC7852-1 Only): Internal 1.5V Voltage Regulator Output. VDD (Pin 39/Pin 28): Internally Generated 3.3V Power Supply Output Pin. Bypass this pin to SGND with a low ESR 2.2µF capacitor. Do not load this pin with external current. PWM output HIGH voltage equals to VDD voltage. If a PWM HIGH voltage between 3.3V and VCC voltage is desirable, it is allowed to bias this pin with external source at the desirable voltage. CLKOUT (Pin 40/Pin 29): Clock Output Pin. PWM1, PWM2, PWM3, PMW4, PWM5, PWM6 (Pins 41, 42, 43, 44, 45, 46/Pins 30, 31, 32, 33, 34, 35): (Top) Gate Signal Outputs. This signal goes to the PWM or top gate input of the external gate driver or integrated driver MOSFET or Power Block. This is a three-state compatible output. PWM output HIGH voltage equals to Vdd pin voltage. PLLIN (Pin 47/Pin 1): External Synchronization Input to Phase Detector Pin. A clock on the pin will synchronize the internal oscillator with the clock on this pin. The PLL compensation network is integrated into the IC. GND (Pin 48/Pin 36): Ground. All small-signal components and compensation components should be connected here. The exposed pad must be soldered to the PCB for rated thermal performance. Exposed pad (Pin 49/Pin 37): Ground. For more information www.analog.com Rev. B LTC7852/LTC7852-1 FUNCTIONAL BLOCK DIAGRAMS LTC7852 FREQ PLLIN CLKOUT PHCFG IMON VDD V1P5 1.5V LDO AMP – + SNSAVG SNSP PLL/SYNC UVLO R VIN + CIN PWM ON Q SWITCH LOGIC RUN 5k + VCC VDD S OSC VCC + OV ICMP VOUT COUT – + SNSP – SNSAVG AMP SLOPE COMPENSATION ILIM SNSN PGOOD ACTIVE CLAMP 1 R ITHD + ITH 0.4625V UV 5µA – RC DIFFAMP HICCUP – EA CC + VCC + – 0.45V SS + – RUN + + 20k + 20k + – VOSNS+ – OV 1.22V 1µA/5µA – 0.5V REF 0.5375V 20k 20k VOSNS– GND SS RUN 78521 BD1 CSS Rev. B For more information www.analog.com 9 LTC7852/LTC7852-1 FUNCTIONAL BLOCK DIAGRAMS LTC7852-1 FREQ PLLIN CLKOUT VDD PHCFG LDO UVLO VCC PLL/SYNC R PWM ON Q VIN CIN SWITCH LOGIC RUN 5k + + VDD S OSC VCC + OV ICMP VOUT COUT – SNSP SLOPE COMPENSATION VCC ILIM 1.5V SNSN PGOOD ACTIVE CLAMP 1 R ITHD + ITH 0.4625V UV 5µA – RC DIFFAMP HICCUP – EA CC + VCC + – 0.45V SS + – RUN + + 20k + 20k + – VOSNS+ – OV 1.22V 1µA/5µA – 0.5V REF 0.5375V 20k 20k VOSNS– GND SS RUN 78521 BD2 CSS Rev. B 10 For more information www.analog.com LTC7852/LTC7852-1 OPERATION Main Control Loop The LTC7852/LTC7852-1 uses an LTC proprietary current sensing, current mode step-down architecture. During normal operation, the top MOSFET is turned on every cycle when the oscillator sets the RS latch, and turned off when the main current comparator, ICMP, resets the RS latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier, EA. The remote sense amplifier (diffamp) produces a signal equal to the differential voltage sensed across the output capacitor divided down by the feedback divider and re-references it to the local IC ground. The error amplifier receives this feedback signal and compares it to the internal 0.5V reference. When the load current increases, it causes a slight decrease in the VOSNS+ pin voltage relative to the 0.5V reference, which in turn causes the ITH voltage to increase until the inductor’s average current equals the new load current. After the top MOSFET has turned off, the bottom MOSFET is turned on until the beginning of the next cycle. The inductor current is allowed to reverse at light loads or under large transient conditions. The main control loop is shut down by pulling the RUN pin low. Releasing RUN allows an internal 1.3µA current source to pull up the RUN pin. When the RUN pin reaches 1.22V, the main control loop is enabled and the IC is powered up. When the RUN pin is low, all functions are kept in a controlled state. Sensing Signal of Very Low DCR (LTC7852) The LTC7852 employs a unique architecture to enhance the signal-to-noise ratio, enabling it to operate with a small sense signal of a very low value inductor DCR, 1mΩ or less. This improves power efficiency, and reduces jitter due to the switching noise which could corrupt the signal. The LTC7852 can sense a DCR value as low as 0.2mΩ with careful PCB layout. Each phase has two negative sense pins, SNSN and SNSAVG, which share the positive sense pin SNSP. These sense pins acquire signals and internally processes them for a 14dB signal-to-noise ratio improvement. In the meantime, the current limit threshold is still a function of the inductor peak current and its DCR value, and can be accurately set from 10mV to 30mV in 5mV steps with the ILIM pin. The filter across the inductor should have a time constant R1 • C1 equal to 1/5 of the time constant of the output inductor L/DCR. The filter at SNSAVG should have a bandwidth of three times larger than SNSP R1 • C1. Driver MOSFET (DrMOS) Current Sensing (LTC7852-1) The LTC7852-1 is dedicated for converters using DrMOS current sensing. The SNSN pins are connected to an internal 1.5V voltage regulator with current sinking and sourcing capability. It serves as a common mode bias for all the DrMOSs’ current sensing differential signals. Shutdown and Start-Up (RUN and SS Pins) The LTC7852/LTC7852-1 can be shut down using the RUN pin. Pulling the RUN pin below 1.14V shuts down the main control loop for the controller and most internal circuits. Releasing the RUN pin allows an internal 1.3µA current to pull up the pin and enable the controller. Alternatively, the RUN pin may be externally pulled up or driven directly by logic. Be careful not to exceed the absolute maximum rating of 6V on this pin. The start-up of the controller’s output voltage, VOUT, is controlled by the voltage on the SS pin. When the voltage on the SS pin is less than the 0.5V internal reference, the LTC7852/LTC7852-1 regulates the VOSNS+ voltage to the SS pin voltage instead of the 0.5V reference. This allows the SS pin to be used to program a soft-start by connecting an external capacitor from the SS pin to GND. The minimum required SS capacitor is 22nF. An internal 5µA pull-up current charges this capacitor, creating a voltage ramp on the SS pin. As the SS voltage rises linearly from 0V to 0.5V (and beyond), the output voltage, VOUT, rises smoothly from its pre-biased value to its final set value. Certain applications can result in the start-up of the converter into a non-zero load voltage, where residual charge is stored on the output capacitor at the onset of converter switching. In order to prevent the output from discharging under these conditions, the bottom MOSFET is disabled until soft-start is greater than VOSNS+. Rev. B For more information www.analog.com 11 LTC7852/LTC7852-1 OPERATION When the RUN pin is pulled low to disable the controller, or when VCC drops below its undervoltage lockout threshold of 4.0V, the SS pin is pulled low by an internal MOSFET. When in undervoltage lockout, the controller is disabled and the external MOSFETs are held off. signals are evenly interleaved. Table 1 shows the detailed information. A 6-phase single output converter can be configured by floating the PHCFG pin, while externally connecting the ITHs, VOSNS+s, VOSNS–s, RUNs, ILIMs and SS pins, respectively. Frequency Selection and Phase-Locked Loop (FREQ and PLLIN Pins) Multichip Operation The selection of switching frequency is a trade-off between efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching losses, but requires larger inductance and/or capacitance to maintain low output ripple voltage. If the PLLIN pin is not being driven by an external clock source, the FREQ pin can be used to program the controller’s operating frequency from 250kHz to 1.2MHz. There is a precision 20µA current flowing out of the FREQ pin so that the user can program the controller’s switching frequency with a single resistor to GND. A curve is provided in the Applications Information section showing the relationship between the voltage on the FREQ pin and switching frequency. A phase-locked loop (PLL) is available on the LTC7852/LTC7852-1 to synchronize the internal oscillator to an external clock source that is connected to the PLLIN pin. The PLL loop filter network is integrated inside the LTC7852/LTC7852-1. The phase-locked loop is capable of locking to any frequency within the range of 250kHz to 1.2MHz. The frequency setting resistor should always be present to set the controller’s initial switching frequency before locking to the external clock. Multiphase Operation For output loads that demand high current, multiple LTC7852/LTC7852-1s can be daisy chained to run out of phase to provide more output current without increasing input and output voltage ripple. The ITH, VOSNS+, VOSNS–, ILIM and SS pins of one phase should be tied to the corresponding pins of the other phases. The PLLIN pin allows the LTC7852/LTC7852-1 to synchronize to the CLKOUT signal of another LTC7852/LTC7852-1, or other external clock source. For the LTC7852/LTC7852-1 synchronized to PLLIN clock signal, the rising edge of PWM1 is lined up with the rising edge of the PLLIN clock. The CLKOUT signal can be connected to the PLLIN pin of the following LTC7852/LTC7852-1 stage to line up both the frequency and the phase of the entire system. In 3+3 mode, the phase difference between PH1 and CLKOUT is 90°. In this mode, a total of 12 phases can be daisy chained to run simultaneously out-of-phase with respect to each other. In 4+2 mode, the phase difference between PH1 and CLKOUT is 225°. With two ICs in this mode, an 8 phase interleaving power stage could be configured. In 5+1 mode, difference between PH1 and CLKOUT is 252°. With two ICs in this mode a 10 phase interleaving power stage could be configured. Table 1. LTC7852/LTC7852-1 provides flexible phase configurations for dual high current outputs. When PHCFG is either grounded, floated or tied to INTVCC, the controller is in 4+2 mode, 3+3 mode and 5+1 mode, respectively. In order to minimize the input and output voltage ripple and increase the power conversion efficiency, the multi-phase PWM OUTPUT 1 OUTPUT 2 3+3 Mode 0° 120° 240° 60° 180° 300° 90° 4+2 Mode 0° 90° 180° 270° 45° 225° 225° 5+1 Mode 0° 72° 144° 216° 288° 252° 252° 6 Phase Mode 0° 120° 240° 60° 180° 300° 90° PH1 PH2 PH3 PH4 PH5 PH6 CLKOUT Figure 1 shows the connections necessary for 8-, 10- or 12-phase operation. Rev. B 12 For more information www.analog.com LTC7852/LTC7852-1 OPERATION PHCFG LTC7852 225, 315, 45, 135 PHCFG LTC7852 CLKOUT PLLIN CLKOUT PLLIN 78521 F01a (a) 8-Phase Configuration VCC 0, 72, 144, 216, 288 PHCFG LTC7852 VCC 252, 324, 36, 108, 180 PHCFG LTC7852 CLKOUT PLLIN CLKOUT VOUT PLLIN LTC7852 90, 150, 210, 270, 330, 30 PHCFG RD1 RD2 CF1 VOSNS+ VOSNS– LTC7852/ LTC7852-1 + DIFFAMP – 78521 F02 Figure 2. Differential Amplifier Connection LTC7852 CLKOUT PLLIN COUT1 10Ω (b) 10-Phase Configuration PHCFG 10Ω COUT2 78521 F01b 0, 60, 120, 180, 240, 300 VOSNS– to the load ground. See Figure 2. The LTC7852/ LTC7852-1 differential amplifier is configured for unity gain, meaning that the difference between VOSNS+ and VOSNS– is translated to its output, relative to GND. The differential amplifier’s output is internally connected to the error amplifier inverting input. Care should be taken to route the VOSNS+ and VOSNS– PCB traces parallel to each other all the way to the remote sensing points on the board. In addition, avoid routing these sensitive traces near any high speed switching nodes in the circuit. Ideally, the VOSNS+ and VOSNS– traces should be shielded by a low impedance ground plane to maintain signal integrity. FEEDBACK DIVIDER 0, 90, 180, 270 Power Good (PGOOD Pin) CLKOUT PLLIN 78521 F01c (c) 12-Phase Configuration Figure 1. Phase Operations Sensing the Output Voltage with a Differential Amplifier The LTC7852/LTC7852-1 includes two low offset, high input impedance, unity-gain, high bandwidth differential amplifiers for applications that require true remote sensing. Sensing the load across the load capacitors directly benefits regulation in high current, low voltage applications, where board interconnection losses can be a significant portion of the total error budget. Connect VOSNS+ to the center tap of the feedback divider across the output load, and The PGOOD pin is connected to the open drain of an internal N-channel MOSFET. The MOSFET turns on and pulls the PGOOD pin low when the VOSNS+ pin voltage is not within ±10% of the 0.5V reference voltage. The PGOOD pin is also pulled low when the RUN pin is below 1.14V or when the LTC7852/LTC7852-1 is in the soft-start phase. When the VOSNS+ pin voltage is within the ±5% regulation window, the MOSFET is turned off and the pin is allowed to be pulled up by an external resistor to a source of up to 6V. The PGOOD pin will flag power good immediately when the VOSNS+ pin is within the regulation window. However, there is an internal 45µs power-bad mask delay when the VOSNS+ goes out of the window. There is an independent PGOOD pin for each channel. For single output configuration, the two PGOOD pins can be tied together. Rev. B For more information www.analog.com 13 LTC7852/LTC7852-1 OPERATION Output Overvoltage Protection Load Current Monitoring An overvoltage comparator, OV, guards against transient overshoots (>10%) as well as other more serious conditions that may overvoltage the output. In such cases, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The LTC7852’s IMON pins outputs a voltage proportional to the load current of the corresponding channel. IMON is referred to the regulated 1.5V common mode voltage at the V1P5 pin. A decoupling capacitor might be placed between IMON and V1P5 for noise decoupling. Please note that the IMON pin is not a low impedance signal source. Minimize the leakage current of circuitry connecting to this pin for best accuracy. The linear transfer function from load current to the IMON signal is: Undervoltage Lockout The LTC7852/LTC7852-1 has two functions that help protect the controller in case of undervoltage conditions. A precision UVLO comparator constantly monitors the VCC voltage to ensure that an adequate PWM voltage is present. It locks out the switching action when VCC is below 4.0V during ramping up. To prevent oscillation when there is a disturbance on the VCC, the UVLO comparator has 200mV of precision hysteresis. The RUN pin can be configured to detect an undervoltage condition of the power stage input voltage as needed. Because the RUN pin has a precision turn-on reference of 1.22V, one can use a resistor divider across the input voltage to turn on the IC when input voltage is high enough. An extra 4µA of current flows out of the RUN pin once the RUN pin voltage passes 1.22V. The RUN comparator itself has about 80mV of hysteresis. One can program additional hysteresis for the RUN comparator by adjusting the values of the resistive divider. Always set the power stage input voltage undervoltage detection threshold higher than the controller UVLO threshold so that the LTC7852/LTC7852-1 is enabled after the power stage. VIMON = 150mV/VILIM • ∑(VSNSAVG)/NPH where: NPH: number of paralleled phases VILIM: maximum sense voltage of the selected ILIM level VSNSAVG: differential sense voltage between SNSP and SNSAVG of each phase For a 6-phase single output converter, tie the IMON1 and IMON2 together; this signal represents the total current of six phases. Rev. B 14 For more information www.analog.com LTC7852/LTC7852-1 APPLICATIONS INFORMATION The Typical Application on the first page of this data sheet is a basic LTC7852 application circuit. The LTC7852 is designed and optimized for use with a very low DCR value by utilizing a novel approach to reduce the noise sensitivity of the sensing signal by a factor of 14dB. DCR sensing is popular because it saves expensive current sensing resistors and is more power efficient, especially in high current applications. However, as the DCR value drops below 1mΩ, the signal-to-noise ratio is low and current sensing is difficult. LTC7852 uses an LTC proprietary technique to solve this issue. In general, external component selection is driven by the load requirement, and begins with the DCR and inductor value. Next, power MOSFETs are selected. Finally, input and output capacitors are selected. LTC7852-1 is designed for use with DrMOS with a current sensing signal. With DrMOS current sensing, the inductor DCR value does not impact the current sensing/current sharing accuracy, and the maximum current limit could be continuously programmed by external sensing circuitry. equal to one-fifth the L/DCR of the inductor. The SNSAVG pin is connected to the second filter with a time constant three times that of R1• C1. Therefore, the switching ripple at SNSAVG is attenuated. Do not float these pins during normal operation. Filter components, especially capacitors, must be placed close to the LTC7852/LTC7852-1, and the sense lines should run close together to a Kelvin connection underneath the current sense element (Figure 3). The LTC7852 is designed to be used with a very low DCR value to sense inductor current, requiring proper care, during layout of the sense lines. Otherwise, the parasitic resistance, capacitance and inductance will degrade the current sense signal integrity, making the programmed current limit unpredictable. As shown in Figure 4, resistor R1 is placed close to the output inductor and R2, C1, C2 are placed close to the IC pins to prevent noise coupling to the sense signal. TO SENSE FILTER, NEXT TO THE CONTROLLER COUT Current Limit Programming The ILIM pin is a 5-level logic input which sets the maximum current limit of the controller. When ILIM is either grounded, floated or tied to VCC, the typical value for the maximum current sense threshold will be 10mV, 20mV or 30mV, respectively. Set ILIM to one-fourth VCC or three-fourths VCC for maximum current sense thresholds of 15mV and 25mV respectively. Please note that the ILIM pin has an internal 500k pull-down to GND and a 500k pull-up to VCC. For the best current limit accuracy, use the highest setting that is applicable to the output requirements. SNSP, SNSN and SNSAVG (LTC7852 only) Pins The SNSP and SNSN pins are the inputs to the current comparators, while the SNSP and SNSAVG pins are the input of an internal amplifier. The differential signal across SNSP and SNSAVG is an averaged value of the signal across SNSP and SNSN. The operating input voltage range is 0V to 2V for all three sense pins. All the sense pins that are connected to the current comparator or the amplifier are high impedance with input bias currents of less than 1µA. The SNSN should be connected directly to VOUT. The SNSP pin connects to the filter that has a R1• C1 time constant INDUCTOR 78521 F03 Figure 3. Sense Lines Placement with Inductor DCR The LTC7852 could also be used like any typical current mode controller by disabling the SNSAVG pin, tying it to VCC. An RSENSE resistor or a RC filter can be used to sense the output inductor signal and connects to the SNSP pin. If the RC filter is used, its time constant, R•C, should be equal to the L/DCR time constant of the output inductor. Inductor DCR Sensing The LTC7852 is specifically designed for high load current applications requiring the highest possible efficiency; it is capable of sensing the signal of an inductor DCR in the sub milliohm range (Figure 4). The DCR is the DC winding resistance of the inductor’s copper, which is often less than 1mΩ for high current inductors. In high current and low output voltage applications, a conduction loss of a high DCR or a sense resistor will cause a significant reduction in power efficiency. For a specific output requirement,choose the inductor with the DCR that satisfies the maximum desirable sense voltage, and uses the relationship of the Rev. B For more information www.analog.com 15 LTC7852/LTC7852-1 APPLICATIONS INFORMATION 5V VIN VLOGIC 2.2µF VCC VDD IN PWM INDUCTOR BOOST LTC4449 TG L VOUT BG LTC7852 SNSP DCR TS R1 C2 SNSAVG C1 R2 SNSN PLACE C1, C2, R2 NEXT TO IC. PLACE R1 NEXT TO INDUCTOR. 78521 F04 Figure 4. Inductor DCR Current Sensing sense pin filters to output inductor characteristics as depicted below. DCR = VSENSE(MAX) IMAX + � IL 2 L/DCR = 5 • R1 • C1= 1.6 • R2• C2 where: VSENSE(MAX): Maximum sense voltage for a given ILIM threshold IMAX: Maximum load current ΔIL: Inductor ripple current Ensure that R1 has a power rating higher than this value. However, DCR sensing eliminates the conduction loss of a sense resistor; it will provide a better efficiency at heavy loads. To maintain a good signal-to-noise ratio for the current sense signal, using a minimum ∆VSENSE of 2mV for duty cycles less than 40% is desirable. The actual ripple voltage will be determined by the following equation: V V –V � VSENSE = OUT • IN OUT VIN R 1C 1 • fOSE DrMOS Current Sensing The LTC7852-1 is designed to work with DrMOS which has built-in current sensing. The SNSN pins are regulated at 1.5V and a 2.2µF ~10µF low ESR ceramic decoupling capacitor to ground is required. L, DCR: Output inductor characteristics R1 • C1: Filter time constant of the SNSN pin Soft-Start R2 • C2: Filter time constant of the SNSAVG pin The LTC7852/LTC7852-1 has the ability to soft-start by itself. A capacitor may be connected to its SS. The controller is in the shutdown state if its RUN pin voltage is below 1.22V. Its SS pin is actively pulled to ground in this shutdown state. If the RUN pin voltage is above 1.22V, the controller powers up. A soft-start current of 5µA then starts to charge the SS soft-start capacitor. Note that soft-start is achieved not by limiting the maximum output current of the controller but by controlling the output ramp voltage according to the ramp rate on the SS pin. The soft-start range is defined to be the voltage range from 0V to 0.5V on the SS pin. The total soft-start time can be calculated as: Typically, C1 and C2 are selected in the range of 0.047μF to 0.47μF. If C1 and C2 are chosen to be 220nF, and an inductor of 250nH with 0.32mΩ DCR is selected, R1 and R2 will be 715Ω and 2.21k respectively. There will be some power loss in R1 that relates to the duty cycle, and will be the most in continuous mode at the maximum input voltage: PLOSS (R1) = ( VIN(MAX) � VOUT ) • VOUT R1 16 C tSOFTSTART = 0.5V • SS 5µA For more information www.analog.com Rev. B LTC7852/LTC7852-1 APPLICATIONS INFORMATION When SS is rising from 0V to 0.4V, the controller disables the bottom MOSFET until ITH rises above 0.8V so that it always starts in discontinuous mode. After SS > 0.4V, the controller is in forced continuous mode, ensuring a clean PGOOD signal. Pre-Biased Output Start-Up There may be situations that require the power supply to start up with a pre-bias on the output capacitors. In this case, it is desirable to start up without discharging that output pre-bias. The LTC7852/LTC7852-1 can safely power up into a pre-biased output without discharging it. The LTC7852/LTC7852-1 accomplishes this by disabling both the top and bottom MOSFETs until the SS pin voltage and the internal soft-start voltage are above the VOSNS+ pin voltage. When VOSNS+ is higher than SS or the internal soft-start voltage, the error amp output is railed low. The control loop would like to turn the bottom MOSFET on, which would discharge the output. Disabling both top and bottom MOSFETs prevents the pre-biased output voltage from being discharged. When SS and the internal soft-start both cross 500mV or VOSNS+, whichever is lower, the top MOSFET is enabled. The bottom MOSFET is enabled later on after ITH rises above 800mV. If the pre-bias is higher than the OV threshold, the bottom gate is turned on immediately to pull the output back into the regulation window. Overcurrent Fault Recovery When the output of the power supply is loaded beyond its preset current limit, the regulated output voltage may collapse depending on the load. The output may be shorted to ground through a very low impedance path or it may be a resistive short, in which case the output will collapse partially, until the load current equals the preset current limit. The controller will continue to source current into the short for 32 switching periods. The amount of current sourced depends on the ILIM pin setting. If the overcurrent fault still exists after 32 switching periods, the controller enters hiccup mode. The ITH pin will be pulled to ground by an internal MOSFET, and therefore both the MOSFETs are off. The SS soft-start capacitor will be discharged by a 2.5µA current. When the SS reaches to ground, the ITH is released and the circuit retries to soft-start, as described in the section Shutdown and Start-Up. The hiccup overcurrent protection is not disabled during soft-start period.The sleep time is estimated by: � 2.7V 3.3V � t SLEEP = C SS � + ÷ � ISS 2.5µA � ITH Fault Conditions: Current Limit The LTC7852/LTC7852-1’s current limiting is not disabled during soft-start. Under short-circuit conditions with very low duty cycles, the LTC7852/LTC7852-1 will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short circuit ripple current is determined by the minimum on-time tON(MIN) of the LTC7852/LTC7852-1(≈40ns with power stage), the input voltage and inductor value: V � IL(SC) = tON(MIN) • IN L The resulting short-circuit current is: � � VSENSE(MAX) 1 ISC = � � � IL(SC) ÷ 2 � R SENSE � SS SW1 SW2 Figure 5. Hiccup Mode Overcurrent Protection and Recovery If the short is removed before the 32 switching period timer expires, the output soft recovers using the internal softstart, thus reducing output overshoot. In the absence of this feature, the output capacitors would have been charged at current limit, and in applications with minimal output capacitance this may have resulted in output overshoot. Rev. B For more information www.analog.com 17 LTC7852/LTC7852-1 APPLICATIONS INFORMATION Inductor Value Calculation Given the desired input and output voltages, the inductor value and operating frequency, fOSC, directly determine the inductor’s peak-to-peak ripple current: �V � V � V OUT IRIPPLE = OUT � IN ÷ VIN � fOSC • L � Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors, and output voltage ripple. Thus, highest efficiency operation is obtained at low frequency with a small ripple current. Achieving this, however, requires a large inductor. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). Note that the largest ripple current occurs at the highest input voltage. To guarantee that ripple current does not exceed a specified maximum, the inductor should be chosen according to: L� VIN � VOUT V • OUT fOSC • IRIPPLE VIN Inductor Core Selection Once the inductance value is determined, the type of inductor must be selected. Core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Power MOSFET and Schottky Diode (Optional) Selection At least two external power MOSFETs need to be selected: One N-channel MOSFET for the top (main) switch and one or more N-channel MOSFET(s) for the bottom (synchronous) switch. The number, type and on-resistance of all MOSFETs selected take into account the voltage step-down ratio as well as the actual position (main or synchronous) in which the MOSFET will be used. A much smaller and much lower input capacitance MOSFET should be used for the top MOSFET in applications that have an output voltage that is less than one-third of the input voltage. In applications where VIN >> VOUT, the top MOSFETs’ onresistance is normally less important for overall efficiency than its input capacitance at operating frequencies above 300kHz. MOSFET manufacturers have designed special purpose devices that provide reasonably low on-resistance with significantly reduced input capacitance for the main switch application in switching regulators. The peak-to-peak MOSFET gate drive levels are set by the bias voltage of the driver, requiring the use of logiclevel threshold MOSFETs in most applications. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic-level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the on-resistance, RDS(ON), input capacitance, input voltage and maximum output current. MOSFET input capacitance is a combination of several components but can be taken from the typical gate charge curve included on most data sheets (Figure 6). The curve is generated by forcing a constant input current into the gate of a common source, current source loaded stage and then plotting the gate voltage versus time. VIN MILLER EFFECT VGS PWM Pins The PWM pins are three-state compatible outputs, designed to drive MOSFET drivers and DrMOSs which do not represent a heavy capacitive load. An external resistor divider may be used to set the voltage to mid-rail while in the high impedance state. a b + VGS QIN + VDS – – 78521 F06 Figure 6. Gate Charge Characteristic Rev. B 18 For more information www.analog.com LTC7852/LTC7852-1 APPLICATIONS INFORMATION The initial slope is the effect of the gate-to-source and the gate-to-drain capacitance. The flat portion of the curve is the result of the Miller multiplication effect of the drain-to-gate capacitance as the drain drops the voltage across the current source load. The upper sloping line is due to the drain-to-gate accumulation capacitance and the gate-to-source capacitance. The Miller charge (the increase in coulombs on the horizontal axis from a to b while the curve is flat) is specified for a given VDS drain voltage, but can be adjusted for different VDS voltages by multiplying the ratio of the application VDS to the curve specified VDS values. A way to estimate the CMILLER term is to take the change in gate charge from points a and b on a manufacturer’s data sheet and divide by the stated VDS voltage specified. CMILLER is the most important selection criterion for determining the transition loss term in the top MOSFET but is not directly specified on MOSFET data sheets. CRSS and COS are specified sometimes but definitions of these parameters are not included. When the controller is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT VIN �V � V � OUT Synchronous Switch Duty Cycle = � IN ÷ VIN � � The power dissipation for the main and synchronous MOSFETs at maximum output current are given by: V 2 PMAIN = OUT (IMAX ) (1+ � ) RDS(ON) + VIN ( VIN ) 2 �IMAX � � ÷(RDR ) (CMILLER ) • � 2 � � � 1 1 � �• f + �� VINTVCC � VTH(MIN) VTH(MIN) �� PSYNC = VIN � VOUT VIN 2 (IMAX ) (1+ � ) RDS(ON) where δ is the temperature dependency of RDS(ON), RDR is the effective top driver resistance (approximately 2Ω at VGS = VMILLER), VIN is the drain potential and the change in drain potential in the particular application. VTH(MIN) is the data sheet specified typical gate threshold voltage specified in the power MOSFET data sheet at the specified drain current. CMILLER is the calculated capacitance using the gate charge curve from the MOSFET data sheet and the technique described. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which peak at the highest input voltage. For VIN < 20V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. An optional Schottky diode across the synchronous MOSFET conducts during the dead time between the conduction of the two large power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead time and requiring a reverserecovery period which could cost as much as several percent in efficiency. A 2A to 8A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition loss due to their larger junction capacitance. MOSFET Driver Selection Gate driver ICs, DrMOSs and power blocks with an interface compatible with the LTC7852/LTC7852-1’s three-state PWM outputs can be used. Always enable the power stage first, before the LTC7852/LTC7852-1 is enabled. Rev. B For more information www.analog.com 19 LTC7852/LTC7852-1 APPLICATIONS INFORMATION CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle (VOUT)/(VIN). To prevent large voltage transients, a low ESR capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: 1/2 I CIN Required IRMS  MAX ��( VOUT ) ( VIN � VOUT )�� VIN  This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturers’ ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet size or height requirements in the design. Due to the high operating frequency of the LTC7852/LTC7852-1, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. Ceramic capacitors are becoming very popular for small designs but several cautions should be observed. X7R, X5R and Y5V are examples of a few of the ceramic materials used as the dielectric layer, and these different dielectrics have very different effect on the capacitance value due to the voltage and temperature conditions applied. Physically, if the capacitance value changes due to applied voltage change, there is a concomitant piezo effect which results in radiating sound! A load that draws varying current at an audible rate may cause an attendant varying input voltage on a ceramic capacitor, resulting in an audible signal. A secondary issue relates to the energy flowing back into a ceramic capacitor whose capacitance value is being reduced by the increasing charge. The voltage can increase at a considerably higher rate than the constant current being supplied because the capacitance value is decreasing as the voltage is increasing! Nevertheless, ceramic capacitors, when properly selected and used, can provide the lowest overall loss due to their extremely low ESR. A small (0.1µF to 1µF) bypass capacitor, CIN, between the chip VIN pin and ground, placed close to the LTC7852/ LTC7852-1, is also suggested. A 2.2Ω to 10Ω resistor placed between CIN and VIN pin provides further isolation. The selection of COUT is driven by the required effective series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering. The steady-state output ripple (∆VOUT) is determined by: � 1 � � VOUT  � IRIPPLE �ESR + ÷ 8fCOUT � �  where f = operating frequency, COUT = output capacitance and ∆IRIPPLE = ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IRIPPLE increases with input voltage. The output ripple will be less than 50mV at maximum VIN with ∆IRIPPLE = 0.4IOUT(MAX) assuming: COUT required ESR < N • RSENSE and COUT > 1 (8f)(R SENSE ) The emergence of very low ESR capacitors in small, surface mount packages makes very small physical implementations possible. The ability to externally compensate the switching regulator loop using the ITH pin allows a much wider selection of output capacitor types. The impedance characteristic of each capacitor type is significantly different than an ideal capacitor and therefore requires accurate modeling or bench evaluation during design. Manufacturers such as Nichicon, Nippon Chemi-Con and Sanyo should be considered for high performance through-hole capacitors. The OS-CON semiconductor dielectric capacitors available from Sanyo and the Panasonic SP surface mount types have a good ESR • size product. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. Ceramic capacitors from AVX, Taiyo Yuden, Murata and TDK offer high capacitance value and very low ESR, especially applicable for low output voltage applications. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available Rev. B 20 For more information www.analog.com LTC7852/LTC7852-1 APPLICATIONS INFORMATION in surface mount configurations. New special polymer surface mount capacitors offer very low ESR also but have much lower capacitive density per unit volume. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. Several excellent choices are the AVX TPS, AVX TPSV, the KEMET T510 series of surface mount tantalums or the Panasonic SP series of surface mount special polymer capacitors available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo POSCAP, Sanyo OS-CON, Nichicon PL series and Sprague 595D series. Consult the manufacturers for other specific recommendations. Differential Amplifier VDD and V1P5 LDO The LTC7852/LTC7852-1 features a true PMOS LDO that supplies power to VDD and V1P5 from the VCC supply. The VDD and V1P5 must be bypassed to ground with a minimum of 2.2µF ceramic capacitor or low ESR electrolytic capacitor. No matter what type of bulk capacitor is used, an additional 0.1µF ceramic capacitor placed directly adjacent to the VDD and GND pins is highly recommended. Please do not load the LDO with an external circuit at the VDD pin. VDD must be within approximately 7% of its targeted value before the RUN pin is released. In addition, when V1P5 is approximately 20% below its regulated value, the controller is kept in shutdown. The LTC7852/LTC7852-1 has true remote voltage sense capability. The sense connections should be returned from the load, back to the differential amplifier’s inputs through a common, tightly coupled pair of PC traces. The differential amplifier rejects common mode signals capacitively or inductively radiated into the feedback PC traces as well as ground loop disturbances. The LTC7852/ LTC7852-1 diffamp has high input impedance on VOSNS+ pin. The output of the diffamp connects to the inverting input of the error amplifier internally. Phase-Locked Loop and Frequency Synchronization Setting Output Voltage The output of the phase detector is a pair of complementary current sources that charge or discharge the internal filter network. There is a precision 20µA current flowing out of the FREQ pin. This allows the user to use a single resistor to GND to set the switching frequency when no external clock is applied to the PLLIN pin. Do not program the FREQ pin voltage below 0.57V. The internal switch between the FREQ pin and the integrated PLL filter network is on, allowing the filter network to be pre-charged at the same voltage as of the FREQ pin. The relationship between the voltage on the FREQ pin and operating frequency is shown in Figure 7 and specified in the Electrical Characteristics table. If an external clock is detected on the PLLIN pin, the internal switch mentioned above turns off and isolates the influence of the FREQ pin. Note that the LTC7852/LTC7852-1 can only be synchronized to an external clock whose frequency is within range of the LTC7852/LTC7852-1’s internal VCO. Do not synchronize to a clock below 250kHz. A simplified block diagram is shown in Figure 8. The LTC7852/LTC7852-1 output voltage is set by an external feedback resistive divider carefully placed across the output, as shown in Figure 2. The regulated output voltage is determined by: � R � VOUT = 0.5V • �1+ D1 ÷ � RD2 � To improve the frequency response, a feedforward capacitor, CF1, may be used. Great care should be taken to route the VOSNS+ line away from noise sources, such as the inductor or the SW line. To minimize the effect of the voltage drop caused by high current flowing through board conductance; connect VOSNS– and VOSNS+ sense lines close to the ground and the load output respectively. The LTC7852/LTC7852-1 has a phase-locked loop (PLL) comprised of an internal voltage-controlled oscillator (VCO) and a phase detector. This allows the turn-on of the top MOSFET to be locked to the rising edge of an external clock signal applied to the PLLIN pin. The phase detector is an edge sensitive digital type that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector does not exhibit false lock to harmonics of the external clock. Rev. B For more information www.analog.com 21 LTC7852/LTC7852-1 APPLICATIONS INFORMATION Using the CLKOUT and PHCFG Pins in Multiphase Applications 1300 FREQUENCY (kHz) 1100 The LTC7852/LTC7852-1 features CLKOUT and PHCFG pins that allow multiple LTC7852/LTC7852-1 ICs to be daisy chained together in multiphase applications. The clock output signal on the CLKOUT pin can be used to synchronize additional ICs in a 8-, 10- or 12-phase power supply solution feeding a single high current output, or even several outputs from the same input supply. 900 700 500 300 100 0.4 0.6 0.8 1.0 1.2 VFREQ (V) 1.4 1.6 The PHCFG pin is used to adjust the phase relationship between six channels, as well as the phase relationship between channel 1 and CLKOUT. The phases are calculated relative to zero degrees, defined as the rising edge of PWM1. Refer to the Applications Information section for more details on how to create multiphase applications. 1.8 78521 F07 Figure 7. Relationship Between Oscillator Frequency and Voltage at the FREQ Pin 3.3V VCC RSET 20µA FREQ EXTERNAL OSCILLATOR PLLIN DIGITAL SYNC PHASE/ FREQUENCY DETECTOR VCO 78521 F08 Figure 8. Phase-Locked Loop Block Diagram If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced continuously from the phase detector output, pulling up the filter network. When the external clock frequency is less than fOSC, current is sunk continuously, pulling down the filter network. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. The voltage on the filter network is adjusted until the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the phase detector output is high impedance and the filter capacitor CLP holds the voltage. Typically, the external clock (on the PLLIN pin) input high threshold is 1.6V, while the input low threshold is 1V. 22 Minimum On-Time Considerations Minimum on-time, tON(MIN), is the smallest time duration that the LTC7852/LTC7852-1 is capable of turning on the top MOSFET. It is determined by internal timing delays, power stage timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: V tON(MIN) < OUT VIN (f) If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the voltage ripple and current ripple will increase. The minimum on-time for the LTC7852/LTC7852-1 is approximately 40ns, with good PCB layout, minimum 30% inductor current ripple and at least 2mV ripple on the current sense signal. The minimum on-time can be affected by PCB switching noise in the voltage and current loop. As the peak sense voltage decreases the minimum on time gradually increases This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. For more information www.analog.com Rev. B LTC7852/LTC7852-1 APPLICATIONS INFORMATION Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC7852/LTC7852-1 circuits: 1) IC VCC current, 2) MOSFET driver current, 3) I2R losses, 4) topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics table. VIN current typically results in a small (1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of CLOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. These items are also illustrated graphically in the layout diagram of Figure 9. Check the following in the PC layout: 1. The VCC, VDD, V1P5 decoupling capacitor should be placed immediately adjacent to the IC between the VCC pin and GND plane. A 1µF ceramic capacitor of the X7R or X5R type is small enough to fit very close to the IC. An additional 4.7µF to 10µF of ceramic, tantalum or other very low ESR capacitance is recommended in order to keep the internal IC supply quiet. 2. Place the feedback divider between the + and – terminals of COUT. Route VOSNS+ and VOSNS– with minimum PC trace spacing from the IC to the feedback divider. 3. Are the SNSAVG, SNSP and SNSN printed circuit traces routed together with minimum PC trace spacing? The filter capacitors between SNSAVG, SNSP and SNSN should be as close as possible to the pins of the IC. 4. Do the (+) plates of CIN connect to the drain of the topside MOSFET as closely as possible? This capacitor provides the pulsed current to the MOSFET. 5. Keep the switching nodes away from sensitive small signal nodes (SNSP, SNSAVG, SNSN, VOSNS+, VOSNS–). Ideally the PWM and switch nodes printed circuit traces should be routed away and separated from the IC and especially the quiet side of the IC. Separate the high dv/ dt traces from sensitive small-signal nodes with ground traces or ground planes. 6. Use a low impedance source such as a logic gate to drive the PLLIN pin and keep the lead as short as possible. Rev. B 24 For more information www.analog.com LTC7852/LTC7852-1 APPLICATIONS INFORMATION RIN RSENSE L1 VIN + SW2 CIN VOUT + SW1 COUT RL 78521 F09 BOLD LINES INDICATE HIGH, SWITCHING CURRENTS. KEEP LINES TO A MINIMUM LENGTH. Figure 9. Branch Current Waveforms 7. The 47pF to 330pF ceramic capacitor between the ITH pin and signal ground should be placed as close as possible to the IC. Figure 9 illustrates all branch currents in a switching regulator. It becomes very clear after studying the current waveforms why it is critical to keep the high switching current paths to a small physical size. High electric and magnetic fields will radiate from these loops just as radio stations transmit signals. The output capacitor ground should return to the negative terminal of the input capacitor and not share a common ground path with any switched current paths. The left half of the circuit gives rise to the noise generated by a switching regulator. The ground terminations of the synchronous MOSFET and Schottky diode should return to the bottom plate(s) of the input capacitor(s) with a short isolated PC trace since very high switched currents are present. External OPTI-LOOP® compensation allows overcompensation for PC layouts which are not optimized but this is not the recommended design procedure. 8. Are the signal and power grounds kept separate? The IC ground pin and the ground return of CINTVCC must return to the combined COUT (–) terminals. The VOSNS+ and ITH traces should be as short as possible. The path formed by the top N-channel MOSFET, Schottky diode and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. 9. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTVCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the GND pin of the IC. Design Example A design example of a 6-phase high current regulator is shown in Figure 10. Assume VIN = 12V(nominal), VIN = 20V(maximum), VOUT = 1.0V, IMAX = 200A, and f = 400kHz. The regulated output voltage is determined by: � R � VOUT = 0.5V • �1+ D1 ÷ � RD2 � Using a 20k 1% resistor from the VFB node to ground, the top feedback resistor is 20k. The frequency is set by biasing the FREQ pin to 0.75V (see Figure 7). The inductance value is based on a 30% maximum ripple current assumption (10A per phase). The highest value of ripple current occurs at the maximum input voltage: L= � VOUT � �1� ÷ f • � IL(MAX) �� VIN(MAX) ÷ � VOUT Rev. B For more information www.analog.com 25 LTC7852/LTC7852-1 APPLICATIONS INFORMATION This design will require 0.23µH. The Wurth 744301025, 0.25µH inductor is chosen. At the nominal input voltage (12V), the ripple current will be: � V VOUT � ÷ � IL(NOM) = OUT ��1� f • L � VIN(NOM) ÷ � The power dissipation on the topside MOSFET can be easily estimated. Choosing an Infineon BSC050NE2LS MOSFET results in: RDS(ON) = 7.1mΩ (max), VMILLER = 2.8V, CMILLER ≅ 108pF. At maximum input voltage with TJ (estimated) = 75°C. PMAIN = It will have 9.2A (28%) ripple. The peak inductor current will be the maximum DC value plus one-half the ripple current, or 38A per phase. VOUT VIN(MAX)f = 1.0V 20V(400kHz) = 124ns DCR sensing is used in this circuit. If C1 and C2 are chosen to be 220nF, based on the chosen 0.25µH inductor with 0.32mΩ DCR, R1 and R2 can be calculated as: � 1 � 1 + (400kHz) � �5V � 2.8V 2.8V � � = 492mW + 467mW = 959mW / phase An Infineon BSC010NE2LS, RDS(ON) = 1.3mΩ (Max), is chosen for the bottom FET. The resulting power loss is: PSYNC = R2 = L/DCR • C2 • 1.6 = 2.22k Choose R1 = 715Ω and R2 = 2.32k. The maximum DCR of the inductor is 0.34mΩ. The VSENSE(MAX) is calculated as: The current limit is chosen to be 15mV. (33.3A)2 [1+ (0.005)(75°C � 25°C)] • 20V � 1.0V (33.3A)2 • 20V [1+ (0.005) • (75° � 25°C)] • 0.0013� R1 = L/DCR • C1 • 5 = 710Ω VSENSE(MAX) = IPEAK • DCRMAX = 12mV 20V � 33.3A � (0.0071� ) + (20V)2 � ÷(2� )(108pF) • � 2 � The minimum on-time occurs at the maximum VIN, and should not be less than 100ns (includes margin): tON(MIN) = 1.0V PSYNC = 1.7W / Phase CIN is chosen for an equivalent RMS current rating of at least 13.7A. COUT is chosen with an equivalent ESR of 4.5mΩ for low output ripple. The output ripple is continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR (∆IL) = 0.0045Ω • 10A = 45mVP-P Further reductions in output voltage ripple can be made by placing a 100µF ceramic capacitor across COUT. Rev. B 26 For more information www.analog.com LTC7852/LTC7852-1 TYPICAL APPLICATIONS VIN 5V TO 13V + L1,2,3 0.25µH 0.32mΩ + 330µF ×9 VCC 4.5V TO 5.5V 4.7µF LTC4449 VCC M1 2.2µF 4.7µF IMON1 V1P5 IMON2 VDD PGOOD1 PGOOD2 PWM1-3 PWM4-6 LTC4449 M3 M2 L4,5,6 0.25µH 0.32mΩ + M4 LTC7852 PLLIN 715Ω 220nF 220nF CLKOUT SNSP4-6 SNSP1-3 330µF ×9 VOUT 1.0V 200A 220nF 220nF 715Ω SNSAVG4-6 SNSAVG1-3 2.32k 2.32k SNSN1-3 SNSN4-6 VOSNS1+ VOSNS2+ VOSNS1– VOSNS2– 20k 20k ITH1 ITH2 SS1 SS2 RUN1 ILIM1 FREQ GND PHCFG ILIM2 RUN2 L1 - L6: Wurth 744301025 M1, M3: BSC050NE2LS M2, M4: BSC010NE2LS 2.49k 150pF x2 0.47µF 5.6nF VCC 37.4k 10k 3.32k 78521 F10 Figure 10. Single Output 6-Phase 1V/200A LTC7852 Converter with Discrete Drivers and MOSFETs Rev. B For more information www.analog.com 27 LTC7852/LTC7852-1 TYPICAL APPLICATIONS VIN 5V TO 13V VOUT1 0.9V 120A + VCC 4.5V TO 5.5V VCC L1-4 0.25µH 0.32mΩ + 2.2µF 4.7µF DrMOS IMON1 V1P5 IMON2 PGOOD1 PGOOD2 PWM1-4 PWM5-6 330µF ×12 PLLIN 715Ω LTC7852 SNSP1-4 220nF VDD 18.7k 220nF 220nF 220nF SNSAVG5-6 2.32k SNSN1-4 SNSN5-6 VOSNS1+ VOSNS2+ VOSNS1– VOSNS2– 28k 20k 3.01k 150pF ITH1 ITH2 SS1 SS2 0.22µF RUN1 ILIM1 FREQ GND PHCFG ILIM2 RUN2 L1 - L6: Wurth 744301025 DrMOS: FDMF5820DC 330µF ×6 715Ω 3.01k 3.3nF + VOUT2 1.2V 60A CLKOUT 2.32k 15k DrMOS SNSP5-6 SNSAVG1-4 L5-6 0.25µH 0.32mΩ 150pF 3.3nF 0.22µF 37.4k 78521 F11 Figure 11. Dual Output 4-Phase 0.9V/120A and 2-Phase 1.2V/60A LTC7852 Converter with DRMOS Rev. B 28 For more information www.analog.com LTC7852/LTC7852-1 PACKAGE DESCRIPTION RHE Package RHE Package 48-Lead GQFN (5mm (5mm ×× 6mm) 6mm) 48-Lead Plastic Plastic GQFN (Reference LTC DWG Rev Ø) Ø) (Reference LTC DWG ## 05-08-1527 05-08-1527 Rev 2.60 ±0.10 3.60 ±0.05 1.90 ±0.10 PACKAGE OUTLINE 5.20 ±0.05 RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 5.00 ±0.10 PIN 1 NOTCH 0.35 × 45° CHAMFER 0.80 REF 1.00 REF 39 48 38 PIN 1 TOP MARK (SEE NOTE 6) 1 0.55 REF 0.25 REF 1.90 ±0.10 6.00 ±0.10 2.60 ±0.10 0.40 BSC 0.85 ±0.10 0.20 ±0.05 14 24 (RHE48) GQFN 0116 REV Ø 24 15 1.10 ±0.10 0.65 ±0.05 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS BOTTOM VIEW—EXPOSED PAD 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.25mm ON ANY SIDE 5. EXPOSED PAD SHALL BE Pd Ni Au PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE Rev. B For more information www.analog.com 29 LTC7852/LTC7852-1 PACKAGE DESCRIPTION UFD Package 36-Lead Plastic QFN (4mm × 5mm) (Reference LTC DWG # 05-08-1575 Rev Ø) 0.70 ±0.05 4.50 ±0.05 3.10 ±0.05 2.80 REF 2.65 ±0.05 3.65 ±0.05 PACKAGE OUTLINE 0.15 REF DETAIL A 0.20 ±0.05 0.40 BSC 3.60 REF 4.10 ±0.05 5.50 ±0.05 0.23 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 ±0.10 (2 SIDES) R = 0.05 TYP 0.75 ±0.05 PIN 1 NOTCH 0.35 × 45° CHAMFER 2.80 REF R = 0.100 TYP 35 36 0.40 ±0.10 PIN 1 TOP MARK (NOTE 6) 28 1 2 5.00 ±0.10 (2 SIDES) 3.60 REF 3.65 ±0.10 DETAIL A 2.65 ±0.10 0.275 REF 0.200 REF 0.00 – 0.05 (UFD36) QFN 0317 REV Ø 18 0.20 ±0.05 0.40 BSC BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGHD-3). 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE Rev. B 30 For more information www.analog.com LTC7852/LTC7852-1 REVISION HISTORY REV DATE DESCRIPTION A 11/19 Changed RUN Pin Note to 5μA PAGE NUMBER 8 Changed Main Control Loop and Shutdown/Start-up Current to 1.3μA 28 B 01/21 Added Pad Labels 3 Rev. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license For is granted implication or otherwise under any patent or patent rights of Analog Devices. more by information www.analog.com 31 LTC7852/LTC7852-1 TYPICAL APPLICATION VCC 4.5V TO 5.5V 4.7µF VIN 5V TO 12V 2.2µF VCC VOUT 0.5V 330A VDD PGOOD1 PGOOD2 PWM6 PWM1-5 CLKOUT VCC VDRV VIN PWM TDA21470 SW REFIN IOUT PGND PLLIN R3 1k LTC7852-1 SNSN SNSN C5 R10 1k C2 0.25µH L1~5 VOSNS1+ VOSNS2– VOSNS1– ITH2 ITH1 SS2 VIN VDRV VCC TDA21470 PWM SW PGND IOUT R1 10k 330µF X15 330µF X15 10nF R8 10k C16 10nF SS1 PWM6 CLKOUT PLLIN LTC7852-1 2.2µF C13 SNSN SNSN SNSP6 SNSP1-5 100pF 6.8nF PGOOD1 PWM1-5 R13 1k R9 1k 1.54k 100pF VCC REFIN 10nF 10k SNSP1-5 VOSNS2+ VDD PGOOD2 0.25µH L6~10 10n SNSP6 RUN2 ILIM2 2.2µF 4.7µF 2.2µF VOSNS1+ VOSNS2+ VOSNS1– VOSNS2– ITH1 ITH2 SS1 FREQ GND PHCFG ILIM1 RUN1 SS2 RUN1 ILIM1 PHCFG GND FREQ C12 0.1µF ILIM2 RUN2 37.4k 37.4K R11 1k VCC R12 3.01k VCC 7852 TA02 L1~L10: Wurth 744309025 Figure 12. Single Output, 10-Phase 0.5V/330A Converter Using LTC7852-1 with DrMOS RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC7851 Quad Output Multiphase Step-Down Voltage Mode DC/DC Operates with DrMOS, Power Blocks or External Drivers/MOSFETs, Controller with Accurate Current Sharing VIN Range Depends on External Components, 4.5V≤ VCC ≤ 5.5V, 0.6V ≤ VOUT ≤ VCC –0.5V LTC3861 Dual, Multiphase Step-Down Voltage Mode DC/DC Controller with Accurate Current Sharing Operates with Power Blocks, DrMOS or External MOSFETs 3V≤ VIN ≤ 24V LTM4650/LTM4650A Dual 25A or Single 50A Step-Down DC/DC µModule Regulator 4.5V ≤ VIN ≤ 15V, 0.6V ≤ VOUT ≤ 1.8V (LTM4650), 5.5V (LTM4650A) 16mm × 16mm × 4.4mm BGA and LGA Packages LTM4678 Dual 25A or Single 50A μModule Regulator with Digital Power System Management 4.5V ≤ VIN ≤16V; 0.5V ≤ VOUT ≤ 3.3V I2C/PMBus Interface, 16mm × 16mm × 5.86mm, BGA Package LTC3774 Dual, Mulitphase Curent Mode Synchronous Step-Down DC/DC Controller for Sub-Milliohm DCR Sensing Operates with DrMOS, Power Blocks or External Drivers/MOSFETs, 4.5V≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 3.5V LTC3875 Dual, 2-Phase, Synchronous Controller with SubMilliohm DCR Sensing and Temperature Compensation 4.75V ≤ VIN ≤38V; 0.6V ≤ VOUT ≤ 3.5V/5V, Excellent current Share when Paralleled LTC3884 Dual Output MultiPhase Step-Down Controller with SubMilliOhm DCR Sensing Current Mode Control and Digital Power System Management 4.5V ≤ VIN ≤ 38V, 0.5V ≤ VOUT (±0.5%) ≤ 5.5V, 70mS Start-Up, I2C/ PMBus Interface, Programmable Analog Loop Compensation, Input Current Sense LTC3882/LTC3882-1 Dual Output Multiphase Step-Down DC/DC Voltage Mode 3V ≤ VIN ≤ 38V, 0.5V ≤ VOUT1,2 ≤ 5.25V, ±0.5% VOUT Accuracy I2C/ Controller with Digital Power System Management PMBus Interface, uses DrMOS or Power Blocks LTC3887/LTC3887-1 Dual Output Multiphase Step-Down DC/DC Controller with Digital Power System Management, 70mS Start-Up 4.5V ≤ VIN ≤ 24V, 0.5V ≤ VOUT0,1 (±0.5%) ≤ 5.5V, 70mS Start-Up, I2C/ PMBus Interface, –1 Version uses DrMOS or Power Blocks LTC4449 High Speed Synchronous N-Channel MOSFET Driver VIN up to 38V, 4V ≤ VCC ≤ 6.5V Adaptive Shoot-Through Protection, 2mm × 3mm DFN-8 Rev. B 32 01/21 www.analog.com For more information www.analog.com  ANALOG DEVICES, INC. 2019–2021