19-3289; Rev 1; 6/05
KIT
ATION
EVALU
LE
B
A
IL
A
AV
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
The MAX8576–MAX8579 synchronous PWM buck controllers use a hysteretic voltage-mode control algorithm
to achieve a fast transient response without requiring
loop compensation. The MAX8576/MAX8577 contain an
internal LDO regulator allowing the controllers to function from only one 3V to 28V input supply. The
MAX8578/MAX8579 do not contain the internal LDO
and require a separate supply to power the IC when the
input supply is higher than 5.5V. The MAX8576–
MAX8579 output voltages are adjustable from 0.6V to
0.9 x VIN at loads up to 15A.
Nominal switching frequency is programmable over the
200kHz to 500kHz range. High-side MOSFET sensing is
used for adjustable hiccup current-limit and short-circuit protection. The MAX8576/MAX8578 can start up
into a precharged output without pulling the output voltage down. The MAX8577/MAX8579 have startup output
overvoltage protection (OVP), and will pull down a
precharged output.
Applications
Features
♦ 3V to 28V Supply Voltage Range
♦ 1.2% Accurate Over Temperature
♦ Adjustable Output Voltage Down to 0.6V
♦ 200kHz to 500kHz Switching Frequency
♦ Adjustable Temperature-Compensated Hiccup
Current Limit
♦ Lossless Peak Current Sensing
♦ Monotonic Startup into Prebias Output
(MAX8576/MAX8578)
♦ Startup Overvoltage Protection
(MAX8577/MAX8579)
♦ Enable/Shutdown
♦ Adjustable Soft-Start
Ordering Information
TEMP RANGE
PIN-PACKAGE
MAX8576EUB
-40°C to +85°C
10 µMAX®
AGP and PCI-Express Power Supplies
MAX8577EUB
-40°C to +85°C
10 µMAX
Graphic-Card Power Supplies
MAX8578EUB
-40°C to +85°C
10 µMAX
Set-Top Boxes
MAX8579EUB
-40°C to +85°C
10 µMAX
Motherboard Power Supplies
PART
Point-of-Load Power Supplies
Typical Operating Circuit
INPUT
UP TO 28V
FB
OCSET
SS
VL
GND
DL
IN
MAX8576
MAX8577
DH
OUTPUT
0.6V TO 0.9 x VIN
LX
BST
µMAX is a registered trademark of Maxim Integrated Products, Inc.
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX8576–MAX8579
General Description
MAX8576–MAX8579
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
ABSOLUTE MAXIMUM RATINGS
IN to GND (MAX8576/MAX8577) ...........................-0.3V to +30V
VL to GND (MAX8576/MAX8577).............................-0.3V to +6V
IN to VL (MAX8576/MAX8577) ...............................-0.3V to +30V
VCC to GND (MAX8578/MAX8579) ..........................-0.3V to +6V
SS to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V
SS to GND (MAX8578/MAX8579)...............-0.3V to (VCC + 0.3)V
DL to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V
DL to GND (MAX8578/MAX8579) ..............-0.3V to (VCC + 0.3)V
BST to GND ............................................................-0.3V to +36V
BST to LX..................................................................-0.3V to +6V
LX to GND .....................-1V (-2.5V for 300mV and an overcurrent event is
detected, DH is immediately set low and four sequential
overcurrent events terminate the run cycle. Once the
run cycle is terminated, the SS capacitor is slowly discharged through the internal 250nA current sink to provide a hiccup current-limit effect. Choosing the proper
value resistor is discussed in the Setting the Current
Limit section.
Switching Frequency
Nominal switching frequency is programmable over the
200kHz to 500kHz range. This allows tradeoffs in efficiency, switching frequency, inductor value, and component size. Faster switching frequency allows for
smaller inductor values but does result in some efficiency loss. Inductor-value calculations are provided in the
Inductor Value section. The switching frequency is
tuned by the selection of the feed-forward capacitor
(CFF). See the Feed-Forward Capacitor section.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation in the MAX8576–MAX8579. When the junction temperature exceeds T J = +160°C, an internal thermal
sensor shuts down the IC, allowing the IC to cool. The
thermal sensor turns the IC on again after the junction
temperature cools to +140°C, resulting in a pulsed output during continuous thermal-overload conditions.
Design Procedures
IN
Setting the Output Voltage
BST
DH
MAX8576–
MAX8579
N
LX
DL
N
Select an output voltage between 0.6V and 0.9 x VIN by
connecting FB to a resistive voltage-divider between LX
and GND (see Figures 2 and 3). Choose R1 for approximately 50µA to 150µA bias current in the resistive
divider. A wide range of resistor values is acceptable,
but a good starting point is to choose R1 as 6.04kΩ.
Then, R3 is given by:
Figure 4. DH Boost Circuit
______________________________________________________________________________________
13
MAX8576–MAX8579
to SS. See Figure 2 for details. The maximum on-resistance of the small external n-channel MOSFET should
be less than 40Ω so that the SS voltage is below 10mV.
MAX8576–MAX8579
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
⎛ VOUT + 0.01V + (RDC × 0.5 × IOUTMAX ) ⎞
R3 = R1 × ⎜
− 1⎟
VFB
⎝
⎠
where VFB = 0.590V, RDC is the DC resistance of the
output inductor, IOUTMAX is the maximum output current. The term 0.01V is to reflect 1/2 of the feedbackthreshold hysteresis.
Inductor Value
The inductor value is bounded by two operating parameters: the switching frequency and the inductor peakto-peak ripple current. The peak-to-peak ripple current
is typically in the range of 20% to 40% of the maximum
output current. The equation below defines the inductance value:
⎛
⎞
VOUT × (VIN − VOUT )
L= ⎜
⎟
⎜ VIN × fS × ILOAD(MAX ) × LIR ⎟
⎝
⎠
where LIR is the ratio of inductor current ripple to DC
load current and fS is the switching frequency. A good
compromise between size, efficiency, and cost is an
LIR of 30%. The selected inductor must have a saturated current rating above the sum of the maximum output
current and half of the peak-to-peak ripple current. The
DC current rating of the inductor must be above the
maximum output current to keep the temperature rise
within the desired range. In addition, the DC resistance
of the inductor must meet the requirement below:
RDC ≤
ΔVOUT
IOUTMAX
where ΔVOUT is the maximum-allowed output-voltage
drop from no load to full load (IOUTMAX).
Setting the Current Limit
Resistor R2 (R7 for the MAX8577/MAX8579) of Figure 2
(Figure 3 for the MAX8577/MAX8579) sets the current
limit and is connected between OCSET and the drain of
the high-side n-channel MOSFET. An internal 50µA
current sink sets the maximum voltage drop across the
high-side n-channel MOSFET relative to VIN. The maximum VDS drop needs to be determined. This is calculated by:
VDS(ON)MAX = IDS(MAX) × RDS(ON)MAX
IDS(MAX) must be equal or greater than the maximum
peak inductor current at the maximum output current.
Use RDS(ON)MAX at the junction temperature of +25°C.
The current limit is temperature compensated.
14
ROCSET is calculated using the VDS(ON)MAX with the
following formula:
VDS(ON)MAX
ROCSET =
50μA
A 0.01µF ceramic capacitor is required in parallel with
ROCSET to decouple high-frequency noise.
MOSFET Selection
The MAX8576–MAX8579 drive two external, logic-level,
n-channel MOSFETs as the circuit switching elements.
The key selection parameters are:
1) On-resistance (RDS(ON)): the lower, the better.
2) Maximum drain-to-source voltage (V DSS): should
be at least 20% higher than the input supply rail at
the high-side MOSFET’s drain.
3) Gate charges (Qg, Qgd, Qgs): the lower, the better.
For a 3.3V input application, choose a MOSFET with a
rated RDS(ON) at VGS = 2.5V. For a 5V input application, choose the MOSFETs with rated RDS(ON) at VGS
≤ 4.5V. For a good compromise between efficiency and
cost, choose the high-side MOSFET (N1) that has conduction losses equal to switching loss at nominal input
voltage and output current. The selected high-side
MOSFET (N1) must have RDS(ON) that satisfies the current-limit-setting condition above. For N2, make sure
that it does not spuriously turn on due to dV/dt caused
by N1 turning on as this results in shoot-through current
degrading the efficiency. MOSFETs with a lower Qgd /
Qgs ratio have higher immunity to dV/dt.
For proper thermal-management design, the power dissipation must be calculated at the desired maximum
operating junction temperature, maximum output current, and worst-case input voltage (for the low-side
MOSFET, worst case is at VIN(MAX); for the high-side
MOSFET, it could be either at VIN(MAX) or VIN(MIN)). N1
and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch;
therefore, major losses are: the channel-conduction
loss (P N2CC ) and the body-diode conduction loss
(PN2DC).
⎛
⎞
V
PN2CC = ⎜1 − OUT ⎟ × ILOAD2 × RDS(ON)
VIN ⎠
⎝
Use RDS(ON) at TJ(MAX).
PN2DC = 2 × ILOAD × VF × t dt × fS
where VF is the body-diode forward-voltage drop, tDT is
the dead time between N1 and N2 switching transitions
(40ns typ), and fS is the switching frequency.
______________________________________________________________________________________
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
⎛V
⎞
PN1CC = ⎜ OUT ⎟ × ILOAD2 × RDS(ON)
⎝ VIN ⎠
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple-current
requirement (IRMS) imposed by the switching currents
defined by the following equation:
IRMS =
ILOAD ×
VOUT × (VIN − VOUT )
VIN
Use RDS(ON) at TJ(MAX).
⎛ Qgs + Qgd ⎞
PN1SW = VIN × ILOAD × ⎜
⎟ × fS
⎝ IGATE ⎠
where IGATE is the average DH driver output-current
capability determined by:
IGATE ≅ 0.5 ×
VL
RDH + RGATE
where RDH is the high-side MOSFET driver’s on-resistance (2Ω typ) and RGATE is the internal gate resistance of the MOSFET (approximately 2Ω).
RGATE
PN1DR = Qg × VGS × fS ×
RGATE + RDH
where VGS is approximately equal to VL.
In addition to the losses above, allow about 20% more
for additional losses due to MOSFET output capacitances and N2 body-diode reverse-recovery charge
dissipated in N1 that exists, but is not well defined in
the MOSFET data sheet. Refer to the MOSFET data
sheet for thermal-resistance specification to calculate
the PC board area needed to maintain the desired maximum operating junction temperature with the above
calculated power dissipations.
To reduce EMI caused by switching noise, add 0.1µF
ceramic capacitor from the high-side switch drain to the
low-side switch source or add resistors in series with
DH and DL to slow down the switching transitions.
However, adding series resistors increases the power
dissipation of the MOSFET, so be sure this does not
overheat the MOSFET.
The minimum load current must exceed the high-side
MOSFET’s maximum leakage current over temperature
if fault conditions are expected.
I RMS has a maximum value when the input voltage
equals twice the output voltage (VIN = 2 x VOUT), so
IRMS(MAX) = ILOAD / 2. Ceramic capacitors are recommended due to their low ESR and ESL at high frequency, with relatively lower cost. Choose a capacitor that
exhibits less than 10°C temperature rise at the maximum
operating RMS current for optimum long-term reliability.
Output Capacitor
The key selection parameters for the output capacitor
are the actual capacitance value, the ESR, the equivalent series inductance (ESL), and the voltage-rating
requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The
output ripple has three components: variations in the
charge stored in the output capacitor, the voltage drop
across the capacitor’s ESR, and the ESL caused by the
current into and out of the capacitor. The maximum output ripple voltage can be estimated by:
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL)
The output voltage ripple as a consequence of the ESR
and output capacitance is:
VRIPPLE(ESR) = IP−P × ESR
VRIPPLE(C) =
IP−P
COUT × fS
⎛V ⎞
VRIPPLE(ESL) = ⎜ IN ⎟ × ESL
⎝ L ⎠
⎛ V − VOUT ⎞ ⎛ VOUT ⎞
IP−P = ⎜ IN
⎟ × ⎜ V ⎟
fS × L
⎝
⎠ ⎝ IN ⎠
where IP-P is the peak-to-peak inductor current (see the
Inductor Value section). These equations are suitable
for initial capacitor selection, but final values should be
______________________________________________________________________________________
15
MAX8576–MAX8579
N1 operates as a duty-cycle control switch and has the
following major losses: the channel-conduction loss
(PN1CC), the VL overlapping switching loss (PN1SW),
and the drive loss (PN1DR). N1 does not have bodydiode conduction loss because the diode never conducts current.
MAX8576–MAX8579
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
chosen based on a prototype or evaluation circuit. As a
general rule, a smaller current ripple results in less output voltage ripple. Since the inductor ripple current is a
factor of the inductor value and input voltage, the output
voltage ripple decreases with larger inductance and
increases with higher input voltages. For reliable and
safe operation, ensure that the capacitor’s voltage and
ripple-current ratings exceed the calculated values.
The response of the MAX8576–MAX8579 to a load
1transient depends on the selected output capacitors.
After a load transient, the output voltage instantly
changes by ESR times ΔILOAD. Before the controller
can respond, the output voltage deviates further
depending on the inductor and output capacitor values. The controller responds immediately as the output
voltage deviates from its regulation limit (see the
Typical Operating Characteristics).
The MAX8576–MAX8579 are compatible with both aluminum electrolytic and ceramic output capacitors. Due
to the limited capacitance of a ceramic capacitor, it is
typically used for a higher switching frequency and
lower output current. Aluminum electrolytic is more
applicable to frequencies up to 300kHz and can support higher output current with its much higher capacitance value.
Due to the much higher ESL and ESR of the aluminum
electrolytic capacitor, an RC filter (R7 and C12 of Figure
2) is required to prevent excessive ESL and ESR ripple
from tripping the feedback threshold prematurely.
MOSFET Snubber Circuit
Fast-switching transitions cause ringing because of
resonating circuit parasitic inductance and capacitance at the switching nodes. This high-frequency ringing occurs at LX’s rising and falling transitions and can
interfere with circuit performance and generate EMI. To
dampen this ringing, a series RC snubber circuit is
added across each switch. Below is the procedure for
selecting the value of the series RC circuit:
1) Connect a scope probe to measure V LX to GND,
and observe the ringing frequency, fR.
2) Find the capacitor value (connected from LX to
GND) that reduces the ringing frequency by half.
The circuit parasitic (CPAR) at LX is then equal to 1/3
the value of the added capacitance above. The circuit
parasitic inductance (LPAR) is calculated by:
LPAR =
16
1
(2πfR )2 × CPAR
The resistor for critical dampening (RSNUB) is equal to
2π x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak voltage excursion.
The capacitor (CSNUB) should be at least 2 to 4 times
the value of CPAR to be effective. The power loss of the
snubber circuit is dissipated in the resistor (PRSNUB)
and can be calculated as:
PRSNUB = CSNUB × (VIN )2 × fSW
where VIN is the input voltage and fSW is the switching
frequency. Choose an RSNUB power rating that meets
the specific application’s derating rule for the power
dissipation calculated.
Feed-Forward Capacitor
The feed-forward capacitor, C8 (Figure 2, MAX8576/
MAX8577 with aluminum electrolytic output capacitor),
or C19 (Figure 3, MAX8578/MAX8579 with ceramic output capacitor), dominantly affects the switching frequency. Choose a ceramic X7R capacitor with a value
given by:
C8 =
⎛ 1
⎛ V
⎞
1
V ⎞
×⎜
− 120ns × IN ⎟ × 49.5 × ⎜1− OUT ⎟
RFB ⎝ FS
VOUT ⎠
VIN ⎠
⎝
C19 =
⎛ V
⎞
⎛ 1
1
V ⎞
×⎜
− 120ns × IN ⎟ × 39.5 × ⎜1− OUT ⎟
RFB ⎝ FS
VOUT ⎠
VIN ⎠
⎝
or
where FS is the desired switching frequency, and RFB
is the parallel combination of the two feedback dividerresistors (R1 and R3 of Figure 2, and R9 and R11 of
Figure 3).
Select the closest standard value to C8 and C19 as
possible.
The output inductor and output capacitor also affect the
switching frequency, but to a much lesser extent.
The equations for C8 and C19 above should yield within ±30% of the desired switching frequency for most
applications. The values of C8 and C19 can be
increased to lower the frequency, or decreased to raise
the frequency for better accuracy.
Application Information
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low
switching losses and clean, stable operation. The
switching power stage requires particular attention.
Follow these guidelines for good PC board layout:
______________________________________________________________________________________
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
5) Place the MOSFET as close as possible to the IC to
minimize trace inductance. If parallel MOSFETs are
used, keep the gate connection to both gates
equal.
6) Connect the drain leads of the power MOSFET to a
large copper area to help cool the device. Refer to
the power MOSFET data sheet for the recommended copper area.
2) For output current > 10A, a four-layer PC board is
recommended. Pour a ground plane in the second
layer underneath the IC to minimize noise coupling.
7) Place the feedback components as close to the IC
pins as possible. The feedback divider-resistor from
FB to the output inductor should be connected
directly to the inductor and not sharing with other
connections to this node.
3) Input, output, and VL capacitors are connected to
the power ground plane with the exception of C12
and C22. These capacitors and all other capacitors
are connected to the analog ground plane.
4) Make the connection from the current-limit setting
resistor directly to the high-side MOSFET’s drain to
minimize the effect of PC board trace resistance
and inductance.
8) Refer to the EV kit for further guidelines.
Suggested External Component Manufacturers
MANUFACTURER
COMPONENT
Central Semiconductor
Panasonic
WEBSITE
PHONE
Diodes
www.centralsemi.com
631-435-1110
Inductors
www.panasonic.com
402-564-3131
Sumida
Inductors
www.sumida.com
847-956-0666
International Rectifier
MOSFETs
www.irf.com
800-341-0392
Kemet
Capacitors
www.kemet.com
864-963-6300
Taiyo Yuden
Capacitors
www.t-yuden.com
408-573-4150
TDK
Capacitors
www.component.tdk.com
888-835-6646
Rubycon
Capacitors
www.rubycon.com
408-467-3864
Pin Configurations
TOP VIEW
FB 1
SS
2
VL
3
GND
4
DL
5
10 OCSET
MAX8576
MAX8577
μMAX
FB 1
10 OCSET
9
IN
SS
2
8
DH
VCC
3
7
LX
GND
4
7
LX
6
BST
DL
5
6
BST
MAX8578
MAX8579
9
EN
8
DH
μMAX
Chip Information
TRANSISTOR COUNT: 2087
PROCESSS: BICMOS
______________________________________________________________________________________
17
MAX8576–MAX8579
1) Place IC decoupling capacitors as close to IC pins
as possible. Place the input ceramic decoupling
capacitor directly across and as close as possible to
the high-side MOSFET’s drain and the low-side
MOSFET’s source. This is to help contain the high
switching current within this small loop.
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
e
10LUMAX.EPS
MAX8576–MAX8579
3V to 28V Input, Low-Cost, Hysteretic
Synchronous Step-Down Controllers
4X S
10
10
H
Ø0.50±0.1
0.6±0.1
1
1
0.6±0.1
BOTTOM VIEW
TOP VIEW
D2
MILLIMETERS
INCHES
MAX
DIM MIN
0.043
A
0.006
0.002
A1
A2
0.030
0.037
0.120
0.116
D1
0.118
D2
0.114
0.120
E1
0.116
0.118
E2
0.114
0.199
H
0.187
L
0.0157 0.0275
L1
0.037 REF
0.0106
b
0.007
e
0.0197 BSC
c
0.0035 0.0078
0.0196 REF
S
α
0°
6°
MAX
MIN
1.10
0.05
0.15
0.75
0.95
2.95
3.05
3.00
2.89
2.95
3.05
2.89
3.00
4.75
5.05
0.40
0.70
0.940 REF
0.270
0.177
0.500 BSC
0.200
0.090
0.498 REF
0°
6°
E2
GAGE PLANE
A2
c
A
b
A1
α
E1
D1
L
L1
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL
DOCUMENT CONTROL NO.
21-0061
REV.
1
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2005 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products, Inc.
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