OP471GS

a High Speed, Low Noise Quad Operational Amplifier OP471 FEATURES Excellent Speed: 8 V/
s Typ Low Noise: 11 nV/÷Hz @ 1 kHz Max Unity-Gain Stable High Gain Bandwidth: 6.5 MHz Typ Low Input Offset Voltage: 0.8 mV Max Low Offset Voltage Drift: 4
V/C Max High Gain: 500 V/mV Min Outstanding CMR: 105 dB Min Industry Standard Quad Pinouts PIN CONFIGURATIONS 14-Lead Hermetic Dip (Y-Suffix) 14-Lead Plastic Dip (P-Suffix) OUT A 1 14 OUT D OUT A 1 14 OUT D –IN A 2 13 –IN D –IN A 2 13 –IN D +IN A 3 12 +IN D +IN A 3 12 +IN D V+ 4 11 V– V+ 4 11 V– GENERAL DESCRIPTION +IN B 5 10 +IN C +IN B 5 10 +IN C The OP471 is a monolithic quad op amp featuring low noise, 11 nV/÷Hz Max @ 1 kHz, excellent speed, 8 V/ms typical, a gain bandwidth of 6.5 MHz, and unity-gain stability. –IN B 6 9 –IN C –IN B 6 9 –IN C OUT B 7 8 OUT C OUT B 7 8 OUT C OP471 The OP471 has an input offset voltage under 0.8 mV and an input offset voltage drift below 4 mV/∞C, guaranteed over the full military temperature range. Open-loop gain of the OP471 is over 500,000 into a 10 kW load ensuring outstanding gain accuracy and linearity. The input bias current is under 25 nA limiting errors due to signal source resistance. The OP471’s CMR of over 105 dB and PSRR of under 5.6 mV/V significantly reduce errors caused by ground noise and power supply fluctuations. OP471 16-Lead SOIC (S-Suffix) OUT A 1 16 OUT D 2 15 –IN D +IN A 3 14 +IN D 13 V– –IN A V+ 4 The OP471 offers excellent amplifier matching which is important for applications such as multiple gain blocks, low-noise instrumentation amplifiers, quad buffers and low-noise active filters. The OP471 conforms to the industry standard 14-lead DIP pinout. It is pin-compatible with the LM148/LM149, HA4741, RM4156, MC33074, TL084 and TL074 quad op amps and can be used to upgrade systems using these devices. OP471 +IN B 5 12 +IN C –IN B 6 11 –IN C OUT B 7 10 NC 8 9 OUT C NC NC = NO CONNECT For applications requiring even lower voltage noise the OP470 with a voltage density of 5 nV/÷Hz Max @ 1 kHz is recommended. V+ BIAS OUT –IN +IN V– Figure 1. Simplified Schematic REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 2002 OP471–SPECIFICATIONS ELECTRICAL CHARACTERISTICS (@ V = 15 V, T = 25C, unless otherwise noted.) S Parameter Symbol Input Offset Voltage VOS Input Offset Current IOS Input Bias Current Conditions A Min OP471E Typ Max OP471F Min Typ Max OP471G Min Typ Max Unit 0.25 0.8 0.5 1.5 1.0 1.8 mV VCM = 0 V 4 10 7 20 12 30 nA IB VCM = 0 V 7 25 15 50 25 60 nA Input Noise Voltage en p-p 0.1 Hz to 10 Hz 250 500 250 500 250 500 nV p–p Input Noise Voltage Density2 en fO = 10 Hz fO = 100 Hz fO = 1 kHz 9 7 6.5 9 7 6.5 9 7 6.5 nV/÷Hz nV/÷Hz nV/÷Hz Input Noise Current Density in fO = 10 Hz fO = 100 Hz fO = 1 kHz 1.7 0.7 0.4 1.7 0.7 0.4 1.7 07 0.4 Large-Signal Voltage Gain AVO V = ± 10 V RL = 10 kW RL = 2 kW 500 350 700 550 300 500 175 275 300 500 175 275 V/mV V/mV Input Voltage Range3 IVR ± 11 ± 12 ± 11 ± 12 ± 11 ± 12 V Output Voltage Swing VO RL ≥ 2 kW ± 12 ± 13 ± 12 ± 13 ± 12 ± 13 V Common-Mode Rejection CMR VCM = ± 11 V 105 120 95 95 dB Power Supply Rejection Ratio PSRR VS = 4.5 V to 18 V Slew Rate SR Supply Current (All Amplifiers) ISY No Load 9.2 Gain Bandwidth Product GBW Av = 10 6.5 6.5 6.5 Channel Separation1 CS VO = 20 V p-p fO = 10 Hz 150 125 150 125 150 dB Input Capacitance CIN 2.6 2.6 2.6 pF Input Resistance Differential-Mode RIN 1.1 1.1 1.1 MW Input Resistance Common-Mode RINCM 11 11 11 GW Settling Time tS 4.5 7.5 4.5 7.5 4.5 7.5 ms ms 1 1 6.5 125 AV = 1 To 0.1% To 0.01 % 16 12 11 5.6 8 11 115 5.6 6.5 16 12 11 17.8 8 9.2 11 pA÷Hz pA÷Hz pA÷Hz 115 5.6 6.5 16 12 11 17.8 8 9.2 mV/V V/ms 11 mA MHz NOTES 1 Guaranteed but not 100% tested. 2 Sample tested. 3 Guaranteed by CMR test. –2– REV. A OP471 (Vs = ±15 V, –25 C £ TA £ 85C for OP471E/F, –40C £ TA £ 85 for OP471G,  ELECTRICAL CHARACTERISTICS unless otherwise noted.) OP471G Min Typ Max Input Offset Voltage VOS 0.3 1.1 0.6 2.0 1.2 Average Input Offset Voltage Drift TCVOS 1 4 2 7 4 Input Offset Current los VCM = 0 V 5 20 8 40 20 50 nA IB VCM = 0 V 13 50 25 70 40 75 nA Large-Signal Voltage Gain Avo VO = ± 10 V RL = 10 kW RL = 2 kW Input Voltage Range* IVR Output Voltage Swing VO Common-Mode Rejection Power Supply Rejection Ratio Supply Current (All Amplifiers) Min OP471F Min Typ Max Symbol Input Bias Current Conditions OP471E Typ Max Parameter 2.5 Unit mV mV/∞C 375 250 600 400 200 400 125 200 200 400 125 200 V/mV ± 11 ± 12 ± 11 ± 12 ± 11 ± 12 V RL ≥ 2 kW ± 12 ± 13 ± 12 ± 13 ± 12 ± 13 V CMR VCM = ± 11 V 100 115 90 90 dB PSRR VS = ± 4.5 V to ± 18 V 3.2 10 18 31.6 18 31.6 mV/V ISY No Load 9.3 11 9.3 11 9.3 11 mA 110 110 *Guaranteed by CMR test. ABSOLUTE MAXIMUM RATINGS 1 Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . ± 1.0 V Differential Input Current2 . . . . . . . . . . . . . . . . . . . . ± 25 mW Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . Supply Voltage Output Short-Circuit Duration . . . . . . . . . . . . . . . Continuous Storage Temperature Range P, Y-Package . . . . . . . . . . . . . . . . . . . . . . –65∞C to +150∞C Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300∞C Junction Temperature (Ti) . . . . . . . . . . . . . –65∞C to +150∞C Operating Temperature Range OP471E, OP471F . . . . . . . . . . . . . . . . . . . –25∞C to +85∞C OP471G . . . . . . . . . . . . . . . . . . . . . . . . . . . –40∞C to +85∞C NOTES 1 Absolute Maximum Ratings apply to packaged parts, unless otherwise noted. 2 The OP471’s inputs are protected by back-to-back diodes. Current limiting resistors are not used in order to achieve low noise performance. If differential voltage exceeds ± 1.0 V, the input current should be limited to ± 25 mA. Package Type JA* JC Unit 14-Lead Hermetic DIP(Y) 94 10 ∞C/W 14-Lead Plastic DIP(P) 76 33 ∞C/W 16-Lead SOIC (S) 88 23 ∞C/W *␪JA is specified for worst-case mounting conditions, i.e., ␪JA is specified for device in socket for TO, CERDIP, PDIP packages; ␪JA is specified for device soldered to printed circuit board for SO packages. ORDERING GUIDE TA = 25∞C VOS MAX (mV) 800 1,500 1,800 1,800 Package Options Operating Temperature Range 14-Lead CERDIP Plastic OP471EY OP471FY* OP471GP OP471GS IND IND XIND XIND *Not for new design. Obsolete April 2002. For military processed devices, please refer to the standard microcircuit drawing (SMD) available at www.dscc.dla.mil/programs/milspec/default.asp 5962-88565022A - OP471ARCMDA 5962-88565023A - OP471ATCMDA 5962-8856502CA - OP471AYMDA CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the OP471 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. A –3– WARNING! ESD SENSITIVE DEVICE OP471–Typical Performance Characteristics 100 10 TA = 25C 20 10 5 4 3 I/F CORNER = 5Hz AT 10Hz 1s 5mV NOISE VOLTAGE – 100nV/DIV 40 30 VOLTAGE NOISE – nV/ Hz VOLTAGE NOISE – nV/ Hz TA = 25C VS = 15V 8 AT 1kHz 6 4 100 90 10 TA = 25C VS = 15V 0% 2 0 1 1 10 100 FREQUENCY – Hz 2 1k 15 8 10 20 INPUT OFFSET VOLTAGE –
V 20 10 I/F CORNER = 5Hz 2 300 200 100 0 –75 –50 1 10 100 FREQUENCY – Hz CHANGE IN OFFSET VOLTAGE –
V VS = 15V 40 30 1 4 6 TIME – Seconds TPC 3. 0.1 Hz to 10 Hz Noise 400 TA = 25C VS = 15V 1k –25 0 25 50 75 TEMPERATURE – C 100 INPUT OFFSET CURRENT – nA 9 10 5 14 12 10 8 6 4 2 0 1 2 3 TIME – Minutes 4 5 TPC 6. Warm-Up Offset Voltage Drift 10 20 15 16 TPC 5. Input Offset Voltage vs. Temperature TPC 4. Current Noise Density vs. Frequency VS = 15V VCM = 0V TA = 25C VS = 15V 18 0 125 10 VS = 15V VCM = 0V TA = 25C VS = 15V 8 INPUT BIAS CURRENT – nA 5 4 3 2 20 TPC 2. Voltage Noise Density vs. Supply Voltage 100 VOLTAGE NOISE – nV/ Hz 10 SUPPLY VOLTAGE – V TPC 1. Voltage Noise Density vs. Frequency INPUT BIAS CURRENT – nA 5 0 7 6 5 4 3 2 9 8 7 6 1 0 –75 –50 –25 0 25 50 75 TEMPERATURE – C 100 125 TPC 7. Input Bias Current vs. Temperature 0 –75 –50 –25 0 25 50 75 TEMPERATURE – C 100 125 TPC 8. Input Offset Current vs. Temperature –4– 5 –12.5 –7.5 –2.5 2.5 7.5 12.5 COMMON-MODE VOLTAGE – V TPC 9. Input Bias Current vs. Common-Mode Voltage REV. A OP471 TA = 25C VS = 15V 100 90 80 70 60 50 40 30 TA = +125C 8 TA = –55C 6 4 VS = 15V 9 TOTAL SUPPLY CURRENT – mA 110 TA = +25C TOTAL SUPPLY CURRENT – mA 120 CMR – dB 10 10 130 8 7 6 5 4 3 20 10 1 100 1k 10k FREQUENCY – Hz 100k 2 1M TPC 10. CMR vs. Frequency –PSR 70 60 50 +PSR 40 30 20 10 0 10 1 100 25 15 140 GAIN 5 PHASE MARGIN = 57 160 0 180 –5 200 220 2 3 4 5 6 7 8 9 10 FREQUENCY – MHz –10 1 TPC 16. Open-Loop Gain, Phase Shift vs. Frequency REV. A 70 60 50 40 30 10 100 1k 75 100 125 60 40 20 0 –20 1k 10k 100k 1M 10M 100M 10k 100k 1M FREQUENCY – Hz 8 VS = 15V GBW 1000 500 0 10M TPC 15. Closed-Loop Gain vs. Frequency 80 1500 0 50 TA = 25C VS = 15V TA = 25C RL = 10k 100 120 10 90 80 2000 OPEN-LOOP GAIN – V/mV PHASE 110 100 TPC 14. Open-Loop Gain vs. Frequency PHASE SHIFT – Degrees OPEN-LOOP GAIN – dB 20 80 TA = 25C VS = 15V FREQUENCY – Hz 80 TA = 25C VS = 15V 25 TPC 12. Total Supply Current vs. Temperature 1 TPC 13. PSR vs. Frequency 0 TPC 11. Total Supply Current vs. Supply Voltage 20 10 0 1k 10k 100k 1M 10M 100M FREQUENCY – Hz –25 TEMPERATURE – C CLOSED-LOOP GAIN – dB PSR – dB 90 80 OPEN-LOOP GAIN – dB 110 100 2 –75 –50 20 SUPPLY VOLTAGE – V 140 130 120 TA = 25C VS = 15V 15 PHASE MARGIN – Degrees 140 130 120 10 5 0 5 10 15 SUPPLY VOLTAGE – V 20 TPC 17. Open-Loop Gain vs. Supply Voltage –5– 70 6 60 4  50 40 –75 –50 –25 2 GAIN-BANDWIDTH PRODUCT – MHz 10 0 0 25 50 75 100 125 150 TEMPERATURE – C TPC 18. Gain-Bandwidth Product, Phase Margin vs. Temperature OP471 TA = 25C VS = 15V THD = 1% 20 16 12 8 TA = 25C VS = 15V 16 POSITIVE SWING 14 12 NEGATIVE SWING 10 8 6 4 4 240 180 120 AV = 100 60 2 10k 100k 1M FREQUENCY – Hz AV = 1 0 100 10M 1k LOAD RESISTANCE –  170 9.0 TA = 25C VS = 15V VO = 20V p-p TO 100kHz 160 CHANNEL SEPARATION – dB –SR 8.0 +SR 7.5 7.0 6.5 10k TPC 20. Maximum Output Voltage vs. Load Resistance TPC 19. Maximum Output Swing vs. Frequency 8.5 150 140 130 120 110 100 90 80 70 60 6.0 0 25 50 75 –75 –50 –25 TEMPERATURE – C 50 10 100 125 TA = 25C VS = 15V AV = 1 90 10 0% TPC 25. Large-Signal Transient Response 1M 10M 1k 10k 100k 1M FREQUENCY – Hz 10M 100M TPC 21. Closed-Loop Output Impedance vs. Frequency 1 TA = 25C VS = 15V VO = 10V p-p RL = 2k 0.1 0.01 AV = 10 AV = 1 0.001 10 100 1k FREQUENCY – Hz 10k TPC 24. Total Harmonic Distortion vs. Frequency TA = 25C VS = 15V AV = 1 90 10 5µs 1k 10k 100k FREQUENCY – Hz 100 0% 5V 100 TPC 23. Channel Separation vs. Frequency TPC 22. Slew Rate vs. Temperature 100 0 100 TOTAL HARMONIC DISTORTION – % 0 1k SLEW RATE – V/
s TA = 25C VS = 15V 300 OUTPUT IMPEDANCE –  24 18 MAXIMUM OUTPUT – V PEAK-TO-PEAK AMPLITUDE – V 360 20 28 0.2µs 50mV TPC 26. Small-Signal Transient Response –6– REV. A OP471 100 5k 500 1/4 OP471 TOTAL NOISE – nV/ Hz V1 20V p-p 50k 50 1/4 OP471 OP11 10 OP400 OP471 V2 OP470 RESISTOR NOISE ONLY CHANNEL SEPARATION = 20 LOG V1 V2 / 1000 1 100 Figure 2. Channel Separation Test Circuit 1k 10k RS – SOURCE RESISTANCE –  100k Figure 4. Total Noise vs. Source Resistance (Including Resistor Noise) at 1 kHz +18V 2 +1V 3 100 6 4 1 A 5 +1V 11 B 7 9 –1V 10 TOTAL NOISE – nV/ Hz –18V 13 C 8 D 12 –1V 14 OP11 OP400 10 OP471 OP470 RESISTOR NOISE ONLY Figure 3. Burn-In Circuit 1 100 APPLICATIONS INFORMATION Voltage and Current Noise The OP471 is a very low-noise quad op amp, exhibiting a typical voltage noise of only 6.5 Hz @ 1 kHz. The low noise characteristic of the OP471 is, in part, achieved by operating the input transistors at high collector currents since the voltage noise is inversely proportional to the square root of the collector current. Current noise, however, is directly proportional to the square root of the collector current. As a result, the outstanding voltage noise performance of the OP471 is gained at the expense of current noise performance which is typical for low noise amplifiers. To obtain the best noise performance in a circuit, it is vital to understand the relationship between voltage noise (en), current noise (in), and resistor noise (et). Total Noise and Source Resistance The total noise of an op amp can be calculated by: En = (e n ) + (inR S ) + (et ) 2 2 2 where: 100k Figure 5. Total Noise vs. Source Resistance (Including Resistor Noise) at 10 Hz Figure 4 shows the relationship between total noise at 1 kHz and source resistance. For RS < 1 kW the total noise is dominated by the voltage noise of the OP471. As RS rises above 1 kW, total noise increases and is dominated by resistor noise rather than by voltage or current noise of the OP471. When RS exceeds 20 kW, current noise of the OP471 becomes the major contributor to total noise. Figure 5 also shows the relationship between total noise and source resistance, but at 10 Hz. Total noise increases more quickly than shown in Figure 4 because current noise is inversely proportional to the square root of frequency. In Figure 5, current noise of the OP471 dominates the total noise when RS > 5 kW. From Figures 4 and 5, it can be seen that to reduce total noise, source resistance must be kept to a minimum. In applications with a high source resistance, the OP400, with lower current noise than the OP471, will provide lower total noise. En = total input referred noise en = op amp voltage noise in = op amp current noise et = source resistance thermal noise RS = source resistance The total noise is referred to the input and at the output would be amplified by the circuit gain. REV. A 1k 10k RS – SOURCE RESISTANCE –  –7– OP471 1000 For reference, typical source resistances of some signal sources are listed in Table I. OP11 PEAK-TO-PEAK NOISE – nV OP400 TABLE I. OP471 100 Device Source Impedance Strain gauge < 500 W Typically used in low-frequency applications. Magnetic tapehead < 1,500 W Low IB very important to reduce self-magnetization problems when direct coupling is used. OP471 IB can be neglected. Magnetic phonograph cartridges < 1,500 W Similar need for low IB in direct coupled applications. OP471 will not introduce any self -magnetization problem. OP470 RESISTOR NOISE ONLY 10 100 100k 1k 10k RS – SOURCE RESISTANCE –  Figure 6. Peak-to-Peak Noise (0.1 Hz to 10 Hz) vs. Source Resistance (Includes Resistor Noise) Figure 6 shows peak-to-peak noise versus source resistance over the 0.1 Hz to 10 Hz range. Once again, at low values of RS, the voltage noise of the OP471 is the major contributor to peak-to-peak noise. Current noise becomes the major contributor as RS increases. The crossover point between the OP471 and the OP400 for peak-to-peak noise is at RS = 17 W. Linear variable < 1,500 W differential transformer Comments Used in rugged servo-feedback applications. Bandwidth of interest is 400 Hz to 5 kHz. *For further information regarding noise calculations, see “Minimization of Noise in Op Amp Applications,” Application Note AN-15. The OP470 is a lower noise version of the OP471, with a typical noise voltage density of 3.2 nV/÷Hz @ 1 kHz. The OP470 offers lower offset voltage and higher gain than the OP471, but is a slower speed device, with a slew rate of 2 V/ms compared to a slew rate of 8 V/ms for the OP471. R3 1.24k R1 5 R2 5 C1 2
F OP471 DUT OP27E R5 909 R4 200 C4 0.22
F R6 600k D1 1N4148 D2 OP15E 1N4148 R9 306k R8 10k R10 65.4k R11 65.4k C3 0.22
F R14 4.99k OP15E R13 5.9k C2 0.032
F eOUT C5 1
F R12 10k GAIN = 50,000 VS = 15V Figure 7. Peak-to-Peak Voltage Noise Test Circuit (0.1 Hz to 10 Hz) –8– REV. A OP471 100 Noise Measurements - Peak-to-Peak Voltage Noise The circuit of Figure 7 is a test setup for measuring peak-to-peak voltage noise. To measure the 500 nV peak-to-peak noise specification of the OP471 in the 0.1 Hz to 10 Hz range, the following precautions must be observed: 80 GAIN – dB 1. The device must be warmed up for at least five minutes. As shown in the warm-up drift curve, the offset voltage typically changes 13 mV due to increasing chip temperature after power-up. In the 10-second measurement interval, these temperature-induced effects can exceed tens-of-nanovolts. 60 40 20 2. For similar reasons, the device must be well-shielded from air currents. Shielding also minimizes thermocouple effects. 0 0.01 3. Sudden motion in the vicinity of the device can also “feedthrough” to increase the observed noise. 4. The test time to measure 0.1 Hz to 10 Hz noise should not exceed 10 seconds. As shown in the noise-tester frequency-response curve of Figure 8, the 0.1 Hz corner is defined by only one pole. The test time of 10 seconds acts as an additional pole to eliminate noise contribution from the frequency band below 0.1 Hz. 0.1 1 FREQUENCY – Hz 10 Figure 8. 0.1 Hz to 10 Hz Peak-to-Peak Voltage Noise Test Circuit Frequency Response Noise Measurement - Noise Voltage Density The circuit of Figure 9 shows a quick and reliable method of measuring the noise voltage density of quad op amps. Each individual amplifier is series connected and is in unity-gain, save the final amplifier which is in a noninverting gain of 101. Since the ac noise voltages of each amplifier are uncorrelated, they add in rms fashion to yield: 5. A noise voltage density test is recommended when measuring noise on a large number of units. A 10 Hz noise voltage density measurement will correlate well with a 0.1 Hz to 10 Hz peak-to-peak noise reading, since both results are determined by the white noise and the location of the 1/f corner frequency. e OUT = 101 Ê e nA 2 + e nB 2 + e nC 2 + e nD 2 ˆ Ë ¯ 6. Power should be supplied to the test circuit by well bypassed, low noise supplies, e.g, batteries. These will minimize output noise introduced through the amplifier supply pins. The OP471 is a monolithic device with four identical amplifiers. The noise voltage density of each individual amplifier will match, giving: e OUT = 101 Ê 4e n 2 ˆ = 101 (2e n ) Ë ¯ R1 100 1/4 OP471 1/4 OP471 1/4 OP471 R2 10k 1/4 OP471 eOUT TO SPECTRUM ANALYZER eOUT (nV Hz) = 101(2en) VS = 15V Figure 9. Noise Voltage Density Test Circuit REV. A 100 –9– OP471 Noise Measurement - Current Noise Density The test circuit shown in Figure 10 can be used to measure current noise density. The formula relating the voltage output to current noise density is: ( 2 Ê e nOUT ˆ Á ˜ - 40nV / Hz Ë G ¯ in = ) 2 adds phase shift in the feedback network and reduces stability. A simple circuit to eliminate this effect is shown in Figure 11. The added components, C1 and R3, decouple the amplifier from the load capacitance and provide additional stability. The values of C1 and R3 shown in Figure 11 are for load capacitances of up to 1,000 pF when used with the OP471. In applications where the OP471’s inverting or noninverting inputs are driven by a low source impedance (under 100 W) or connected to ground, if V+ is applied before V–, or when V– is disconnected, excessive parasitic currents will flow. RS where: Most applications use dual tracking supplies and with the device supply pins properly bypassed, power-up will not present a problem. A source resistance of at least 100 W in series with all inputs (Figure 11) will limit the parasitic currents to a safe level if V– is disconnected. It should be noted that any source resistance, even 100 W, adds noise to the circuit. Where noise is required to be kept at a minimum, a germanium or Schottky diode can be used to clamp the V– pin and eliminate the parasitic current flow instead of using series limiting resistors. For most applications, only one diode clamp is required per board or system. G = gain of 10,000 RS = 100 kW source resistance Capacative Load Driving and Power Supply Considerations The OP471 is unity-gain stable and is capable of driving large capacitive loads without oscillating. Nonetheless, good supply bypassing is highly recommended. Proper supply bypassing reduces problems caused by supply line noise and improves the capacitive load driving capability of the OP471. R3 1.24k R1 5 Rf R2 100k OP471 DUT en OUT TO SPECTRUM ANALYZER OP27E OP471 R5 8.06k R4 200 8V/
s GAIN = 10,000 VS = 15V Figure 10. Current Noise Density Test Circuit V+ C2 10
F + R2 C1 200pF R1 OP471 100* C4 10
F + VOUT CL 1000pF C5 0.1
F * *SEE TEXT R3 50 Unity-Gain Buffer Applications When Rf £ 100 W and the input is driven with a fast, large signal pulse (>1 V), the output waveform will look as shown in Figure 12. During the fast feedthrough-like portion of the output, the input protection diodes effectively short the output to the input, and a current, limited only by the output short-circuit protection, will be drawn by the signal generator. With Rf ≥ 500 W, the output is capable of handling the current requirements (IL £ 20 mA at 10 V); the amplifier will stay in its active mode and a smooth transition will occur. C3 0.1
F VIN Figure 12. Pulsed Operation When Rf > 3 kW, a pole created by Rf and the amplifier’s input capacitance (2.6 pF) creates additional phase shift and reduces phase margin. A small capacitor (20 pF to 50 pF) in parallel with Rf helps eliminate this problem. APPLICATIONS Low Noise Amplifier V– PLACE SUPPLY DECOUPLING CAPACITORS AT OP471 Figure 11. Driving Large Capacitive Loads In the standard feedback amplifier, the op amp’s output resistance combines with the load capacitance to form a lowpass filter that A simple method of reducing amplifier noise by paralleling amplifiers is shown in Figure 13. Amplifier noise, depicted in Figure 14, is around 5 nV/÷Hz @ 1 kHz (R.T.I.). Gain for each paralleled amplifier and the entire circuit is 100. The 200 W resistors limit circulating currents and provide an effective output resistance of 50 W. The amplifier is stable with a 10 nF capacitive load and can supply up to 30 mA of output drive. –10– REV. A OP471 High-Speed Differential Line Driver NOISE DENSITY – 0.58nV/ Hz/DIV REFERRED TO INPUT The circuit of Figure 15 is a unique line driver widely used in professional audio applications. With ± 18 V supplies, the line driver can deliver a differential signal of 30 V p-p into a 1.5 kW load. The output of the differential line driver looks exactly like a transformer. Either output can be shorted to ground without changing the circuit gain of 5, so the amplifier can easily be set for inverting, noninverting, or differential operation. The line driver can drive unbalanced loads, like a true transformer. +15V VIN 90 10 0% R3 200 1/4 OP471E R1 50 100 Figure 14. Noise Density of Low-Noise Amplifier, G = 100 R2 5k –15V R6 200 1/4 OP471E R4 50 R4 10k R5 5k R7 50 VOUT = 100VIN R8 10k R2 2k R9 200 1/4 OP471E 1/4 OP471 R6 2k R8 5k R12 200 R10 50 R3 2k R13 10k R9 10k R12 1k R10 50 1/4 OP471 R5 10k R11 5k Figure 15. High-Speed Differential Line Driver Figure 13. Low-Noise Amplifier High-Output Amplifier The amplifier shown in Figure 16 is capable of driving 20 V p-p into a floating 400 W load. Design of the amplifier is based on a bridge configuration. A1 amplifies the input signal and drives the load with the help of A2. Amplifier A3 is a unity-gain inverter which drives the load with help from A4. Gain of the high output amplifier with the component values shown is 10, but can easily be changed by varying R1 or R2. +15V C1 10
F + R5 5k C2 0.1
F R2 9k R6 5k R1 1k 1/4 OP471E A1 R3 50 R7 50 VIN C3 0.1
F 1/4 OP471E A2 C4 10
F + R4 50 RL R8 50 –15V Figure 16. High-Output Amplifier REV. A –OUT R14 1k R7 2k IN R1 10k 1/4 OP471E R11 50 1/4 OP471 –11– 1/4 OP471E A4 1/4 OP471E A3 +OUT OP471 Quad Programmable Gain Amplifier The combination of the quad OP471 and the DAC8408, a quad 8-bit CMOS DAC, creates a space-saving quad programmable gain amplifier. The digital code present at the DAC, which is easily set by a microprocessor, determines the ratio between the fixed DAC feedback resistor and the impedance the DAC ladder presents to the op amp feedback loop. Gain of each amplifier is: where n equals the decimal equivalent of the 8-bit digital code present at the DAC. If the digital code present at the DAC consists of all zeros, the feedback loop will be open causing the op amp output to saturate. The 20 MW resistors placed in parallel with the DAC feedback loop eliminates this problem with a very small reduction in gain accuracy. VOUT 256 = – VIN n VDD DAC-8408ET RFBA VINA VREF A IOUT1A R1 20M DAC A +15V 1/4 OP470E VOUTA IOUT2A/2B –15V RFBB VINB VREFB IOUT1B DAC B R2 20M 1/4 OP470E VOUTB 1/4 OP470E VOUTC 1/4 OP470E VOUTD RFBC VINC VREF C IOUT1C R3 20M DAC C IOUT2C/2D RFBD VIND VREF D DAC D DAC DATA BUS PINS 9 (LSB) – 16 (MSB) IOUT1D R4 20M DGND Figure 17. Quad Programmable Gain Amplifier –12– REV. A OP471 Low Phase Error Amplifier R2 = R1 R2 The simple amplifier depicted in Figure 18 utilizes monolithic matched operational amplifiers and a few resistors to substantially reduce phase error compared to conventional amplifier designs. At a given gain, the frequency range for a specified phase accuracy is over a decade greater than for a standard single op amp amplifier. R2 K1 1/4 OP471E A2 The low phase error amplifier performs second-order frequency compensation through the response of op amp A2 in the feedback loop of A1. Both op amps must be extremely well matched in frequency response. At low frequencies, the A1 feedback loop forces V2/(K1 + 1) = VIN. The A2 feedback loop forces Vo/(K1 +1) = V2/(K1 + 1) yielding an overall transfer function of VO/VIN = K1 + 1. The dc gain is determined by the resistor divider at the output, VO, and is not directly affected by the resistor divider around A2. Note that similar to a conventional single op amp amplifier, the dc gain is set by resistor ratios only. Minimum gain for the low phase error amplifier is 10. 1/4 OP471E A1 VIN V2 R1 R1 K1 VO VO = (K1 + 1) V IN ASSUME: A1 AND A2 ARE MATCHED.  AO (s) = sT Figure 18. Low Phase Error Amplifier Figure 19 compares the phase error performance of the low phase error amplifier with a conventional single op amp amplifier and a cascaded two-stage amplifier. The low phase error amplifier shows a much lower phase error, particularly for frequencies where ␻/␤␻T < 0.1. For example, phase error of –0.1∞ occurs at 0.002 ␻/␤␻T for the single op amp amplifier, but at 0.11 ␻/␤␻T for the low phase error amplifier. 0 PHASE SHIFT – Degrees –1 For more detailed information on the low phase error amplifier, see Application Note AN-107. –2 SINGLE OP AMP (CONVENTIONAL DESIGN) –3 CASCADED (TWO STAGES) –4 –5 LOW-PHASE ERROR AMPLIFIER –6 –7 0.001 0.005 0.01 0.05 0.1 FREQUENCY RATIO – 1/, /T 0.5 Figure 19. Phase Error Comparison REV. A –13– 1 OP471 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 14-Lead PDIP Package (N-14) 0.795 (20.19) 0.725 (18.42) 14 8 0.280 (7.11) 0.240 (6.10) 7 1 PIN 1 0.100 (2.54) BSC 0.060 (1.52) 0.015 (0.38) 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 0.210 (5.33) MAX 0.130 (3.30) 0.160 (4.06) MIN 0.115 (2.93) 0.022 (0.558) 0.070 (1.77) SEATING PLANE 0.014 (0.356) 0.045 (1.15) 0.015 (0.381) 0.008 (0.204) 14-Lead CERDIP Package (Q-14) 0.005 (0.13) MIN 0.098 (2.49) MAX 14 8 PIN 1 1 7 0.100 (2.54) BSC 0.785 (19.94) MAX 0.200 (5.08) MAX 0.310 (7.87) 0.220 (5.59) 0.320 (8.13) 0.290 (7.37) 0.060 (1.52) 0.015 (0.38) 0.150 (3.81) MIN 0.070 (1.78) SEATING PLANE 0.030 (0.76) 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.014 (0.36) 15 0 0.015 (0.38) 0.008 (0.20) 16-Lead SOIC Package (R-16) 0.4133 (10.50) 0.3977 (10.00) 9 16 0.2992 (7.60) 0.2914 (7.40) PIN 1 0.4193 (10.65) 0.3937 (10.00) 8 1 0.050 (1.27) BSC 0.0118 (0.30) 0.0040 (0.10) 0.1043 (2.65) 0.0926 (2.35) 8 0.0192 (0.49) SEATING 0 0.0125 (0.32) 0.0138 (0.35) PLANE 0.0091 (0.23) –14– 0.0291 (0.74) 45 0.0098 (0.25) 0.0500 (1.27) 0.0157 (0.40) REV. A OP471 Revision History Location Page Data Sheet changed from REV. 0 to REV. A. Edits to FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Edits to ELECTRICAL CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Deleted DICE CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Deleted WAFER TEST CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 REV. A –15– –16– PRINTED IN U.S.A. C00307–0–4/02(A)