A6271-1
Automotive, High-Current LED Controller
FEATURES AND BENEFITS
DESCRIPTION
The A6271-1 is a DC-DC converter controller, providing a
programmable constant-current output for driving high-power
LEDs in series. The controller is based on a programmable
fixed-frequency, peak current-mode control architecture. The
DC-DC converter can be configured in a myriad of different
switching configurations including boost, buck-boost, buck
(ground-referenced switch), and SEPIC.
•
•
•
•
•
•
•
Automotive AEC-Q100 qualified
Constant-current LED drive
4.2 to 50 V supply
53.3 V maximum LED string voltage
Boost, buck-boost, buck, and SEPIC switching converters
Programmable switching frequency 70 to 700 kHz
PWM-controlled PMOS driver allows accurate LED
current control at low duty cycles
• Dimming via external PWM, internal PWM and/or
analog dimming
• Frequency dither scheme for effective spread spectrum to
reduce EMI
• Comprehensive fault protection and fault flag
PACKAGE:
The A6271-1 provides a cost-effective solution using an
external logic-level MOSFET and minimum additional external
components. The maximum LED current is set with a single
external sense resistor and can be accurately modulated using
a current reference input (analog control). External PWM
dimming is possible via the PWMIN input, which also provides
a shutdown mode. As an alternative, an internal PWM dimming
circuit can be used by programming the PWMIN and DR pins.
Either PWM scheme controls the PWMOUT output which
drives an external p-channel MOSFET connected in series
with the LED string. This MOSFET is also used to isolate the
load during certain fault conditions, including output shorts
to ground.
16-Pin eTSSOP (LP)
with Exposed Thermal Pad
Continued on next page...
APPLICATIONS
• Automotive high-power LED lighting systems
• Fog lights, reversing lights, daytime running lights,
position lights, headlights
Not to scale
L1
56 µH
D2
VBAT
C1
4.7 µF
C2
4.7 µF
R1
2.7 kΩ
C3
1 µF
C7
47 nF
VIN
VREG
LP
LN
R6
1.35 Ω
R5
150 Ω
PWMOUT
M2
GND
FAULTn
DR
PWM Control
A6271-1
OVUV
PWMIN
IREF
C4
22 nF
R7
4.3 kΩ
SG
SP
OSC
R2
73.2 kΩ
DITH
R3
220 kΩ
COMP
R4
39 Ω
C5
680 nF
R8
12 Ω
LED1
C8
4.4 µF
M1
R11
240 kΩ
LED14
R9
1.3 kΩ
C6
22 pF
R10
75 mΩ
GND
Figure 1: Boost Switcher Driving 14 LEDs at 150 mA
A6271-1-DS, Rev. 3
MCO-0000277
July 9, 2018
A6271-1
Automotive, High-Current LED Controller
DESCRIPTION (continued)
The A6271-1 has been carefully designed to minimize
electromagnetic emissions through distributed decoupling and an
externally programmable frequency dither circuit configured for
the EMI specification CISPR 25. It is also possible to program the
fundamental switching frequency below 150 kHz where most EMI
standards begin.
features to protect the IC, the LED driver system, and the LED
string against faults. Fixed-output overvoltage protection ensures
no maximum voltage rating violations, even under a single point
failure of the programmable-output overvoltage protection circuit.
Other protection features include: LED overload (boost), output
undervoltage (buck or buck-boost), input supply (VIN) undervoltage,
5 V regulator (VREG) output undervoltage, high-side supply (PWM
PMOS) undervoltage, and thermal protection.
The A6271-1 has a comprehensive set of integrated protection
SELECTION GUIDE
Part Number
Packing [1]
Package
A6271KLPTR-T-1
4000 pieces per 13-in. reel
16-pin eTSSOP with exposed thermal pad
[1] Contact Allegro™
for additional packing options.
ABSOLUTE MAXIMUM RATINGS [2]
Characteristic
VIN
Symbol
Notes
VIN
Rating
Unit
–0.3 to 55
V
PWMOUT, LP, LN, OVUV
–0.3 to 58
V
OSC, DITH, COMP, FAULTn, SG, SP,
IREF, PWMIN, DR, VREG
–0.3 to 6.5
V
–0.5 to 0.5
V
TJ(max)
150
°C
Tstg
–55 to 150
°C
LP
VLP
Maximum Continuous Junction
Temperature
Storage Temperature Range
[2] With
With respect to LN
respect to GND.
THERMAL CHARACTERISTICS
Characteristic
eTSSOP Package
Symbol
RθJA
RθJC
[3] Additional
Test Conditions [3]
Value
Unit
4-layer PCB based on JEDEC standard JESD51-7
34
°C/W
2-layer PCB with 3.8in2 of copper area each side
43
°C/W
Junction to thermal pad
2
°C/W
thermal information available on the Allegro website.
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
2
A6271-1
Automotive, High-Current LED Controller
Table of Contents
Features and Benefits........................................................ 1
Description......................................................................... 1
Package............................................................................. 1
Applications....................................................................... 1
Selection Guide................................................................. 2
Absolute Maximum Ratings............................................... 2
Thermal Characteristics..................................................... 2
Pinout Diagrams and Terminal List.................................... 4
Functional Block Diagram.................................................. 5
Electrical Characteristics................................................... 6
Functional Description....................................................... 9
Circuit Operation................................................................ 9
Converter...................................................................... 9
External Pulse-Width Modulation Dimming........................ 9
Internal Pulse-Width Modulation Dimming........................ 9
Analog Dimming........................................................... 10
Soft-Start..................................................................... 10
LED Current-Sense Resistor...........................................11
Sleep Mode..................................................................11
5 V Regulator, VREG.....................................................11
Oscillator..................................................................... 12
Frequency Dithering..................................................... 12
Protection.................................................................... 12
Component Selection...................................................... 16
Inductor.......................................................................... 16
Boost Inductor Selection............................................... 16
Switch Current Sense................................................... 16
Slope Compensation.................................................... 17
Control Loop Compensation.......................................... 17
Low-Side Switching MOSFET........................................ 18
Recirculation Diode...................................................... 19
High-Side PWM MOSFET............................................. 19
Output Capacitor.......................................................... 20
Input Capacitor............................................................ 20
Layout............................................................................ 20
Reducing EMI.................................................................. 21
Snubber...................................................................... 21
Input Filter................................................................... 21
Frequency Dithering..................................................... 22
Application Circuits.......................................................... 24
Package Outline Drawing................................................ 36
Appendix A: Special Notes on A6271-1 Operating
in SEPIC Mode...........................................................A-1
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
3
A6271-1
Automotive, High-Current LED Controller
PINOUT DIAGRAM AND TERMINAL LIST TABLE
COMP
1
16 VIN
IREF
2
15 LP
FAULTn
3
14 LN
OSC
4
13 PWMOUT
DITH
5
PAD
12 OVUV
DR
6
11 GND
PWMIN
7
10 SP
VREG
8
9
SG
Package LP, 16-Pin eTSSOP Pinout Diagram
Terminal List Table
Symbol
Number
Function
COMP
1
Compensation pin for output of GM error amplifier.
IREF
2
Analog dimming input. With a capacitor connected to this pin, provides a soft-start period when coming out of
sleep mode.
FAULTn
3
Open drain. Logic low indicates detection of a fault.
Faults include: LED overload (boost), output undervoltage (buck or buck-boost), output overvoltage,
programmable overvoltage, input supply (VIN) undervoltage, 5 V Regulator (VREG) output undervoltage.
OSC
4
Oscillator input for setting switching frequency and for external synchronization.
DITH
5
Dither frequency range set. Connect resistor from this pin to GND. Connect to VREG if not used.
DR
6
A voltage applied to this pin programs the duty cycle of PWM internal mode.
PWMIN
7
Used for either putting the device into sleep mode or analog dimming control. Can also be used for external or
internal PWM control.
VREG
8
5 V regulator output. Connect filter capacitor from VREG to GND.
SG
9
Switch gate drive output.
SP
10
Switch current sense and slope compensation.
GND
11
Ground.
OVUV
12
Programmable-output overvoltage and undervoltage protection input.
PWMOUT
13
PWM gate drive for external p-channel MOSFET (active low).
LN
14
LED current sense -ve.
LP
15
LED current sense +ve.
VIN
16
Main supply.
PAD
–
Exposed pad provides both electrical contact to the ground and good thermal contact to the PCB. This pad
must be soldered to the ground plane preferably by multiple through-hole vias.
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
4
A6271-1
Automotive, High-Current LED Controller
RSL
VBAT
PWMOUT
5V
Linear
Reg
VREG
CA
LP
–
+
–8 V
wrt LP
–
CB
+
–
+
–
+
CD
LN
–
OVUV
VIN
+
AA
Prog
Output OV
Output OV
VREG
FAULTn
Internal
Linear
Reg
Output UV
Fault
Block
+
–+
External
Analog
Dimming
IREF
Analog
DIM/
Soft
Start
–
+
–CE
AB
PWM
Overload
–
CF
+
R
Q
SP
+
AC
RSLOPE
–
PWMIN
SG
S
PWM
External
PWM
Dimming
VIN UVLO
VREG UVLO
Prog Output OV
Output OV
Thermal
–
RSS
CG
+
Osc
DR
GND
Dither
Internal PWM
Generator
A6271-1
COMP
Internal
PWM
Dimming
Enable
OSC
DITH
Internal PWM
Duty Cycle
External Sync
Figure 2: Functional Block Diagram
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
5
A6271-1
Automotive, High-Current LED Controller
ELECTRICAL CHARACTERISTICS: Valid at TJ = –40°C to 150°C, VIN = 5 to 45 V, unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
VIN
VIN undervoltage turn off plus VIN undervoltage
hysteresis cleared; PWMOUT undervoltage turnon cleared
4.2
–
50
V
IINQ
SG Open Circuit
–
3.5
5
mA
IINS
PWMIN = GND > disable time
–
6
20
µA
IREG = 0 to 2 mA, VIN ≥ 5.3 V
4.85
5.04
5.15
V
IREG = 2 mA, VIN = 5 V
4.65
–
–
V
IREG = 2 mA, VIN = 9 to 45 V,
TJ = –40°C to 125°C
4.95
5.05
5.15
V
70
–
–
mA
30
–
ns
SUPPLY and REFERENCE
VIN Functional Operating Range [2]
VIN Quiescent Current
VREG Output Voltage
VREG
VREG Output Voltage [3]
VREG
VREG Current Limit
IREGCL
GATE OUTPUT DRIVE
Turn-On Time
Turn-Off Time
Minimum Off-Time
Pull-Up On-Resistance
Pull-Down On-Resistance
tr
CLOAD = 1 nF, 20% to 80%
–
tf
CLOAD = 1 nF, 80% to 20%
–
30
–
ns
–
135
165
ns
TJ = 25°C, IGHx = –100 mA
–
1.7
–
Ω
TJ = 150°C, IGHx = –100 mA
–
–
3.6
Ω
TJ = 25°C, IGLx = 100 mA
–
0.75
–
Ω
toff(MIN)
RDS(on)UP
RDS(on)DN
TJ = 150°C, IGLx = 100 mA
–
–
2
Ω
–
VREG
V
–
0.1
V
Output High Voltage
VSGH
ISG = –100 µA
VREG –
0.1
Output Low Voltage
VSGL
ISG = 100 µA
–
LOGIC INPUTS AND OUTPUTS
FAULTn Output (Open Drain)
VOL
IOL = 1 mA, fault asserted
–
–
0.4
V
FAULTn Output Leakage Current [1]
IOH
VO = 5.5 V, fault not asserted
–1
–
1
µA
PWMIN Low Voltage
VPWMINL
–
–
0.3
V
PWMIN High Voltage
VPWMINH
2
–
–
V
VIhys
150
180
–
mV
IPWMSLEEP
–
–1.5
–
µA
–
500
–
kHz
Input Hysteresis
PWMIN Sleep Pull-Up Current [1]
OSCILLATOR
ROSC = 51 kΩ
Oscillator Frequency
fOSC
315
350
385
kHz
Oscillator Frequency Range [3]
fOSC
70
–
700
kHz
OSC Input Low Voltage
VOIL
–
–
0.8
V
OSC Input High Voltage
VOIH
2
–
–
V
OSC Watchdog Period
tOSWD
17
–
–
µs
ROSC = 73.4 kΩ
Between successive rising edges
Continued on the next page…
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
6
A6271-1
Automotive, High-Current LED Controller
ELECTRICAL CHARACTERISTICS (continued): Valid at TJ = –40°C to 150°C, VIN = 5 to 45 V, unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
LED CURRENT SENSE
Input Bias Current LN
ILN
VLP = VLN = 12 V
–
5
–
µA
Input Bias Current LP
ILP
VLP = VLN = 12 V
–
200
–
µA
200
204
208
mV
5
–
53.3
V
Differential Sense Voltage
VIDL
Input Common-Mode Range
PWMIN = high, VIDL = VLP – VLN, IREF > 1.2 V
VCMLH
VLP = VLN
tDISAN
PWMIN = low
24.8
29
35
ms
VIREF = 0.5 V
–
102
–
mV
ANALOG DIMMING
Disable Time
Differential Sense Voltage
VIDL
VIREF = 0.25 V
47
51
55
mV
IREF Maximum Voltage
VIREFMAX
Corresponds to sense voltage = 200 mV
–
1
–
V
IREF Minimum Voltage
VIREFMIN
Corresponds to sense voltage = 0 mV
–
0
–
V
PWM DIMMING: INTERNAL AND EXTERNAL
PWMIN to LED Turn-On Time
tDIMON
CL = 2 nF between PWMOUT and LN
–
270
–
ns
PWMIN to LED Turn-Off Time
tDIMOFF
CL = 2 nF between PWMOUT and LN
–
210
–
ns
PWMOUT Low Voltage
VPWMLO
LED on, PWMOUT wrt LP, VIN = 10 V
–9
–
–6.5
V
IPULLUP
PWMIN = low, PWMOUT wrt LP = 0 V
–
–25
–
mA
PWMIN = high, PWMOUT wrt LP = –8 V
–
50
–
mA
24.8
29
35
ms
–
1000
–
Hz
Peak Pull-Up
Current [1]
Peak Pull-Down Current
IPULLDOWN
PWM DIMMING: EXTERNAL
Disable Time
tDISEPWM
PWMIN = low
PWM DIMMING: INTERNAL
Maximum PWM Dimming Frequency
fPWM
Minimum PWM Dimming Frequency
fPWM
PWM Dimming Frequency
fPWM
DPWM90
PWM Duty Cycle
VDR = 3.24 V, fPWM = 200 Hz
DPWM5
VDR = 180 mV, fPWM = 200 Hz
DPWM0
VDR = 0 V, fPWM = 200 Hz
VDRDCMAX
Disable Time
–
200
–
Hz
180
200
220
Hz
TJ = 25°C
87
90
93
%
TJ = 150°C
–
90
–
%
TJ = 25°C
4.5
5
5.5
%
TJ = 150°C
–
5
–
%
TJ = 25°C
–
0.3
–
%
70 kΩ between PWMIN and GND
Minimum voltage on DR for 100% duty cycle
–
3.6
–
V
12.4
14.5
17.5
ms
Coming out of sleep mode
–
–1
–
µA
Voltage at IREF pin for 100% LED current
–
1
–
V
–
100
–
mV
–20
–
–
µA
tDISIPWM
PWMIN = low
ISOURCE
VRAMPUP
SOFT-START
Startup Ramp Up Source Current [1]
Ramp Up Threshold
Ramp Down Threshold
VRAMPDOWN Voltage at IREF pin for 10% LED current
SWITCH CURRENT SENSE AND AMPLIFIER
Input Bias Current [1]
Switch Current Overload Threshold
Voltage Gain
IBIASS
Voltage [3]
VSP = 300 mV, RSLOPE = 1.5 kΩ
VIDS
370
400
440
mV
ACS
–
2.25
–
V/V
Continued on the next page…
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
7
A6271-1
Automotive, High-Current LED Controller
ELECTRICAL CHARACTERISTICS (continued): Valid at TJ = –40°C to 150°C, VIN = 5 to 45 V, unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
–116
–
–93
µA
AVEA
–
62
–
dB
SLOPE COMPENSATION
Peak Current [1]
ISLOPE
Sawtooth current waveform added to currentsense input (SP)
GM AMPLIFIER
Open Loop DC Gain
Transconductance
gmCOMP
1100
1500
1900
µA/V
COMP Sink Current [1]
ICOMP
–
290
–
µA
COMP Source Current [1]
ICOMP
–
–105
–
µA
ILCOMP
–
±200
–
nA
COMP Leakage
Current [1]
DITHER GENERATOR
Dither Modulation Frequency
fDITH
7.9
9.6
11.2
kHz
Maximum Switching Frequency
fOSCMAX
ROSC = 72 kΩ, RDITH = 110 kΩ
348
400
452
kHz
Minimum Switching Frequency
fOSCMIN
ROSC = 72 kΩ, RDITH = 110 kΩ
261
300
339
kHz
PROTECTION FEATURES
Fault Blank Timer [4]
VIN Undervoltage Turn-Off
tFB
VINUV
VIN Undervoltage Hysteresis
DVINUV
VREG Undervoltage Turn-Off
VREGUV
VREG Undervoltage Hysteresis
Startup
–
3
–
ms
Decreasing VIN , IREG = 2 mA
3.9
–
4.2
V
250
–
380
mV
Decreasing VREG
3.25
–
3.5
V
–
300
–
mV
DVREGUV
LED Overcurrent Threshold
VOCLED
LP wrt LN
260
320
380
mV
Fixed-Output Overvoltage Threshold
VFOOV
Monitored at LP pin with respect to GND
53.3
55.5
57
V
Programmable-Output Overvoltage Threshold
VPOOV
OVUV wrt LP
–1.24
–1.1
–1
V
Output Undervoltage Threshold
VOUV
OVUV wrt LP
–110
–85
–60
mV
Switch Current Overload Period
tSCOP
Inner loop switch current
–
64
–
clock
cycles
LED Overcurrent Period
tOPI
–
2
–
clock
cycles
LED Output Undervoltage Period
tOPV
–
30
–
clock
cycles
Hiccup Shutdown Period
tHIC
LED overcurrent, or output undervoltage, or
overvoltage, or switch overload
22
26.5
31.75
ms
–
–
6
V
PWMOUT Undervoltage Turn-On
VPWMUVON
Measured at LP wrt GND
PWMOUT Undervoltage Turn-Off
VPWMUVOFF
Measured at LP wrt GND
3.7
–
5.8
V
Overtemperature Shutdown Threshold
TJF
Temperature increasing
155
170
–
°C
Overtemperature Hysteresis
DTJ
Recovery = TJF – DTJ
–
20
–
°C
[1] For
input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.
Function is correct, but some parameters may not meet specification.
[3] Parameters guaranteed by design and characterization.
[4] Fault blank timer only enabled for either output undervoltage or switch current overload.
[2]
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
8
A6271-1
Automotive, High-Current LED Controller
FUNCTIONAL DESCRIPTION
The A6271-1 is a DC-DC converter controller designed to drive
series-connected high-power LEDs in automotive applications. The A6271-1 can be configured in a variety of switching topologies, including: boost, buck-boost, SEPIC, and buck
(ground-referenced switch). For each switching configuration, the
appropriate loop compensation (COMP) and slope compensation
(SLOPE) passive components are selected for optimal performance.
The A6271-1 integrates all the necessary control elements to
provide a cost-effective solution using an external logic-level,
n-channel MOSFET (switching device), p-channel MOSFET
(PWM device), and minimum additional external passive components. The maximum LED current is set with a single external
sense resistor and can be accurately modulated using a current
reference input (analog control). Direct PWM control is possible
via the PWMIN input, which also provides a shutdown mode.
Circuit Operation
CONVERTER
The controller is based on a fixed-frequency, peak current-mode
control architecture. There are two loops within the controller.
The inner loop, formed by the amplifier AC (refer to Functional
Block Diagram), the slope generator, the comparator, CF, and the
RS bistable, controls the inductor current as measured through
the switching MOSFET by the sense resistor RSS. The outer loop,
formed by the amplifier AA and the integrating GM amplifier
AB, controls the average LED current by providing the current
demand signal for the inner loop.
The LED current is measured by the sense resistor, RSL, and is
averaged and amplified to a level where it is compared to the
internal reference current to produce an error signal at the output
of the GM amplifier, AB. This error signal is effectively the current demand signal and determines the amount of energy transferred to the LEDs on a cycle-by-cycle basis via the inner loop.
increases, causing the current demand signal to decrease. This
reduces the amount of energy transferred to the LED load by terminating the switch current sooner and reducing the LED current.
EXTERNAL PULSE-WIDTH MODULATION DIMMING
An external logic signal can be applied to PWMIN pin to control
the on/off of LED current. Average brightness of the LED is
directly proportional to the duty cycle of the control signal. This
technique is commonly known as PWM dimming.
During PWM operation, when PWMIN is pulled low, the LED
stack PWMOUT is pulled high with respect to LP, turning off
the external p-channel MOSFET, isolating the LED string. In
addition, the GM output (amplifier AB) is ‘parked’ (COMP components disconnected) at the new level and the gate drive (SG)
is disabled. As the output capacitance is isolated from the LED
string, there is no loss of charge.
When PWMIN goes high impedance, or is pulled high, the
COMP components are reconnected (with the previous ‘parked’
value’), the gate drive (SG) is enabled, PWMOUT is pulled to
around –8 V with respect to LP turning on the external MOSFET
and allowing current to flow through the LED string.
INTERNAL PULSE-WIDTH MODULATION DIMMING
Where an external PWM signal is not available, the internal
PWM generator can be used for controlling the LED brightness.
A resistor connected between the PWMIN pin and GND sets the
PWM frequency according to the following formula:
RFREQ =
14,000
fPWM
where RFREQ is in kΩ and fPWM is in Hz.
The duty cycle is controlled by applying a voltage to the DR pin.
The VREG can be used for the supply voltage and a potential
divider can be used to set the DR voltage. An additional resistor
can be added in parallel via a MOSFET switch between DR and
GND to change the duty cycle between two levels.
The control loops work together as follows: at the beginning
of each oscillator cycle, the bistable is set and the switching
MOSFET is on. The switch current builds up due to the voltage developed across the inductor, and when the corresponding
signal produced at the output of amplifier AC reaches the current
demand level on the output of amplifier AB, the bistable is reset
and the switching MOSFET is turned off. The cycle is repeated
on the next oscillator cycle.
So, for example, with a DR voltage = 1.8 V, the programmed duty
cycle = 50%.
If the current through the LEDs increases, the output of AA
In terms of the control of the external MOSFET via the PWM-
The relationship between the DR voltage and the duty cycle is as
follows:
PWM Duty cycle (%) = 27.81 × DR voltage
Allegro MicroSystems, LLC
955 Perimeter Road
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9
A6271-1
Automotive, High-Current LED Controller
OUT pin, the control is identical as the external PWM scheme. If
no internal PWM is required, the DR pin should be connected to
VREG. Also note that whenever an external PWM signal is applied
to the PWMIN pin, it overrides the internal PWM operation.
to the following formula:
When using the internal PWM scheme, an n-channel MOSFET is
required to open the ground connection of the resistor connected
between PWMIN and GND to ensure that startup occurs. The
gate of the MOSFET is connected to VREG as shown in Figure
11 and Figure 13, or to an external control signal as shown in Figure 12 and Figure 14.
where tsoft is the desired soft-start period.
As the PWMIN input has a pull-up of only 1.1 µA in sleep mode,
it is essential that the zero gate voltage, drain current (leakage) of
the MOSFET does not exceed this number at maximum ambient
temperatures.
ANALOG DIMMING
The IREF pin can then be used for full analog control. The LED
current can be linearly adjusted from zero to full (100%) LED
current (ILED) by changing the IREF pin from 0 to ≥ 1 V.
This feature is useful in applications where PWM control is either
not required or not available and the LEDs require some dynamic
correction for brightness adjustment.
Csoft =
-6
tsoft × 1 × 10
1.2
If analog dimming is applied, the equivalent current source from
this circuit will add to the internal 1 µA source current on the
IREF node. Generally speaking, when using analog dimming via
VREG and a potential divider, no soft-start or negligible soft-start
is provided as shown in the example below. References are taken
from Figure 9 on page 23:
1 µA
R5
20 kΩ
VREG
(5 V)
R7
10 kΩ
C6
22 nF
IREF
Pin
Figure 3: Charging Of Soft-Start Cap During Power Up
From the above diagram, VREG, R5, and R7 can be simplified
using Norton’s Theorem.
Analog dimming can be used along with either pulse-widthmodulation technique, internal or external. This is useful for
applications where some color correction is required along with
brightness control.
The equivalent resistance can be found:
Soft-start can be provided via the analog dim signal when either
coming out of sleep or hiccup mode. The internal 1 µA internal
source current on the IREF node can be overridden by applying
a ramp signal to IREF. The soft-start duration is controlled by the
signal on IREF as it is ramped from 0 to 1 V.
The current source can be found:
RT =
20 × 10
= 6.67 kΩ
20 + 10
Isource=
5
= 750 µA
RT
If no soft-start is required, the IREF pin should be connected to
VREG.
SOFT-START
When the A6271-1 comes out of sleep mode, soft-start is required
to bring the output voltage up in a controlled open-loop fashion.
This minimizes the possibility of the control loop saturating during the startup phase and subsequent output voltage overshoot,
which can induce high transient peak currents in the LED string
prior to the loop being brought back into linear control.
The soft-start period can be programmed by the selection of the
appropriate capacitor between IREF pin and GND pin according
1 µA
750 µA
RT
6.67 kΩ
C6
22 nF
IREF
Pin
Figure 4: Equivalent Circuit Of The Previous Diagram
From the above schematic, it is clear that the 750 µA current
source will dominate and almost no soft-start will be provided. In
this particular case, the only option is to resize C6, or increase the
values of R5 and R7, or both.
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A6271-1
Automotive, High-Current LED Controller
LED CURRENT-SENSE RESISTOR
5 V REGULATOR, VREG
The LED current is programmed by the LED sense resistor, RSL,
according to:
To provide a filtered output and to ensure the regulator is stable,
a 1 µF ceramic capacitor is required to be connected between
VREG and GND. The ceramic type should be a quality type such
as X5R, X7R, or X8R.
ILED =
VIDL
RSL
where the loop typically regulates VIDL to 200 mV when in either
internal or external PWM modes.
The power loss of the resistor should be taken into account to
ensure the correct package size is selected.
The power loss of the LED current-sense resistor, RSL is:
P = ILED2 × RSL
It is advisable to insert a 150 Ω resistor in series with the LN pin,
as shown below, to protect the internal ESD structures between
LN and LP under certain fault conditions. The 150 Ω value is
selected as a balance between limiting the fault current and minimizing the LED current error caused by the bias current flowing into the LN pin. A small cap (such as 47 nF) may be added
between LP and LN pin. It helps to filter switching noises from
current sense signal.
The 5 V regulator is sized for driving the external switching
MOSFET. However, it can be used for functions that require
minimal current, e.g. pulling up the FAULTn output and providing a reference for the DR, the IREF pin, or both.
To check the load that the MOSFET provides, it is necessary to
check the total gate charge required for a 5 V drive. This can be
derived from the gate charge, QG, versus gate drive voltage, VGS,
from the MOSFET datasheet. Once the gate charge is found, the
regulator load current can be determined:
ILOAD = (QG × fSW )+ Iexternal
where Iexternal is the additional circuitry added to the VREG
output.
The ILOAD should not exceed the VREG external current limit
(IREGCL).
750
700
LP
LN
47 nF
600
RSL
150 Ω
Figure 5: Recommended Current Sensing Network
SLEEP MODE
If PWMIN is held low for longer than the disable time, tDIS1 or
tDIS2, then the A6271-1 will shut down and put the majority of the
circuitry into a low-power sleep mode.
When internal PWM dimming is used, the disable time, tDIS1, is
14.5 ms.
When either external PWM dimming or analog dimming is used,
the disable time, tDIS2 is 29 ms.
Oscillator Frequency, FOSC (kHz)
A6271-1
650
550
500
450
400
350
300
250
200
150
100
50
0
50
100
150
200
250
300
350
400
Oscillator Resistor Value, ROSC (kΩ)
Figure 6: ROSC Required for a Particular Oscillator
Frequency
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A6271-1
Automotive, High-Current LED Controller
OSCILLATOR
The main oscillator may be configured as a clock source or
it may be driven by an external clock signal. The oscillator is
designed to run between 70 and 700 kHz.
Δf as a percentage of the delta with respect to the oscillator frequency is (50 / 350) × 100% = 14.3%.
Therefore, RDITH can be found from:
f = ±22 ×
When the oscillator is configured as a clock source, the frequency
is programmed via an external resistor between OSC pin and
GND pin. The appropriate resistor can be found:
ROSC =
25,690
fOSC
where ROSC is in kΩ and fOSC is in kHz.
Figure 4 shows the resulting ROSC for various frequencies.
When the OSC pin is driven by an external clock source, a number of A6271-1s can be synchronized together. If the clock period
is greater than or equal to 17 µs, a watchdog circuit causes the
running frequency to default to the internal oscillator, which runs
at 350 kHz.
If the oscillator pin goes either open circuit or short circuits to
GND, the running frequency defaults to 350 kHz.
FREQUENCY DITHERING
To assist in minimizing EMI emissions, the main oscillator can
be dithered so that the energy is spread over a defined frequency
band. The defined frequency band is effectively the minimum
and maximum switching frequency selected. This frequency is
varied above and below the selected oscillator frequency and is
set via a resistor connected between Dither pin and GND pin. The
frequency band can be selected as follows:
Δ f = ±22 ×
ROSC
RDITH
where Δf is a plus/minus percentage change with respect to the
oscillator frequency.
For example, if an oscillator frequency of 350 kHz and a dithered frequency band of ±50 kHz was selected, given a minimum
switching frequency of 300 kHz and a maximum switching
frequency of 400 kHz, the ROSC and RDITHER can be found:
25,690
ROSC =
fOSC
ROSC =
25,690
= 73.4 kΩ , say 72 kΩ
350
RDITH = 22 ×
ROSC
RDITH
72
= 110 k
14.3
The switching frequency is modulated at a rate of 10 kHz via a
triangular waveform. This means in one modulation cycle, the
switching frequency varies linearly from a minimum to a maximum to a minimum again.
If the dither feature is not required, the DITH pin should be tied
to VREG.
PROTECTION
The A6271-1 includes a number of safety features to ensure the
controller, the external power components, and the LED string
are protected.
The Fault Flag becomes active (pulled low) for any fault.
When the device recovers from a fault, a soft-start is performed
unless analog dimming is selected and the DR pin is tied to
VREG.
After the LED current has returned to regulation for 1 ms, the
FAULT pin is then released. This allows the FAULT flag to be
used as a ‘Power Good’ signal.
Note: If IREF pin is held below 0.1 V (corresponding to 64 counts: low-side MOSFET (SG) and PWM MOSFET (PWMOUT) off and FAULTn active, hiccup
period, then auto-restart with soft-start.
Note: fault blanked for 3 ms during startup.
LED Overcurrent
Low-side MOSFET (SG) and PWM MOSFET (PWMOUT) immediately off and FAULTn active, hiccup period
after 2 counts, then auto-restart with soft-start.
Output Undervoltage
PWM MOSFET (PWMOUT) immediately off. FAULTn active. Low-side MOSFET (SG) continues switching for
30 cycles. If fault still exist after 30 cycles, IC stops switching and enters hiccup period, then auto-restart with
soft-start. Note: this undervoltage fault detection is blanked for 3 ms during startup.
Fixed-Output Overvoltage
Low-side MOSFET (SG) off and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active, hiccup
period after 1 count, then auto-restart with soft-start.
Programmable-Output Overvoltage
Low-side MOSFET (SG) off and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active, hiccup
period after 1 count, then auto-restart with soft-start.
Input Undervoltage
Low-side MOSFET (SG) and PWM MOSFET (PWMOUT) immediately turns off and FAULTn active assuming
there is sufficient drive to the flag. Once input voltage is above the VIN undervoltage threshold, plus hysteresis,
auto-restart with soft-start occurs.
VREG Undervoltage
Low-side MOSFET (SG) immediately turns off and FAULTn active assuming there is sufficient drive to the flag.
Once VREG voltage is above the VREG undervoltage threshold, plus hysteresis, then auto-restart.
Thermal Shutdown
Low-side MOSFET (SG) immediately turns off and FAULTn active. Auto-restart with soft start occurs after the
temperature drops below the overtemperature minus hysteresis level.
PWMOUT Undervoltage
PWM MOSFET (PWMOUT) off immediately and FAULTn active. Low-side MOSFET (SG) continues switching
to pump up output voltage. Once LP rises above VPWMUVON, PWMOUT is re-activated.
OSC Pin Fault
The oscillator will switch to default frequency of 350 kHz.
COMP Short to GND
Force regulator to minimum duty cycle.
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A6271-1
Automotive, High-Current LED Controller
COMPONENT SELECTION
Inductor
The maximum average inductor current can be determined:
The main factor in selecting the inductor value is to target a certain ripple current to ensure the peak current-mode control works
correctly. A reasonable figure is a peak-to-peak ripple current
of around 15% of the average inductor current. The maximum
inductor current occurs at minimum input voltage and maximum
duty cycle.
BOOST INDUCTOR SELECTION
The maximum duty cycle can be found:
VLED + (Vf – VIN(MIN) )
DMAX =
VLED + Vf
where VLED is the LED output voltage, Vf is the forward voltage drop of the recirculation diode, and VIN(MIN) is the minimum
input voltage.
The maximum average inductor current can be determined:
ILED
IAVE =
(1 – DMAX )
The ripple current, ΔI = 0.15 × IAVE .
The minimum inductance can now be found:
L=
(VLED + Vf – VIN(MIN) )× (1 – DMAX )
ΔI × fSW
where fSW is the switching frequency.
The peak current in the inductor is:
ILPK = IAVE +
ΔI
2
BUCK-BOOST INDUCTOR SELECTION
The maximum duty cycle can be found:
VLED + Vf
DMAX =
VLED + Vf + VIN(MIN)
where VLED is the LED output voltage, Vf is the forward voltage drop of the recirculation diode, and VIN(MIN) is the minimum
input voltage.
IAVE =
ILED
1 – DMAX
The ripple current, ΔI = 0.15 × IAVE .
The minimum inductance can now be found:
L=
VIN(MIN)× DMAX
ΔI × fSW
where fSW is the switching frequency.
The peak current in the inductor is:
ILPK = IAVE +
ΔI
2
When selecting an inductor from manufacturers’ datasheets, there
are often two current ratings given:
1. Saturation current. This is the current level that causes the
inductance to drop by between 10 and 40% depending on the
manufacturer.
The saturation current should be greater than the peak current, ILPK, with some margin to allow for overload conditions.
2. RMS or average current. This is the current level that determines a certain temperature rise in the inductor with a given
ambient temperature. This is normally presented as a single
figure: operating temperature.
The RMS or average inductor current rating should be greater
than the estimated maximum average current, IAVE.
Recommended inductor manufacturers:
• Coilcraft: MSS1278T or MSS1078T Range
• TDK: SLF12575 type H
SWITCH CURRENT SENSE
The switch current sense of the ‘inner loop’ is measured by the
external sense resistor, RSS, and the switch sense amplifier, AC.
As well as providing the peak current information to determine
the duty cycle, it also provides pulse-by-pulse current limiting
through the switching MOSFET and slope compensation to prevent subharmonic oscillations at duty cycles greater than 50%.
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A6271-1
Automotive, High-Current LED Controller
The current limit of the inner loop is set by the input limit of the
sense amplifier, VIDS, the maximum switch current that has been
determined, and the effects of the slope compensation have to be
taken into account. The operating duty cycle has to be calculated
at maximum load and minimum operating input voltage. The
amount of slope compensation can be calculated for this operating point and can then be added to the actual current-sense signal
to determine the maximum signal amplitude before cycle-bycycle current limiting takes effect. Refer to Slope Compensation
Section to find diL /dt then diSLOPE / dt.
RSS =
1.2×
(
0.32
diSLOPE DMAX
×
ILP +
FSW
dt
(
))
Note that the minimum value of VIDS is used with an additional
20% to allow for margin.
ILP is the peak current in the inductor.
MOSFET, is used to program the appropriate voltage level to
scale the slope compensation for correct use with the appropriate
topology and set up conditions that have been adopted.
Boost Slope Resistor
The inductor down slope is:
diL
VLED + Vf – VIN(MIN)
=
dt
L
Buck-Boost Slope Resistor
The inductor down slope is:
diL
VLED + Vf
=
dt
L
The optimum down slope as illustrated by Ridley can be found
from:
(
Boost RSS Power Loss
Using the DMAX and IAVE from the boost part of the inductor section, the power loss of RSS can be found:
Ploss = IAVE2 × DMAX × RSS
Buck-Boost RSS Power Loss
Using the DMAX and IAVE from the buck-boost part of the inductor section, the power loss of RSS can be found:
Ploss = IAVE2 × DMAX × RSS
Resistor manufacturers typically derate the devices from an
ambient temperature of around 70°C. The power rating including
derating of the sense resistor should exceed the maximum power
loss at maximum ambient temperature.
SLOPE COMPENSATION
Slope compensation can be added to the MOSFET current-sense
signal on pin SP to prevent subharmonic oscillations where the
peak-to-average control error becomes increasingly larger at duty
cycles in excess of 50%. A current source is provided at the SP
pin as a sawtooth from 0 to 100 µA. An external resistor, RSLOPE,
connected between the SP pin and the source connection of the
)
diSLOPE diL
0.18
=
× 1–
dt
dt
DMAX
The power loss of the switch current-sense resistor, RSS, can be
found:
The slope compensation resistor can be found:
diSLOPE
× RSS
dt
RSLOPE=
–6
–6
100 × 10 × 1 × 10 × fSW
where RSLOPE is in ohms (Ω).
CONTROL LOOP COMPENSATION
The recommended way of closing the control loop is to remove
the influence of the right-hand plane zero (RHPZ) in both boost
and buck-boost topologies. The reason for this is that the RHPZ
increases the gain by 20 dB/decade and at the same time introduces a 90-degree phase lag.
The minimum frequency that the RHPZ occurs at is:
For boost mode:
fRHPZ=
VLED × (1 – DMAX )
2
2 × π × L × ILED
For buck-boost mode:
2
fRHPZ=
VLED × (1 – DMAX )
2 × π × L × ILED × DMAX
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A6271-1
Automotive, High-Current LED Controller
It is recommended that the 0 dB crossover point is approximately:
fRHPZ
fCROSS =
5
The frequency of the first GM amplifier pole can be found:
fp1 =
With effective peak current-mode control, it can be assumed that
the second power pole is pushed high enough in the frequency
domain to have no influence on the overall loop response. It is
reasonable to assume the overall loop response is effectively a
single pole set by the GM amplifier (COMP node). The error
amp zero is set at the same frequency as the output power pole to
ensure the loop is closed at a rate of 20 dB/decade.
The open-loop DC gain of the system can be found:
Boost:
DC Gain =
RSS ×
(( VI ) + (n × R
LED
dyn
LED
Buck-Boost:
DC Gain =
( VI )
) + R ))
5 × 1,259 × RSL × (1 – DMAX ) ×
5 × 1,259 × RSL × (1 – DMAX) ×
RSS ×
(( VI ) + D
LED
LED
MAX
LED
LED
SL
( VI )
Capacitor on the output of the GM amplifier (COMP node)
required to achieve the above pole position:
–6
Ccomp =
750 × 10
2 × π × fp1 × 1,258
The frequency position of the power stage pole and the GM
amplifier zero is:
Boost:
fp2 and fz1 =
VLED + ILED × ((n × Rdyn ) + RSL )
2 × π × VLED × COUT × ((n × Rdyn ) + RSL )
Buck-Boost:
fp2 and fz1 =
VLED + DMAX × ILED × ((n × Rdyn ) + RSL )
2 × π × VLED × COUT × ((n × Rdyn ) + RSL )
The resistor (Rcomp) in series with the compensation capacitor
(Ccomp) on the COMP node can be found:
LED
LED
1
2 × π × RC × DC Gain
)
× ((n × Rdyn ) + RSL )
where n = number of LEDs and Rdyn = LED dynamic resistance.
Note that the LED dynamic resistance may be given in the LED
datasheet. If it is not, it can be derived by a simple measurement.
Set up a power supply with a current limit at the operating point
(ILED1). Apply the current to an individual LED and measure
the voltage drop (VLED1). Change the current limit by a small
amount, say 5% (ILED2), and measure the voltage drop (VLED2).
The dynamic resistance can be estimated:
VLED1 – VLED2
Rdyn =
ILED1 – I LED2
The RC constant required to achieve 0 dB with a slope of 20 dB/
decade at the crossover frequency, fCROSS:
1
RC =
2 × π × fCROSS
Rcomp =
1
2 × π × fp2 × Ccomp
LOW-SIDE SWITCHING MOSFET
A logic-level n-channel MOSFET is used as the switch for the
DC-DC converter.
In the boost configuration, the maximum voltage across the
drain-source connection is:
VDS = VLED + Vf
In the buck-boost configuration, the maximum voltage across the
drain-source connection is:
VDS = VLED + Vf + VINMAX
The actual rating of the MOSFET selected should be greater than
the maximum voltage plus some margin. It is recommended that
the minimum margin should be no less than 20% of the maximum voltage.
In the case of buck-boost mode, the maximum rating should factor in load-dump conditions.
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A6271-1
Automotive, High-Current LED Controller
In terms of the current rating, the MOSFET is generally selected
for a low RDS rating to minimize the power dissipation. This
means the current rating is well in excess of the actual maximum
current used in the application.
The power loss in the MOSFET is determined by the static loss
and the switching losses.
Static Loss
Using the DMAX and IAVE from the boost or buck-boost part of
the inductor section, the power loss of RDS can be found:
Ploss = IAVE2 × DMAX × RDS
Note that the RDS figures are generally presented at 25°C room
ambients. The actual RDS can be determined by considering the
normalized RDS versus temperature graph.
Another consideration of the static loss is cold-crank situations.
It is important to ensure the gate-drive amplitude (derived from
VREG) at the minimum input voltage provides sufficient drive
that the RDS does not increase by much, therefore minimizing any
increase in losses. A good quality logic-level MOSFET should
have good RDS performance at drive voltages of less than 4 V.
The VREG load can be determined by estimating the gate losses.
From the MOSFET datasheet, the total gate charge can estimated
with a gate drive of 5 V using the appropriate graph. In addition,
any other circuitry that VREG is powering should also be factored. The current drawn from VREG due to the MOSFET drive
can be determined:
VREGMOSFETload = QTOTALGate × fSW
Switching Losses
The switching losses in the MOSFET are determined by the
length of time of the Miller region. To minimize conducted and
radiated EMI emissions, this region is deliberately extended by
adding series resistance between the gate drive (SG) and the gate
of the device. It is assumed that the turn-off loss is similar to the
turn-on loss.
In the case of the boost converter, the switching loss:
Pswitch = (VLED + Vf ) × IAVE × tmiller × fSW
In the case of the buck boost converter, the switching loss:
Pswitch = (VLED + Vf + VIN(MIN) ) × IAVE × tmiller × fSW
RECIRCULATION DIODE
The diode should have a low forward voltage to reduce conduction losses and a low capacitance to reduce switching losses
and minimize EMI. Schottky diodes can provide both features
if carefully selected. The forward voltage drop is a natural
advantage for Schottky diodes and reduces as the current rating
increases. However, as the current rating increases, the diode
capacitance also increases, so the optimum selection is usually
the lowest current rating above the required maximum, in this
case ILPK.
In the boost configuration, the maximum reverse voltage across
the diode is:
VRRM = VLED + Vf
In the buck-boost configuration, the maximum reverse voltage
across the diode is:
VRRM = VLED + Vf + VIN(MAX)
The actual rating of the diode selected should be greater than the
maximum voltage plus some margin. It is recommended that the
minimum margin should be no less than 20% of the maximum
voltage. In the case of buck-boost mode, the maximum rating
should factor in load-dump conditions.
HIGH-SIDE PWM MOSFET
A p-channel MOSFET is used as the PWM switch for the LED
stack.
In both boost and buck-boost modes, the maximum voltage
across the drain-source connection is VLED. The actual rating
of the MOSFET selected should be greater than the maximum
voltage plus some margin. It is recommended that the minimum
margin should be no less than 20% of the maximum voltage.
The power loss of this MOSFET is dominated by the static loss.
The switching losses can largely be ignored as the PWM frequencies are relatively low.
The power loss of the MOSFET RDS can be found:
Ploss = ILED2 × RDS
The gate drive for the PWM MOSFET is derived from the LED
output rail (LP pin). In boost and buck-boost modes, this node is
boosted with respect to the input voltage (VIN), so there should be
sufficient negative gate drive.
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A6271-1
Automotive, High-Current LED Controller
In other operating modes such as buck, where the output voltage is less than the input voltage, it may be necessary to use low
threshold p-channel MOSFETs to ensure adequate overdrive during cold-crank situations.
OUTPUT CAPACITOR
There are several points to consider when selecting the output
capacitor.
Due to the switching topology used, the ripple current for this
circuit is high since the output capacitor provides the LED current when the DC-DC converter switch is active in both boost
and buck-boost modes. The capacitor is then recharged each time
the inductor passes energy to the output. The ripple current on the
output capacitor will be equal to the peak inductor current. The
corresponding output ripple can be derived from the amount of
charge transferred to the output during the switch on time.
To minimize heating effects and voltage ripple, the equivalent
series resistance (ESR) and the equivalent series inductance (ESL)
should be kept as low as possible. This can be achieved by multilayer ceramic chip (MLCC) capacitors. To reduce performance
variation over temperature, low drift types such as X7R and X5R
should be used.
The value of the output capacitor will typically be in the range
of 3.3 to 10 µF, and it should be rated above the maximum LED
stack voltage, VLED. There is an E-field effect with ceramic
capacitors that causes the capacitance to fall at elevated voltages.
It is therefore recommended that a good margin is selected to
minimize this effect.
One potential issue of ceramic capacitors is audible noise during
pulse-width modulation (PWM). This is caused by the piezoelectric effect of the ceramic substrate. To minimize the effects
of this, it is recommended to use multiple physically smaller
capacitors. If this is still an issue, it is recommended that either
low-impedance electrolytic or polymer capacitors be used.
INPUT CAPACITOR
The function of the input filter capacitor is to provide a lowimpedance shunt path for the current drawn by the A6271-1 when
the switching MOSFET turns on. The objective is to minimize
the ripple current reflected back into the source supply. This
approach helps to minimize conducted emissions into the power
source. Additional line impedance in the form of chokes can be
added to improve the emissions further.
In a correctly designed system, with a quality capacitor or capacitors positioned adjacent to the power train circuitry, these capacitors should supply the ripple current.
The amount of capacitance required at the input is dictated by the
EMI performance. This is usually distributed with series ferrite
beads and either differential-mode chokes, or common-mode
chokes, or both.
Layout
The following layout guidelines should be followed to ensure
satisfactory electrical and EMI performance.
Ground planes should be used on as many layers as possible. This
is essential in minimizing ‘ground bounce’ (differential voltage
across the ground connection). ‘Ground bounce’ can lead to radiated noise which can then be picked up on both input and output
connections and manifest as common-mode noise. Any ground
planes on different layers should be connected using multiple vias
in an attempt to minimize ground impedances. The ground tab
under the A6271-1 should also have multiple vias connecting to
the ground plane or planes.
The drain connection of the switching MOSFET, PWM MOSFET, and cathode terminal of the recirculation diode are used for
thermal heatsinking. It is advised to use sufficient copper around
these connections on the component layer of the PCB only. The
areas directly under these connections on the PCB should form
part of the ground plane. The reason for restricting the copper
area on these nodes is because they can radiate noise due to the
nature of the dv/dt and di/dt power signals that appear.
The area of the switching power loops should be minimized as
much as possible. In addition, the trace connections should be as
wide as possible to minimize parasitic leakage inductances, but at
the same time not compromising the power loop area. There are
two power loops:
Loop 1: formed by the input filter, main switching MOSFET,
power inductor, and inner loop sense resistor.
Loop 2: formed by the power inductor, recirculation diode,
LED sense resistor, PWM MOSFET, and the output capacitor or
capacitors.
Where practical, keep input or output filter magnetics as far away
from the power-switching inductor (L1) as possible. This is to
avoid or at least minimize the effects of magnetic crosstalk.
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A6271-1
Automotive, High-Current LED Controller
One of the major noise contributors is the switching MOSFET
(M1). Slowing down the gate drive without compromising the
thermal solution will help to minimize noise.
RO =
Lleak
Cleak
To comply with CISPR 25, a common-mode choke is typically
required as part of the input filter.
RO can be selected as the damping resistor. Typically, either an
0805 or 0603 resistor case size is adequate.
Reducing EMI
INPUT FILTER
It is essential that good layout practice as defined in the Layout
Section should be adopted. The following techniques are also
recommended.
The selection of the components that form the input filter
depends on the noise that is present in the system in terms of the
frequency and whether it is common mode or differential mode.
In addition, the common automotive standards that exist define
onerous specification limit lines for the emissions in the AM band
(approximately 530 kHz to 1.7 MHz) and the FM band (approximately 70 to 108 MHz). Some consideration must be given to
these frequency bands to understand how to filter these regions.
SNUBBER
Adding a low-loss R-C snubber network between the drain of
the main switching MOSFET and ground helps to suppress the
resonant ringing on the switching node. The process for selecting
these components involves some ‘trial and error’ on the actual
printed circuit board.
Step 1: Measure the voltage resonance frequency on the LX
node.
Step 2: Add an additional capacitance between LX and ground
until the resonant frequency is halved. Note that this capacitance
should be around 1 nF.
Step 3: Two equations with two unknowns are now obtained:
FRES =
1
2 × π × Lleak × Cleak
1
FRES
=
2 × π × Lleak × Cnew
2
where Cnew = Cleak + Cadd.
Cadd = additional capacitance added.
Cleak = parasitic capacitance.
Lleak = parasitic inductance.
Now to halve the frequency, Cleak + Cadd = 4 x Cleak
At the lower frequencies, below a few 10s of MHz, the noise is
generally dominated by differential noise with some common
mode noise. At frequencies above a few 10s of MHz, the noise is
dominated by common mode noise with some differential noise.
To address the differential noise, a differntial inductor can be
used along with differential capacitance to form an L-C filter.
One problem in using standard differential inductors is that the
self-resonance frequency (SRF) is typically in the region of a few
10s of MHz, even with a modest few microhenries (µH) (note:
the higher the inductance, the lower the SRF). This means that
above the self-resonant frequency points, these components actually amplify the noise and make matters worse.
Some differential-mode inductive filtering is always necessary.
Ferrite beads can be used for this function. Although ferrite beads
are designed to act as a lossy resistor at particular frequency
bands, they do have an inherent inductive element which can be
in the region of several µH. The inductance of a ferrite bead can
be extracted from the reactance information graph (refer to Figure 7). At a particular frequency, the reactance can be found and
then the inductance can be derived.
With Cleak solved, Lleak can also be solved.
As a single ferrite bead may not be effective enough, a two-stage
ferrite bead filter approach can be taken. These components,
along with input differential-mode capacitors, can form L-C filter
stages. For the best result, the first L-C filter should be placed as
close to the power stage as possible.
The characteristic impedance of the parasitic components can be
found:
At higher frequencies, the majority of the noise problems is
associated with common-mode noise. This noise is induced by
Therefore,
Cadd
Cleak =
3
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21
A6271-1
Automotive, High-Current LED Controller
ground-referenced differential noise radiating through the ground
plane. This noise can be picked up on the input stage forming
common-mode noise on the positive and negative power supply
connections to the battery. Even with excellent layout, this noise
is always present.
All filter capacitors should be a quality ceramic: X7R or X8R.
FREQUENCY DITHERING
Further improvements to the differential-mode performance can
be made by the use of frequency dithering techniques.
10000
The A6271-1 contains a dither circuit which changes the switching oscillator frequency on a cycle-by-cycle basis across a
defined frequency band.
1000
Impedance (Ω)
away from the switching circuit, the more likely it will be to filter
this noise.
As the noise in a switcher is typically narrow-band noise, both
the peak and average signals are similar in amplitude. When a
frequency dither scheme is introduced, it ‘spreads’ the noise,
converting it from narrow band into broad band. While the peak
noise reduces across the majority of the spectrum, the reduction
in the average measurement is a lot more effective. This effect
is particularly helpful in the conducted emissions for the average
measurement between 76 and 108 MHz. This region is unusual,
as typically the peak and average limit lines track one another,
but in this area, while the peak limit line increases, the average
limit line reduces (refer to Figure 10 below). Note the red limit
lines are peak and the pink limit lines are average.
100
10
1
90
1
10
100
1000
Frequency (MHz)
Z (comm)
80
Z (diff)
60
Figure 9: Common-Mode (CM) Choke Impedance
Another important consideration is the relative positioning of
the common-mode choke with respect to the switching inductor.
Magnetic crosstalk can occur between these components which
can degrade the effectiveness of the CM choke. Even the use of
a magnetically-screened switching inductor is not sufficient to
avoid this problem. It is important to physically separate these
two components as far as possible. Another advantage of a good
physical separation is that any ‘ground bounce’ induced CM
noise will couple onto the input power traces. The strength of the
coupling will reduce with distance. The further the CM choke is
50
dBµV
To address this problem, a common-mode inductor is required.
This inductor is selected to present a high impedance around the
FM region. An example of a common-mode (CM) choke with a
high impedance around the FM region is shown below in Figure 9.
Limits
55025 P5
55025 A5
70
Transducer
UniRFPro
40
30
Traces
PK+
AV
20
10
0
-10
150 kHz
1 MHz
10 MHz
108 MHz
Figure 10: CISPR 25 Class 5 Limit Lines
The reason that the average limit line is relatively low (making
it challenging to pass) is that average weighted signals have an
adverse effect on FM radio signals.
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A6271-1
Automotive, High-Current LED Controller
Minimizing the noise at the high end of the conducted emission frequency spectrum also benefits the radiated noise at this
frequency band and above.
Note that a modulation frequency of 10 kHz was chosen, since
this aligns with the resolution bandwidth of the measurement
receiver as defined in CISPR 25. The resolution bandwidth is
effectively the ‘measurement window’ at each measurement step.
Another consideration when optimizing the frequency dithering
is the depth of frequency. This is the maximum and minimum
switching frequency that the converter operates, effectively the
‘spread range’. The wider the spread range is, the more effective
the dithering is. However, there is a trade-off with switching
losses and sizing of the power inductor in terms of inductance
value and the corresponding physical size.
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A6271-1
Automotive, High-Current LED Controller
APPLICATION CIRCUITS
L1
12 µH
D2
VBAT
C2
4.7 µF
C1
4.7 µF
C3
100 nF
D1
C9
47 nF
C4
100 nF
R13
470 mΩ
R11
150 Ω
R14
2.7 Ω
R12 10 kΩ
VREG
C5
1 µF
R1
2.7 kΩ
A6271-1
M4
SG
SP
R5
20 kΩ
IREF
Analog
Control
C6
22 nF
OSC
R8
100 kΩ
R6
10 kΩ
DITH
R9
220 kΩ
LED1
OVUV
PWMIN
M1
R7
10 kΩ
PWMOUT
R15
4.3 kΩ
DR
R4
68 kΩ
M2
LN
FAULTn
R3
10 kΩ
Low = no
PWM (100%)
LP
GND
R2
270 kΩ
High = internal
PWM ON (5%)
VIN
COMP
R10
39 Ω
C7
470 nF
R16
0Ω
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
M3
LED12
R21
220 kΩ
R17
1.2 kΩ
R18
R19
R20
C8 10 kΩ 68 mΩ 68 mΩ
22 pF
GND
Figure 11: Boost Driving 12 LEDs at 500 mA, Switching Frequency 250 kHz
Internal PWM (5%) and/or analog dimming, no soft-start, and frequency dither on.
Table 2: Application Circuit 1 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
470 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1
200 mA, 40 V Schottky diode
ST / BAT54
D2
10 A, 60 V Schottky diode
Vishay / SS10P6
L1
22 µH, high current shielded
construction
Vishay / IHLP-5050FDER220M-5A
M1,M2
N-channel signal MOSFET
2N7000 or equivalent
M3
N-channel 50 A, 100 V MOSFET
Vishay / SQD50N10
Continued on next page...
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A6271-1
Automotive, High-Current LED Controller
Table 2: Application Circuit 1 Bill of Materials (continued)
Reference
Description
M4
P-channel 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Heatsink for M3 (TO-252)
AAVID Thermalloy / 573100D00010G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
270 kΩ, 1%, 0603 or 0805
R3,R6,R12,R18
10 kΩ, 1%, 0603 or 0805
R4
68 kΩ, 1%, 0603 or 0805
R5
20 kΩ, 1%, 0603 or 0805
R7
10 kΩ, potentiometer
R8
100 kΩ, 1%, 0603 or 0805
R9
220 kΩ, 1%, 0603 or 0805
R10
39 Ω, 1%, 0603 or 0805
R11
150 Ω, 1%, 0603 or 0805
R13
470 mΩ, 1%, 0805 or 1206
R14
2.7 Ω, 1%, 0603 or 0805
R15
4.3 kΩ, 1%, 0603 or 0805
R16
0 Ω, 0603 or 0805
R17
1.2 kΩ, 1%, 0603 or 0805
R19,R20
68 mΩ, 1%, 2010
R21
220 kΩ, 1%, 0603 or 0805
Manufacturer/Part Number
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A6271-1
Automotive, High-Current LED Controller
L1
56 µH
D2
VBAT
C1
4.7 µF
C2
4.7 µF
C3
100 nF
D1
C9
47 nF
C4
100 nF
R7
2.7 Ω
R5
150 Ω
R8
2.7 Ω
R6 10 kΩ
R1
2.7 kΩ
C5
1 µF
VREG
VIN
LP
LN
PWMOUT
M2
GND
FAULTn
R9
4.3 kΩ
DR
External PWM
Drive Signal
Important Note:
If PWMIN pin is left floating,
LEDs will turn on at 100%
duty cycle.
A6271-1
OVUV
SG
PWMIN
SP
IREF
C6
22 nF
OSC
R2
73.2 kΩ
DITH
R3
220 kΩ
LED1
COMP
R4
39 Ω
C7
680 nF
R10
12 Ω
C10
C11
C12
2.2 µF 2.2 µF 2.2 µF
M1
R15
240 kΩ
R11
1.3 kΩ
R12
R13
10 kΩ 150 mΩ
C8
22 pF
LED14
R14
150 mΩ
GND
Figure 12: Boost Driving 14 LEDs at 150 mA, Switching Frequency 350 kHz
External PWM, no analog dimming, soft-start, and frequency dither on.
Table 3: Application Circuit 2 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
680 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
C10,C11,C12
2.2 µF, ceramic capacitor, X7R, 100 V
D1
200 mA, 40 V Schottky diode
ST / BAT54
D2
2 A, 100 V Schottky diode
Vishay, ST / SS2H10
L1
56 µH, power inductor shielded
construction
Coilcraft / MSS1048T-563ML
M1
N-channel, 30 A, 100 V MOSFET
NXP / PSMN038-100YLX
TDK, MuRata
Continued on next page...
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26
A6271-1
Automotive, High-Current LED Controller
Table 3: Application Circuit 2 Bill of Materials (continued)
Reference
Description
Manufacturer/Part Number
M2
P-channel 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
73.2 kΩ, 1%, 0603 or 0805
R3
220 kΩ, 1%, 0603 or 0805
R4
39 Ω, 1%, 0603 or 0805
R5
150 Ω, 1%, 0603 or 0805
R6,R12
10 kΩ, 1%, 0603 or 0805
R7,R8
2.7 Ω, 1%, 0805 or 1206
R9
4.3 kΩ, 1%, 0603 or 0805
R10
12 Ω, 1%, 0603 or 0805
R11
1.3 kΩ, 1%, 0603 or 0805
R13,R14
150 mΩ, 1%, 1206
R15
240 kΩ, 1%, 0603 or 0805
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A6271-1
Automotive, High-Current LED Controller
L1
12 µH
D2
VBAT
C1
4.7 µF
C2
4.7 µF
C3
100 nF
D1
C9
47 nF
C4
100 nF
R11
270 mΩ
R9
150 Ω
R12
270 mΩ
R10 10 kΩ
VREG
C5
1 µF
R1
2.7 kΩ
LN
PWMOUT
M4
FAULTn
LED1
DR
R3
10 kΩ
R4
68 kΩ
A6271-1
SG
SP
IREF
M2
C6
22 nF
OSC
R6
100 kΩ
DITH
R7
220 kΩ
R6
10 kΩ
R13
4.3 kΩ
OVUV
PWMIN
M1
Low = no
PWM (100%)
LP
GND
R2
270 kΩ
High = internal
PWM ON (5%)
VIN
COMP
R8
39 Ω
C7
220 nF
R14
0Ω
M3
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
LED5
R19
84.5 kΩ
R15
1 kΩ
R16
R17
R18
10 kΩ 62 mΩ 62 mΩ
C8
22 pF
GND
Figure 13: Buck-Boost Driving 5 LEDs at 1.5 A, Switching Frequency 250 kHz
Internal PWM (5%), no analog dimming, soft-start, and frequency dither on.
Table 4: Application Circuit 3 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
220 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1
200 mA, 40 V Schottky diode
ST / BAT54
D2
10 A, 60 V Schottky diode
Vishay / SS10P6
L1
12 µH, high current shielded
construction
Vishay / IHLP-5050FDER120M-5A
M1,M2
N-channel signal MOSFET
2N7000 or equivalent
M3
N-channel 50 A, 100 V MOSFET
Vishay / SQD50N10
Continued on next page...
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A6271-1
Automotive, High-Current LED Controller
Table 4: Application Circuit 3 Bill of Materials (continued)
Reference
Description
M4
P-channel 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Heatsink for M3 (TO-252)
AAVID Thermalloy / 573100D00010G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
270 kΩ, 1%, 0603 or 0805
R3,R5,R10,R16
10 kΩ, 1%, 0603 or 0805
R4
68 kΩ, 1%, 0603 or 0805
R6
100 kΩ, 1%, 0603 or 0805
R7
220 kΩ, 1%, 0603 or 0805
R8
39 Ω, 1%, 0603 or 0805
R9
150 Ω, 1%, 0603 or 0805
R11, R12
270 mΩ, 1%, 0805 or 1206
R13
4.3 kΩ, 1%, 0603 or 0805
R14
0 Ω, 0603 or 0805
R15
1 kΩ, 1%, 0603 or 0805
R17,R18
62 mΩ, 1%, 2010
R19
84.5 kΩ, 1%, 0603 or 0805
Manufacturer/Part Number
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A6271-1
Automotive, High-Current LED Controller
L1
47 µH
D2
VBAT
C1
4.7 µF
C2
4.7 µF
C3
100 nF
D1
C4
47 nF
C4
100 nF
R8
1Ω
R6
150 Ω
R9
1Ω
R7 10 kΩ
R1
2.7 kΩ
C5
1 µF
VIN
VREG
LP
LN
PWMOUT
M2
GND
FAULTn
DR
External PWM
Drive Signal
A6271-1
OVUV
SG
PWMIN
SP
R2
20 kΩ
R3
10 kΩ
IREF
Analog
Control
C6
22 nF
OSC
R4
100 kΩ
DITH
R10
4.3 kΩ
COMP
R5
27 Ω
C7
470 nF
R11
24 Ω
M1
LED1
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
LED4
R16
68 kΩ
R12
1 kΩ
R13
R14
10 kΩ 180 mΩ
C8
22 pF
R15
180 mΩ
GND
Figure 14: Buck-Boost Driving 4 LEDs at 400 mA, Switching Frequency 250 kHz
External PWM and/or analog dimming, no soft-start, and frequency dither off.
Table 5: Application Circuit 4 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
470 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1
200 mA, 40 V Schottky diode
ST / BAT54
D2
2 A, 60 V Schottky diode
ON Semiconductor / MBRS260T3
L1
47 µH, power inductor shielded
construction
Coilcraft / MSS1048T-473ML
M1
N-channel, 30 A, 100 V MOSFET
NXP / PSMN038-100YLX
M2
P-channel 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Continued on next page...
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A6271-1
Automotive, High-Current LED Controller
Table 5: Application Circuit 4 Bill of Materials (continued)
Reference
Description
R1
2.7 kΩ, 1%, 0603 or 0805
R2
20 kΩ, 1%, 0603 or 0805
R3
10 kΩ, potentiometer
R4
100 kΩ, 1%, 0603 or 0805
R5
27 Ω, 1%, 0603 or 0805
R6
150 Ω, 1%, 0603 or 0805
R7, R13
10 kΩ, 1%, 0603 or 0805
R8, R9
1 Ω, 1%, 0805 or 1206
R10
4.3 kΩ, 1%, 0603 or 0805
R11
24 Ω, 1%, 0603 or 0805
R12
1 kΩ, 1%, 0603 or 0805
R14, R15
180 mΩ, 1%, 1206
R16
68 kΩ, 1%, 0603 or 0805
Manufacturer/Part Number
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A6271-1
DRL
D1
Park
D3
Automotive, High-Current LED Controller
22 µH
L1
D2
C1
4.7 µF
C2
4.7 µF
C3
100 nF
C9
47 nF
C4
100 nF
R13
470 mΩ
R11
150 Ω
R14
2.7 Ω
R12
C5
1 µF
R1
2.7 kΩ
DR
R4
M2
PWMOUT
A6271-1
10 kΩ
M4
IREF
R7
Analog
Control
SG
SP
R5
20 kΩ
C6
22 nF
OSC
R8
100 kΩ
R6
10 kΩ
DITH
R9
220 kΩ
R15
4.3 kΩ
OVUV
PWMIN
68 kΩ
10 kΩ
LN
FAULTn
270 kΩ
R3
10 kΩ
M1
LP
GND
R2
R22
20 kΩ
VIN
VREG
COMP
R10
39 Ω
C7
470 nF
R16
0Ω
M3
LED1
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
LED12
R21
220 kΩ
R17
1.2 kΩ
R18
10 kΩ
C8
22 pF
R19
68 mΩ
R20
68 mΩ
GND
Figure 15: Park / DRL Application (Park-Dominant)
Boost Driving 12 LEDs at 500 mA, Switching Frequency 250 kHz.
Park Mode: Internal PWM (5%) and analog dimming, no soft start, and frequency dither on.
Daylight Running Light (DRL) Mode: 100% LED current and analog dimming, no soft start, and frequency dither on.
Battery voltage applied either at Park terminal or DRL terminal. Note that Park Mode is dominant.
Table 6: Application Circuit 5 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
470 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1, D3
3 A, 100 V Schottky diode
ON Semiconductor / NRVBS3100T3G
D2
10 A, 60 V Schottky diode
Vishay / SS10P6
L1
22 µH, high current shielded
construction
Vishay / IHLP-5050FDER220M-5A
M1, M2
N-channel signal MOSFET
2N7000 or equivalent
Continued on next page...
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A6271-1
Automotive, High-Current LED Controller
Table 6: Application Circuit 5 Bill of Materials (continued)
Reference
Description
Manufacturer/Part Number
M3
N-channel, 50 A, 100 V MOSFET
Vishay / SQD50N10
M4
P-channel, 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Heatsink for M3 (TO-252)
AAVID Thermalloy / 573100D00010G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
270 kΩ, 1%, 0603 or 0805
R3,R6,R12,R18
10 kΩ, 1%, 0603 or 0805
R4
68 kΩ, 1%, 0603 or 0805
R5, R22
20 kΩ, 1%, 0603 or 0805
R7
10 kΩ, potentiometer
R8
100 kΩ, 1%, 0603 or 0805
R9
220 kΩ, 1%, 0805 or 1206
R10
39 Ω, 1%, 0603 or 0805
R11
150 Ω, 1%, 0603 or 0805
R13
470 mΩ, 1%, 0805 or 1206
R14
2.7 Ω, 1%, 0603 or 0805
R15
4.3 kΩ, 1%, 0603 or 0805
R16
0 Ω, 0603 or 0805
R17
1.2 Ω, 1%, 0603 or 0805
R19, R20
68 mΩ, 1%, 2010
R21
220 kΩ, 1%, 0603 or 0805
Allegro MicroSystems, LLC
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33
A6271-1
Park
D1
DRL
D3
Automotive, High-Current LED Controller
L1
D2
C1
4.7 µF
C2
4.7 µF
C3
100 nF
R21
20 kΩ
D5
22 µH
C9
47 nF
C4
100 nF
R12
470 mΩ
R10
150 Ω
R13
2.7 Ω
R11
R5
100 kΩ
IREF
R6
10 kΩ
R22
100 kΩ
M1
D4
5.1 V
C5
1 µF
R1
2.7 kΩ
LP
LN
PWMOUT
10 kΩ
GND
M4
FAULTn
R2
DR
130 kΩ R3
10 kΩ
R4
M2
VIN
VREG
A6271-1
OVUV
SG
PWMIN
68 kΩ
SP
IREF
C6
22 nF
OSC
R7
100 kΩ
DITH
R8
220 kΩ
R14
4.3 kΩ
COMP
R9
39 Ω
C7
470 nF
R15
0Ω
LED1
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
M3
LED12
R20
220 kΩ
R16
1.2 kΩ
R17
10 kΩ
C8
22 pF
R18
68 mΩ
R19
68 mΩ
GND
Figure 16: Park / DRL Application (DRL-Dominant)
Boost Driving 12 LEDs at 500 mA, Switching Frequency 250 kHz.
Park Mode: Internal PWM (10%) and analog dimming at 45% of target level, no soft start, and frequency dither on.
Daylight Running Light (DRL) Mode: 100% LED current, no analog dimming, no soft start, and frequency dither on.
Battery voltage applied either at Park terminal or DRL terminal. Note that DRL Mode is dominant.
Table 7: Application Circuit 6 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
470 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1, D3
3 A, 100 V Schottky diode
D2
10 A, 60 V Schottky diode
Vishay / SS10P6
D4
5.1 V, Zener diode
ON Semiconductor / SZBZX84C5V1
D5
Signal diode
1N4148WS
L1
22 µH, high current shielded
construction
Vishay / IHLP-5050FDER220M-5A
ON Semiconductor / NRVBS3100T3G
Continued on next page...
Allegro MicroSystems, LLC
955 Perimeter Road
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www.allegromicro.com
34
A6271-1
Automotive, High-Current LED Controller
Table 7: Application Circuit 6 Bill of Materials (continued)
Reference
Description
Manufacturer/Part Number
M1, M2
N-channel signal MOSFET
ON Semiconductor, IR / NTR4003N
M3
N-channel, 50 A, 100 V MOSFET
Vishay / SQD50N10
M4
P-channel, 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Heatsink for M3 (TO-252)
AAVID Thermalloy / 573100D00010G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
130 kΩ, 1%, 0603 or 0805
R3,R6,R11,R17
10 kΩ, 1%, 0603 or 0805
R4
68 kΩ, 1%, 0603 or 0805
R5, R7, R22
100 kΩ, 1%, 0603 or 0805
R8
220 kΩ, 1%, 0805 or 1206
R9
39 Ω, 1%, 0603 or 0805
R10
150 Ω, 1%, 0603 or 0805
R12
470 mΩ, 1%, 0805 or 1206
R13
2.7 Ω, 1%, 0603 or 0805
R14
4.3 kΩ, 1%, 0603 or 0805
R15
0 Ω, 0603 or 0805
R16
1.2 kΩ, 1%, 0603 or 0805
R18, R19
68 mΩ, 1%, 2010
R20
220 kΩ, 1%, 0603 or 0805
R21
110 kΩ, 1%, 0603 or 0805
Allegro MicroSystems, LLC
955 Perimeter Road
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35
A6271-1
Automotive, High-Current LED Controller
PACKAGE OUTLINE DRAWING
For Reference Only – Not for Tooling Use
(Reference MO-153 ABT)
Dimensions in millimeters. NOT TO SCALE
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
0.65
0.45
8º
0º
5.00 ±0.10
16
16
0.20
0.09
1.70
B
3 NOM
4.40 ±0.10
6.40 ±0.20
A
1
3.00
0.60 ±0.15
1.00 REF
2
3 NOM
1 2
0.25 BSC
Branded Face
C
16X
0.10 C
0.30
0.19
6.10
3.00
SEATING PLANE
GAUGE PLANE
C
SEATING
PLANE
PCB Layout Reference View
1.20 MAX
0.65 BSC
NNNNNNN
0.15
0.00
YYWW
LLLL
A Terminal #1 mark area
B Exposed thermal pad (bottom surface); dimensions may vary with device
C Reference land pattern layout (reference IPC7351 SOP65P640X110-17M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
D Branding scale and appearance at supplier discretion
1
D
Standard Branding Reference View
N = Device part number
= Supplier emblem
Y = Last two digits of year of manufacture
W= Week of manufacture
L = Characters 5-8 of lot number
Figure 17: Package LP, 16-Pin eTSSOP with Exposed Thermal Pad
Allegro MicroSystems, LLC
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36
A6271-1
Automotive, High-Current LED Controller
Revision Table
Number
Date
Description
–
May 5, 2017
1
January 2, 2018
Initial release
Updated Figures 1, 11, 12, 13, 15, 16, and A-2; added Table of Contents
2
January 24, 2018
Updated Ramp Up and Ramp Down Threshold test conditions (page 7), and Protection section (page 12-13)
3
July 9, 2018
Note added to Figure 12 (page 26).
Copyright ©2018, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of
Allegro’s product can reasonably be expected to cause bodily harm.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
Copies of this document are considered uncontrolled documents.
For the latest version of this document, visit our website:
www.allegromicro.com
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37
A6271-1
Automotive, High-Current LED Controller
APPENDIX A: SPECIAL NOTES ON A6271-1 OPERATING IN SEPIC MODE
In automotive lighting applications, SEPIC (Single-Ended
Primary-Inductor Converter) is an attractive alternative to the traditional Boost and Buck-Boost topologies. As can be seen in the
following comparison diagram, the SEPIC configuration looks
very similar to the Boost configuration, except for the additional
inductor L2 and DC-blocking capacitor CFLY.
L2
VOUT = VIN × [ 1/(1-D)]
VIN
D
L
Battery
VOUT = VIN × [ D/(1-D)]
VOUT
VIN
VIN
C FLY
L1
VIN
LP
R SL
LN
D2
PWMOUT
C REG
R2
C REG
M2
A6271-1
M1
C OUT
SG
GND
PWMOUT
VREG
A6271-1
PWMIN
LN
VOUT
M2
M1
VOUT
LP
R SL
VREG
D1
C IN
Battery
C IN
L1 & L2 can be either
separate or integrated
PWMIN
SP
GND
SP
R SS
Boost Configuration
C OUT
SG
R SS
SEPIC Configuration
Figure A-1: Comparison of A6271-1 in Boost and SEPIC configurations
The main advantages of using SEPIC configuration are:
• VOUT (VLED) can be either higher or lower than VIN. This
allows the converter to handle both Cold-Crank (VIN down to
~5 V) and Load-Dump (VIN up to 40 V) situations.
• There is a DC-blocking capacitor between VIN and VOUT. This
naturally protects the power supply against Output-Shorted-toGND fault.
On the other hand, several limitations of SEPIC must be considered, namely:
• Need to use either two inductors, or one dual-winding coupled
inductor.
• The switching MOSFET and diode are subjected to higher
voltage stress of VIN + VLED. This means higher switching
loses compared to Boost configuration. Note that in BuckBoost configuration the same voltage stress applies.
• At power up, a ‘bleeder’ circuit is needed to help the IC
bypass undervoltage protection (since initially the output is at
0 V).
• A minimum output voltage of around 6 V (two WLEDs in
series) is required, for the PWMOUT gate driver to turn on the
external PMOS.
Allegro MicroSystems, LLC
955 Perimeter Road
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A-1
A6271-1
Automotive, High-Current LED Controller
The next application diagram shows the A6271-1 configured in
SEPIC mode, for driving 5 LEDs at 1.5 A.
L2 12 µH
C13
4.7 µF
L1 & L2 can be either
separate or integrated
VBAT
C1
4.7 µF
C2
4.7 µF
C3
100 nF
R20
1 kΩ
D3
D1
C9
47 nF
C4
100 nF
R11
270 mΩ
VREG
C5
1 µF
VIN
LP
LN
PWMOUT
M4
GND
FAULTn
LED1
DR
R3
10 kΩ
R4
68 kΩ
A6271-1
SG
SP
IREF
M2
C6
22 nF
OSC
R6
100 kΩ
DITH
R7
220 kΩ
R6
10 kΩ
R13
4.3 kΩ
OVUV
PWMIN
M1
Low = no
PWM (100%)
R12
270 mΩ
R10 10 kΩ
R2
270 kΩ
High = internal
PWM ON (5%)
VOUT
R9
150 Ω
VOUT
R1
2.7 kΩ
D2
L1 12 µH
COMP
R8
39 Ω
C7
220 nF
R14
0Ω
C10
C11
C12
4.7 µF 4.7 µF 4.7 µF
M3
LED5
R19
84.5 kΩ
R15
1 kΩ
R16
R17
R18
10 kΩ 62 mΩ 62 mΩ
C8
22 pF
GND
Figure A-2: SEPIC Driving 5 LEDs at 1.5 A, Switching Frequency 250 kHz
Internal PWM (5%), no analog dimming, soft-start, and frequency dither on.
Table 8: Application Circuit 7 Bill of Materials
Reference
Description
Manufacturer/Part Number
C1,C2,C10,C11,C12,C13
4.7 µF, ceramic capacitor, X7R, 50 V
TDK, MuRata
C3,C4
100 nF, ceramic capacitor, X7R, 50 V
C5
1 µF, ceramic capacitor, X7R, 16 V
C6
22 nF, ceramic capacitor, X7R, 50 V
C7
220 nF, ceramic capacitor, X7R, 50 V
C8
22 pF, ceramic capacitor, X7R, 50 V
C9
47 nF, ceramic capacitor, X7R, 16 V
D1, D3
200 mA, 40 V Schottky diode
ST / BAT54
D2
10 A, 60 V Schottky diode
Vishay / SS10P6
L1, L2
12 µH, high-current shielded construction
Vishay / IHLP-5050FDER120M-5A
M1,M2
N-channel signal MOSFET
2N7000 or equivalent
M3
N-channel 50 A, 100 V MOSFET
Vishay / SQD50N10
Continued on next page...
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
A-2
A6271-1
Automotive, High-Current LED Controller
Table 8: Application Circuit 7 Bill of Materials (continued)
Reference
Description
M4
P-channel 15 A, 100 V MOSFET
Infineon / SPD15P10PL G
Heatsink for M3 (TO-252)
AAVID Thermalloy / 573100D00010G
R1
2.7 kΩ, 1%, 0603 or 0805
R2
270 kΩ, 1%, 0603 or 0805
R3,R5,R10,R16
10 kΩ, 1%, 0603 or 0805
R4
68 kΩ, 1%, 0603 or 0805
R6
100 kΩ, 1%, 0603 or 0805
R7
220 kΩ, 1%, 0603 or 0805
R8
39 Ω, 1%, 0603 or 0805
R9
150 Ω, 1%, 0603 or 0805
R11,R12
270 mΩ, 1%, 0805 or 1206
R13
4.3 kΩ, 1%, 0603 or 0805
R14
0 Ω, 0603 or 0805
R15
1 kΩ, 1%, 0603 or 0805
R17,R18
62 mΩ, 1%, 2010
R19
84.5 kΩ, 1%, 0603 or 0805
R20
1 kΩ, 1%, 0603 or 0805
Manufacturer/Part Number
Allegro MicroSystems, LLC
955 Perimeter Road
Manchester, NH 03103-3353 U.S.A.
www.allegromicro.com
A-3
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