AAT2510IWP-AA-T1

Dual 400mA, 1MHz Step-Down DC-DC Converter General Description The AAT2510 is a member of AnalogicTech's Total Power Management IC™ (TPMIC™) product family. It is comprised of two 1MHz step-down converters designed to minimize external component size and cost. The input voltage ranges from 2.7V to 5.5V. The output voltage ranges from 0.6V to the maximum applied input voltage and is either fixed or externally adjustable. Peak current mode control with internal compensation provides a stable converter with low ESR ceramic output capacitors for extremely low output ripple. Each channel has a low 25µA quiescent operating current, which is critical for maintaining high efficiency at light load. For maximum battery life, each converter's highside P-channel MOSFET conducts continuously when the input voltage approaches dropout (100% duty cycle operation). Both regulators have independent input and enable inputs. The AAT2510 is available in a thermally-enhanced 12-pin TDFN33 package, and is rated over the -40°C to +85°C temperature range. AAT2510 Features • • • • • • • • • • • • • • • • • SystemPower™ Up to 96% Efficiency 25µA Quiescent Current Per Channel VIN Range: 2.7V to 5.5V Fixed VOUT Range: 0.6V to VIN Adjustable VOUT Range: 0.6V to 2.5V Output Current: 400mA Low RDS(ON) 0.4Ω Integrated Power Switches Low Drop Out 100% Duty Cycle 1.0MHz Switching Frequency Shutdown Current 0.6V Output TA = 25°C 1 597 250 0.7 600 0.2 615 0.2 1.5 1.0 140 15 VEN(L) VEN(H) IEN 0.6 VIN = VFB = 5.5V 1.4 -1.0 1.0 V V µA 1. The AAT2510 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured by design, characterization, and correlation with statistical process controls. 2. For adjustable version with higher than 2.5V output, please consult your AnalogicTech representative. 4 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter Typical Channel Characteristics Efficiency vs. Load (VOUT = 2.5V; L = 4.7μH) VIN = 3.3V 100 2.0 AAT2510 (VOUT = 2.5V; L = 4.7μH) Load Regulation VIN = 3.0V Efficiency (%) 90 Output Error (%) 1.0 80 VIN = 3.6V VIN = 3.0V 0.0 70 -1.0 VIN = 3.3V VIN = 3.6V 60 0.1 1.0 10 100 1000 -2.0 0.1 1.0 10 100 100 0 Output Current (mA) Output Current (mA) Efficiency vs. Load (VOUT = 1.8V; L = 4.7μH) VIN = 3.6V 100 90 2.0 (VOUT = 1.8V; L = 4.7μH) DC Regulation Output Error (%) VIN = 2.7V Efficiency (%) 1.0 VIN = 4.2V 80 70 60 50 VIN = 4.2V 0.0 VIN = 2.7V -1.0 VIN = 3.6V 0.1 1.0 10 100 1000 -2.0 0.1 1.0 10 100 100 0 Output Current (mA) Output Current (mA) Frequency vs. Input Voltage (VOUT = 1.8V) 1.0 2.0 Output Voltage Error vs. Temperature (VIN = 3.6V; VO = 1.5V) Frequency Variation (%) Output Error (%) 2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5 0.5 0.0 -0.5 -1.0 -1.5 -2.0 1.0 0.0 -1.0 -2.0 -40 -20 0 20 40 60 80 100 Input Voltage (V) Temperature (°C) 2510.2006.05.1.10 5 Dual 400mA, 1MHz Step-Down DC-DC Converter Typical Channel Characteristics Switching Frequency vs. Temperature (VIN = 3.6V; VO = 1.5V) 0.20 35 AAT2510 Quiescent Current vs. Input Voltage (VO = 1.8V) 85°C 30 Variation (%) 0.10 Supply Current (μA) 25°C 25 0.00 -0.10 20 -0.20 -40 15 -20 0 20 40 60 80 100 2.5 -40°C 3.0 3.5 4.0 4.5 5.0 5.5 6.0 Temperature (°C) Input Voltage (V) P-Channel RDS(ON) vs. Input Voltage 750 700 650 Load Transient Response (30mA - 300mA; VIN = 3.6V; VOUT = 1.8V; C1 = 10μF) 2.0 1.4 1.2 1.0 300mA 30mA 0.8 0.6 0.4 0.2 0.0 -0.2 Load and Inductor Current (200mA/div) (bottom) Output Voltage (top) (V) 5.0 5.5 6.0 RDS(ON) (mΩ) 120°C 100°C 1.9 1.8 1.7 1.6 1.5 1.4 1.3 600 550 500 450 400 350 300 2.5 3.0 3.5 4.0 4.5 25°C 85°C 1.2 Input Voltage (V) Time (25μs/div) N-Channel RDS(ON) vs. Input Voltage Output Voltage (AC Coupled) (top) (V) 750 700 650 120°C 0.1 0.0 -0.1 -0.2 -0.3 -0.4 -0.5 -0.6 -0.7 Load Transient Response (30mA - 300mA; VIN = 3.6V; VOUT = 1.8V; C1 = 10μF; C4 = 100pF; see Figure 2) 1.4 1.2 300mA 30mA 1.0 0.8 0.6 0.4 0.2 0.0 -0.2 Load and Inductor Current (200mA/div) (bottom) RDS(ON) (mΩ) 600 550 500 450 400 350 300 2.5 3.0 3.5 25°C 85°C 100°C 4.0 4.5 5.0 5.5 6.0 Time (25μs/div) Input Voltage (V) 6 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter Typical Channel Characteristics Load Transient Response (30mA - 300mA; VIN = 3.6V; VOUT = 1.8V; C1 = 4.7μF) 2.0 1.9 1.4 1.2 1.0 300mA 30mA 0.8 0.6 0.4 0.2 0.0 -0.2 1.90 1.85 AAT2510 Line Transient (VOUT = 1.8V @ 400mA) 7.0 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 Load and Inductor Current (200mA/div) (bottom) Output Voltage (top) (V) 1.8 1.7 1.6 1.5 1.4 1.3 1.2 Output Voltage (top) (V) 1.80 1.75 1.70 1.65 1.60 1.55 1.50 Input Voltage (bottom) (V) Time (25μs/div) Time (25μs/div) Line Regulation (VOUT = 1.8V) 0.1 0.05 0 -0.05 -0.1 -0.15 -0.2 -0.25 -0.3 -0.35 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 4.0 3.0 2.0 1.0 0.0 -1.0 -2.0 -3.0 -4.0 Soft Start (VIN = 3.6V; VOUT = 1.8V; 400mA) Enable and Output Voltage (top) (V) 3.5 3.0 IOUT = 100mA IOUT = 10mA IOUT = 400mA Accuracy (%) Inductor Current (bottom) (A) 2.5 2.0 1.5 1.0 0.5 0.0 -0.5 Input Voltage (V) 250μs/div Output Ripple (VIN = 3.6V; VOUT = 1.8V; 400mA) Output Voltage (AC Coupled) (top) (mV) 40 20 0 -20 -40 -60 -80 -100 -120 0.9 0.8 Inductor Current (bottom) (A) 0.7 0.6 0.5 0.4 0.3 0.2 0.1 Time (250ns/div) 2510.2006.05.1.10 7 Dual 400mA, 1MHz Step-Down DC-DC Converter Functional Block Diagram FB1 VIN1 AAT2510 Err. Amp. Comp. DH Logic Voltage Reference Control Logic LX1 DL EN1 GND1 SGND1 FB2 See Note VIN2 Err. Amp. Comp. DH Logic Voltage Reference LX2 DL EN2 See Note Control Logic GND2 SGND2 Note: Internal resistor divider included for ≥1.2V versions. For low voltage versions, the feedback pin is tied directly to the error amplifier input. Operation Device Summary The AAT2510 is a constant frequency peak current mode PWM converter with internal compensation. Each channel has independent input, enable, feedback, and ground pins with non-synchronized 1MHz clocks. Both converters are designed to operate with an input voltage range of 2.7V to 5.5V. The output voltage ranges from 0.6V to the input voltage for the internally fixed version and up to 2.5V for the externally adjustable version. The 0.6V fixed model shown in Figure 1 is also the adjustable version and is externally programmable with a resistive divider as shown in Figure 2. The converter MOSFET power stage is sized for 400mA load capability with up to 96% efficiency. Light load efficiency exceeds 80% at a 500µA load. 8 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter VIN = 2.7 - 5.5V C3 10μF 12 1 AAT2510 VIN U1 AAT2510 VIN1 EN1 LX1 FB1 VIN2 EN2 LX2 FB2 9 4 8 5 6 7 C8 0.1μF C3 10μF 12 U1 AAT2510 VIN1 EN1 LX1 FB1 SGND1 GND1 VIN2 EN2 LX2 FB2 SGND2 GND2 9 4 8 5 6 7 1 C8 0.1μF 2.5V L2 10μH C5 100pF R3 187k C2 10μF 1.8V L2 4.7μH 1.8V at 400mA 2.5V at 400mA L1 4.7μH 11 2 3 L1 C4 100pF R1 118k 4.7μH 11 2 3 10 SGND1 SGND2 GND1 GND2 C1 4.7μF 10 C2 4.7μF C1 10μF R2 59.0k R4 59.0k L1, L2 Sumida CDRH3D16-4R7 C1, C2 Murata GRM219R61A475KE19 C3 Murata GRM21BR60J106KE19 Figure 1: AAT2510 Fixed Output. Figure 2: AAT2510 Adjustable Output with Enhanced Transient Response. Soft Start The AAT2510 soft-start control prevents output voltage overshoot and limits inrush current when either the input power or the enable input is applied. When pulled low, the enable input forces the converter into a low-power, non-switching state with a bias current of less than 1µA. MOSFET is turned on continuously for 100% duty cycle. At 100% duty cycle, the output voltage tracks the input voltage minus the I*R drop of the high side P-channel MOSFET RDS(ON). Low Supply The under-voltage lockout (UVLO) feature guarantees sufficient VIN bias and proper operation of all internal circuitry prior to activation. Low Dropout Operation For conditions where the input voltage drops to the output voltage level, the converter duty cycle increases to 100%. As 100% duty cycle is approached, the minimum off-time initially forces the high side on-time to exceed the 1MHz clock cycle and reduce the effective switching frequency. Once the input drops below the level where the output can be regulated, the high side P-channel Fault Protection For overload conditions, the peak inductor current is limited. Thermal protection disables switching when the internal dissipation or ambient temperature becomes excessive. The junction over-temperature threshold is 140°C with 15°C of hysteresis. 2510.2006.05.1.10 9 Dual 400mA, 1MHz Step-Down DC-DC Converter Applications Information Inductor Selection The step-down converter uses peak current mode control with slope compensation to maintain stability for duty cycles greater than 50%. The output inductor value must be selected so the inductor current down slope meets the internal slope compensation requirements. The internal slope compensation for the adjustable and low-voltage fixed versions of the AAT2510 is 0.24A/µsec. This equates to a slope compensation that is 75% of the inductor current down slope for a 1.5V output and 4.7µH inductor. AAT2510 Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive losses due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. The 4.7µH CDRH3D16 series inductor selected from Sumida has a 105mΩ DCR and a 900mA DC current rating. At full load, the inductor DC loss is 17mW which gives a 2.8% loss in efficiency for a 400mA 1.5V output. m= 0.75 ⋅ VO 0.75 ⋅ 1.5V A = = 0.24 L 4.7μH μsec Input Capacitor Select a 4.7µF to 10µF X7R or X5R ceramic capacitor for the input. To estimate the required input capacitor size, determine the acceptable input ripple level (VPP) and solve for C. The calculated value varies with input voltage and is a maximum when VIN is double the output voltage. This is the internal slope compensation for the adjustable (0.6V) version or low-voltage fixed version. When externally programming the 0.6V version to a 2.5V output, the calculated inductance would be 7.5µH. L= 0.75V 0.75 ⋅ VO μsec ≈ 3 A ⋅ VO = m 0.24A /μsec CIN = V⎞ VO ⎛ ⋅ 1- O VIN ⎝ VIN ⎠ ⎛ VPP ⎞ - ESR ⋅ FS ⎝ IO ⎠ μsec =3 ⋅ 2.5V = 7.5μH A In this case, a standard 10µH value is selected. For high-voltage fixed versions (2.5V and above), m = 0.48A/µsec. Table 1 displays inductor values for the AAT2510 fixed and adjustable options. This equation provides an estimate for the input capacitor required for a single channel. Configuration 0.6V Adjustable With External Resistive Divider Fixed Output Output Voltage 0.6V to 2.0V 2.5V 0.6V to 2.0V 2.5V to 3.3V Inductor 4.7µH 10µH 4.7µH 4.7µH Slope Compensation 0.24A/µsec 0.24A/µsec 0.24A/µsec 0.48A/µsec Table 1: Inductor Values. 10 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter The equation below solves for input capacitor size for both channels. It makes the worst-case assumptions that both converters are operating at 50% duty cycle and are synchronized. AAT2510 capacitor RMS current ripple are a maximum at 50% duty cycle. The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the AAT2510. Low ESR/ESL X7R and X5R ceramic capacitors are ideal for this function. To minimize the stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C3 and C8) can be seen in the evaluation board layout in Figure 4. Since decoupling must be as close to the input pins as possible, it is necessary to use two decoupling capacitors. C3 provides the bulk capacitance required for both converters, while C8 is a high frequency bypass capacitor for the second channel (see C3 and C8 placement in Figure 4). A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low ESR ceramic input capacitor, can create a high Q network that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short printed circuit board trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead inductance cannot be reduced to a level that does not affect converter performance, a high ESR tantalum or aluminum electrolytic capacitor should be placed in parallel with the low ESR, ESL bypass ceramic capacitor. This dampens the high Q network and stabilizes the system. CIN = 1 ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO1 + IO2 ⎠ Because the AAT2510 channels will generally operate at different duty cycles and are not synchronized, the actual ripple will vary and be less than the ripple (VPP) used to solve for the input capacitor in the equation above. Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the proper value. For example, the capacitance of a 10µF 6.3V X5R ceramic capacitor with 5V DC applied is actually about 6µF. The maximum input capacitor RMS current is: IRMS = IO1 · ⎛ ⎝ VO1 ⎛ V⎞ · 1 - O1 ⎞ + IO2 · ⎛ VIN ⎝ VIN ⎠ ⎠ ⎝ VO2 ⎛ V⎞ · 1 - O2 ⎞ VIN ⎝ VIN ⎠ ⎠ The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current of both converters combined. IRMS(MAX) = IO1(MAX) + IO2(MAX) 2 This equation also makes the worst-case assumption that both converters are operating at 50% duty cycle and are synchronized. Since the converters are not synchronized and are not both operating at 50% duty cycle, the actual RMS current will always be less than this. Losses associated with the input ceramic capacitor are typically minimal. The term VIN · ⎝1 - VIN ⎠ appears in both the input voltage ripple and input capacitor RMS current equations. It is a maximum when VO is twice VIN. This is why the input voltage ripple and the input VO ⎛ VO ⎞ Output Capacitor The output capacitor limits the output ripple and provides holdup during large load transitions. A 4.7µF to 10µF X5R or X7R ceramic capacitor typically provides sufficient bulk capacitance to stabi- 2510.2006.05.1.10 11 Dual 400mA, 1MHz Step-Down DC-DC Converter lize the output during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. The output voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current the ceramic output capacitor alone supplies the load current until the loop responds. As the loop responds, the inductor current increases to match the load current demand. This typically takes two to three switching cycles and can be estimated by: AAT2510 Adjustable Output Resistor Selection For applications requiring an adjustable output voltage, the 0.6V version can be programmed externally. Resistors R1 through R4 of Figure 2 program the output to regulate at a voltage higher than 0.6V. To limit the bias current required for the external feedback resistor string, the minimum suggested value for R2 and R4 is 59kΩ. Although a larger value will reduce the quiescent current, it will also increase the impedance of the feedback node, making it more sensitive to external noise and interference. Table 2 summarizes the resistor values for various output voltages with R2 and R4 set to either 59kΩ for good noise immunity or 221kΩ for reduced no load input current. ⎛ VOUT ⎞ ⎛ 1.5V ⎞ R1 = V -1 · R2 = 0.6V - 1 · 59kΩ = 88.5kΩ ⎝ REF ⎠ ⎝ ⎠ COUT = 3 · ΔILOAD VDROOP · FS Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the minimum output capacitor value to 4.7µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, and gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: · (VIN(MAX) - VOUT) V IRMS(MAX) = · OUT L · F · VIN(MAX) 2· 3 1 The adjustable version of the AAT2510 in combination with an external feedforward capacitor (C4 and C5 of Figure 2) delivers enhanced transient response for extreme pulsed load applications. The addition of the feedforward capacitor typically requires a larger output capacitor (C1 and C2) for stability. R2, R4 = 59kΩ VOUT (V) 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 R2, R4 = 221kΩ R1, R3 (kΩ) 75 113 150 187 221 261 301 332 442 464 523 715 R1, R3 (kΩ) 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 Dissipation due to the RMS current in the ceramic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot spot temperature. Table 2: Adjustable Resistor Values For Use With 0.6V Version. 12 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter Thermal Calculations There are three types of losses associated with the AAT2510 converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the RDS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified form of the dual converter losses is given by: AAT2510 Given the total losses, the maximum junction temperature can be derived from the θJA for the TDFN33-12 package which is 50°C/W. TJ(MAX) = PTOTAL · ΘJA + TAMB PCB Layout The following guidelines should be used to insure a proper layout. PTOTAL = IO12 · (RDSON(HS) · VO1 + RDSON(LS) · [VIN -VO1]) VIN IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN + + (tsw · F · [IO1 + IO2] + 2 · IQ) · VIN IQ is the AAT2510 quiescent current for one channel and tsw is used to estimate the full load switching losses. For the condition where channel one is in dropout at 100% duty cycle, the total device dissipation reduces to: PTOTAL = IO12 · RDSON(HS) IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN + + (tsw · F · IO2 + 2 · IQ) · VIN Since RDS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. 1. Due to the pin placement of VIN for both converters, proper decoupling is not possible with just one input capacitor. The large input capacitor C3 should connect as closely as possible to VP and GND, as shown in Figure 4. The additional input bypass capacitor C8 is necessary for proper high frequency decoupling of the second converter. 2. The output capacitor and inductor should be connected as closely as possible. The connection of the inductor to the LX pin should also be as short as possible. 3. The feedback trace should be separate from any power trace and connect as closely as possible to the load point. Sensing along a high-current load trace will degrade DC load regulation. If external feedback resistors are used, they should be placed as closely as possible to the FB pin. This prevents noise from being coupled into the high impedance feedback node. 4. The resistance of the trace from the load return to GND should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal signal ground and the power ground. 5. For good thermal coupling, PCB vias are required from the pad for the TDFN paddle to the ground plane. The via diameter should be 0.3mm to 0.33mm and positioned on a 1.2 mm grid. 2510.2006.05.1.10 13 Dual 400mA, 1MHz Step-Down DC-DC Converter Design Example Specifications VO1 = 2.5V @ 400mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VO2 = 1.8V @ 400mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VIN FS = 2.7V to 4.2V (3.6V nominal) = 1.0 MHz AAT2510 TAMB = 85°C 2.5V VO1 Output Inductor L1 = 3 μsec μsec ⋅ VO1 = 3 ⋅ 2.5V = 7.5μH A A (see Table 1) For Sumida inductor CDRH3D16, 10µH, DCR = 210mΩ. ⎛ 2.5V⎞ VO ⎛ V⎞ 2.5V ⋅ 1 - O1 = ⋅ ⎝1 = 100mA VIN ⎠ 10μH ⋅ 1.0MHz 4.2V⎠ L1 ⋅ F ⎝ ΔI1 = IPK1 = IO1 + ΔI1 = 0.4A + 0.05A = 0.45A 2 PL1 = IO12 ⋅ DCR = 0.4A2 ⋅ 210mΩ = 34mW 1.8V VO2 Output Inductor L2 = 3 μsec μsec ⋅ VO2 = 3 ⋅ 1.8V = 5.4μH (see Table 1) A A For Sumida inductor CDRH3D16, 4.7µH, DCR = 105mΩ. ⎛ 1.8V ⎞ VO2 ⎛ V⎞ 1.8V ⋅ 1 - O2 = ⋅ 1= 218mA VIN ⎠ 4.7μH ⋅ 1.0MHz ⎝ 4.2V⎠ L⋅F ⎝ ΔI2 = IPK2 = IO2 + ΔI2 = 0.4A + 0.11A = 0.51A 2 PL2 = IO22 ⋅ DCR = 0.4A2 ⋅ 105mΩ = 17mW 14 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter 2.5V Output Capacitor COUT = 3 · ΔILOAD 3 · 0.3A = = 4.5μF 0.2V · 1MHz VDROOP · FS (VOUT) · (VIN(MAX) - VOUT) 1 2.5V · (4.2V - 2.5V) · = 29mArms = L · F · VIN(MAX) 2 · 3 10μH · 1MHz · 4.2V 2· 3 1 · AAT2510 IRMS(MAX) = Pesr = esr · IRMS2 = 5mΩ · (29mA)2 = 4.2μW 1.8V Output Capacitor COUT = 3 · ΔILOAD 3 · 0.3A = = 4.5μF 0.2V · 1MHz VDROOP · FS (VOUT) · (VIN(MAX) - VOUT) 1 1.8V · (4.2V - 1.8V) · = 63mArms = L · F · VIN(MAX) 2 · 3 4.7μH · 1.0MHz · 4.2V 2· 3 1 · IRMS(MAX) = Pesr = esr · IRMS2 = 5mΩ · (63mA)2 = 20μW Input Capacitor Input Ripple VPP = 25mV. CIN = 1 ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO1 + IO2 ⎠ = 1 = 9.5μF ⎛ 25mV ⎞ - 5mΩ · 4 · 1MHz ⎝ 0.8A ⎠ IRMS(MAX) = IO1 + IO2 = 0.4Arms 2 P = esr · IRMS2 = 5mΩ · (0.4A)2 = 0.8mW 2510.2006.05.1.10 15 Dual 400mA, 1MHz Step-Down DC-DC Converter AAT2510 Losses AAT2510 The maximum dissipation occurs at dropout where VIN = 2.7V. All values assume an ambient temperature of 85°C and a junction temperature of 120°C. PTOTAL = IO12 · (RDSON(HS) · VO1 + RDSON(LS) · (VIN -VO1)) + IO22 · (RDSON(HS) · VO2 + RDSON(LS) · (VIN -VO2)) VIN + (tsw · F · IO2 + 2 · IQ) · VIN = 0.42 · (0.725Ω · 2.5V + 0.7Ω · (2.7V - 2.5V)) + 0.42 · (0.725Ω · 1.8V + 0.7Ω · (2.7V - 1.8V)) 2.7V + 5ns · 1MHz · 0.4A + 60μA) · 2.7V = 240mW TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 240mW = 97°C Output 1 Enable 123 VIN C41 R1 see Table 3 U1 AAT2510 1 2 3 LX1 12 11 10 9 8 7 EN1 FB1 SGND1 EN2 FB2 SGND2 VIN1 LX1 GND1 VIN2 LX2 GND2 L1 see Table 3 C3 10μF LX2 L2 see Table 3 C11 4.7μF C8 0.1μF C7 0.01μF C21 4.7μF GND C6 0.01μF VO2 VO1 C51 R3 see Table 3 4 5 6 R4 59.0k GND R2 59.0k 321 Output 2 Enable Figure 3: AAT2510 Evaluation Board Schematic. 1. For enhanced transient configuration C5, C4 = 100pF and C1, C2 = 10µF. 16 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter Adjustable Version (0.6V device) VOUT (V) 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 AAT2510 R2, R4 = 59kΩ R1, R3 (kΩ) 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 R2, R4 = 221kΩ1 R1, R3 (kΩ) 75.0 113 150 187 221 261 301 332 442 464 523 715 L1, L2 (µH) 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 or 6.8 10 Fixed Version VOUT (V) 0.6-3.3V R2, R4 Not Used R1, R3 (kΩ) 0 L1, L2 (µH) 4.7 Table 3: Evaluation Board Component Values. Figure 4: AAT2510 Evaluation Board Top Side. Figure 5: AAT2510 Evaluation Board Bottom Side. 1. For reduced quiescent current, R2 and R4 = 221kΩ. 2510.2006.05.1.10 17 Dual 400mA, 1MHz Step-Down DC-DC Converter Inductance (µH) 4.7 10 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 4.7 AAT2510 Manufacturer Sumida Sumida MuRata MuRata MuRata Coilcraft Coilcraft Coiltronics Coiltronics Coiltronics Coiltronics Part Number CDRH3D16-4R7 CDRH3D16-100 LQH32CN4R7M23 LQH32CN4R7M33 LQH32CN4R7M53 LPO6610-472 LPO3310-472 SDRC10-4R7 SDR10-4R7 SD3118-4R7 SD18-4R7 Max DC Current (A) 0.90 0.55 0.45 0.65 0.65 1.10 0.80 1.53 1.30 0.98 1.77 DCR (Ω) 0.11 0.21 0.20 0.15 0.15 0.20 0.27 0.117 0.122 0.122 0.082 Size (mm) LxWxH 3.8x3.8x1.8 3.8x3.8x1.8 2.5x3.2x2.0 2.5x3.2x2.0 2.5x3.2x1.55 5.5x6.6x1.0 3.3x3.3x1.0 4.5x3.6x1.0 5.7x4.4x1.0 3.1x3.1x1.85 5.2x5.2x1.8 Type Shielded Shielded Non-Shielded Non-Shielded Non-Shielded 1mm 1mm 1mm Shielded 1mm Shielded Shielded Shielded Table 4: Typical Surface Mount Inductors. Manufacturer MuRata MuRata MuRata Part Number GRM219R61A475KE19 GRM21BR60J106KE19 GRM21BR60J226ME39 Value 4.7µF 10uF 22uF Voltage 10V 6.3V 6.3V Temp. Co. X5R X5R X5R Case 0805 0805 0805 Table 5: Surface Mount Capacitors. 18 2510.2006.05.1.10 Dual 400mA, 1MHz Step-Down DC-DC Converter Ordering Information Package TDFN33-12 TDFN33-12 TDFN33-12 TDFN33-12 TDFN33-12 AAT2510 Voltage Channel 1 Channel 2 Marking1 OBXYY PNXYY PEXYY OTXYY QJXYY Part Number (Tape and Reel)2 AAT2510IWP-AA-T1 AAT2510IWP-AW-T1 AAT2510IWP-IE-T1 AAT2510IWP-IG-T1 AAT2510IWP-IH-T1 0.6V 0.6V 1.8V 1.8V 1.8V 0.6V 3.3V 1.2V 1.5V 1.6V All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. Legend Voltage Adjustable (0.6V) 0.9 1.2 1.5 1.8 1.9 2.5 2.6 2.7 2.8 2.85 2.9 3.0 3.3 4.2 Code A B E G I Y N O P Q R S T W C 1. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD. 2510.2006.05.1.10 19 Dual 400mA, 1MHz Step-Down DC-DC Converter Package Information TDFN33-12 AAT2510 Index Area (D/2 x E/2) Detail "B" 3.00 ± 0.05 2.40 ± 0.05 0.3 ± 0.10 0.16 0.375 ± 0.125 0.075 ± 0.075 0.1 REF Top View Bottom View Pin 1 Indicator (optional) 7.5° ± 7.5° + 0.05 0.8 -0.20 0.229 ± 0.051 0.05 ± 0.05 Option A: C0.30 (4x) max Chamfered corner Option B: R0.30 (4x) max Round corner Detail "B" Side View Detail "A" All dimensions in millimeters. © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service without notice. Customers are advised to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. AnalogicTech warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with AnalogicTech’s standard warranty. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders. Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 20 2510.2006.05.1.10 0.23 ± 0.05 0.45 ± 0.05 Detail "A" 3.00 ± 0.05 1.70 ± 0.05