AOZ1014
EZBuck™ 5A Simple Buck Regulator
General Description
Features
The AOZ1014 is a high efficiency, simple to use, 5A buck
regulator. The AOZ1014 works from a 4.5V to 16V input
voltage range, and provides up to 5A of continuous
output current with an output voltage adjustable down to
0.8V.
●
4.5V to 16V operating input voltage range
●
32mΩ internal PFET switch for high efficiency: up to
95%
●
Internal soft start
●
Output voltage adjustable to 0.8V
●
5A continuous output current
●
Fixed 500kHz PWM operation
●
Cycle-by-cycle current limit
The AOZ1014 comes in SO-8 and DFN-8 packages and
is rated over a -40°C to +85°C ambient temperature
range.
●
w
e
n
f or
●
Point of load DC/DC conversion
●
PCIe graphics cards
●
Set top boxes
●
DVD drives and HDD
●
LCD panels
●
Cable modems
●
Telecom/networking/datacom equipment
●
●
No
m
o
c
e
tr
m
s.
n
g
i
des
Short-circuit protection
Thermal shutdown
d
e
d
n
e
Applications
Small size SO-8 and DFN-8 packages
Typical Application
VIN
C1
22μF
C6
NC
VIN
From μPC
VOUT
3.3V
L1 3.3μF
U1
EN
AOZ1014
LX
C4
NU
COMP
RC
CC
R1
C2
10μF
FB
C5
1000pF
AGND
Rs
20Ω
GND
D1
C3
100μF
R2
10kΩ
Cs
1nF
Figure 1. 3.3V/5A Buck Down Regulator
Rev. 1.2 October 2009
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Page 1 of 19
AOZ1014
Ordering Information
Part Number
Ambient Temperature Range
Package
Environmental
AOZ1014AI*
-40°C to +85°C
SO-8
RoHS
AOZ1014AIL*
-40°C to +85°C
SO-8
Green
AOZ1014DI*
-40°C to +85°C
DFN-8
Green
* Not recommended for new designs. Replacement part is AOZ1094.
AOS Green Products use reduced levels of Halogens, and are also RoHS compliant.
Please visit www.aosmd.com/web/quality/rohs_compliant.jsp for additional information.
Pin Configuration
VIN
1
8
LX
PGND
2
7
LX
AGND
3
6
EN
VIN
FB
4
5
COMP
1
8
LX
7
LX
6
EN
5
COMP
LX
PGND
2
AGND
3
FB
4
AGND
SO-8
4x5 DFN
(Top View)
(Top View)
Pin Description
Pin Number
Pin Name
Pin Function
1
VIN
2
PGND
Power ground. Electrically needs to be connected to AGND.
3
AGND
Reference connection for controller section. Also used as thermal connection for controller
section. Electrically needs to be connected to PGND.
4
FB
The FB pin is used to determine the output voltage via a resistor divider between the output
and GND.
5
COMP
6
EN
The enable pin is active high. Connect EN pin to VIN if not used. Do not leave the EN pin floating.
7, 8
LX
PWM output connection to inductor. Thermal connection for output stage.
Rev. 1.2 October 2009
Supply voltage input. When VIN rises above the UVLO threshold the device starts up.
External loop compensation pin.
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Page 2 of 19
AOZ1014
Block Diagram
VIN
UVLO
& POR
EN
Internal
+5V
5V LDO
Regulator
OTP
+
ISen
–
Reference
& Bias
Softstart
Q1
ILimit
+
0.8V
+
EAmp
FB
–
–
PWM
Comp
PWM
Control
Logic
+
Level
Shifter
+
FET
Driver
LX
LX
COMP
500kHz
Oscillator
AGND
PGND
Absolute Maximum Ratings
Recommend Operating Ratings
Exceeding the Absolute Maximum ratings may damage the
device.
The device is not guaranteed to operate beyond the Maximum
Operating Ratings.
Parameter
Supply Voltage (VIN)
Rating
Parameter
18V
Supply Voltage (VIN)
LX to AGND
-0.7V to VIN+0.3V
Output Voltage Range
EN to AGND
-0.3V to VIN+0.3V
Ambient Temperature (TA)
FB to AGND
-0.3V to 6V
COMP to AGND
-0.3V to 6V
PGND to AGND
-0.3V to +0.3V
Junction Temperature (TJ)
+150°C
Storage Temperature (TS)
-65°C to +150°C
Rev. 1.2 October 2009
Package Thermal Resistance (ΘJA)(1)
SO-8
DFN-8
Rating
4.5V to 16V
0.8V to VIN
-40°C to +85°C
82°C/W
50°C/W
Note:
1. The value of ΘJA is measured with the device mounted on 1-in2 FR-4
board with 2oz. Copper, in a still air environment with TA = 25°C. The
value in any given application depends on the user's specific board
design.
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Page 3 of 19
AOZ1014
Electrical Characteristics
TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified(2)
Symbol
VIN
Parameter
Conditions
Supply Voltage
Min.
Typ.
4.5
Max.
Units
16
V
4.00
3.70
Input Under-Voltage Lockout Threshold
VIN Rising
VIN Falling
Supply Current (Quiescent)
IOUT = 0, VFB = 1.2V, VEN > 1.2V
2
3
mA
IOFF
Shutdown Supply Current
VEN = 0V
3
20
µA
VFB
Feedback Voltage
0.8
0.818
VUVLO
IIN
0.782
V
V
Load Regulation
0.5
%
Line Regulation
1
%
IFB
Feedback Voltage Input Current
VEN
EN Input Threshold
VHYS
EN Input Hysteresis
200
Off Threshold
On Threshold
0.6
2.0
nA
V
mV
100
MODULATOR
Frequency
350
DMAX
Maximum Duty Cycle
100
DMIN
Minimum Duty Cycle
fO
500
600
kHz
%
6
%
Error Amplifier Voltage Gain
500
V/V
Error Amplifier Transconductance
200
µA / V
PROTECTION
ILIM
Current Limit
Over-Temperature Shutdown Limit
tSS
6
TJ Rising
TJ Falling
Soft Start Interval
8
A
145
100
°C
4
ms
OUTPUT STAGE
High-Side Switch On-Resistance
VIN = 12V
VIN = 5V
25
41
32
55
mΩ
Note:
2. Specification in BOLD indicate an ambient temperature range of -40°C to +85°C. These specifications are guaranteed by design.
Rev. 1.2 October 2009
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Page 4 of 19
AOZ1014
Typical Performance Characteristics
Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified.
Light Load (DCM) Operation
Full Load (CCM) Operation
Vin
ripple
50mV/div
Vin
ripple
0.1V/div
Vo
ripple
50mV/div
Vo
ripple
50mV/div
IL
2A/div
IL
2A/div
VLX
10V/div
VLX
10V/div
1μs/div
1μs/div
Startup to Full Load
Full Load to Turnoff
Vin
5V/div
Vin
5V/div
Vo
1V/div
Vo
1V/div
lin
1A/div
lin
1A/div
1ms/div
1ms/div
50% to 100% Load Transient
Light Load to Turnoff
Vo
ripple
0.1V/div
Vin
5V/div
Vo
1V/div
lo
2A/div
100μs/div
Rev. 1.2 October 2009
lin
1A/div
1s/div
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Page 5 of 19
AOZ1014
Typical Performance Characteristics (Continued)
Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified.
Short Circuit Protection
Short Circuit Recovery
Vo
2V/div
Vo
2V/div
IL
2A/div
IL
2A/div
100μs/div
1ms/div
AOZ1014 Efficiency
Efficiency (VIN = 12V) vs. Load Current
100
8.0V OUTPUT
Efficieny (%)
95
5.0V OUTPUT
90
3.3V OUTPUT
85
80
75
0
0.5
1
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
Load Current (A)
Rev. 1.2 October 2009
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Page 6 of 19
AOZ1014
Thermal de-rating curves for SO-8 package part under typical input and output conditions. Circuit of Figure 1.
25°C ambient temperature and natural convection (air speed < 50LFM) unless otherwise specified.
Derating Curve at 5V Input
Derating Curve at 12V Input
6
6
1.8V OUTPUT
5.0V OUTPUT
4
3
2
1
0
25
1.8V OUTPUT
5
3.3V OUTPUT
Output Current (IO)
Output Current (IO)
5
4
5.0V OUTPUT
3.3V OUTPUT
8.0V OUTPUT
3
2
1
35
45
55
65
75
0
25
85
35
Ambient Temperature (TA)
45
55
65
75
85
Ambient Temperature (TA)
Thermal de-rating curves for DFN-8 package part under typical input and output conditions. Circuit of Figure 1.
25°C ambient temperature and natural convection (air speed < 50LFM) unless otherwise specified.
Derating Curve at 5V Input
Derating Curve at 12V Input
6
6
5
Output Current (IO)
Output Current (IO)
5
1.8V OUTPUT
3.3V OUTPUT
4
5.0V OUTPUT
3
2
1
0
25
8.0V OUTPUT
1.8V OUTPUT
4
3.3V OUTPUT
5.0V OUTPUT
3
2
1
35
45
55
65
75
85
0
25
Ambient Temperature (TA)
Rev. 1.2 October 2009
35
45
55
65
75
85
Ambient Temperature (TA)
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Page 7 of 19
AOZ1014
Detailed Description
The AOZ1014 is a current-mode step down regulator
with integrated high side PMOS switch and a low side
freewheeling Schottky diode. It operates from a 4.5V to
16V input voltage range and supplies up to 5A of load
current. The duty cycle can be adjusted from 6% to 100%
allowing a wide range of output voltage. Features include
enable control, Power-On Reset, input under voltage
lockout, fixed internal soft-start and thermal shut down.
The AOZ1014 is available in SO-8 and thermally
enhanced DFN-8 package.
Enable and Soft Start
The AOZ1014 has internal soft start feature to limit
in-rush current and ensure the output voltage ramps up
smoothly to regulation voltage. A soft start process
begins when the input voltage rises to 4.0V and voltage
on EN pin is HIGH. In soft start process, the output
voltage is ramped to regulation voltage in typically 4ms.
The 4ms soft start time is set internally.
The EN pin of the AOZ1014 is active high. Connect the
EN pin to VIN if enable function is not used. Pulling EN to
ground will disable the AOZ1014. Do not leave it open.
The voltage on EN pin must be above 2.0V to enable the
AOZ1014. When voltage on EN pin falls below 0.6V, the
AOZ1014 is disabled. If an application circuit requires the
AOZ1014 to be disabled, an open drain or open collector
circuit should be used to interface to EN pin.
Steady-State Operation
Under steady-state conditions, the converter operates
in fixed frequency and Continuous-Conduction Mode
(CCM).
The AOZ1014 integrates an internal P-MOSFET as the
high-side switch. Inductor current is sensed by amplifying
the voltage drop across the drain to source of the high
side power MOSFET. Output voltage is divided down by
the external voltage divider at the FB pin. The difference
of the FB pin voltage and reference is amplified by the
internal transconductance error amplifier. The error voltage, which shows on the COMP pin, is compared against
the current signal, which is sum of inductor current signal
and ramp compensation signal, at PWM comparator
input. If the current signal is less than the error voltage,
the internal high-side switch is on. The inductor current
flows from the input through the inductor to the output.
When the current signal exceeds the error voltage, the
high-side switch is off. The inductor current is freewheeling through the external Schottky diode to output.
Rev. 1.2 October 2009
The AOZ1014 uses a P-Channel MOSFET as the high
side switch. It saves the bootstrap capacitor normally
seen in a circuit which is using an NMOS switch. It allows
100% turn-on of the upper switch to achieve linear regulation mode of operation. The minimum voltage drop from
VIN to VO is the load current times DC resistance of
MOSFET plus DC resistance of buck inductor. It can be
calculated by equation below:
V O_MAX = V IN – I O × ( R DS ( ON ) + R inductor )
where;
VO_MAX is the maximum output voltage,
VIN is the input voltage from 4.5V to 16V,
IO is the output current from 0A to 5A,
RDS(ON) is the on resistance of internal MOSFET, the value is
between 25mΩ and 55mΩ depending on input voltage and
junction temperature, and
Rinductor is the inductor DC resistance.
Switching Frequency
The AOZ1014 switching frequency is fixed and set by an
internal oscillator. The practical switching frequency
could range from 350kHz to 600kHz due to device
variation.
Output Voltage Programming
Output voltage can be set by feeding back the output
to the FB pin with a resistor divider network. In the
application circuit shown in Figure 1. The resistor divider
network includes R1 and R2. Usually, a design is started
by picking a fixed R2 value and calculating the required
R1 with equation below:
R 1⎞
⎛
V O = 0.8 × ⎜ 1 + -------⎟
R 2⎠
⎝
Some standard values of R1 and R2 for the most commonly used output voltage values are listed in Table 1.
Table 1.
R1 (kΩ)
VO (V)
R2 (kΩ)
0.8
1.0
Open
1.2
4.99
10
1.5
10
11.5
1.8
12.7
10.2
2.5
21.5
10
3.3
31.6
10
5.0
52.3
10
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Page 8 of 19
AOZ1014
The combination of R1 and R2 should be large enough to
avoid drawing excessive current from the output, which
will cause power loss.
Thermal Protection
Since the switch duty cycle can be as high as 100%, the
maximum output voltage can be set as high as the input
voltage minus the voltage drop on upper PMOS and
inductor.
An internal temperature sensor monitors the junction
temperature. It shuts down the internal control circuit and
high side PMOS if the junction temperature exceeds
145°C. The regulator will restart automatically under the
control of soft-start circuit when the junction temperature
decreases to 100°C.
Protection Features
Application Information
The AOZ1014 has multiple protection features to prevent
system circuit damage under abnormal conditions.
The basic AOZ1014 application circuit is shown in
Figure 1. Component selection is explained below.
Input Capacitor
Over Current Protection (OCP)
The sensed inductor current signal is also used for
over current protection. Since the AOZ1014 employs
peak current mode control, the COMP pin voltage is
proportional to the peak inductor current. The COMP pin
voltage is limited to be between 0.4V and 2.5V internally.
The peak inductor current is automatically limited cycle
by cycle.
The cycle by cycle current limit threshold is set between
6A and 8A. When the load current reaches the current
limit threshold, the cycle by cycle current limit circuit turns
off the high side switch immediately to terminate the
current duty cycle. The inductor current stops rising. The
cycle by cycle current limit protection directly limits
inductor peak current. The average inductor current is
also limited due to the limitation on peak inductor current.
When the cycle by cycle current limit circuit is triggered,
the output voltage drops as the duty cycle is decreasing.
The AOZ1014 has internal short circuit protection to
protect itself from catastrophic failure under output short
circuit conditions. The FB pin voltage is proportional to
the output voltage. Whenever FB pin voltage is below
0.2V, the short circuit protection circuit is triggered.
As a result, the converter is shut down and hiccups at a
frequency equal to 1/8 of normal switching frequency.
The converter will start up via a soft start once the short
circuit condition disappears. In short circuit protection
mode, the inductor average current is greatly reduced
because of the low hiccup frequency.
Power-On Reset (POR)
The input capacitor must be connected to the VIN pin and
PGND pin of the AOZ1014 to maintain steady input voltage and filter out the pulsing input current. The voltage
rating of input capacitor must be greater than maximum
input voltage plus ripple voltage.
The input ripple voltage can be approximated by the
equation below:
VO ⎞ VO
IO
⎛
ΔV IN = ----------------- × ⎜ 1 – ---------⎟ × --------f × C IN ⎝
V IN⎠ V IN
Since the input current is discontinuous in a buck
converter, the current stress on the input capacitor is
another concern when selecting the capacitor. For a
buck circuit, the RMS value of input capacitor current can
be calculated by:
VO ⎛
VO ⎞
- ⎜ 1 – --------⎟
I CIN_RMS = I O × -------V IN ⎝
V IN⎠
If let m equal the conversion ratio:
VO
-------- = m
V IN
The relationship between the input capacitor RMS current and voltage conversion ratio is calculated and shown
in Figure 2 on the next page. It can be seen that when VO
is half of VIN, CIN is under the worst current stress. The
worst current stress on CIN is 0.5 x IO .
A power-on reset circuit monitors the input voltage.
When the input voltage exceeds 4V, the converter starts
operation. When input voltage falls below 3.7V, the
converter shuts down.
Rev. 1.2 October 2009
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Page 9 of 19
AOZ1014
The peak inductor current is:
0.5
ΔI L
I Lpeak = I O + -------2
0.4
High inductance gives low inductor ripple current but
requires a larger size inductor to avoid saturation. Low
ripple current reduces inductor core losses. Low ripple
current also reduces RMS current through the inductor
and switches, which results in less conduction loss.
Usually, peak to peak ripple current on inductor is
designed to be 20% to 30% of output current.
ICIN_RMS(m) 0.3
IO
0.2
0.1
0
0
0.5
m
1
Figure 2. ICIN vs. Voltage Conversion Ratio
For reliable operation and best performance, the input
capacitors must have current rating higher than ICIN_RMS
at the worst operating conditions. Ceramic capacitors are
preferred for input capacitors because of their low
ESR and high ripple current rating. Depending on the
application circuits, other low ESR tantalum capacitors
or aluminum electrolytic capacitors may also be used.
When selecting ceramic capacitors, X5R or X7R type
dielectric ceramic capacitors are preferred for their better
temperature and voltage characteristics. Note that the
ripple current rating from capacitor manufacturers is
based on certain amount of life time. Further de-rating
may be necessary for practical design requirement.
When selecting the inductor, make sure it is able to
handle the peak current at the highest operating temperature without saturation.
The inductor takes the highest current in a buck circuit.
The conduction loss on the inductor needs to be checked
for thermal and efficiency requirements.
Surface mount inductors in different shape and styles are
available from Coilcraft, Elytone and Murata. Shielded
inductors are small and radiate less EMI noise, but they
cost more than unshielded inductors. The choice
depends on EMI requirement, price and size.
Table 2 lists some inductors for typical output voltage
design.
Output Capacitor
Inductor
The inductor is used to supply constant current to the
output when it is driven by a switching voltage. For a
given input and output voltage, inductance and switching
frequency together decide the inductor ripple current,
which is:
VO ⎛
VO ⎞
-⎟
ΔI L = ----------- × ⎜ 1 – -------f×L ⎝
V IN⎠
The output capacitor is selected based on the DC output
voltage rating, output ripple voltage specification, and
ripple current rating.
The selected output capacitor must have a higher rated
voltage specification than the maximum desired output
voltage including ripple. De-rating needs to be considered for long term reliability.
Output ripple voltage specification is another important
factor for selecting the output capacitor. In a buck
converter circuit, output ripple voltage is determined by
Table 2. Typical Inductors
Vout
L1
Manufacture
5.0V
Shielded, 4.7µH, MSS1278-472MLD
Shielded, 4.7µH, MSS1260-472MLD
Coilcraft
3.3V
Un-shielded, 3.3µH, DO3316P-332MLD
Coilcraft
Shielded, 3.3µH, DO1260-332NXD
Coilcraft
Shielded, 3.3µH, ET553-3R3
ELYTONE
Shield, 2.2µH, ET553-2R2
ELYTONE
Un-shielded, 2.2µH, DO3316P-222MLD
Coilcraft
Shielded, 2.2µH, MSS1260-222NXD
Coilcraft
1.8V
Rev. 1.2 October 2009
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Coilcraft
Page 10 of 19
AOZ1014
inductor value, switching frequency, output capacitor
value and ESR. It can be calculated by the equation
below:
input voltage, and the current rating should be greater
than the maximum load current.
Loop Compensation
1
ΔV O = ΔI L × ⎛ ESR CO + -------------------------⎞
⎝
8×f×C ⎠
The AOZ1014 employs peak current mode control for
easy use and fast transient response. Peak current mode
control eliminates the double pole effect of the output
L&C filter. It greatly simplifies the compensation loop
design.
O
where,
CO is output capacitor value, and
ESRCO is the equivalent series resistance of the output
capacitor.
When low ESR ceramic capacitor is used as output
capacitor, the impedance of the capacitor at the switching frequency dominates. Output ripple is mainly caused
by capacitor value and inductor ripple current. The output
ripple voltage calculation can be simplified to:
1
ΔV O = ΔI L × ------------------------8×f×C
With peak current mode control, the buck power stage
can be simplified to be a one-pole and one-zero system
in frequency domain. The pole is dominant pole and can
be calculated by:
1
f P1 = ----------------------------------2π × C O × R L
O
If the impedance of ESR at switching frequency
dominates, the output ripple voltage is mainly decided
by capacitor ESR and inductor ripple current. The output
ripple voltage calculation can be further simplified to:
ΔV O = ΔI L × ESR CO
The zero is a ESR zero due to output capacitor and its
ESR. It is can be calculated by:
1
f Z1 = -----------------------------------------------2π × C O × ESR CO
where;
CO is the output filter capacitor,
For lower output ripple voltage across the entire operating temperature range, X5R or X7R dielectric type of
ceramic, or other low ESR tantalum capacitor or aluminum electrolytic capacitor may also be used as output
capacitors.
In a buck converter, output capacitor current is continuous. The RMS current of output capacitor is decided
by the peak to peak inductor ripple current. It can be
calculated by:
ΔI L
I CO_RMS = ---------12
Usually, the ripple current rating of the output capacitor
is a smaller issue because of the low current stress.
When the buck inductor is selected to be very small
and inductor ripple current is high, output capacitor could
be overstressed.
ESRCO is the equivalent series resistance of output capacitor.
The compensation design is actually to shape the
converter close loop transfer function to get desired gain
and phase. Several different types of compensation
networks can be used for AOZ1014. For most cases, a
series capacitor and resistor network connected to the
COMP pin sets the pole-zero and is adequate for a stable
high-bandwidth control loop.
In the AOZ1014, FB pin and COMP pin are the inverting
input and the output of internal transconductance error
amplifier. A series R and C compensation network
connected to COMP provides one pole and one zero.
The pole is:
G EA
f P2 = ------------------------------------------2π × C C × G VEA
where;
Schottky Diode Selection
The external freewheeling diode supplies the current to
the inductor when the high side PMOS switch is off. To
reduce the losses due to the forward voltage drop and
recovery of diode, Schottky diode is recommended to
use. The maximum reverse voltage rating of the chosen
Schottky diode should be greater than the maximum
Rev. 1.2 October 2009
RL is load resistor value, and
GEA is the error amplifier transconductance, which is 200 x 10-6
A/V,
GVEA is the error amplifier voltage gain, which is 500 V/V, and
CC is compensation capacitor.
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Page 11 of 19
AOZ1014
The zero given by the external compensation network,
capacitor CC and resistor RC ,is located at:
1
f Z2 = ----------------------------------2π × C C × R C
Table 3 lists the values for a typical output voltage design
when output is 44µF ceramics capacitor.
To design the compensation circuit, a target crossover
frequency fC for close loop must be selected. The system
crossover frequency is where control loop has unity gain.
The crossover frequency is also called the converter
bandwidth. Generally a higher bandwidth means faster
response to load transient. However, the bandwidth
should not be too high because of system stability
concerns. When designing the compensation loop,
converter stability under all line and load condition must
be considered.
Usually, it is recommended to set the bandwidth to be
less than 1/10 of switching frequency. The AOZ1014
operates at a fixed switching frequency range from
350kHz to 600kHz. The recommended crossover
frequency is less than 30kHz.
f C = 30kHz
The strategy for choosing RC and CC is to set the cross
over frequency with RC and set the compensator zero
with CC. Using selected crossover frequency, fC, to
calculate RC:
VO
2π × C O
R C = f C × ---------- × ----------------------------V FB G EA × G CS
where;
fC is the desired crossover frequency,
VFB is 0.8V,
GEA is the error amplifier transconductance, which is 200 x 10-6
A/V, and
GCS is the current sense circuit transconductance, which is
9.02 A/V.
The compensation capacitor CC and resistor RC together
make a zero. This zero is put somewhere close to the
dominate pole, fP1, but lower than 1/5 of the selected
crossover frequency. CC can is selected by:
1.5
C C = ----------------------------------2π × R C × f P1
The previous equation can also be simplified to:
CO × RL
C C = --------------------RC
Rev. 1.2 October 2009
An easy-to-use application software which helps to
design and simulate the compensation loop can be found
at www.aosmd.com.
Table 3.
VOUT
L1
RC
CC
1.8V
2.2µH
51.1kΩ
1.0nF
3.3V
3.3µH
20kΩ
1.0nF
5V
5.6µH
31.6kΩ
1.0nF
8V
10µH
49.9kΩ
1.0nF
Thermal Management and Layout
Consideration
In the AOZ1014 buck regulator circuit, high pulsing current flows through two circuit loops. The first loop starts
from the input capacitors, to the VIN pin, to the LX pins, to
the filter inductor, to the output capacitor and load, and
then returns to the input capacitor through ground.
Current flows in the first loop when the high side switch is
on. The second loop starts from inductor, to the output
capacitors and load, to the anode of Schottky diode, to
the cathode of Schottky diode. Current flows in the
second loop when the low side diode is on.
In PCB layout, minimizing the two loops area reduces the
noise of this circuit and improves efficiency. A ground
plane is strongly recommended to connect input capacitor, output capacitor, and PGND pin of the AOZ1014.
In the AOZ1014 buck regulator circuit, the two major
power dissipating components are the AOZ1014, the
Schottky diode, and output inductor. The total power
dissipation of converter circuit can be measured by input
power minus output power.
P total_loss = V IN × I IN – V O × I O
The power dissipation in Schottky can be approximately
calculated as:
P diode_loss = IO × ( 1 – D ) × V FW_Schottky
where;
VFW_Schottky is the Schottky diode forward voltage drop.
The power dissipation of inductor can be approximately
calculated by output current and DCR of inductor.
P inductor_loss = IO2 × R inductor × 1.1
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Page 12 of 19
AOZ1014
The actual junction temperature can be calculated
with power dissipation in the AOZ1014 and thermal
impedance from junction to ambient is:
T junction = ( P total_loss – P inductor_loss ) × Θ JA
The maximum junction temperature of AOZ1014 is
145°C, which limits the maximum load current capability.
Please see the thermal de-rating curves for maximum
load current of the AOZ1014 under different ambient
temperatures.
The thermal performance of the AOZ1014 is strongly
affected by the PCB layout. Extra care should be taken
by users during the design process to ensure that the IC
will operate under the recommended environmental
conditions.
Figure 3. AOZ1014 (SO-8) PCB Layout
Several layout tips are listed below for the best electric
and thermal performance. Figure 3 below illustrates a
PCB layout example as a reference.
1. Do not use thermal relief connection to the VIN and
the PGND pin. Pour a maximized copper area to
the PGND pin and the VIN pin to help thermal
dissipation.
2. Input capacitor should be connected to the VIN pin
and the PGND pin as close as possible.
3. A ground plane is preferred. If a ground plane is not
used, separate PGND from AGND and connect them
only at one point to avoid the PGND pin noise
coupling to the AGND pin.
Figure 4. AOZ1014 (DFN-8) PCB Layout
4. Make the current trace from LX pins to L to Co to the
PGND as short as possible.
5. Pour copper plane on all unused board area and
connect it to stable DC nodes, like VIN, GND, or
VOUT.
6. The two LX pins are connected to the internal PFET
drain. They are low resistance thermal conduction
path and most noisy switching node. Connect a
copper plane to the LX pin to help thermal
dissipation. This copper plane should not be too
large otherwise switching noise may be coupled to
other parts of the circuit.
7. Keep sensitive signal traces away from the LX pins.
8. For the DFN package, thermal pad must be soldered
to the PCB metal. When multiple layer PCB is used,
4 to 6 thermal vias should be placed on the thermal
pad and connected to PCB metal on other layers to
help thermal dissipation.
Rev. 1.2 October 2009
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Page 13 of 19
AOZ1014
Package Dimensions, SO-8L
D
Gauge Plane
Seating Plane
e
0.25
8
L
E
E1
h x 45°
1
C
θ
7° (4x)
A2 A
0.1
b
A1
Dimensions in millimeters
2.20
5.74
1.27
0.80
Unit: mm
Symbols
A
Min.
1.35
A1
A2
Dimensions in inches
Max.
1.75
0.25
1.65
Symbols
A
Min.
0.053
Nom.
0.065
Max.
0.069
0.10
1.25
Nom.
1.65
—
1.50
A1
A2
0.004
0.049
—
0.059
0.010
0.065
b
c
D
0.31
0.17
4.80
—
—
4.90
0.51
0.25
5.00
b
c
D
0.012
0.007
0.189
—
—
0.193
0.020
0.010
0.197
E1
e
E
3.80
3.90
4.00
1.27 BSC
0.150
h
L
0.25
0.40
6.00
—
—
6.20
0.50
1.27
E1
e
E
h
L
0.010
0.016
—
—
0.020
0.050
θ
0°
—
8°
θ
0°
—
8°
5.80
0.154 0.157
0.050 BSC
0.228 0.236 0.244
Notes:
1. All dimensions are in millimeters.
2. Dimensions are inclusive of plating
3. Package body sizes exclude mold flash and gate burrs. Mold flash at the non-lead sides should be less than 6 mils.
4. Dimension L is measured in gauge plane.
5. Controlling dimension is millimeter, converted inch dimensions are not necessarily exact.
Rev. 1.2 October 2009
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Page 14 of 19
AOZ1014
Tape and Reel Dimensions, SO-8L
SO-8 Carrier Tape
P1
D1
See Note 3
P2
T
See Note 5
E1
E2
E
See Note 3
B0
K0
A0
D0
P0
Feeding Direction
Unit: mm
Package
SO-8
(12mm)
A0
6.40
±0.10
B0
5.20
±0.10
K0
2.10
±0.10
D0
1.60
±0.10
D1
1.50
±0.10
E
12.00
±0.10
SO-8 Reel
E1
1.75
±0.10
E2
5.50
±0.10
P0
8.00
±0.10
P1
4.00
±0.10
P2
2.00
±0.10
T
0.25
±0.10
W1
S
G
N
M
K
V
R
H
W
N
Tape Size Reel Size
M
W
12mm
ø330
ø330.00 ø97.00 13.00
±0.10 ±0.30
±0.50
W1
17.40
±1.00
H
K
ø13.00
10.60
+0.50/-0.20
S
2.00
±0.50
G
—
R
—
V
—
SO-8 Tape
Leader/Trailer
& Orientation
Trailer Tape
300mm min. or
75 empty pockets
Rev. 1.2 October 2009
Components Tape
Orientation in Pocket
www.aosmd.com
Leader Tape
500mm min. or
125 empty pockets
Page 15 of 19
AOZ1014
Package Dimensions, DFN 5x4
D
A
Pin #1 IDA
D/2
B
e
1
L
E/2
R
aaa C
E
E3
E2
Index Area
(D/2 x E/2)
D2
aaa C
ccc C
A3
D3
L1
Seating C
Plane
A
ddd C
A1
b
bbb
CAB
Dimensions in millimeters
Recommended Land Pattern
2.125
1.775
0.6
2.7
0.8
2.2
0.5
0.95
Unit: mm
Symbols
A
Min.
0.80
A1
A3
0.00
b
D
Nom.
0.90
Dimensions in inches
Symbols
A
Min.
0.031
0.02
0.05
0.20 REF
A1
A3
0.000
0.001 0.002
0.008 REF
0.35
0.40
0.45
5.00 BSC
b
D
0.014
0.016 0.018
0.197 BSC
D2
D3
E
1.975
1.625
2.125 2.225
1.775 1.875
4.00 BSC
D2
D3
E
0.078
0.064
0.084 0.088
0.070 0.074
0.157 BSC
E2
E3
2.500
2.050
2.750
2.300
E2
E3
0.098
0.081
e
L
L1
0.600
0.400
0.95 BSC
0.700 0.800
0.500 0.600
e
L
L1
0.024
0.016
R
aaa
bbb
ccc
ddd
–
–
–
–
0.30 REF
0.15
0.10
0.10
0.08
R
aaa
bbb
ccc
ddd
–
–
–
–
2.650
2.200
Max.
1.00
–
–
–
–
Nom.
0.035
0.104
0.087
Max.
0.039
0.108
0.091
0.037 BSC
0.028 0.031
0.020 0.024
0.012 REF
0.006
0.004
0.004
0.003
–
–
–
–
Notes:
1. Dimensions and tolerancing conform to ASME Y14.5M-1994.
2. All dimensions are in millimeters.
3. The location of the terminal #1 identifier and terminal numbering convention conforms to JEDEC publication 95 SP-002.
4. Dimension b applies to metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. If the terminal has the
optional radius on the other end of the terminal, the dimension b should not be measured in that radius area.
5. Coplanarity applies to the terminals and all other bottom surface metallization.
6. Drawing shown are for illustration only.
Rev. 1.2 October 2009
www.aosmd.com
Page 16 of 19
AOZ1014
Tape Dimensions, DFN 5x4
Tape
R0
0.
.40
20
T
D1
E1
E2
D0
E
B0
Feeding
Direction
K0
P0
A0
Unit: mm
Package
A0
B0
K0
D0
D1
E
E1
E2
P0
P1
P2
T
DFN 5x4
(12 mm)
5.30
±0.10
4.30
±0.10
1.20
±0.10
1.50
Min.
Typ.
1.50
+0.10 / –0
12.00
±0.30
1.75
±0.10
5.50
±0.10
8.00
±0.10
4.00
±0.20
2.00
±0.10
0.30
±0.05
Leader/Trailer and Orientation
Trailer Tape
(300mm Min.)
Rev. 1.2 October 2009
Components Tape
Orientation in Pocket
www.aosmd.com
Leader Tape
(500mm Min.)
Page 17 of 19
AOZ1014
II
R1
59
Reel Dimensions, DFN 5x4
I
R1
6.0±1
21
M
R1
I
27
Zoom In
R6
R1
P
5
R5
B
W1
III
Zoom In
3-1.8
0.05
II
ø1
/4
3-ø1
.9
±0
"
A
ø2
.0
A A
N=ø100±2
3-
3-
/8"
Zoom In
ø9
6±
0.2
5
1.8
6.0
1.8
6.45±0.05
8.00
6.2
ø2
2.20
1.
8.9±0.1
14 REF
0.00
0
5.0
ø13.0
R1.10
R3.10
C
1.8
12 REF
11.90
ø86
.0±0
10°
41.5 REF
43.00
44.5±0.1
44.5±0.1
.95
R3
4.0
6.10
VIEW: C
3-
8.0±0.1
ø3
"
16
ø3
/
3-
38°
40°
10.0
EF
8R
46.0±0.1
R0.5
.1
3.3
6.50
R4
R1
2.00
ø9
20
ø17.0
A
0.00
-0.05
/1
2.00
6.50
0.80
3.00
2.5
1.80
+0.05
6"
8.000.00
10.71
6°
Rev. 1.2 October 2009
www.aosmd.com
Page 18 of 19
AOZ1014
AOZ1014 Package Marking
SO-8 Package
AOZ1014AI
AOZ1014AIL (Underlined, Halogen Free)
Z1014AI
FAYWLT
Z1014AI
Part Number
Code
Assembly Lot
Code
Fab & Assembly
Location
Part Number
Code
FAYWLT
Assembly Lot
Code
Fab & Assembly
Location
Year & Week Code
Year & Week Code
DFN-8 Package
Z1014DI
FAYWLT
Part Number Code
Assembly Lot Code
Fab & Assembly Location
Year & Week Code
This data sheet contains preliminary data; supplementary data may be published at a later date.
Alpha & Omega Semiconductor reserves the right to make changes at any time without notice.
LIFE SUPPORT POLICY
ALPHA & OMEGA SEMICONDUCTOR PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL
COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS.
As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant into
the body or (b) support or sustain life, and (c) whose
failure to perform when properly used in accordance
with instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of
the user.
Rev. 1.2 October 2009
2. A critical component in any component of a life
support, device, or system whose failure to perform can
be reasonably expected to cause the failure of the life
support device or system, or to affect its safety or
effectiveness.
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Page 19 of 19
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