HCPL-7800A/HCPL-7800
Isolation Amplifer
Datasheet
Lead (Pb) Free
RoHS 6 fully
compliant
RoHS 6 fully compliant options available;
-xxxE denotes a lead-free product
Description
Features
The HCPL-7800(A) isolation amplifier family was designed
for current sensing in electronic motor drives. In a typical
implementation, motor currents flow through an external
resistor and the resulting analog voltage drop is sensed
by the HCPL-7800(A). A differential output voltage is
created on the other side of the HCPL-7800(A) optical
isolation barrier. This differential output voltage is proportional to the motor current and can be converted to
a single-ended signal by using an op-amp as shown in
the recommended application circuit. Since commonmode voltage swings of several hundred volts in tens of
nanoseconds are common in modern switching inverter
motor drives, the HCPL-7800(A) was designed to ignore
very high common-mode transient slew rates (of at least
10 kV/µs).
• 15 kV/µs Common-Mode Rejection at VCM = 1000 V
• Compact, Auto-Insertable Standard 8-pin DIP Package
• 0.00025 V/V/°C Gain Drift vs. Temperature
• 0.3 mV Input Offset Voltage
• 100 kHz Bandwidth
• 0.004% Nonlinearity
• Worldwide Safety Approval: UL 1577 (3750 Vrms/1 min.)
and CSA, IEC/EN/DIN EN 60747-5-2
• Advanced Sigma-Delta (Σ−∆) A/D Converter Technology
The high CMR capability of the HCPL-7800(A) isolation
amplifier provides the precision and stability needed to
accurately monitor motor current in high noise motor
control environ-ments, providing for smoother control
(less “torque ripple”) in various types of motor control
applications.
The product can also be used for general analog signal
isolation applications requiring high accuracy, stability,
and linearity under similarly severe noise con-ditions.
For general applications, we recommend the HCPL-7800
(gain tolerance of ±3%). For precision applications Avago
Technologies offers the HCPL-7800A with part-to-part
gain tolerance of ±1%. The HCPL-7800(A) utilizes sigma
delta (Σ−∆) analog-to-digital converter technology,
chopper stabilized amplifiers, and a fully differential
circuit topology.
Together, these features deliver unequaled isolationmode noise rejection, as well as excellent offset and
gain accuracy and stability over time and temperature.
This performance is delivered in a compact, auto-insertable, industry standard 8-pin DIP package that meets
worldwide regulatory safety standards. (A gull-wing
surface mount option #300 is also available).
• Fully Differential Circuit Topology
Applications
• Motor Phase and Rail Current Sensing
• Inverter Current Sensing
• Switched Mode Power Supply Signal Isolation
• General Purpose Current Sensing and Monitoring
• General Purpose Analog Signal Isolation
Functional Diagram
IDD1
IDD2
VDD1
1
VIN+
2
+
+
7 VOUT+
VIN-
3
-
-
6 VOUT-
GND1 4
SHIELD
8 VDD2
5 GND2
NOTE: A 0.1 μF bypass capacitor must be connected
between pins 1 and 4 and between pins 5 and 8.
CAUTION: It is advised that normal static precautions be taken in handling and assembly
of this component to prevent damage and /or degradation which may be induced by ESD.
Ordering Information
HCPL-7800A/HCPL-7800 is UL Recognized with 3750 Vrms for 1 minute per UL1577.
Option
Part number
RoHS
Compliant
Non-RoHS
Compliant
-000E
No option
-300E
#300
-500E
#500
HCPL-7800A
HCPL-7800
Surface
Mount
Package
300 mil
DIP-8
Gull
Wing
X
X
X
X
Tape
& Reel
IEC/EN/DIN EN
60747-5-2
Quantity
X
X
50 per tube
X
50 per tube
X
1000 per reel
To order, choose a part number from the part number column and combine with the desired option from the option
column to form an order entry.
Example 1:
HCPL-7800A-500E to order product of Gull Wing Surface Mount package in Tape and Reel packaging with
IEC/EN/DIN EN 60747-5-2 Safety Approval in RoHS compliant.
Example 2:
HCPL-7800 to order product of 300 mil DIP package in tube packaging and non-RoHS compliant.
Option datasheets are available. Contact your Avago sales representative or authorized distributor for information.
Remarks: The notation ‘#XXX’ is used for existing products, while (new) products launched since 15th July 2001 and
RoHS compliant option will use ‘-XXXE’.
Package Outline Drawings
Standard DIP Package
9.80 ± 0.25
(0.386 ± 0.010)
8
7
6
DIMENSIONS IN MILLIMETERS AND (INCHES).
5
DATE CODE
A 7800
NOTE:
FLOATING LEAD PROTRUSION IS 0.5 mm (20 mils) MAX.
YYWW
1
1.19 (0.047) MAX.
3.56 ± 0.13
(0.140 ± 0.005)
2
3
4
7.62 ± 0.25
(0.300 ± 0.010)
1.78 (0.070) MAX.
6.35 ± 0.25
(0.250 ± 0.010)
4.70 (0.185) MAX.
0.51 (0.020) MIN.
2.92 (0.115) MIN.
1.080 ± 0.320
(0.043 ± 0.013)
0.65 (0.025) MAX.
5˚ TYP.
0.20 (0.008)
0.33 (0.013)
2.54 ± 0.25
(0.100 ± 0.010)
Note:
Initial or continued variation in the color of the HCPL-7800(A)’s white mold compound is normal and does not affect device performance or
reliability.
Gull Wing Surface Mount Option 300
LAND PATTERN RECOMMENDATION
9.80 ± 0.25
(0.386 ± 0.010)
8
7
6
1.016 (0.040)
5
A 7800
6.350 ± 0.25
(0.250 ± 0.010)
YYWW
1
2
3
10.9 (0.430)
4
2.0 (0.080)
1.27 (0.050)
9.65 ± 0.25
(0.380 ± 0.010)
1.780
(0.070)
MAX.
1.19
(0.047)
MAX.
7.62 ± 0.25
(0.300 ± 0.010)
0.20 (0.008)
0.33 (0.013)
3.56 ± 0.13
(0.140 ± 0.005)
1.080 ± 0.320
(0.043 ± 0.013)
0.635 ± 0.25
(0.025 ± 0.010)
2.54
(0.100)
BSC
DIMENSIONS IN MILLIMETERS (INCHES).
TOLERANCES (UNLESS OTHERWISE SPECIFIED):
0.635 ± 0.130
(0.025 ± 0.005)
xx.xx = 0.01
xx.xxx = 0.005
NOTE: FLOATING LEAD PROTRUSION IS 0.5 mm (20 mils) MAX.
12˚ NOM.
LEAD COPLANARITY
MAXIMUM: 0.102 (0.004)
Maximum Solder Reflow Thermal Profile
300
PREHEATING RATE 3˚C + 1˚C/–0.5˚C/SEC.
REFLOW HEATING RATE 2.5˚C ± 0.5˚C/SEC.
TEMPERATURE (˚C)
200
PEAK
TEMP.
245˚C
PEAK
TEMP.
240˚C
2.5˚C ± 0.5˚C/SEC.
SOLDERING
TIME
200˚C
30
SEC.
160˚C
150˚C
140˚C
PEAK
TEMP.
230˚C
30
SEC.
3˚C + 1˚C/–0.5˚C
100
PREHEATING TIME
150˚C, 90 + 30 SEC.
50 SEC.
TIGHT
TYPICAL
LOOSE
0
0
50
ROOM TEMPERATURE
100
150
TIME (SECONDS)
Note: Use of non-chlorine-activated fluxes is highly recommended.
Recommended Pb-Free IR Profile
TEMPERATURE (˚C)
tp
Tp
217 ˚C
TL
Tsmax
Tsmin
260 +0/-5 ˚C
TIME WITHIN 5 ˚C of ACTUAL
PEAK TEMPERATURE
20-40 SEC.
RAMP-UP
3 ˚C/SEC. MAX.
150 - 200 ˚C
ts
PREHEAT
60 to 180 SEC.
RAMP-DOWN
6 ˚C/SEC. MAX.
tL
60 to 150 SEC.
25
t 25 ˚C to PEAK
TIME (SECONDS)
NOTES:
THE TIME FROM 25 ˚C to PEAK TEMPERATURE = 8 MINUTES MAX.
Tsmax = 200 ˚C, Tsmin = 150 ˚C
Note: Use of non-chlorine-activated fluxes is highly recommended.
200
250
Regulatory Information
The HCPL-7800(A) has been approved by the following organizations:
IEC/EN/DIN EN 60747-5-2
UL
Approved under:
IEC 60747-5-2:1997 + A1:2002
EN 60747-5-2:2001 + A1:2002
DIN EN 60747-5-2 (VDE 0884 Teil 2): 2003-01.
Approved under UL 1577, component
recognition program up to VISO = 3750 Vrms.
CSA
Approved under CSA Component Acceptance
Notice #5, File CA 88324.
IEC/EN/DIN EN 60747-5-2 Insulation Characteristics[1]
Description
Symbol
Characteristic
Installation classification per DIN VDE 0110/1.89, Table 1
for rated mains voltage 300 Vrms
for rated mains voltage 600 Vrms
I-IV
I-III
Climatic Classification
55/100/21
Pollution Degree (DIN VDE 0110/1.89)
2
Maximum Working Insulation Voltage
Unit
VIORM
891
VPEAK
VPR
1670
VPEAK
VPR
1336
VPEAK
VIOTM
6000
VPEAK
Safety-limiting values—maximum values
allowed in the event of a failure.
Case Temperature
Input Current[3]
Output Power[3]
TS
IS,INPUT
PS,OUTPUT
175
400
600
°C
mA
mW
Insulation Resistance at TS, VIO = 500 V
RS
>109
Ω
Input to Output Test Voltage, Method b[2]
Input to Output Test Voltage, Method a[2]
VIORM x 1.5 = VPR, Type and Sample Test,
tm = 60 sec, Partial discharge < 5 pC
Highest Allowable Overvoltage
(Transient Overvoltage tini = 10 sec)
Notes:
1. Insulation characteristics are guaranteed only within the safety maximum ratings which
must be ensured by protective circuits within the application. Surface Mount Classification is
Class A in accordance with CECC00802.
2. Refer to the optocoupler section of the Isolation and Control Components Designer’s Catalog, under Product Safety Regulations section, (IEC/EN/DIN EN 60747-5-2) for a detailed description of Method a and Method b partial discharge test profiles.
3. Refer to the following figure for dependence of PS and IS on ambient temperature.
800
OUTPUT POWER - PS, INPUT CURRENT - IS
VIORM x 1.875 = VPR, 100% Production Test with
tm = 1 sec, Partial discharge < 5 pC
P S (mW)
700
I S (mA)
600
500
400
300
200
100
0
0
25
50
75
100
125
150
o
TA - CASE TEMPERATURE - C
175
200
Insulation and Safety Related Specifications
Parameter
Symbol
Value
Unit
Conditions
Minimum External Air Gap
(Clearance)
L(101)
7.4
mm
Measured from input terminals to output
terminals, shortest distance through air.
Minimum External Tracking
(Creepage)
L(102)
8.0
mm
Measured from input terminals to output
terminals, shortest distance path along body.
0.5
mm
Through insulation distance conductor to
conductor, usually the straight line distance
thickness between the emitter and detector.
>175
Volts
DIN IEC 112/VDE 0303 Part 1
Minimum Internal Plastic Gap
(Internal Clearance)
Tracking Resistance
(Comparative Tracking Index)
CTI
Isolation Group
III a
Material Group
(DIN VDE 0110, 1/89, Table 1)
Absolute Maximum Ratings
Parameter
Symbol
Min.
Max.
Unit
Storage Temperature
TS
-55
125
°C
Operating Temperature
TA
- 40
100
Supply Voltage
VDD1, VDD2
0
5.5
Steady-State Input Voltage
2 Second Transient Input Voltage
VIN+, VIN-
-2.0
-6.0
VDD1 +0.5
Output Voltage
VOUT
-0.5
VDD2 +0.5
Solder Reflow Temperature Profile
See Maximum Solder Reflow Thermal Profile Section
Note
V
Recommended Operating Conditions
Parameter
Symbol
Min.
Max.
Unit
Ambient Operating Temperature
TA
-40
85
°C
Supply Voltage
VDD1, VDD2
4.5
5.5
V
Input Voltage (accurate and linear)
VIN+, VIN-
-200
200
mV
Input Voltage (functional)
VIN+, VIN-
-2
2
V
Note
1
DC Electrical Specifications
Unless otherwise noted, all typicals and figures are at the nominal operating conditions of VIN+ = 0, VIN- = 0 V, VDD1 =
VDD2 = 5 V and TA = 25°C; all Min./Max. specifications are within the Recommended Operating Conditions.
Parameter
Symbol
Min.
Typ.
Max.
Unit
Test Conditions
Fig.
Input Offset Voltage
VOS
-2.0
0.3
2.0
mV
TA = 25°C
1,2
-3.0
Magnitude of Input Offset
Change vs. Temperature
|DVOS/DTA|
Gain (HCPL-7800A)
G1
Gain (HCPL-7800)
G3
3.0
-40°C < TA < +85°C,
-4.5 V < (VDD1, VDD2) < 5.5 V
3.0
10.0
µV/°C
7.92
8.00
8.08
V/V
7.76
8.00
8.24
Magnitude of VOUT
|DG/DTA|
Gain Change vs.Temperature
0.00025
VOUT 200 mV Nonlinearity
NL200
0.0037
Magnitude of VOUT
200 mV Nonlinearity
Change vs. Temperature
|dNL200/dT|
0.0002
VOUT 100 mV Nonlinearity
NL100
0.0027
Maximum Input Voltage
before VOUT Clipping
|VIN+|MAX
308.0
Input Supply Current
IDD1
10.86
16.0
Output Supply Current
IDD2
11.56
16.0
Input Current
IIN+
-0.5
5.0
Magnitude of Input
Bias Current vs.
Temperature Coefficient
|dIIN/dT|
0.45
nA/°C
Output Low Voltage
VOL
1.29
V
Output High Voltage
VOH
Output Common-Mode
Voltage
VOCM
Output Short-Circuit
Current
|IOSC|
18.6
mA
Equivalent Input Impedance
RIN
500
VOUT Output Resistance
ROUT
15
kW
W
Input DC Common-Mode
Rejection Ratio
CMRRIN
76
dB
2.545
2
4,5,6
3
%
4
-200 mV < VIN+ < 200 mV
7,8
5
% / °C
0.2
%
-100 mV < VIN+ < 100 mV
mV
3.80
2.2
-200 mV < VIN+ < 200 mV,
TA = 25°C,
3
V/V/°C
0.35
Note
mA
6
9
VIN+ = 400 mV
10
VIN+ = -400 mV
µA
7
8
11
9
10
V
2.8
V
11
12
AC Electrical Specifications
Unless otherwise noted, all typicals and figures are at the nominal operating conditions of VIN+ = 0, VIN- = 0 V, VDD1 =
VDD2 = 5 V and TA = 25°C; all Min./Max. specifications are within the Recommended Operating Conditions.
Parameter
Symbol
Min.
Typ.
Unit
Test Conditions
Fig.
VOUT Bandwidth
(-3 dB) sine wave.
BW
50
100
kHz
VIN+ = 200 mVpk-pk
12,13
VOUT Noise
NOUT
31.5
mVrms
VIN+ = 0.0 V
VIN to VOUT
Signal Delay
(50 – 10%)
tPD10
2.03
3.3
µs
14,15
VIN to VOUT
Signal Delay
(50 – 50%)
tPD50
3.47
5.6
VIN+ = 0 mV to 150 mV
step.
Measured at output of
MC34081 on Figure 15.
VIN to VOUT
Signal Delay
(50 – 90%)
tPD90
4.99
9.9
VOUT
Rise/ Fall Time
(10 – 90%)
tR/F
2.96
6.6
Common Mode
Transient
Immunity
CMTI
15.0
kV/µs
VCM = 1 kV, TA = 25°C
16
Power Supply
Rejection
PSR
170
mVrms
With recommended
application circuit.
10.0
Max.
Note
13
14
15
Package Characteristics
Parameter
Symbol
Min.
Input-Output
Momentary Withstand
Voltage
VISO
3750
Resistance
(Input-Output)
RI-O
Capacitance
(Input-Output)
CI-O
Typ.
Max.
Unit
Test Condition
Fig.
Note
Vrms
RH < 50%,
t = 1 min.
TA = 25°C
16,17
>109
Ω
VI-O = 500 VDC
18
1.2
pF
ƒ = 1 MHz
18
Notes:
General Note: Typical values represent the mean value of all characterization units at the nominal operating conditions. Typical drift
specifications are determined by calculating the rate of change of the
specified parameter versus the drift pa-rameter (at nominal operating conditions) for each characterization unit, and then averaging the
individual unit rates. The corresponding drift figures are normalized
to the nominal operating conditions and show how much drift occurs
as the par-ticular drift parameter is varied from its nominal value, with
all other parameters held at their nominal operating values. Note that
the typical drift specifications in the tables below may differ from the
slopes of the mean curves shown in the corresponding figures.
1. Avago Technologies recommends operation with VIN- = 0 V (tied to
GND1). Limiting VIN+ to 100 mV will improve DC nonlinearity and
nonlinearity drift. If VIN- is brought above VDD1 – 2 V, an internal test
mode may be activated. This test mode is for testing LED coupling
and is not intended for customer use.
2. This is the Absolute Value of Input Offset Change vs. Temperature.
3. Gain is defined as the slope of the best-fit line of differential output
voltage (VOUT+–VOUT- ) vs. differential input voltage (VIN+–VIN-) over
the specified input range.
4. This is the Absolute Value of Gain Change vs. Temperature.
5. Nonlinearity is defined as half of the peak-to-peak output deviation
from the best-fit gain line, expressed as a percentage of the fullscale differential output voltage.
6. NL100 is the nonlinearity specified over an input voltage range of
±100 mV.
7. The input supply current decreases as the differential input voltage
(VIN+–VIN-) decreases.
8. The maximum specified output supply current occurs when the
differential input voltage (VIN+–VIN-) = -200 mV, the maximum recommended operat-ing input voltage. However, the out-put supply current will continue to rise for differential input voltages up to
approximately -300 mV, beyond which the output supply current
remains constant.
9. Because of the switched-capacitor nature of the input sigma-delta
con-verter, time-averaged values are shown.
10. When the differential input signal exceeds approximately 308 mV,
the outputs will limit at the typical values shown.
11. Short circuit current is the amount of output current generated
when either output is shorted to VDD2 or ground.
12. CMRR is defined as the ratio of the differential signal
gain (signal applied differentially between pins 2 and 3)
to the common-mode gain (input pins tied together and the signal
applied to both inputs at the same time), expressed in dB.
13. Output noise comes from two primary sources: chopper noise
and sigma-delta quantization noise. Chopper noise results from
chopper stabilization of the output op-amps. It occurs at a specific
frequency (typically 400 kHz at room temperature), and is not attenuated by the internal output filter. A filter circuit can be easily
added to the external post-amplifier to reduce the total rms output
noise. The internal output filter does eliminate most, but not all, of
the sigma-delta quantization noise. The magnitude of the output
quantization noise is very small at lower frequencies (below 10 kHz)
and increases with increasing frequency.
14. CMTI (Common Mode Transient Immunity or CMR, Common Mode
Rejection) is tested by applying an exponentially rising/falling voltage step on pin 4 (GND1) with respect to pin 5 (GND2). The rise time
of the test waveform is set to approximately 50 ns. The amplitude
of the step is adjusted until the differential output (VOUT+–VOUT-)
exhibits more than a 200 mV deviation from the average output
voltage for more than 1µs. The HCPL-7800(A) will continue to function if more than 10 kV/µs common mode slopes are applied, as
long as the breakdown voltage limitations are observed.
15. Data sheet value is the differential amplitude of the transient at the
output of the HCPL-7800(A) when a 1 Vpk-pk, 1 MHz square wave
with 40 ns rise and fall times is applied to both VDD1 and VDD2.
16. In accordance with UL 1577, each optocoupler is proof tested by applying an insulation test voltage ≥4500 Vrms for 1 second (leakage
detection current limit, II-O ≤ 5 µA). This test is performed before
the 100% production test for partial discharge (method b) shown in
IEC/EN/DIN EN 60747-5-2 Insulation Characteristic Table.
17. The Input-Output Momentary Withstand Voltage is a dielectric
voltage rating that should not be interpreted as an input-output
continuous voltage rating. For the continuous voltage rating refer
to the IEC/EN/DIN EN 60747-5-2 insulation characteristics table and
your equipment level safety specification.
18. This is a two-terminal measurement: pins 1–4 are shorted together
and pins 5–8 are shorted together.
VDD1
VDD2
+15 V
0.1 µF
1
8
0.1 µF
2
10 K
7
+
HCPL-7800
0.1 µF
3
6
4
5
VOUT
10 K
-
0.47
µF
AD624CD
GAIN = 100
0.1 µF
0.47
µF
-15 V
Figure 1. Input Offset Voltage Test Circuit.
0.6
0.5
0.4
0.3
-25
5
35
65
95
125
vs. VDD1
0.38
8.03
vs. VDD2
0.37
G - GAIN - V/V
0.7
0.2
-55
8.035
0.39
VOS - INPUT OFFSET VOLTAGE - mV
VOS - INPUT OFFSET VOLTAGE - mV
0.8
0.36
0.35
TA - TEMPERATURE - ˚C
8.02
8.015
0.34
0.33
4.5
8.025
4.75
5.0
5.25
8.01
-55 -35 -15 5
5.5
Figure 2. Input Offset Voltage vs. Temperature.
Figure 3. Input Offset vs. Supply.
VDD1
VDD2
25 45 65 85 105 125
TA - TEMPERATURE - ¡C
VDD - SUPPLY VOLTAGE - V
Figure 4. Gain vs. Temperature.
+15 V
+15 V
0.1 µF
1
8
0.1 µF
VIN
404
0.1 µF
2
7
10 K
+
HCPL-7800
13.2
3
6
4
5
10 K
0.47
µF
+
VOUT
-
0.01 µF
AD624CD
GAIN = 4
AD624CD
GAIN = 10
0.1 µF
-15 V
10 K
0.47
µF
Figure 5. Gain and Nonlinearity Test Circuit.
-
0.1 µF
0.47
µF
-15 V
10
0.1 µF
8.032
0.03
0.005
NL - NONLINEARITY - %
vs. VDD1
8.026
vs. VDD2
8.024
4.5
4.75
5.0
5.25
0.02
0.015
0.01
0
-55
5.5
-25
IDD - SUPPLY CURRENT - mA
VO - OUTPUT VOLTAGE - V
2.6
1.8
VOP
VOR
0.1
0.3
95
0.002
4.5
125
7
IDD1
IDD2
-0.3
-0.1
0.1
0.3
100
1000
10000
FREQUENCY (Hz)
Figure 12. Gain vs. Frequency.
100000
-3
-4
-100
-150
-200
-300
-0.4
-0.2
0
0.2
0.4
0.6
Figure 11. Input Current vs. Input Voltage.
PD - PROPAGATION DELAY - µS
PHASE - DEGREES
-50
-250
10
-2
5.5
0
-3
5.5
VIN - INPUT VOLTAGE - V
50
-2
5.25
-1
-5
-0.6
0.5
Figure 10. Supply Current vs. Input Voltage.
0
5.0
Figure 8. Nonlinearity vs. Supply.
VIN - INPUT VOLTAGE - V
1
-1
4.75
VDD - SUPPLY VOLTAGE - V
10
4
-0.5
0.5
Figure 9. Output Voltage vs. Input Voltage.
GAIN - dB
65
0
VIN - INPUT VOLTAGE - V
11
35
13
3.4
-4
5
Figure 7. Nonlinearity vs. Temperature.
4.2
-0.1
vs. VDD1
TA - TEMPERATURE - ¡C
Figure 6. Gain vs. Supply.
-0.3
0.003
vs. VDD2
VDD - SUPPLY VOLTAGE - V
1.0
-0.5
0.004
0.005
IIN - INPUT CURRENT - µA
G - GAIN - V/V
8.028
NL - NONLINEARITY - %
0.025
8.03
10
100
1000
10000
FREQUENCY (Hz)
Figure 13. Phase vs. Frequency.
100000
4.7
Tpd 10
Tpd 50
Tpd 90
Trise
3.9
3.1
2.3
1.5
-55
-25
5
35
65
95
125
TA - TEMPERATURE - ˚C
Figure 14. Propagation Delay vs. Temperature.
10 K
VDD1
VDD2
+15 V
0.1 µF
1
0.1 µF
8
0.1 µF
2
VIN
2K
7
HCPL-7800
0.01 µF
3
6
4
5
VOUT
2K
+
MC34081
0.1 µF
10 K
-15 V
VIN IMPEDANCE LESS THAN 10 Ω.
Figure 15. Propagation Delay Test Circuits.
10 K
150 pF
VDD2
78L05
+15 V
IN OUT
0.1
µF
0.1
µF
1
0.1 µF
8
0.1 µF
2
2K
7
HCPL-7800
9V
3
6
4
5
-
VCM
Figure 16. CMTI Test Circuits.
12
VOUT
+
MC34081
0.1 µF
10 K
150
pF
PULSE GEN.
+
2K
-15 V
Application Information
Power Supplies and Bypassing
The recommended supply con-nections are shown in
Figure 17. A floating power supply (which in many applications could be the same supply that is used to drive
the high-side power transistor) is regulated to 5 V using
a simple zener diode (D1); the value of resistor R4 should
be chosen to supply sufficient current from the existing
floating supply. The voltage from the current sensing
resistor (Rsense) is applied to the input of the HCPL7800(A) through an RC anti-aliasing filter (R2 and C2).
Although the application circuit is relatively simple, a few
recommendations should be followed to ensure optimal
performance.
The power supply for the HCPL -7800(A) is most often
obtained from the same supply used to power the
power transistor gate drive circuit. If a dedicated supply
is required, in many cases it is possible to add an additional winding on an existing transformer. Otherwise,
some sort of simple isolated supply can be used, such as
a line powered transformer or a high-frequency DC-DC
converter.
An inexpensive 78L05 three-terminal regulator can also
be used to reduce the floating supply voltage to 5 V. To
help attenuate high-frequency power supply noise or
ripple, a resistor or inductor can be used in series with
the input of the regulator to form a low-pass filter with
the regulator’s input bypass capacitor.
+
HV+
GATE DRIVE
CIRCUIT
FLOATING
POWER
SUPPLY
***
-
D1
5.1 V
C1
0.1 µF
R2
39 Ω
MOTOR
***
+ R1 RSENSE
***
HV-
Figure 17. Recommended Supply and Sense Resistor Connections.
13
C2
0.01 µF
HCPL-7800
As shown in Figure 18, 0.1 µF bypass capacitors (C1, C2)
should be located as close as possible to the pins of the
HCPL-7800(A). The bypass capacitors are required because
of the high-speed digital nature of the signals inside the
HCPL-7800(A). A 0.01 µF bypass capacitor (C2) is also recommended at the input due to the switched-capacitor
nature of the input circuit. The input bypass capacitor
also forms part of the anti-aliasing filter, which is recommended to prevent high-frequency noise from aliasing
down to lower frequencies and interfering with the input
signal. The input filter also performs an important reliability function—it reduces transient spikes from ESD events
flowing through the current sensing resistor.
PC Board Layout
The design of the printed circuit board (PCB) should follow
good layout practices, such as keeping bypass capacitors
close to the supply pins, keeping output signals away
from input signals, the use of ground and power planes,
etc. In addition, the layout of the PCB can also affect the
isolation transient immunity (CMTI) of the HCPL-7800(A),
due primarily to stray capacitive coupling between the
input and the output circuits. To obtain optimal CMTI
performance, the layout of the PC board should minimize
any stray coupling by maintaining the maximum possible
distance between the input and output sides of the circuit
and ensuring that any ground or power plane on the PC
board does not pass directly below or extend much wider
than the body of the HCPL-7800(A).
POSITIVE
FLOATING
SUPPLY
HV+
C5
150 pF
GATE DRIVE
CIRCUIT
R3
***
10.0 K
U1
78L05
IN
+5 V
+15 V
C8
0.1 µF
OUT
C1
C2
0.1
µF
0.1
µF
R5
68
1
8
2
7
C4
0.1 µF
C3
0.01
µF
U2
6
3
R1
2.00 K
R2
U3
+ MC34081
2.00 K
MOTOR
***
+
-
4
5
RSENSE
HCPL-7800
***
HV-
Figure 18: Recommended Application Circuit.
C2
R5
C4
C3
TO VDD1
TO RSENSE+
TO RSENSE-
Figure 19. Example Printed Circuit Board Layout.
14
TO VDD2
VOUT+
VOUT-
C7
C6
150 pF
R4
10.0 K
0.1 µF
-15 V
VOUT
Current Sensing Resistors
The current sensing resistor should have low resistance (to
minimize power dissipation), low inductance (to minimize
di/dt induced voltage spikes which could adversely affect
operation), and reasonable tolerance (to maintain overall
circuit accuracy). Choosing a particular value for the
resistor is usually a compro-mise between minimizing
power dissipation and maximizing accu-racy. Smaller
sense resistance decreases power dissipation, while larger
sense resistance can improve circuit accuracy by utilizing
the full input range of the HCPL -7800(A).
MOTOR OUTPUT POWER - HORSEPOWER
The first step in selecting a sense resistor is determining
how much current the resistor will be sensing. The graph
in Figure 20 shows the RMS current in each phase of a
three-phase induction motor as a function of average
motor output power (in horsepower, hp) and motor
drive supply voltage. The maximum value of the sense
re-sistor is determined by the current being measured
and the maxi-mum recommended input voltage of the
isolation amplifier. The maximum sense resistance can
be calculated by taking the maxi-mum recommended
input voltage and dividing by the peak current that the
sense resistor should see during normal operation. For
example, if a motor will have a maximum RMS current
of 10 A and can experience up to 50% overloads during
normal op-eration, then the peak current is 21.1 A (=10 x
1.414 x 1.5). Assuming a maximum input voltage of 200
mV, the maximum value of sense resistance in this case
would be about 10 mΩ.
40
440 V
380 V
220 V
120 V
35
30
25
20
15
5
0
5
10
15
20 25
30 35
MOTOR PHASE CURRENT - A (rms)
Figure 20. Motor Output Horsepower vs. Motor Phase Current and Supply
The maximum average power dissipation in the sense
resistor can also be easily calculated by multiplying the
sense resistance times the square of the maximum RMS
current, which is about 1 W in the previous example. If
the power dissipation in the sense resistor is too high, the
resistance can be decreased below the maximum value
to decrease power dissipation. The minimum value of the
sense resistor is limited by precision and accuracy requirements of the design. As the resistance value is reduced,
the output voltage across the resistor is also reduced,
which means that the offset and noise, which are fixed,
15
When sensing currents large enough to cause significant
heating of the sense resistor, the temperature coefficient
(tempco) of the resistor can introduce nonlinearity due to
the signal dependent temperature rise of the resistor. The
effect increases as the resistor-to-ambient thermal resistance increases. This effect can be minimized by reducing
the thermal resistance of the current sensing resistor or
by using a resistor with a lower tempco. Lowering the
thermal resistance can be accomplished by repositioning
the current sensing resistor on the PC board, by using
larger PC board traces to carry away more heat, or by
using a heat sink.
For a two-terminal current sensing resistor, as the value
of resistance decreases, the re-sistance of the leads
become a significant percentage of the total resistance.
This has two primary effects on resistor accuracy. First,
the effective resistance of the sense resistor can become
dependent on factors such as how long the leads are, how
they are bent, how far they are inserted into the board,
and how far solder wicks up the leads during assembly
(these issues will be discussed in more detail shortly).
Second, the leads are typically made from a material, such
as copper, which has a much higher tempco than the
material from which the resistive element itself is made,
resulting in a higher tempco overall.
Both of these effects are eliminated when a four-terminal
current sensing resistor is used. A four- terminal resistor
has two additional terminals that are Kelvin-connected
directly across the resistive element itself; these two
terminals are used to monitor the voltage across the
resistive element while the other two terminals are used
to carry the load current. Because of the Kelvin connection,
any voltage drops across the leads carrying the load current
should have no impact on the measured voltage.
10
0
become a larger percentage of the signal amp-litude. The
selected value of the sense resistor will fall somewhere
between the minimum and maximum values, depending
on the particular requirements of a specific design.
When laying out a PC board for the current sensing
resistors, a couple of points should be kept in mind. The
Kelvin connections to the resistor should be brought
together under the body of the resistor and then run very
close to each other to the input of the HCPL-7800(A); this
minimizes the loop area of the connection and reduces
the possibility of stray magnetic fields from interfering
with the measured signal. If the sense resistor is not
located on the same PC board as the HCPL-7800(A) circuit,
a tightly twisted pair of wires can accomplish the same
thing.
Also, multiple layers of the PC board can be used to
increase current carrying capacity. Numerous platedthrough vias should surround each non-Kelvin terminal of
the sense resistor to help distribute the current between
the layers of the PC board. The PC board should use 2 or
4 oz. copper for the layers, resulting in a current carrying
capacity in excess of 20 A. Making the current carrying
traces on the PC board fairly large can also improve the
sense resistor’s power dissipation capability by acting as a
heat sink. Liberal use of vias where the load current enters
and exits the PC board is also recommended.
Output Side
Note: Please refer to Avago Technologies Application Note 1078 for
additional information on using Isolation Amplifiers.
In addition, the op-amp should also have enough
bandwidth and slew rate so that it does not adversely
affect the response speed of the overall circuit. The postamplifier circuit includes a pair of capacitors (C5 and C6)
that form a single-pole low-pass filter; these capacitors
allow the bandwidth of the post-amp to be adjusted
independently of the gain and are useful for reducing
the output noise from the isola-tion amplifier. Many
different op-amps could be used in the circuit, including:
MC34082A (Motorola), TLO32A, TLO52A, and TLC277
(Texas Instruments), LF412A (National Semiconductor).
Sense Resistor Connections
The recommended method for connecting the HCPL7800(A) to the current sensing resistor is shown in Figure
18. VIN+ (pin 2 of the HPCL-7800(A)) is connected to the
positive terminal of the sense resistor, while VIN- (pin
3) is shorted to GND1 (pin 4), with the power-supply
return path functioning as the sense line to the negative
terminal of the current sense resistor. This allows a single
pair of wires or PC board traces to connect the HCPL7800(A) circuit to the sense resistor. By referencing the
input circuit to the negative side of the sense resistor,
any load current induced noise transients on the resistor
are seen as a common-mode signal and will not interfere
with the current-sense signal. This is important because
the large load currents flowing through the motor drive,
along with the parasitic inductances inherent in the
wiring of the circuit, can generate both noise spikes and
offsets that are relatively large compared to the small
voltages that are being measured across the current
sensing resistor.
If the same power supply is used both for the gate
drive circuit and for the current sensing circuit, it is very
important that the connection from GND1 of the HCPL7800(A) to the sense resistor be the only return path for
supply current to the gate drive power supply in order
to eliminate potential ground loop problems. The only
direct connection between the HCPL-7800(A) circuit
and the gate drive circuit should be the positive power
supply line.
16
The op-amp used in the external post-amplifier circuit
should be of sufficiently high precision so that it does not
contribute a significant amount of offset or offset drift
relative to the contribution from the isolation amplifier.
Generally, op-amps with bipolar input stages exhibit
better offset performance than op-amps with JFET or
MOSFET input stages.
The gain-setting resistors in the post-amp should have a
tolerance of 1% or better to ensure adequate CMRR and
adequate gain toler-ance for the overall circuit. Resistor
networks can be used that have much better ratio tolerances than can be achieved using discrete resistors. A
resistor network also reduces the total number of components for the circuit as well as the required board space.
FREQUENTLY ASKED QUESTIONS ABOUT
THE HCPL-7800(A)
1. THE BASICS
1.1: Why should I use the HCPL-7800(A) for sensing current when Hall-effect sensors are available which don’t
need an isolated supply voltage?
Available in an auto-insertable, 8-pin DIP package, the
HCPL-7800(A) is smaller than and has better linearity,
offset vs. temperature and Common Mode Rejection
(CMR) performance than most Hall-effect sensors. Additionally, often the required input-side power supply
can be derived from the same supply that powers the
gate-drive optocoupler.
2. SENSE RESISTOR AND INPUT FILTER
2.1: Where do I get 10 mΩ resistors? I have never seen one
that low.
Although less common than values above 10 Ω, there
are quite a few manufacturers of resistors suitable for
measuring currents up to 50 A when combined with
the HCPL-7800(A). Example product information may be
found at Dale’s web site (http://www.vishay.com/vishay/
dale) and Isotek’s web site (http://www.isotekcorp.com).
2.2: Should I connect both inputs across the sense resistor
instead of grounding VIN- directly to pin 4?
This is not necessary, but it will work. If you do, be sure
to use an RC filter on both pin 2 (VIN+) and pin 3 (VIN-) to
limit the input voltage at both pads.
2.3: Do I really need an RC filter on the input? What is it
for? Are other values of R and C okay?
The input anti-aliasing filter (R=39 Ω, C=0.01 µF) shown
in the typical application circuit is recommended for
filtering fast switching voltage transients from the input
signal. (This helps to attenuate higher signal frequencies
which could otherwise alias with the input sampling rate
and cause higher input offset voltage.)
Some issues to keep in mind using different filter resistors
or capacitors are:
1. Filter resistor: Input bias current for pins 2 and 3: This
is on the order of 500 nA. If you are using a single
filter resistor in series with pin 2 but not pin 3 the IxR
drop across this resistor will add to the offset error of
the device. As long as this IR drop is small compared
to the input offset voltage there should not be a
problem. If larger-valued resistors are used in series,
it is better to put half of the resistance in series with
pin 2 and half the resistance in series with pin 3. In
this case, the offset voltage is due mainly to resistor
mismatch (typically less than 1% of the resistance
design value) multiplied by the input bias.
17
2. Filter resistor: The equivalent input resistance for
HCPL-7800(A) is around 500 kΩ. It is therefore best
to ensure that the filter resistance is not a significant
percentage of this value; otherwise the offset voltage
will be increased through the resistor divider effect.
[As an example, if Rfilt = 5.5 kΩ, then VOS = (Vin * 1%)
= 2 mV for a maximum 200 mV input and VOS will
vary with respect to Vin.]
3. The input bandwidth is changed as a result of this
different R-C filter configuration. In fact this is one
of the main reasons for changing the input-filter R-C
time constant.
4. Filter capacitance: The input capacitance of the
HCPL-7800(A) is approximately 1.5 pF. For proper
operation the switching input-side sampling
capacitors must be charged from a relatively fixed
(low impedance) voltage source. Therefore, if a filter
capacitor is used it is best for this capacitor to be a
few orders of magnitude greater than the CINPUT (A
value of at least 100 pF works well.)
2.4: How do I ensure that the HCPL-7800(A) is not destroyed as a result of short circuit conditions which cause
voltage drops across the sense resistor that exceed the ratings of the HCPL-7800(A)’s inputs?
Select the sense resistor so that it will have less than 5 V
drop when short circuits occur. The only other requirement is to shut down the drive before the sense resistor
is damaged or its solder joints melt. This ensures that the
input of the HCPL-7800(A) can not be damaged by sense
resistors going open-circuit.
3. ISOLATION AND INSULATION
3.1: How many volts will the HCPL-7800(A) withstand?
The momentary (1 minute) withstand voltage is 3750 V
rms per UL 1577 and CSA Component Acceptance Notice
#5.
4. ACCURACY
4.1: Can the signal to noise ratio be improved?
Yes. Some noise energy exists beyond the 100 kHz
bandwidth of the HCPL-7800(A). Additional filtering
using different filter R,C values in the post-amplifier
application circuit can be used to improve the signal
to noise ratio. For example, by using values of R3 = R4
= 10 kΩ, C5 = C6 = 470 pF in the application circuit
the rms output noise will be cut roughly by a factor of
2. In applications needing only a few kHz bandwidth
even better noise performance can be obtained. The
noise spectral density is roughly 500 nV/š Hz below
20 kHz (input referred).
4.2: Does the gain change if the internal LED light output
degrades with time?
No. The LED is used only to transmit a digital pattern.
Avago Technologies has accounted for LED degradation
in the design of the product to ensure long life.
5. POWER SUPPLIES AND START-UP
5.1: What are the output voltages before the input side
power supply is turned on?
VO+ is close to 1.29 V and VO- is close to 3.80 V. This is
equivalent to the output response at the condition that
LED is completely off.
5.2: How long does the HCPL-7800(A) take to begin working properly after power-up?
Within 1 ms after VDD1 and VDD2 powered the device
starts to work. But it takes longer time for output to settle
down completely. In case of the offset measurement
while both inputs are tied to ground there is initially VOS
adjustment (about 60 ms). The output completely settles
down in 100 ms after device powering up.
6. MISCELLANEOUS
6.1: How does the HCPL-7800(A) measure negative signals
with only a +5 V supply?
The inputs have a series resistor for protection against
large negative inputs. Normal signals are no more than
200 mV in amplitude. Such signals do not forward bias
any junctions sufficiently to interfere with accurate
operation of the switched capacitor input circuit.
For product information and a complete list of distributors, please go to our web site: www.avagotech.com
Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies Limited in the United States and other countries.
Data subject to change. Copyright © 2005-2008 Avago Technologies Limited. All rights reserved. Obsoletes 5989-2161EN
AV02-0410EN - May 26, 2008
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