AL8871Q
60V BUCK-BOOST LED CONTROLLER
Description
Pin Assignments
The AL8871Q is an LED driver, controller IC for driving external
MOSFETs to drive high-current LEDs. The buck-boost topology
controller enables it to efficiently control the current through seriesconnected LEDs. The 60V capability enables it to be used in a wide
range of applications and drive in excess of 15 LEDs in series.
(Top View)
The AL8871Q is a modified hysteretic controller using a patentpending control scheme providing high-output current accuracy.
High-accuracy dimming is achieved through DC control and highfrequency PWM control.
CTRL
1
16
GI
REF
2
15
PWM
NTC
3
14
FAULT
COMP
4
13
CS
EP
The AL8871Q uses two pins for fault diagnosis. A flag output
highlights a fault, while the multi-level status pin gives further
information on the exact fault.
STATUS
5
12
VIN
SGND
6
11
BST
PGND
7
10
DRV
N/C
8
9
N/C
TSSOP-16EP
The AL8871Q has been qualified to AEC-Q100 Grade 1 and is
automotive compliant supporting PPAPs.
Features
Applications
•
•
•
•
•
•
AEC-Q100 Grade 1 Qualified
Wide Input Voltage Range: 5V to 60V
Operating Frequency Up to 1MHz
Analog Dimming Range: 10% to 100%
1000:1 PWM Dimming Resolution at 500 Hz
High Temperature Control of LED Current Using TCTRL
•
•
•
•
•
•
•
•
•
•
Fault Reporting for Abnormal Operations
Overtemperature Shutdown
Available in Thermally Enhanced TSSOP-16EP Package
Totally Lead-Free & Fully RoHS Compliant (Notes 1 & 2)
Halogen- and Antimony-Free. “Green” Device (Note 3)
For automotive applications requiring specific change
control (i.e. parts qualified to AEC-Q100/101/200, PPAP
capable, and manufactured in IATF 16949 certified
facilities), please contact us or your local Diodes
representative. https://www.diodes.com/quality/productdefinitions
Notes:
Automotive Daytime Running Lights
Automotive Head Lamps
Automotive Fog Lamps
Automotive Interior Lamps
1. No purposely added lead. Fully EU Directive 2002/95/EC (RoHS), 2011/65/EU (RoHS 2) & 2015/863/EU (RoHS 3) compliant.
2. See https://www.diodes.com/quality/lead-free/ for more information about Diodes Incorporated’s definitions of Halogen- and Antimony-free, "Green" and
Lead-free.
3. Halogen- and Antimony-free "Green” products are defined as those which contain 12V, DRV is clamped internally to prevent it
exceeding 15V. Below 12V, the minimum DRV pin voltage is 2.5V below VBST.
15. DRV is switched to PGND by an NMOS transistor.
16. If tON exceeds tSTALL, the device forces DRV low to turn off the external switch and then initiate a restart cycle. During this phase, CTRL is
grounded internally, and the COMP pin is switched to its nominal operating voltage before operation is allowed to resume. Restart cycles are
repeated automatically until the operating conditions are such that normal operation can be sustained. If tOFF exceeds tSTALL, the switch remains
off until normal operation is possible.
AL8871Q
Document number: DS42987 Rev. 2 - 2
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AL8871Q
Typical Characteristics
3
Supply Current (mA)
2.5
2
1.5
1
0.5
0
5
10
15
20
25
30
35
40
45
50
55
60
20
35
50
65
80
Junction Temperature (°C)
95
110
125
Supply Voltage (V)
1.252
Reference Voltage (V)
1.251
1.250
1.249
1.248
1.247
-40
-25
-10
5
Duty Cycle L=33µH Rs=150mΩ R9=120kΩ R10=36kΩ
100%
1 LED
4 LEDs
7 LEDs
10 LEDs
13 LEDs
16 LEDs
Duty Cycle
80%
2 LEDs
5 LEDs
8 LEDs
11 LEDs
14 LEDs
3 LEDs
6 LEDs
9 LEDs
12 LEDs
15 LEDs
60%
40%
20%
0%
0
AL8871Q
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5
10
15
20
25
30 35
VIN (V)
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40
45
50
55
60
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AL8871Q
Typical Characteristics—Analog/PWM Dimming
1400
LED Current
300
LED Current (mA)
250
1200
1000
200
800
Switching
Frequency
150
600
100
400
TA = 25°C
VAUX = VIN = 24V
8 LEDs, GI = 0.23
L = 33µH, RS = 300mΩ
50
0
0
0.25
0.5
0.75
ADJ Voltage (V)
1
Switching Frequency (kHz)
350
200
0
1.25
1500
LED Current (mA)
1250
1000
750
500
TA = 25°C fPWM = 100Hz
VIN = VAUX = 24V
L = 33µH, RS = 150mΩ
250
0
0%
20%
40%
60%
PWM Duty Cycle
80%
100%
1000
1250
Typical Characteristics—Thermal Dimming
LED Current Dimming Factor
100%
80%
60%
40%
20%
0%
0
AL8871Q
Document number: DS42987 Rev. 2 - 2
250
500
750
TADJ Pin Voltage [mV]
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AL8871Q
Typical Characteristics (RS = 150mΩ, ILED = 350mA, GIRATIO = 0.23)
0.45
TA = 25°C, VAUX = VIN
L = 33µH,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
LED Current (A)
0.40
0.35
0.30
4 LEDs
6 LEDs
8 LEDs
10 LEDs
12 LEDs
14 LEDs
16 LEDs
0.25
0.20
0.15
5
6
7
8
9
10
11
12
VIN (V)
13
14
15
16
17
18
700
9 LEDs
8 LEDs
7 LEDs
6 LEDs
5 LEDs
Switching Frequency (kHz)
600
500
4 LEDs
2/3 LEDs
1 LED
400
300
200
TA = 25°C, VAUX = VIN
L = 33µH,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
100
0
5
8
11
Vin (V)
14
17
20
90%
5 LEDs
85%
6 LEDs
80%
Efficiency
3 LEDs
9 LEDs
75%
4 LEDs
2 LEDs
70%
1 LED
65%
7 LEDs
60%
TA = 25°C, VAUX = VIN
L = 33µH,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
8 LEDs
55%
50%
5
AL8871Q
Document number: DS42987 Rev. 2 - 2
8
11
Vin (V)
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17
20
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AL8871Q
Typical Characteristics (RS = 150mΩ, ILED = 350mA, GIRATIO = 0.23)
0.45
0.43
TA = 25°C, VAUX = VIN,
8 LEDs,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
L = 33µH
0.41
LED Current (A)
0.39
L = 68µH
0.37
0.35
0.33
L = 100µH
0.31
0.29
0.27
0.25
5
6
7
8
9
10
11
12
VIN (V)
13
14
15
16
17
18
500
450
L = 33µH
Switching Frequency (kHz)
400
350
L = 68µH
300
250
200
L = 100µH
150
TA = 25°C, VAUX = VIN
8 LEDs,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF 8 LEDs
L = 68µH
100
50
0
5
6
7
8
9
10
11
12
VIN (V)
13
14
15
16
17
18
17
18
100%
L = 100µH
90%
Efficiency
80%
70%
TA = 25°C, VAUX = VIN
8 LEDs,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
60%
50%
40%
5
AL8871Q
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8
9
10
11
12
VIN (V)
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14
15
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AL8871Q
Typical Characteristics (8 LEDs, L = 33uH, GIRATIO = 0.23)
0.60
0.55
ILED = 500mA
LED Current (A)
0.50
0.45
0.40
ILED = 350mA
0.35
0.30
TA = 25°C, VAUX = VIN
8 LEDs,
L = 33µH,
8 LEDs
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
0.25
0.20
ILED = 150mA
0.15
0.10
5
6
7
8
9
10
11
12 13
VIN (V)
14
15
16
17
18
19
20
800
ILED = 150mA
Switching Frequency (kHz)
700
600
500
ILED = 350mA
400
300
ILED = 500mA
TA = 25°C, VAUX = VIN
8 LEDs,
L = 33µH
R9 = 120kΩ, R10 = 36kΩ
CIN = 100µF
200
100
0
5
6
7
8
9
10
11
12
13
14
15
16
17
18
VIN (V)
100%
90%
Efficiency
80%
ILED = 150mA
ILED = 500mA
70%
TA = 25°C, VAUX = VIN
8 LEDs,
L = 33µH,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF
60%
50%
ILED = 350mA
40%
5
AL8871Q
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7
8
9
10
11
12
VIN (V)
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14
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AL8871Q
Typical Characteristics—Bootstrap Performance
0.45
0.43
TA = 25°C, L = 33µH
RS = 150mΩ, R9 = 120kΩ
R10 = 36kΩ, VAUX = VIN
Output=8LEDs
0.41
0.39
ILED (A)
0.37
0.35
0.33
0.31
Without bootstrap
0.29
With bootstrap
0.27
0.25
5
6
7
8
9
10
800
11 12
VIN (V)
13
15
16
17
18
Without bootstrap
700
With bootstrap
600
Switching Frequency (kHz)
14
500
400
300
TA = 25°C, L = 33µH
RS = 150mΩ, R9 = 120kΩ
R10 = 36kΩ, VAUX = VIN
Output=8LEDs
200
100
0
5
6
7
8
9
10
100%
11 12 13
VIN (V)
14
15
16
17
18
90%
80%
Efficiency(%)
70%
Without bootstrap
60%
With bootstrap
50%
40%
TA = 25°C, L = 33µH
RS = 150mΩ, R9 = 120kΩ
R10 = 36kΩ, VAUX = VIN
Output=8LEDs
30%
20%
10%
0%
5
AL8871Q
Document number: DS42987 Rev. 2 - 2
6
7
8
9
10
11 12
VIN (V)
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14
15
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AL8871Q
Application Information
The AL8871Q is a high-accuracy, hysteretic-inductive buck/boost/buck-boost controller designed to be used with an external NMOS switch for
current-driving single or multiple series-connected LEDs. The device can be configured to operate in buck, boost, or buck-boost modes by suitable
configuration of the external components as shown in the schematics.
Device Description
Resistor Rs connected between the two inputs of a current monitor within the control loop block senses the coil current. An output from the controlloop drives the input of a comparator, which drives the gate of the external NMOS switch transistor Q1 via the internal gate driver. When the
switch is on, the drain voltage of Q1 is near zero. Current flows from VIN, via Rs, coil, and switch to ground. This current ramps up until an upper
threshold value is reached (see Figure 3). At this point DRV goes low, the switch is turned off, and the drain voltage increases to the load voltage
VLEDS plus the forward voltage of D1 plus VIN.
+11~15V typ.
tOFF
Gate Voltage
tON
0V
VLEDS+VF+VIN
Q1 Drain
Voltage
0V
Ipk
Inductor
Current
0A
Sense
Voltage
VIN-VISM
Mean=225mV*GI_ADJ/(1-D)
LED Current
0A
Figure 3. Operating Waveforms
Current flows via Rs, coil, D1, and LED back to VIN. When the coil current ramps down to a lower threshold value, DRV goes high, the switch is
turned on again, and the cycle of events repeats, which results in continuous oscillation.
The feedback loop adjusts the NMOS switch duty cycle to stabilize the LED current in response to changes in external conditions, including input
voltage and load voltage. Loop compensation is achieved by a single external capacitor C2 connected between COMP and SGND. Note that in
reality, a load capacitor COUT is used, so the LED current waveform shown is smoothed.
The average current in the sense resistor and coil, IRS, is equal to the average of the maximum and minimum threshold currents, and the ripple
current (hysteresis) is equal to the difference between the thresholds.
The average current in the LED, ILED, is always less than IRS. The feedback control loop adjusts the switch duty cycle, D, to achieve a set point at
the sense resistor. This controls IRS. During the interval tOFF, the coil current flows through D1 and the LED load. During tON, the coil current flows
through Q1—not the LEDs. Therefore, the set point is modified by D using a gating function to control ILED indirectly. In order to compensate
internally for the effect of the gating function, a control factor GI_CTRL is used. GI_CTRL is set by a pair of external resistors RGI1 (R10) and RGI2
(R9) (see Figure 4). This allows the sense voltage to be adjusted to an optimum level for power efficiency without significant error in the LED
controlled current.
GI_CTRL = �
AL8871Q
Document number: DS42987 Rev. 2 - 2
RGI1
RGI1 +RGI2
�
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Equation 1
August 2020
© Diodes Incorporated
AL8871Q
Application Information
(continued)
The control loop sets the duty cycle, so the sense resistor current is:
IRS = �
0.225
RS
��
GI_CTRL
1-D
��
VCTRL
VREF
�
Equation 2
IRS equals the coil current. The coil is only connected to the switch and the Schottky diode. The Schottky diode passes the LED current; therefore,
the average LED current is the coil current multiplied by the Schottky diode duty cycle, 1-D.
ILED = IRS �1-D� = �
0.225
RS
� GI_CTRL �
VCTRL
VREF
�
Equation 3
This shows that the LED current depends on the CTRL pin voltage, the reference voltage, and three resistor values (RS, RGI1, and RGI2). It is
independent of the input and output voltages.
If the CTRL pin is connected to the REF pin, it simplifies to:
ILED = �
0.225
RS
� GI_CTRL
Equation 4
Now ILED is dependent only on the three resistor values.
Considering power dissipation and accuracy, it is useful to know how the mean sense voltage varies with input voltage and other parameters.
VRS = IRS RS = 0.225
This shows that the sense voltage varies with duty cycle.
�
GI_CTRL
1-D
� �
VCTRL
VREF
�
Equation 5
Application Circuit Design
External component selection is driven by the characteristics of the load and the input supply because this will determine the kind of topology is
used for the system. Component selection begins with the current setting procedure, the inductor/frequency setting, and the MOSFET selection.
Finally after selecting the freewheeling diode and the output capacitor (if required), the application section covers the PWM dimming and thermal
feedback. The full procedure is greatly accelerated by the web calculator spreadsheet, which includes fully automated component selection and is
available on the Diodes website; however, the full calculation is also given here.
Please note the following particular feature of the web calculator. The GI ratio can be set for automatic calculation, or it can be fixed at a chosen
value. When optimizing a design, it is best first to optimize for the chosen voltage range of most interest using the automatic setting. In order to
subsequently evaluate performance of the circuit over a wider input voltage range, fix the GI ratio in the calculator input field, and set the desired
input voltage range.
Some components depend upon the switching frequency and the duty cycle. The switching frequency is regulated by the AL8871Q to a large
extent depending upon conditions. This is discussed later in this document when dealing with coil selection.
Duty Cycle Calculation
The duty cycle is a function of the input and output voltages. Approximately, the MOSFET switching duty cycle is:
D≈
VOUT
VOUT +VIN
Equation 6
Because D must always be a positive number less than 1, these equations show that VOUT > or = or < VIN. This allows topology selection for the
required voltage range. The more exact equation used in the web calculator is:
Where:
D≈
VOUT +VF +(IIN +IOUT )(RS +RCOIL )
VOUT +VIN +VF −VDSON
Equation 7
VF = Schottky diode forward voltage, estimated for the expected coil current, ICOIL
VDSON = MOSFET drain source voltage in the ON condition (dependent on RDSON and drain current = ICOIL)
RCOIL = DC winding resistance of L1
AL8871Q
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AL8871Q
Application Information
(continued)
The additional terms are relatively small, so the exact equations only make a significant difference at lower operating voltages at the input and
output, for example, low input voltage or a small number of LEDs connected in series. The estimates of VF and VDSON depend on the coil current.
The mean coil current, ICOIL is calculated as follows:
ICOIL = IIN + ILED
Equation 8
ILED is the target LED current and is already known. IIN is calculated with some accuracy later but can be estimated now from the electrical power
efficiency. If the expected efficiency is roughly 90%, the output power POUT is 90% of the input power, PIN, and the coil current is estimated as
follows:
POUT ≈ 0.9 PIN or ILED × N × VLED ≈ 0.9 IIN × VIN
Where N is the number of LEDs connected in series, and VLED is the forward voltage drop of a single LED at ILED.
So,
IIN ≈
ILED ×𝑁×VLED
0.9VIN
Equation 9
Equation 9 can now be used to find ICOIL in Equation 8, which can then be used to estimate the small terms in Equation 7. This completes the
calculation of duty cycle.
An initial estimate of duty cycle is required before choosing a coil. In Equation 7, the following approximations are recommended:
VF = 0.5V
IIN × (RS + RCOIL) = 0.5V
IOUT × (RS + RCOIL) = 0.5V
VDSON = 0.1V
(IIN + IOUT)(RS + RCOIL) = 1.1V
Then Equation 7 becomes:
Setting the LED Current
D≈
VOUT +1.6
VOUT +VIN +0.4
Equation 7a
The LED current requirement determines the choice of the sense resistor Rs. This also depends on the voltage on the CTRL pin and the voltage
on the GI pin according to the topology required.
The CTRL pin can be connected directly to the internal 1.25V reference (VREF) to define the nominal 100% LED current. The CTRL pin can also
be driven with an external DC voltage between 125mV and 1.25V to adjust the LED current proportionally between 10% and 100% of the nominal
value.
The divider ratio GI_CTRL is set less than 0.65V typically for optimized operation. This 0.65V threshold varies in proportion to VCTRL.
CTRL and GI are high-impedance inputs within their normal operating voltage ranges. An internal 1.3V clamp protects the device against
excessive input voltage and limits the maximum output current to approximately 4% above the maximum current set by VREF if the maximum
input voltage is exceeded.
AL8871Q
Document number: DS42987 Rev. 2 - 2
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AL8871Q
Application Information
(continued)
The LED current depends upon the resistors, RS, RGI1, and RGI2 as in
Equations 1 and 3. There is more than one degree of freedom. That is to say,
there is not a unique solution. From Equation 3,
RS
VIN
RS =
�
0.225
ILED
� GI_CTRL �
VCTRL
VREF
�
Equation 10
CTRL
If CTRL is connected to REF, this becomes:
RS = �
0.225
ILED
� GI_CTRL
CS
REF
RGI2
GI
Equation 11
RGI1
SGND
GI_CTRL is given by Equation 1, repeated here for convenience:
RGI1
GI_CTRL = �
�
RGI1 +RGI2
Figure 4. Setting LED Current
Note that from considerations of AL8871Q input bias current, the recommended limits for RGI1 are:
22kΩ < RGI1 < 100kΩ
Equation 12
The additional degree of freedom allows the selection of GI_CTRL within limits, but this can affect overall performance a little. As mentioned
above, the working voltage range at the GI pin is restricted. The permitted range of GI_CTRL is
0.2 < GI_CTRL < 0.5
Equation 13
The mean voltage across the sense resistor is
VRS = ICOIL RS
Equation 14
Note that if GI_CTRL is made larger, these equations show that RS is increased and VRS is increased. Therefore, for the same coil current, the
dissipation in RS is increased. So, in some cases, it is better to minimize GI_CTRL. However, consider Equation 5. If CTRL is connected to REF,
this becomes:
GI_CTRL
VRS = 0.225 �
�
1-D
This shows that VRS becomes smaller than 225mV if GI_CTRL < 1 - D. If D is also small, VRS can become too small. For example if D = 0.2, and
GI_CTRL is the minimum value of 0.2, VRS becomes 0.225 × 0.2 / 0.8 = 56.25 mV. This increases the LED current error due to small offsets in
the system, such as mV drop in the copper printed wiring circuit, or offset uncertainty in the AL8871Q. Now if GI_CTRL is increased to 0.4 or 0.5,
VRS is increased to a value greater than 100mV.
This gives small enough ILED error for most practical purposes. Satisfactory operation will be obtained if VRS is more than about 80mV. This
means GI_CTRL should be greater than (1-DMIN) × 80/225 = (1- DMIN) × 0.355.
There is also a maximum limit on VRS, which gives a maximum limit for GI_CTRL. If VRS exceeds approximately 300mV, or 133% of 225mV, the
STATUS output can indicate an overcurrent condition. This will happen for larger DMAX. Therefore, together with the requirement of Equation 9,
the recommended range for GI_CTRL is
0.355 (1-DMIN) < GI_CTRL < 1.33 ( 1-DMAX)
Equation 15
An optimum compromise for GI_CTRL has been suggested. For example.
GI_CTRLAUTO = 1 - DMAX
Equation 16
This value has been used for the Automatic setting of the web calculator. If 1-DMAX is less than 0.2, GI_CTRL is set to 0.2. If 1- DMAX is greater
than 0.5 then GI_CTRL is set to 0.5.
AL8871Q
Document number: DS42987 Rev. 2 - 2
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AL8871Q
Application Information
(continued)
Once GI_CTRL is selected, a value of RGI1 can be selected from Equation 8. Then RGI2 is calculated as follows, rearranging Equation 1:
RGI2 = RGI1
�
1-GI_CTRL
GI_CTRL
�
Equation 17
For example driving 12 LEDS at a current of 350mA from a 12V supply, each LED has a forward voltage of 3.2V at 350mA, so
Vout = 3.2 × 12 = 38.4V. The duty cycle is approximately
�VOUT -VIN �
VOUT
=
�38.4-12�
38.4
= 0.6875
From Equation 12, we set GI_CTRL to 1 – D = 0.3125.
IF RGI1 = 33kΩ, then from Equation 3, RGI2 = 33000 × (1 -0.3125) / 0.3125 = 72.6kΩ. Select the preferred value RGI2 = 75kΩ. Now GI_CTRL is
adjusted to the new value using Equation 1.
RGI1
33k
GI_CTRL = �
=0.305
� =
RGI1 +RGI2
33k +75k
Now calculate Rs from Equation 6. Assume CTRL is connected to REF.
0.225
VCTRL
0.225
� GI_CTRL �
� =
* 0.305 = 0.196 Ω
RS = �
VREF
ILED
0.35
A preferred value of RS = 0.2Ω gives the desired LED current with an error of 2% due to the preferred value selection.
Table 1 shows typical resistor values used to determine the GI_CTRL ratio with E24 series resistors.
Table 1
GI ratio
RGI1
RG2
0.2
30kΩ
120kΩ
0.25
33kΩ
100kΩ
0.3
39kΩ
91kΩ
0.35
30kΩ
56kΩ
0.4
100kΩ
150kΩ
0.45
51kΩ
62kΩ
0.5
30kΩ
30kΩ
This completes the LED current setting.
AL8871Q
Document number: DS42987 Rev. 2 - 2
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AL8871Q
Application Information
(continued)
Inductor Selection and Frequency Control
The selection of the inductor coil, L1, requires knowledge of the switching frequency and current ripple and depends on the duty cycle to some
extent. In the hysteretic converter, the frequency depends upon the input and output voltages and the switching thresholds of the current monitor.
The peak-to-peak coil current is adjusted by the AL8871Q to control the frequency to a fixed value. This is done by controlling the switching
thresholds within particular limits. This effectively reduces much of the overall frequency range for a given input voltage range. Where the input
voltage range is not excessive, the frequency is regulated to approximately 390kHz. This is helpful in terms of EMC and other system
requirements. Figure 5 shows practical results of switching frequency driving eight LEDs at 350mA.
500
450
L = 33µH
Switching Frequency (kHz)
400
350
300
L = 68µH
250
200
L = 100µH
150
TA = 25°C, VAUX = VIN
8 LEDs,
RS = 150mΩ,
R9 = 120kΩ, R10 = 36kΩ,
CIN = 100µF 8 LEDs
L = 68µH
100
50
0
5
6
7
8
9
10
11
12
VIN (V)
13
14
15
16
17
18
Figure 5. Frequency vs. VIN LED Driver with 350mA LED Current and Various Inductor Values
For larger input voltage variation, or when the choice of coil inductance is not optimum, the switching frequency can depart from the regulated
value, but the regulation of LED current remains successful. If desired, the frequency can to some extent be increased by using a smaller
inductor, or decreased using a larger inductor. The web calculator evaluates the frequency across the input voltage range and the effect of this
upon power efficiency and junction temperatures.
Determination of the input voltage range for which the frequency is regulated may be required. This calculation is very involved and is not given
here. However, the performance in this respect can be evaluated within the web calculator for the chosen inductance.
The inductance is given as follows in terms of peak-to-peak ripple current in the coil, ΔIL, and the MOSFET on time, tON.
L1 = {𝑉𝐼𝑁 − (𝐼𝐼𝑁 + 𝐼𝑂𝑈𝑇 )(𝑅𝐷𝑆𝑂𝑁 + 𝑅𝐶𝑂𝐼𝐿 + 𝑅𝑆 )} ×
𝑡𝑂𝑁
∆𝐼𝐿
Equation 18
Therefore In order to calculate L1, IIN, tON, and ΔIL must be found. The effects of the resistances are small and are estimated.IIN is estimated
from Equation 9. tON is related to switching frequency, f, and duty cycle, D, as follows:
t ON =
D
Equation 19
f
As the regulated frequency is known, and D from Equation 7, or the approximation Equation 7a, is found, this calculation of tON is possible.
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Application Information
(continued)
The AL8871Q sets the ripple current, ΔIL, to between nominally 10% and 30% of the mean coil current, ICOIL, which is found from Equation 8.
The device adjusts the ripple current within this range in order to regulate the switching frequency. A ΔIL value of 20% of ICOIL must be used to
find an inductance, which is optimized for the input voltage range. The range of ripple current control is also modulated by other circuit
parameters as follows.
∆𝐼𝐿𝑀𝐴𝑋 = {0.06 + 0.24(
∆𝐼𝐿𝑀𝐼𝑁 = {0.02 + 0.08(
𝑉𝐴𝐷𝐽
𝑉𝐴𝐷𝐽
𝑉𝑅𝐸𝐹
∆𝐼𝐿𝑀𝐼𝐷 = {0.04 + 0.16(
𝐺𝐼_𝐴𝐷𝐽
)} ×
𝐺𝐼_𝐴𝐷𝐽
)} ×
𝑉𝐴𝐷𝐽
𝑉𝑅𝐸𝐹
1−𝐷
)} ×
𝑉𝑅𝐸𝐹
1−𝐷
𝐺𝐼_𝐴𝐷𝐽
1−𝐷
× 𝐼𝐶𝑂𝐼𝐿
× 𝐼𝐶𝑂𝐼𝐿
Equation 20
× 𝐼𝐶𝑂𝐼𝐿
If ADJ is connected to REF, this simplifies to
∆𝐼𝐿𝑀𝐴𝑋 = 0.3 ×
∆𝐼𝐿𝑀𝐼𝑁 = 0.1 ×
∆𝐼𝐿𝑀𝐼𝐷 = 0.2 ×
1−𝐷
𝐺𝐼_𝐴𝐷𝐽
1−𝐷
𝐺𝐼_𝐴𝐷𝐽
× 𝐼𝐶𝑂𝐼𝐿
× 𝐼𝐶𝑂𝐼𝐿
Equation 20a
1−𝐷
×𝐼
𝐺𝐼_𝐴𝐷𝐽 𝐶𝑂𝐼𝐿
Where ΔILMID is the value of we must use in Equation 18. The inductance value is now established.
The chosen coil must saturate at a current greater than the peak sensed current. This saturation current is the DC current for which the
inductance has decreased by 10% compared to the low current value.
Assuming ±10% ripple current, the peak current can be found from Equation 8, which is adjusted for ripple current:
ICOILPEAK = 1.1 IINMAX + ILED
Equation 21
Where IINMAX is the value of IIN at minimum VIN. The mean current rating is also a factor, but normally the saturation current is the limiting factor.
LED Current Dimming
The AL8871Q has three dimming methods for reducing the average LED current:
1. DC dimming using the CTRL pin
2. PWM dimming using the PWM pin
3. DC dimming for thermal protection using the NTC pin
DC or Analog Dimming
The AL8871Q has a clamp on the CTRL pin to prevent overdriving of the LED current, which results in applying the maximum voltage to internal
circuitry as the reference voltage. This provides a 10:1 dynamic range of DC LED current adjustment.
The equation for DC dimming of the LED current is approximately:
VCTRL
�
ILED_DIM =ILED_NOM �
VREF
Where:
•
•
ILED_DIM is the dimmed LED current
ILED_NOM is the LED current with VCTRL = 1.25V
One consequence of DC dimming is as the CTRL pin voltage reduces, the sense voltage also be reduces, which has an impact on accuracy and
switching frequency especially at lower CTRL pin voltages.
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Application Information
(continued)
PWM Output Current Control & Dimming
The AL8871Q has a dedicated PWM dimming input that allows a wide-dimming frequency range from 100Hz to 1kHz with up to 1000:1 resolution;
however, higher dimming frequencies can be used at the expense of dimming dynamic range and accuracy.
Typically, for a PWM frequency of 1kHz, the error on the current linearity is lower than 5%; in particular, the accuracy is better than 1% for PWM
from 5% to 100%. For a PWM frequency of 100Hz, the error on the current linearity is lower than 2.5%; it becomes negligible for PWM greater
than 5%.
400
350
300
Io(mA)
250
200
150
fPWM=100Hz
fPWM=500Hz
fPWM=1kHz
100
50
0
0
10
20
30
40
50
60
70
80
90
100
PWM Duty(%)
Figure 7. LED Current Linearity and Accuracy with PWM Dimming
The PWM pin is designed to be driven by both 3.3V and 5V logic levels and, as such, does not require open collector/drain drive. It can also be
driven by an open drain/collector transistor. In this case the designer can either use the internal pull-up network or an external pull-up network in
order to speed-up PWM transitions.
LED current can be adjusted digitally, by applying a low frequency
PWM logic signal to the PWM pin to turn the controller on and off.
This will produce an average output current proportional to the
duty cycle of the control signal. During PWM operation, the device
remains powered up and only the output switch is gated by the
control signal.
The PWM signal can achieve extremely high LED current
resolution. In fact, dimming down from 100% to 0, a minimum
pulse width of 2µs can be achieved resulting in very high accuracy.
While the maximum recommended pulse is for the PWM signal is
10ms.
2µs
< 10 ms
Gate
0V
PWM
< 10 ms
0V
2µs
Figure 8. PWM Dimming Minimum and Maximum Pulse
The device can be put in standby by taking the PWM pin to ground, or pulling it to a voltage below 0.4V with a suitable open collector NPN or open
drain NMOS transistor, for a time exceeding 15ms (nominal). In the shutdown state, most of the circuitry inside the device is switched off and
residual quiescent current will be typically 90µA. In particular, the Status pin will go down to GND while the FLAG and REF pins will stay at their
nominal values.
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Application Information
(continued)
STATUS
Standby
state
0V
PWM
0V
~15ms
Figure 9. Stand-by State from PWM Signal
Thermal Control of LED Current
For thermal control of the LEDs, the AL8871Q monitors the voltage on the NTC pin and reduces output current if the voltage on this pin falls
below 625mV. An external NTC thermistor and resistor can therefore be connected as shown below to set the voltage on the NTC pin to 625mV
at the required temperature threshold. This will give 100% LED current below the threshold temperature and a falling current above it as shown
in the graph. The temperature threshold can be altered by adjusting the value of RTH and/or the thermistor to suit the requirements of the chosen
LED.
The Thermal Control feature can be disabled by connecting NTC directly to REF.
Here is a simple procedure to design the thermal feedback circuit:
1)
Select the temperature threshold TTHRESHOLD at which the current must start to decrease
2)
Select the Thermistor TH1 (both resistive value at +25˚C and beta)
3)
Select the value of the resistor RTH as RTH = TH at TTHRESHOLD
ILED
RTH
100%
TH1
REF
NTC
10%
70˚C
85˚C
TLED
Thermal network response:
RTH = 1.8kΩ and TH1=10kΩ (beta =3900)
Figure 10. Thermal Feedback Network
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AL8871Q
Applications Information (Cont.)
The thermistor resistance, RT, at a temperature of T degrees Kelvin is given by
1
T
B� -
R T = RR e
Where:
1
�
TR
RR is the thermistor resistance at the reference temperature, TR
TR is the reference temperature, in Kelvin, normally 273 + 25 = 298K (+25°C)
B is the “beta” value of the thermistor.
For example:
1)
Temperature threshold TTHRESHOLD = 273 + 70 = 343K (+70˚C)
2)
TH1 = 10kΩ at +25˚C and B = 3900
3)
RTH = RT at TTHRESHOLD = 1.8kΩ
RT = 1.8kΩ @ 70˚C
FLAG/STATUS Outputs
The FLAG/STATUS outputs provide a warning of extreme operating or fault conditions. FLAG is an open-drain logic output, which is normally off,
but switches low to indicate that a warning, or fault condition exists. STATUS is a DAC output, which is normally high (4.5V), but switches to a
lower voltage to indicate the nature of the warning/fault.
Conditions monitored, the method of detection and the nominal STATUS output voltage are given in the following table (Note 17):
Table 2
Severity
(Note 18)
Monitored
Parameters
FLAG
Nominal STATUS Voltage
-
-
H
4.5V
1
VAUX < 5.0V
L
4.5V
2
VIN < 5.6V
L
< 3.6V
Output Current out of Regulation (Note
19)
2
VSHP outside normal voltage
range
L
3.6V
Driver Stalled with Switch ‘on’, or ‘off’
(Note 20)
2
tON, or tOFF > 100µs
L
3.6V
Device Temperature above Maximum
Recommended Operating Value
3
TJ > +125°C
L
1.8V
Sense Resistor Current IRS above
Specified Maximum
4
VSENSE > 0.3V
L
0.9V
Warning/Fault Condition
Normal Operation
Supply Undervoltage
Notes:
17. These STATUS pin voltages apply for an input voltage, VIN, of 7.5V < VIN < 60V. Below 7.5V the STATUS pin voltage levels reduce and therefore
may not report the correct status. For 5.4V < VIN < 7.5V the flag pin still reports an error by going low. At low VIN in Boost and Buck-boost modes an
overcurrent status may be indicated when operating at high boost ratios - this due to the feedback loop increasing the sense voltage.
18. Severity 1 denotes lowest severity.
19. This warning will be indicated if the output power demand is higher than the available input power; the loop may not be able to maintain regulation.
20. This warning will be indicated if the gate pin stays at the same level for greater than 100µs (e.g. the output transistor cannot pass enough current to
reach the upper switching threshold).
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FLAG VOLTAGE
Applications Information (Cont.)
VREF
0V
4.5V
Normal
Operations
VAUX
UVLO
STATUS VOLTAGE
3.6V
- VIN UVLO
- STALL
- OUT of REG
2.7V
1.8V
Over
Temperature
0.9V
Over
Current
0A
0
3
2
1
4
SEVERITY
Figure 11. Status levels
In the event of more than one fault/warning condition occurring, the higher severity condition will take precedence. E.g. ‘Excessive coil current’
and ‘Out of regulation’ occurring together will produce an output of 0.9V on the STATUS pin.
If VCTR>1.7V, VSENSE may be greater than the excess coil current threshold in normal operation and an error will be reported. Hence, STATUS
and FLAG are only guaranteed for VCTR
