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LM3402, LM3402HV
SNVS450F – SEPTEMBER 2006 – REVISED OCTOBER 2015
LM3402/HV 0.5-A Constant Current Buck Regulator for Driving High Power LEDs
1 Features
3 Description
•
•
•
•
•
•
•
The LM3402/HV are monolithic switching regulators
designed to deliver constant currents to high power
LEDs. Ideal for automotive, industrial, and general
lighting applications, they contain a high-side Nchannel MOSFET switch with a current limit of 735
mA (typical) for step-down (Buck) regulators.
Hysteretic control with controlled ON-time coupled
with an external resistor allow the converter output
voltage to adjust as needed to deliver a constant
current to series and series - parallel connected
arrays of LEDs of varying number and type, LED
dimming by pulse width modulation (PWM),
broken/open LED protection, low-power shutdown
and thermal shutdown complete the feature set.
1
•
•
•
Integrated 0.5-A N-channel MOSFET
VIN Range from 6 V to 42 V (LM3402)
VIN Range from 6 V to 75 V (LM3402HV)
500 mA Output Current Over Temperature
Cycle-by-Cycle Current Limit
No Control Loop Compensation Required
Separate PWM Dimming and Low Power
Shutdown
Supports All Ceramic-Output Capacitors and
Capacitor-Less Outputs
Thermal Shutdown Protection
VSSOP, SO PowerPAD™ Packages
Device Information(1)
2 Applications
•
•
•
•
•
PART NUMBER
LED Drivers
Constant Current Source
Automotive Lighting
General Illumination
Industrial Lighting
LM3402/HV
PACKAGE
BODY SIZE (NOM)
VSSOP (8)
3.00 mm × 3.00 mm
HSOP (8)
4.89 mm × 3.90 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Diagram
CB
L1
VIN
VIN
CIN
BOOT
SW
RON
D1
IF
RON
LM3402/02HV
CS
RSNS
DIM
GND
VCC
CF
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM3402, LM3402HV
SNVS450F – SEPTEMBER 2006 – REVISED OCTOBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
5
5
6
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 10
7.3 Feature Description................................................. 10
7.4 Device Functional Modes........................................ 13
8
Application and Implementation ........................ 15
8.1 Application Information............................................ 15
8.2 Typical Application .................................................. 23
9 Power Supply Recommendations...................... 29
10 Layout................................................................... 29
10.1 Layout Guidelines ................................................. 29
10.2 Layout Example .................................................... 30
11 Device and Documentation Support ................. 31
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Related Links ........................................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
31
31
31
31
31
31
12 Mechanical, Packaging, and Orderable
Information ........................................................... 31
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision E (April 2013) to Revision F
•
Page
Added ESD Ratings and Timing Requirements tables, Feature Description section, Device Functional Modes,
Application and Implementation section, Power Supply Recommendations section, Layout section, Device and
Documentation Support section, and Mechanical, Packaging, and Orderable Information section ..................................... 1
Changes from Revision D (May 2013) to Revision E
•
2
Page
Changed layout of National Data Sheet to TI format ............................................................................................................. 1
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5 Pin Configuration and Functions
DGK Package
8-Pin VSSOP
Top View
1
DDA Package
8-Pin HSOP
Top View
8
VIN
SW
2
7
BOOT
VCC
DIM
RON
3
1
6
4
2
5
GND
3
CS
4
VIN
SW
BOOT
DIM
GND
DAP
VCC
RON
CS
8
7
6
5
Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
1
SW
O
Switch pin. Connect this pin to the output inductor and Schottky diode.
2
BOOT
O
MOSFET drive bootstrap pin. Connect a 10-nF ceramic capacitor from this pin to SW.
3
DIM
I
Input for PWM dimming. Connect a logic-level PWM signal to this pin to enable and disable the power
MOSFET and reduce the average light output of the LED array.
4
GND
—
5
CS
I
Current sense feedback pin. Set the current through the LED array by connecting a resistor from this pin to
ground.
6
RON
I
ON-time control pin. A resistor connected from this pin to VIN sets the regulator controlled ON-time.
7
VCC
O
Output of the internal 7-V linear regulator. Bypass this pin to ground with a minimum 0.1-µF ceramic capacitor
with X5R or X7R dielectric.
8
VIN
I
Input voltage pin. Nominal operating input range for this pin is 6 V to 42 V (LM3402) or 6 V to 75 V
(LM3402HV).
DAP
—
Ground pin. Connect this pin to system ground.
Thermal Pad. Connect to ground. Place 4-6 vias from DAP to bottom layer ground plane.
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SNVS450F – SEPTEMBER 2006 – REVISED OCTOBER 2015
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
VIN to GND
BOOT to GND
SW to GND
MIN
MAX
LM3402
–0.3
45
LM3402HV
–0.3
76
LM3402
–0.3
59
LM3402HV
–0.3
90
LM3402
UNIT
V
V
–1.5
V
LM3402HV
–1.5
LM3402
–0.3
45
LM3402HV
–0.3
76
BOOT to SW
–0.3
14
V
VCC to GND
–0.3
14
V
DIM to GND
–0.3
7
V
CS to GND
–0.3
7
V
RON to GND
–0.3
7
V
Lead temperature (soldering, 10 s)
260
°C
Infrared/convection reflow (15 s)
235
°C
150
°C
125
°C
BOOT to VCC
Soldering information
Junction temperature
Storage temperature
(1)
(2)
–65
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
6.2 ESD Ratings
VALUE
UNIT
LM3402
V(ESD)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±1000
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±1000
V
LM3402HV
V(ESD)
(1)
(2)
Electrostatic discharge
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VIN
MIN
MAX
LM3402
6
42
LM3402HV
6
75
–40
125
Junction temperature
4
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UNIT
V
°C
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6.4 Thermal Information
LM3402, LM3402HV
THERMAL METRIC (1)
DDA (HSOP)
DGK (VSSOP)
8 PINS
8 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
45.6
154.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
53.1
48.2
°C/W
RθJB
Junction-to-board thermal resistance
25.8
74.2
°C/W
ψJT
Junction-to-top characterization parameter
7.6
4.3
°C/W
ψJB
Junction-to-board characterization parameter
25.7
72.9
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.2
N/A
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
6.5 Electrical Characteristics
VIN = 24 V unless otherwise indicated. Typical values apply over –40°C ≤ TJ ≤ 125°C; minimum and maximum values apply
over the full operating temperature range (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
194
200
206
mV
REGULATION AND OVERVOLTAGE COMPARATORS
VREF-REG
CS Regulation Threshold
CS Decreasing, SW turns on
VREF-0V
CS Overvoltage Threshold
CS Increasing, SW turns off
300
mV
ICS
CS Bias Current
CS = 0 V
0.1
µA
VSD-TH
Shutdown Threshold
RON / SD Increasing
VSD-HYS
Shutdown Hysteresis
RON / SD Decreasing
Minimum OFF-time
CS = 0 V
SHUTDOWN
0.3
0.7
1.05
V
40
mV
300
ns
OFF TIMER
tOFF-MIN
INTERNAL REGULATOR
VCC-REG
VCC Regulated Output
VIN-DO
VIN - VCC Dropout
ICC = 5 mA, 6 V < VIN < 8 V
6.6
VCC-BP-TH
VCC Bypass Threshold
VIN Increasing
8.8
V
VCC-BP-HYS
VCC Bypass Hysteresis
VIN Decreasing
225
mV
VCC-Z-6
VCC-Z-8
VCC Output Impedance
(0 mA < ICC < 5 mA)
VCC-Z-24
(2)
7
7.4
300
VIN = 6 V
55
VIN = 8 V
50
VIN = 24 V
0.4
VIN = 24 V, VCC = 0 V
16
V
mV
Ω
VCC-LIM
VCC Current Limit
VCC-UV-TH
VCC Undervoltage Lock-out
Threshold
mA
VCC Increasing
5.25
V
VCC-UV-HYS
VCC Undervoltage Lock-out
Hysteresis
VCC Decreasing
150
mV
VCC-UV-DLY
VCC Undervoltage Lock-out Filter
Delay
100-mV Overdrive
3
µs
IIN-OP
IIN Operating Current
Non-switching, CS = 0 V
IIN-SD
IIN Shutdown Current
RON / SD = 0 V
600
900
µA
90
180
µA
735
940
mA
CURRENT LIMIT
ILIM
Current Limit Threshold
530
DIM COMPARATOR
VIH
Logic High
DIM Increasing
VIL
Logic Low
DIM Decreasing
IDIM-PU
DIM Pullup Current
DIM = 1.5 V
(1)
(2)
2.2
V
0.8
75
V
µA
Typical specifications represent the most likely parametric norm at 25°C operation.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
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Electrical Characteristics (continued)
VIN = 24 V unless otherwise indicated. Typical values apply over –40°C ≤ TJ ≤ 125°C; minimum and maximum values apply
over the full operating temperature range (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
0.7
1.5
Ω
3
4
V
N-MOSFET AND DRIVER
RDS-ON
Buck Switch ON Resistance
ISW = 200 mA, BOOT-SW = 6.3
V
VDR-UVLO
BOOT Undervoltage Lockout
Threshold
BOOT–SW Increasing
VDR-HYS
BOOT Undervoltage Lockout
Hysteresis
BOOT–SW Decreasing
1.7
400
mV
THERMAL SHUTDOWN
TSD
Thermal Shutdown Threshold
165
°C
TSD-HYS
Thermal Shutdown Hysteresis
25
°C
6.6 Timing Requirements
tON-1
tON-2
6
ON-time 1, VIN = 10 V, RON = 200 kΩ
MIN
NOM
MAX
2.1
2.75
3.4
ON-time 2, VIN = 40 V, RON = 200 kΩ
LM3402
490
650
810
ON-time 2, VIN = 70 V, RON = 200 kΩ
LM3402HV
290
380
470
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UNIT
µs
ns
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6.7 Typical Characteristics
Figure 1. VREF vs Temperature (VIN = 24 V)
Figure 2. VREF vs VIN, LM3402 (TA = 25°C)
Figure 3. VREF vs VIN, LM3402HV (TA = 25°C)
Figure 4. Current Limit vs Temperature (VIN = 24 V)
Figure 5. Current Limit vs VIN, LM3402 (TA = 25°C)
Figure 6. Current Limit vs VIN, LM3402HV (TA = 25°C)
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Typical Characteristics (continued)
8
Figure 7. TON vs VIN, RON = 100 kΩ (TA = 25°C)
Figure 8. TON vs VIN (TA = 25°C)
Figure 9. TON vs VIN, (TA = 25°C)
Figure 10. TON vs RON, LM3402 (TA = 25°C)
Figure 11. TON vs RON, LM3402HV (TA = 25°C)
Figure 12. VCC vs VIN (TA = 25°C)
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Typical Characteristics (continued)
Figure 13. VO-MAX vs fSW, LM3402 (TA = 25°C)
Figure 14. VO-MIN vs fSW, LM3402 (TA = 25°C)
Figure 15. VO-MAX vs fSW, LM3402HV (TA = 25°C)
Figure 16. VO-MIN vs fSW, LM3402HV (TA = 25°C)
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7 Detailed Description
7.1 Overview
The LM3402 and LM3402HV are buck regulators with a wide input voltage range, low voltage reference, and a
fast output enable/disable function. These features combine to make them ideal for use as a constant current
source for LEDs with forward currents as high as 500 mA. The controlled ON-time (COT) architecture is a
combination of hysteretic mode control and a one-shot ON-timer that varies inversely with input voltage.
Hysteretic operation eliminates the need for small-signal control loop compensation. When the converter runs in
continuous conduction mode (CCM) the controlled ON-time maintains a constant switching frequency over the
range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output
overvoltage protection round out the functions of the LM3402/HV.
7.2 Functional Block Diagram
7V BIAS
REGULATOR
VIN
VIN
SENSE
VCC
UVLO
BYPASS
SWITCH
0.7V
VCC
THERMAL
SHUTDOWN
+
300 ns MIN
OFF TIMER
Complete
ON TIMER
RON
RON
Complete
5V
BOOT
Start
Start
GATE DRIVE
UVLO
75 PA
DIM
1.5V
0.2V
+
+
-
SD
VIN
LEVEL
SHIFT
LOGIC
CS
SW
0.3V
+
-
GND
CURRENT
LIMIT OFF
TIMER
+
-
0.735A
BUCK
SWITCH
CURRENT
SENSE
7.3 Feature Description
7.3.1 Controlled ON-time Overview
Figure 17 shows the feedback system used to control the current through an array of LEDs. A voltage signal,
VSNS, is created as the LED current flows through the current setting resistor, RSNS, to ground. VSNS is fed back
to the CS pin, where it is compared against a 200-mV reference, VREF. The ON-comparator turns on the power
MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled ON-time, tON, set by an
external resistor, RON, and by the input voltage, VIN. ON-time is governed using Equation 1.
RON
tON = 1.34 x 10-10 x
VIN
(1)
10
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Feature Description (continued)
At the conclusion of tON the power MOSFET turns off for a minimum OFF-time, tOFF-MIN, of 300 ns. Once tOFF-MIN
is complete the CS comparator compares VSNS and VREF again, waiting to begin the next cycle.
VO
LED 1
IF
LM3402/02HV
LED n
CS
Comparator
One-shot
+
+
-
VF
VSNS
CS
VREF
VF
IF
RSNS
Figure 17. Comparator and One-Shot
The LM3402/HV regulators should be operated in continuous conduction mode (CCM), where inductor current
stays positive throughout the switching cycle. During steady-state operation in the CCM, the converter maintains
a constant switching frequency, which can be selected using Equation 2.
fSW =
VO
1.34 x 10-10 x RON
VO = n x VF + 200 mV
(2)
(3)
VF = forward voltage of each LED, n = number of LEDs in series
7.3.2 Average LED Current Accuracy
The COT architecture regulates the valley of ΔVSNS, the AC portion of VSNS. To determine the average LED
current (which is also the average inductor current) the valley inductor current is calculated using Equation 4.
0.2 VO x tSNS
IL-MIN =
L
RSNS
(4)
In Equation 4 tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The
average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ΔiL:
IF = IL = IL-MIN + ΔiL / 2
(5)
Detailed information for the calculation of ΔiL is given in the Application and Implementation section.
7.3.3 Maximum Output Voltage
The 300 ns minimum off-time limits on the maximum duty cycle of the converter, DMAX, and in turn ,the maximum
output voltage VO(MAX) is determined by Equation 6:
tON
DMAX =
tON + tOFF-MIN
VO(max) = DMAX x VIN
(6)
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Feature Description (continued)
The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the
maximum forward voltage of the LEDs used, VF(MAX), using the expression:
VO(max) - 200 mV
nMAX =
VF(MAX)
(7)
At low switching frequency the maximum duty cycle and output voltage are higher, allowing the LM3402/HV to
regulate output voltages that are nearly equal to input voltage. The following equation relates switching frequency
to maximum output voltage.
TSW - 300 ns
VO(MAX) = VIN x
TSW
TSW = 1/fSW
(8)
7.3.4 Minimum Output Voltage
The minimum recommended ON-time for the LM3402/HV is 300 ns. This lower limit for tON determines the
minimum duty cycle and output voltage that can be regulated based on input voltage and switching frequency.
The relationship is determined by the following equation:
VO(MIN) = VIN x
300 ns
TSW
(9)
7.3.5 High Voltage Bias Regulator
The LM3402/HV contains an internal linear regulator with a 7-V output, connected between the VIN and the VCC
pins. The VCC pin should be bypassed to the GND pin with a 0.1-µF ceramic capacitor connected as close as
possible to the pins of the device. VCC tracks VIN until VIN reaches 8.8 V (typical) and then regulates at 7 V as
VIN increases. Operation begins when VCC crosses 5.25 V.
7.3.6 Internal MOSFET and Driver
The LM3402/HV features an internal power MOSFET as well as a floating driver connected from the SW pin to
the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 3 nC. The
high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and an
external 10 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off. When
the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC voltage
minus the diode drop to drive the MOSFET when its source voltage is equal to VIN.
7.3.7 Fast Shutdown for PWM Dimming
The DIM pin of the LM3402/HV is a TTL logic compatible input for low-frequency PWM dimming of the LED. A
logic low (less than 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array.
While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active to minimize
the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2 V). A 75 µA
(typical) pullup current ensures that the LM3402/HV is on when DIM pin is open circuited, eliminating the need
for a pullup resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and
fall time and the delay from activation of the DIM pin to the response of the internal power MOSFET. In general,
fDIM should be at least one order of magnitude lower than the steady state switching frequency to prevent
aliasing.
7.3.8 Peak Current Limit
The current limit comparator of the LM3402/HV will engage whenever the power MOSFET current (equal to the
inductor current while the MOSFET is on) exceeds 735 mA (typical). The power MOSFET is disabled for a cooldown time that is 10x the steady-state ON-time. At the conclusion of this cool-down time the system re-starts. If
the current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power
hiccup mode, minimizing thermal stress on the LM3402/HV and the external circuit components.
12
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Feature Description (continued)
7.3.9 Overvoltage and Overcurrent Comparator
The CS pin includes an output overvoltage/overcurrent comparator that will disable the power MOSFET
whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current
overshoot is limited to 300 mV / RSNS by this comparator during transients.
The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of
an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a
current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 18
shows a method using a Zener diode, Z1, and Zener limiting resistor, RZ, to limit output voltage to the reverse
breakdown voltage of Z1 plus 200 mV. The Zener diode reverse breakdown voltage, VZ, must be greater than
the maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ.
As discussed in the Maximum Output Voltage section, there is a limit to how high VO can rise during an output
open-circuit that is always less than VIN. If no output capacitor is used, the output stage of the LM3402/HV is
capable of withstanding VO(MAX) indefinitely, however the voltage at the output end of the inductor will oscillate
and can go above VIN or less than 0 V. A small (typically 10 nF) capacitor across the LED array dampens this
oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long as
CO is rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal stress
is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED failure.
CB
VIN
VIN
CIN
BOOT
L1
SW
RON
D1
Z1
RON
LM3402/02HV
RZ
CS
RSNS
DIM
GND
VCC
CF
Figure 18. Output Open Circuit Protection
7.4 Device Functional Modes
7.4.1 Low Power Shutdown
The LM3402/HV can be switched to a low power state (IIN-SD = 90 µA) by grounding the RON pin with a signallevel MOSFET as shown in Figure 19. Low power MOSFETs like the 2N7000, 2N3904, or equivalent are
recommended devices for putting the LM3402/HV into low power shutdown. Logic gates can also be used to shut
down the LM3402/HV as long as the logic low voltage is below the over temperature minimum threshold of 0.3V.
Noise filter circuitry on the RON pin can cause a few pulses with a longer ON-time than normal after RON is
grounded or released. In these cases the OVP/OCP comparator will ensure that the peak inductor or LED current
does not exceed 300 mV / RSNS.
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Device Functional Modes (continued)
CB
L1
VIN
VIN
CIN
BOOT
SW
RON
D1
RON
IF
LM3402/02HV
ON/OFF
Q1
2N7000 or
equivalent
CS
RSNS
DIM
GND
VCC
CF
Figure 19. Low Power Shutdown
7.4.2 Thermal Shutdown
Internal thermal shutdown circuitry is provided to protect the device in the event that the maximum junction
temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values
typical). During thermal shutdown the MOSFET and driver are disabled.
14
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Switching Frequency
Switching frequency is selected based on the tradeoffs between efficiency (better at low frequency), solution
size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower
frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The ON-time of the
LM3402/HV can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. The
maximum switching frequency is limited only by the minimum ON-time requirement.
8.1.2 LED Ripple Current
Selection of the ripple current, ΔiF, through the LED array is analogous to the selection of output ripple voltage in
a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC
output voltage, LED manufacturers generally recommend values for ΔiF ranging from ±5% to ±20% of IF. Higher
LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all.
The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires
more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower
ripple current are a reduction in heating in the LED itself and greater range of the average LED current before
the current limit of the LED or the driving circuitry is reached.
8.1.3 Buck Converters Without Output Capacitors
The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to
the load during the entire switching cycle. By definition an inductor will control the rate of change of current that
flows through it, and this control overcurrent ripple forms the basis for component selection in both voltage
regulators and current regulators. A current regulator such as the LED driver for which the LM3402/HV was
designed focuses on the control of the current through the load, not the voltage across it. A constant current
regulator is free of load current transients, and has no need of output capacitance to supply the load and
maintain output voltage. Referring to the Typical Application Diagram, the inductor and LED can form a single
series chain, sharing the same current. When no output capacitor is used, the same equations that govern
inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a controlled ON-time converter such as
LM3402/HV the ripple current is described by the following expression:
VIN - VO
tON
'iL = 'iF =
L
(10)
A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal-to-noise ratio (SNR).
The CS pin ripple voltage, ΔVSNS, is described by the following:
ΔVSNS = ΔiF x RSNS
(11)
8.1.4 Buck Converters With Output Capacitors
A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while
keeping the same average current through both the inductor and the LED array. This technique is demonstrated
in Typical Application. With this topology the output inductance can be lowered, making the magnetics smaller
and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor value,
improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak current
limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL can be
even if ΔiF is made very small. A parallel output capacitor is also useful in applications where the inductor or
input voltage tolerance is poor. Adding a capacitor that reduces ΔiF to well below the target provides headroom
for changes in inductance or VIN that might otherwise push the peak LED ripple current too high.
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Application Information (continued)
Figure 20 shows the equivalent impedances presented to the inductor current ripple when an output capacitor,
CO, and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor
ripple current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS
comparator.
'iL
CO
rD
'iC
'iF
ESR
'iL
RSNS
Figure 20. LED and CO Ripple Current
To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED
dynamic resistance is not always specified on the manufacturer’s data sheet, but it can be calculated as the
inverse slope of the LED’s VF vs. IF curve. Dividing VF by IF will give an incorrect value that is 5x to 10x too high.
Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one device
multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without Output
Capacitors. The following equations can then be used to estimate ΔiF when using a parallel capacitor:
'iL
'iF =
rD
1+
ZC
1
ZC = ESR +
2S x fSW x CO
(12)
The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal.
Small values of CO that do not significantly reduce ΔiF can also be used to control EMI generated by the
switching action of the LM3402/HV. EMI reduction becomes more important as the length of the connections
between the LED and the rest of the circuit increase.
8.1.5 Input Capacitors
Input capacitors at the VIN pin of the LM3402/HV are selected using requirements for minimum capacitance and
rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power
MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. Switching converters
such as the LM3402/HV have a negative input impedance due to the decrease in input current as input voltage
increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes called
‘power supply interaction’) if the magnitude of the negative input impedance is greater the the input filter
impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative
resistance; however this requires accurate calculation of the input voltage source inductance and resistance,
quantities which can be difficult to determine. An alternative method to select the minimum input capacitance,
CIN(MIN), is to select the maximum voltage ripple which can be tolerated. This value,ΔvIN(MAX), is equal to the
change in voltage across CIN during the converter ON-time, when CIN supplies the load current. CIN(MIN) can be
selected with the following:
CIN(MIN) =
16
IF x tON
'VIN(MAX)
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Application Information (continued)
A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input
capacitance of 2× the CIN(MIN) value is recommended for all LM3402/HV circuits. To determine the rms current
rating, the following formula can be used:
IIN(rms) = IF x D(1 - D)
(14)
Ceramic capacitors are the best choice for the input to the LM3402/HV due to their high ripple current rating, low
ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention
must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or more of their
capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature. A DC voltage
rating equal to twice the expected maximum input voltage is recommended. In addition, the minimum quality
dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is preferred.
8.1.6 Recirculating Diode
The LM3402/HV is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical
Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1
is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the
maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching
converters have some ringing at the switching node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average current, ID, calculated as:
ID = (1 – D) × IF
(15)
This calculation should be done at the maximum expected input voltage. The overall converter efficiency
becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the
load current for an increasing percentage of the time. This power dissipation can be calculated by checking the
typical diode forward voltage, VD, from the I-V curve on the product data sheet and then multiplying it by ID.
Diode data sheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to
estimate the operating die temperature of the Schottky. Multiplying the power dissipation (PD = ID x VD) by θJA
gives the temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature
below the operational maximum.
8.1.7 LED Current During DIM Mode
The LM3402 contains high speed MOSFET gate drive circuitry that switches the main internal power MOSFET
between ON and OFF states. This circuitry uses current derived from the VCC regulator to charge the MOSFET
during turnon, then dumps current from the MOSFET gate to the source (the SW pin) during turnoff. As shown in
the block diagram, the MOSFET drive circuitry contains a gate drive undervoltage lockout (UVLO) circuit that
ensures the MOSFET remains off when there is inadequate VCC voltage for proper operation of the driver. This
watchdog circuitry is always running including during DIM and shutdown modes, and supplies a small amount of
current from VCC to SW. Because the SW pin is connected directly to the LEDs through the buck inductor, this
current returns to ground through the LEDs. The amount of current sourced is a function of the SW voltage, as
shown in Figure 21.
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Application Information (continued)
25
SW CURRENT (PA)
20
15
10
5
0
0
1
2
3
4
5
6
SW VOLTAGE (V)
Figure 21. LED Current From SW Pin
Though most power LEDs are designed to run at several hundred milliamps, some can be seen to glow with a
faint light at extremely low current levels, as low as a couple microamps in some instances. In lab testing, the
forward voltage was found to be approximately 2 V for LEDs that exhibited visible light at these low current
levels. For LEDs that did not show light emission at very low current levels, the forward voltage was found to be
around 900 mV. It is important to remember that the forward voltage is also temperature dependent, decreasing
at higher temperatures. Consequently, with a maximum Vcc voltage of 7.4 V, current will be observed in the
LEDs if the total stack voltage is less than about 6 V at a forward current of several microamps. No current is
observed if the stack voltage is above 6 V, as shown in Figure 21. The need for absolute darkness during DIM
mode is also application dependent. It will not affect regular PWM dimming operation.
The fix for this issue is extremely simple. Place a resistor from the SW pin to ground according to Table 1.
Table 1. LED Resistor Values
18
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NUMBER OF LEDs
RESISTOR VALUE (kΩ)
1
20
2
50
3
90
4
150
5
200
>5
300
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The luminaire designer should ensure that the suggested resistor is effective in eliminating the off-state light
output. A combination of calculations based on LED manufacturer data and lab measurements over temperature
will ensure the best design.
8.1.8 Transient Protection Considerations
Considerations must be made when external sources, loads or connections are made to the switching converter
circuit due to the possibility of Electrostatic Discharge (ESD) or Electric Over Stress (EOS) events occurring and
damaging the integrated circuit (IC) device. All IC device pins contain Zener based clamping structures that are
meant to clamp ESD. ESD events are very low energy events, typically less than 5 µJ (microjoules). Any event
that transfers more energy than this may damage the ESD structure. Damage is typically represented as a short
from the pin to ground as the extreme localized heat of the ESD / EOS event causes the aluminum metal on the
chip to melt, causing the short. This situation is common to all integrated circuits and not just unique to the
LM340X device.
8.1.8.1 CS Pin Protection
When hot swapping in a load (for example, test points, load boards, LED stack), any residual charge on the load
will be immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in
Figure 22 below. The EOS event due to the residual charge from the load is represented as VTRANSIENT.
From measurements, we know that the 8V ESD structure on the CS pin can typically withstand 25mA of direct
current (DC). Adding a 1-kΩ resistor in series with the CS pin, shown in Figure 23, results in the majority of the
transient energy to pass through the discrete sense resistor rather than the device. The series resistor limits the
peak current that can flow during a transient event, thus protecting the CS pin. With the 1-kΩ resistor shown, a
33-V, 49-A transient on the LED return connector terminal could be absorbed as calculated by:
V = 25 mA × 1 kΩ + 8 V = 33 V
I = 33 V / 0.67 Ω = 49 A
(16)
(17)
This is an extremely high energy event, so the protection measures previously described should be adequate to
solve this issue.
LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
8V
~ 0.675
GND
Figure 22. CS Pin, Transient Path
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LM3402
SW
Module
Connector
Module
Connector
CS
VTRANSIENT
1 k5
8V
~ 0.675
GND
Figure 23. CS Pin, Transient Path With Protection
Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The
reason for this is twofold: (1) the CS pin has about 20 pF of inherent capacitance inside it which causes a slight
delay (20 ns for a 1-kΩ series resistor), and (2) the comparator that is watching the voltage at the CS pin uses a
pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100 nA, which will
cause a 0.1-mV change in the 200-mV threshold. These are both very minor changes and are well understood.
The shift in current can either be neglected or taken into consideration by changing the current sense resistance
slightly.
8.1.8.2 CS Pin Protection With OVP
When designing output overvoltage protection into the switching converter circuit using a Zener diode, transient
protection on the CS pin requires additional consideration. As shown in Figure 24, adding a Zener diode from the
output to the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient
energy to be passed through the CS pin’s ESD structure thereby damaging it.
Adding an additional series resistor to the CS pin as shown in Figure 25 will result in the majority of the transient
energy to pass through the sense resistor, thereby protecting the LM340X device.
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LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
1 k5
8V
~ 0.675
GND
Figure 24. CS Pin With OVP, Transient Path
LM3402
SW
Module
Connector
Module
Connector
VTRANSIENT
CS
1 k5
5005
8V
~ 0.675
GND
Figure 25. CS Pin With OVP, Transient Path with Protection
8.1.8.3 VIN Pin Protection
The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80 V.
Any transient that exceeds this voltage may damage the device. Although transient absorption is usually present
at the front end of a switching converter circuit, damage to the VIN pin can still occur.
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When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes)
the circuit board trace inductance as shown in Figure 26. The excited trace inductance then resonates with the
input capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in
excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating
voltage at the VIN pin exceeds the 80-V breakdown voltage of the ESD structure, the ESD structure will activate
and then snap-back to a lower voltage due to its inherent design. If this lower snap-back voltage is less than the
applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the device
being damaged.
An additional TVS or small Zener diode should be placed as close as possible to the VIN pins of each device on
the board, in parallel with the input capacitor as shown in Figure 27. A minor amount of series resistance in the
input line would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often
used as inrush limiters instead.
LM3402
Board Trace
Inductance
VIN
Module
Connector
80V
TVS
VIN
CIN
GND
Module
Connector
Figure 26. VIN Pin With Typical Input Protection
22
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LM3402
Board Trace
Inductance
VIN
Module
Connector
80V
VIN
TVS
CIN
TVS or
smaller
zener diode
GND
Module
Connector
Figure 27. VIN Pin With Additional Input Protection
8.1.8.4 General Comments Regarding Other Pins
Any pin that goes off-board through a connector should have series resistance of at least 1 kΩ to 10 kΩ in series
with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or
cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used
should not be left floating. They should instead be tied to GND or to an appropriate voltage through resistance.
8.2 Typical Application
The first example circuit will guide the user through component selection for an architectural accent lighting
application. A regulated DC voltage input of 24 V ±10% will power a single 1-W white LED at a forward current of
350 mA ±5%. The typical forward voltage of a 1-W InGaN LED is 3.5 V, hence the estimated average output
voltage will be 3.7 V. The objective of this application is to place the complete current regulator and LED in the
compact space formerly occupied by an MR16 halogen light bulb. (The LED will be on a separate metal-core
PCB.) Switching frequency will be as fast as the 300-ns tON limit allows, with the emphasis on space savings
over efficiency. Efficiency cannot be ignored, however, because the confined space with little air-flow requires a
maximum temperature rise of 40°C in each circuit component. A complete bill of materials can be found in
Table 2.
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Typical Application (continued)
CB
L1
VIN = 24V
VIN
BOOT
SW
RON
CIN
D1
RON
CO
LED1
IF = 350 mA
LM3402/02HV
CS
DIM
RSNS
VCC
GND
CF
Figure 28. Schematic for Design Example 1
8.2.1 Design Requirements
Use the following parameters for this design example:
1. VIN = 24 V ± 10%
2. ILED = 350 mA
3. VLED = 3.5 V, 1 LED
8.2.2 Detailed Design Procedure
Table 2 shows the body of materials for this design example.
Table 2. BOM for Design Example 1
ID
PART NUMBER
TYPE
SIZE
PARAMETERS
QTY
VENDOR
U1
LM3402
LED Driver
VSSOP-8
40 V, 0.5 A
1
NSC
L1
SLF7045T-330MR82
Inductor
7 × 7 × 4.5 mm
33 µH, 0.82 A, 96
mΩ
1
TDK
D1
CMHSH5-4
Schottky Diode
SOD-123
40 V, 0.5 A
1
Central Semi
Cf
VJ0805Y104KXXAT
Capacitor
0805
100 nF 10%
1
Vishay
Cb
VJ0805Y103KXXAT
Capacitor
0805
10n F 10%
1
Vishay
Cin
C3216X7R1H105M
Capacitor
1206
1 µF 50 V
1
TDK
Co
C2012X7R1A225M
Capacitor
0805
2.2 µF 10 V
1
TDK
Rsns
ERJ6BQFR75V
Resistor
0805
0.75 Ω 1%
1
Panasonic
Ron
CRCW08055902F
Resistor
0805
59 kΩ 1%
1
Vishay
8.2.2.1 RON and tON
To select RON the expression relating tON to input voltage from the Controlled ON-time Overview section can be
rewritten as Equation 18
tON x VIN
RON =
1.34 x 10-10
(18)
Minimum ON-time occurs at the maximum VIN, which is 24 V × 110% = 26.4 V. RON is therefore calculated using
Equation 19.
RON = (300 × 10–9 × 26.4) / 1.34 × 10–10 = 59105 Ω
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The closest 1% tolerance resistor is 59 kΩ. The switching frequency of the circuit can then be found using the
equation relating RON to fSW:
fSW = 3.7 / (59000 × 1.34 × 10–10) = 468 kHz
(20)
8.2.2.2 Using an Output Capacitor
The inductor will be the largest component used in this design. Because the application does not require any
PWM dimming, an output capacitor can be used to greatly reduce the inductance needed without worry of
slowing the potential PWM dimming frequency. The total solution size is reduced by using an output capacitor
and small inductor as opposed to one large inductor.
8.2.2.3 Output Inductor
Knowing that an output capacitor will be used, the inductor can be selected for a larger current ripple. The
desired maximum value for ΔiL is ±30%, or 0.6 × 350 mA = 210 mAP-P. Minimum inductance is selected at the
maximum input voltage. Re-arranging the equation for current ripple selection yields the following:
LMIN =
VIN(MAX) - VO
'iL
x tON
(21)
(22)
LMIN = [(26.4 – 3.7) × 300 × 10–9] / (0.6 × 0.35) = 32.4 µH
The closest standard inductor value is 33 µH. Off-the-shelf inductors rated at 33 µH are available from many
magnetics manufacturers.
Inductor data sheets should contain three specifications that are used to select the inductor. The first of these is
the average current rating, which for a buck regulator is equal to the average load current, or IF. The average
current rating is given by a specified temperature rise in the inductor, normally 40°C. For this example, the
average current rating should be greater than 350 mA to ensure that heat from the inductor does not reduce the
lifetime of the LED or cause the LM3402 to enter thermal shutdown.
The second specification is the tolerance of the inductance itself, typically ±10% to ±30% of the rated inductance.
In this example, designers use an inductor with a tolerance of ±20%. With this tolerance the typical, minimum,
and maximum inductor current ripples can be calculated using Equation 23 to Equation 25.
ΔiL(TYP) = [(26.4 – 3.7) × 300 × 10–9] / 33 × 10–6 = 206 mAP-P
ΔiL(MIN) = [(26.4 – 3.7) × 300 × 10–9] / 39.6 × 10–6 = 172 mAP-P
ΔiL(MAX) = [(26.4 – 3.7) × 300 × 10–9] / 26.4 × 10–6 = 258 mAP-P
(23)
(24)
(25)
The third specification for an inductor is the peak current rating, normally given as the point at which the
inductance drops off by a given percentage due to saturation of the core. The worst-case peak current occurs at
maximum input voltage and at minimum inductance, and can be determined with:
IL(PEAK) = IF +
'iL(MAX)
2
(26)
(27)
IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA
For this example the peak current rating of the inductor should be greater than 479 mA. In the case of a short
circuit across the LED array, the LM3402 will continue to deliver rated current through the short but will reduce
the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition is equal to:
ΔiL(LED-SHORT) = [(26.4 – 0.2) × 300 × 10–9] / 26.4 × 10–6 = 298 mAP-P IL(PEAK) = 0.35 + 0.149 = 499 mA
(28)
In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit
will engage at a typical peak current of 735 mA. To prevent inductor saturation during these short circuits the
inductor’s peak current rating must be above 735 mA. The device selected is an off-the-shelf inductor rated 33
µH ±20% with a DCR of 96 mΩ and a peak current rating of 0.82 A. The physical dimensions of this inductor are
7 × 7 × 4.5 mm.
8.2.2.3.1 RSNS
The current sensing resistor value can be determined by re-arranging the expression for average LED current
from the LED Current Accuracy section:
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RSNS =
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0.2 x L
IF x L + VO x tSNS - VIN - VO x tON
2
(29)
(30)
RSNS = 0.74Ω, tSNS = 220 ns
Sub-1-Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.75-Ω resistor will give the best accuracy of
the average LED current. To determine the resistor size the power dissipation can be calculated using
Equation 31.
PSNS = (IF)2 × RSNS PSNS = 0.352 × 0.75 = 92 mW
(31)
Standard 0805 size resistors are rated to 125 mW and will be suitable for this application.
To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is rearranged to
yield the following:
ZC =
'iF
'iL - 'iF
x rD
(32)
The target tolerance for LED ripple current is ±5% or 10%P-P = 35 mAP-P, and the LED data sheet gives a typical
value for rD of 1 Ω at 350 mA. The required capacitor impedance to reduce the worst-case inductor ripple current
of 258 mAP-P is therefore:
ZC = [0.035 / (0.258 - 0.035] × 1 = 0.157 Ω
(33)
A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 468 kHz:
CO = 1/(2 × π × 0.157 × 4.68 × 105) = 2.18 µF
(34)
This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series
inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 2.2 µF. The capacitor used
should be rated to 10 V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors
with these specifications in the 0805 case size. A typical value for ESR of 1 mΩ can be read from the curve of
impedance vs frequency in the product data sheet.
8.2.2.4 Input Capacitor
Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 240 mV. The minimum
required capacitance is:
CIN(MIN) = (0.35 × 300 × 10–9) / 0.24 = 438 nF
(35)
In expectation that more capacitance will be needed to prevent power supply interaction a 1-µF ceramic
capacitor rated to 50 V with X7R dielectric in a 1206 case size will be used. In this case, input rms current is:
IIN-RMS = 0.35 × Sqrt(0.154 × 0.846) = 126 mA
(36)
Ripple current ratings for 1206 size ceramic capacitors are typically higher than 1 A, more than enough for this
design.
8.2.2.5 Recirculating Diode
The first parameter for D1 which must be determined is the reverse voltage rating. Schottky diodes are available
at reverse ratings of 30 V and 40 V, often in the same package, with the same forward current rating. To account
for ringing a 40-V Schottky will be used.
The next parameters to be determined are the forward current rating and case size. In this example the low duty
cycle (D = 3.7 / 24 = 15%) requires the recirculating diode D1 to carry the load current much longer than the
internal power MOSFET of the LM3402. The estimated average diode current is:
ID = 0.35 × 0.85 = 298 mA
26
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Schottky diodes are available at forward current ratings of 0.5 A, however the current rating often assumes a
25°C ambient temperature and does not take into account the application restrictions on temperature rise. A
diode rated for higher current may be needed to keep the temperature rise below 40°C. To determine the proper
case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations
section. VD for a small case size such as SOD-123 in a 40 V, 0.5-A Schottky diode at 350 mA is approximately
0.4 V and the θJA is 206°C/W. Power dissipation and temperature rise can be calculated as:
PD = 0.298 × 0.4 = 119 mW TRISE = 0.119 × 206 = 24.5°C
(38)
According to these calculations the SOD-123 diode will meet the requirements. Heating and dissipation are
among the factors most difficult to predict in converter design. If possible, a footprint should be used that is
capable of accepting both SOD-123 and a larger case size, such as SMA. A larger diode with a higher forward
current rating will generally have a lower forward voltage, reducing dissipation, as well as having a lower θJA,
reducing temperature rise.
8.2.2.5.1 CB and CF
The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25-V rating is
appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100-nF ceramic
capacitor, also with X7R dielectric and a 25-V rating.
8.2.2.6 Efficiency
To estimate the electrical efficiency of this example the power dissipation in each current carrying element can
be calculated and summed. This term should not be confused with the optical efficacy of the circuit, which
depends upon the LEDs themselves.
Total output power, PO, is calculated as:
PO = IF × VO = 0.35 × 3.7 = 1.295 W
(39)
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 × RDSON) × D = (0.352 × 1.5) × 0.154 = 28 mW
(40)
Gate charging and VCC loss, PG, in the gate drive and linear regulator:
PG = (IIN-OP + fSW × QG) × VIN PG = (600 × 10–6 + 468000 × 3 × 10–9) × 24 = 48 mW
(41)
Switching loss, PS, in the internal MOSFET:
PS = 0.5 × VIN × IF × (tR + tF) × fSW PS = 0.5 × 24 × 0.35 × (40 × 10-9) × 468000 = 78 mW
(42)
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 × ESR = (0.126)2 × 0.006 = 0.1 mW (negligible)
(43)
DCR loss, PL, in the inductor
PL = IF2 × DCR = 0.352 × 0.096 = 11.8 mW
(44)
Recirculating diode loss, PD = 119 mW
Current Sense Resistor Loss, PSNS = 92 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 1.295 / (1.295 + 0.377) = 77%
8.2.2.7 Die Temperature
TLM3402 = (PC + PG + PS) × θJA TLM3402 = (0.028 + 0.05 + 0.078) × 200 = 31°C
Copyright © 2006–2015, Texas Instruments Incorporated
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(45)
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8.2.3 Application Curves
Top = SW
Middle = DIM
Bottom = LED
Current
Figure 29. PWM Dimming, DIM Rising
28
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Top = SW
Middle = DIM
Bottom = LED
Current
Figure 30. PWM Dimming, DIM Falling
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM3402 LM3402HV
LM3402, LM3402HV
www.ti.com
SNVS450F – SEPTEMBER 2006 – REVISED OCTOBER 2015
9 Power Supply Recommendations
Use any DC output power supply with a maximum voltage high enough for the application. The power supply
should have a minimum current limit of at least 500 mA.
10 Layout
10.1 Layout Guidelines
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and
minimum generation of unwanted EMI.
10.1.1 Compact Layout
Parasitic inductance can be reduced by keeping the power path components close together and keeping the
area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In
particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all
three components without excessive heating from the current it carries. The LM3402/HV operates in two distinct
cycles whose high current paths are shown in Figure 22:
+
-
Figure 31. Buck Converter Current Loops
The dark grey, inner loop represents the high current path during the MOSFET ON-time. The light grey, outer
loop represents the high current path during the off-time.
10.1.2 Ground Plane and Shape Routing
The diagram of Figure 22 is also useful for analyzing the flow of continuous current vs the flow of pulsating
currents. The circuit paths with current flow during both the ON-time and off-time are considered to be continuous
current, while those that carry current during the ON-time or off-time only are pulsating currents. Preference in
routing should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit
EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane
with less risk of injecting noise into other circuits. The path between the input source and the input capacitor and
the path between the recirculating diode and the LEDs/current sense resistor are examples of continuous current
paths. In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating
current. This path should be routed with a short, thick shape, preferably on the component side of the PCB.
Multiple vias in parallel should be used right at the pad of the input capacitor to connect the component side
shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed
by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CB close to the
SW and BOOT pins.
10.1.3 Current Sensing
The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should
be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible
to the CS and GND pins of the IC.
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Layout Guidelines (continued)
10.1.4 Remote LED Arrays
In some applications the LED or LED array can be far away (several inches or more) from the LM3402/HV, or on
a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or
separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce the
effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should remain
on the same PCB, close to the LM3402/HV.
10.2 Layout Example
GND
LED+
D1
L1
CIN
+
SW
VIN
CB
LED-
BOOT
VCC
CF
RSNS
DIM
VIN
RON
RON
GND
CS
THERMAL/POWER VIA
Figure 32. Layout Recommendation
30
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SNVS450F – SEPTEMBER 2006 – REVISED OCTOBER 2015
11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 3. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
LM3402
Click here
Click here
Click here
Click here
Click here
LM3402HV
Click here
Click here
Click here
Click here
Click here
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
PowerPAD is a trademark of Texas Instruments .
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM3402HVMM/NOPB
ACTIVE
VSSOP
DGK
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SNFB
LM3402HVMMX/NOPB
ACTIVE
VSSOP
DGK
8
3500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SNFB
LM3402HVMR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
RoHS & Green
SN
Level-3-260C-168 HR
L3402
HVMR
LM3402HVMRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-3-260C-168 HR
L3402
HVMR
LM3402MM/NOPB
ACTIVE
VSSOP
DGK
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SNEB
LM3402MMX/NOPB
ACTIVE
VSSOP
DGK
8
3500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SNEB
LM3402MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
RoHS & Green
SN
Level-3-260C-168 HR
L3402
MR
LM3402MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-3-260C-168 HR
L3402
MR
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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