ADC1173CIMTCX/NOPB

ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 ADC1173 8-Bit, 3-Volt, 15MSPS, 33mW A/D Converter Check for Samples: ADC1173 FEATURES DESCRIPTION • • • • The ADC1173 is a low power, 15 MSPS analog-todigital converter that digitizes signals to 8 bits while consuming just 33 mW of power (typ). The ADC1173 uses a unique architecture that achieves 7.6 Effective Bits. Output formatting is straight binary coding. 1 2 Internal Sample-and-Hold Function Single +3V Operation Internal Reference Bias Resistors Industry Standard Pinout APPLICATIONS • • • • • • • • Video Digitization Digital Still Cameras Set Top Boxes Camcorders Personal Computer Video Digital Television CCD Imaging Electro-Optics The excellent DC and AC characteristics of this device, together with its low power consumption and +3V single supply operation, make it ideally suited for many video, imaging and communications applications, including use in portable equipment. Furthermore, the ADC1173 is resistant to latch-up and the outputs are short-circuit proof. The top and bottom of the ADC1173's reference ladder is available for connections, enabling a wide range of input possibilities. The ADC1173 is offered in a 24-pin TSSOP package and is designed to operate over the -40°C to +75°C commercial temperature range. KEY SPECIFICATIONS • • • • • • • • Resolution 8 Bits Maximum Sampling Frequency 15 MSPS (min) THD −54 dB (typ) DNL ±0.85 LSB (max) ENOB at 3.58 MHz Input 7.6 Bits (typ) Differential Phase 0.5 Degree (max) Differential Gain 1.5% (typ) Power Consumption 33 mW (typ) (excluding reference current) PIN CONFIGURATION Figure 1. 24-Pin TSSOP (Top View) See PW Package 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1999–2013, Texas Instruments Incorporated ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com BLOCK DIAGRAM 2 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 PIN DESCRIPTIONS AND EQUIVALENT CIRCUITS Pin No. Symbol Equivalent Circuit Description 19 VIN Analog signal input. Conversion range is VRB to VRT. 16 VRTS Reference Top Bias with internal pull-up resistor. Short this pin to VRT to self bias the reference ladder. 17 VRT Analog Input that is the high (top) side of the reference ladder of the ADC. Nominal range is 1.0V to AVDD. Voltage on VRT and VRB inputs define the VIN conversion range. Bypass well. For more information, see REFERENCE INPUTS. 23 VRB Analog Input that is the low (bottom) side of the reference ladder of the ADC. Nominal range is 0V to 2.0V. Voltage on VRT and VRB inputs define the VIN conversion range. Bypass well. For more information, see REFERENCE INPUTS. 22 VRBS Reference Bottom Bias with internal pull down resistor. Short to VRB to self bias the reference ladder. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 3 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com PIN DESCRIPTIONS AND EQUIVALENT CIRCUITS (continued) Pin No. Symbol Equivalent Circuit Description 1 OE CMOS/TTL compatible Digital input that, when low, enables the digital outputs of the ADC1173. When high, the outputs are in a high impedance state. 12 CLK CMOS/TTL compatible digital clock Input. VIN is sampled on the falling edge of CLK input. 3 thru 10 D0-D7 Conversion data digital Output pins. D0 is the LSB, D7 is the MSB. Valid data is output just after the rising edge of the CLK input. These pins are enabled by bringing the OE pin low. 11, 13 DVDD Positive digital supply pin. Connect to a clean, quiet voltage source of +3V. AVDD and DVDD should have a common source and be separately bypassed with a 10µF capacitor and a 0.1µF ceramic chip capacitor. For more information, see POWER SUPPLY CONSIDERATIONS. 2, 24 DVSS The ground return for the digital supply. AVSS and DVSS should be connected together close to the ADC1173. 14, 15, 18 AVDD Positive analog supply pin. Connected to a clean, quiet voltage source of +3V. AVDD and DVDD should have a common source and be separately bypassed with a 10 µF capacitor and a 0.1 µF ceramic chip capacitor. For more information, see POWER SUPPLY CONSIDERATIONS. 20, 21 AVSS The ground return for the analog supply. AVSS and DVSS should be connected together close to the ADC1173 package. These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 4 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 ABSOLUTE MAXIMUM RATINGS (1) (2) (3) AVDD, DVDD 6.5V −0.3V to 6.5V Voltage on Any Pin VRT, VRB AVDD to VSS −0.5 to (AVDD + 0.5V) CLK, OE Voltage Digital Output Voltage Input Current DVSS to DVDD (4) ±25mA Package Input Current (4) ±50mA Package Dissipation at 25°C ESD Susceptibility (6) See Human Body Model 2000V Machine Model 200V Soldering Temp., Infrared, 10 sec. 300°C −65°C to +150°C Storage Temperature (1) (2) (3) (4) (5) (6) (5) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may degrade when the device is not operated under the listed test conditions. All voltages are measured with respect to GND = AVSS = DVSS = 0V, unless otherwise specified. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and specifications. When the input voltage at any pin exceeds the power supplies (that is, less than AVSS or DVSS, or greater than AVDD or DVDD), the current at that pin should be limited to 25 mA. The 50 mA maximum package input current rating limits the number of pins that can safely exceed the power supplies with an input current of 25 mA to two. The absolute maximum junction temperatures (TJmax) for this device is 150°C. The maximum allowable power dissipation is dictated by TJmax, the junction-to-ambient thermal resistance θJA, and the ambient temperature, TA, and can be calculated using the formula PDMAX = (TJmax - TA )/θJA. The power dissipation of this device under normal operation will typically be much lower than that required to raise the junction temperature enough to be a problem. The values for maximum power dissipation listed above will be reached only when the ADC1173 is operated in a severe fault condition (e.g. when input or output pins are driven beyond the power supply voltages, or the power supply polarity is reversed). Obviously, such conditions should always be avoided. Human body model is 100 pF capacitor discharged through a 1.5kΩ resistor. Machine model is 220 pF discharged through ZERO Ω. OPERATING RATINGS (1) (2) −40°C ≤ TA ≤ +75°C Temperature Range AVDD, DVDD +2.7V to +3.6V |AVSS -DVSS| 0V to 100 mV VRT 1.0V to AVDD VRB 0V to 2.0V VRT - VRB 1.0V to 2.8V VIN Voltage Range (1) (2) VRB to VRT Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may degrade when the device is not operated under the listed test conditions. All voltages are measured with respect to GND = AVSS = DVSS = 0V, unless otherwise specified. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 5 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com CONVERTER ELECTRICAL CHARACTERISTICS The following specifications apply for AVDD = DVDD = +3.0VDC, OE = 0V, VRT = +2.0V, VRB = 0V, CL = 20 pF, fCLK = 15MHz at 50% duty cycle. Boldface limits apply for TA = TMIN to TMAX; all other limits TA = 25°C (1) (2) Symbol Parameter Conditions Typical (3) Limits Units DC Accuracy INL Integral Non Linearity ±0.5 ±1.3 LSB ( max) DNL Differential Non Linearity ±0.4 ±0.85 LSB ( max) 0 (max) Missing Codes EOT Top Offset −12 mV EOB Bottom Offset +1.0 mV Video Accuracy DP Differential Phase Error fin = 3.58 MHz sine wave 0.5 Degree DG Differential Gain Error fin = 3.58 MHz sine wave 1.5 % Analog Input and Reference Characteristics V (min) V (max) Input Range CIN VIN Input Capacitance RIN Input Resistance >1 MΩ BW Analog Input Bandwidth 120 MHz RRT Top Reference Resistor 360 RREF Reference Ladder Resistance RRB Bottom Reference Resistor IREF Reference Ladder Current 2.0 VRB VRT VIN VIN = 1.5V + 0.7Vrms VRT to VRB 4 11 300 pF Ω 200 Ω (min) 400 Ω (max) 90 Ω VRT =VRTS, VRB =VRBS 4.2 mA VRT =VRTS,VRB =AVSS 4.8 VRT VRT connected to VRTS Reference Top Self Bias Voltage VRB connected to VRBS VRB Reference Bottom Self Bias Voltage (1) (CLK LOW) (CLK HIGH) VRT connected to VRTS VRB connected to VRBS 1.56 0.36 mA 1.45 1.65 V (min) V (max) 0.32 V (min) 0.40 V (max) The analog inputs are protected as shown below. Input voltage magnitudes up to 6.5V or to 500 mV below GND will not damage this device. However, errors in the A/D conversion can occur if the input goes above VDD or below GND by more than 50 mV. As an example, if AVDD is 2.7VDC, the full-scale input voltage must be ≤2.75VDC to ensure accurate conversions. addf addf (2) (3) 6 To ensure accuracy, it is required that AVDD and DVDD be well bypassed. Each supply pin must be decoupled with separate bypass capacitors. Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are ensured to TI's AOQL (Average Outgoing Quality Level). Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 CONVERTER ELECTRICAL CHARACTERISTICS (continued) The following specifications apply for AVDD = DVDD = +3.0VDC, OE = 0V, VRT = +2.0V, VRB = 0V, CL = 20 pF, fCLK = 15MHz at 50% duty cycle. Boldface limits apply for TA = TMIN to TMAX; all other limits TA = 25°C (1) (2) Symbol Parameter VRTS VRBS Self Bias Voltage Delta VRT - VRB Reference Voltage Delta Conditions Typical VRT connected to VRTS, VRB connected to VRBS 1.2 VRT connected to VRTS, VRB connected to VSS 1.38 2 (3) Limits Units 1.1 1.3 µA (min) µA (max) V 1.0 V (min) VA V (max) Power Supply Characteristics IADD Analog Supply Current DVDD = AVDD = 3.6V 6.8 IDDD Digital Supply Current DVDD = AVDD = 3.6V 2.3 DVDD AVDD = 3.6V, 9.1 IAVDD + IDVDD Total Operating Current Power Consumption DVDD = AVDD = 3.6V, CLK Low (4) DVDD = AVDD = 3.6V mA mA 11.4 mA 41 mW 5.8 33 mA CLK, OE Digital Input Characteristics VIH Logical High Input Voltage DVDD = AVDD = 3.6V 2.2 V (min) VIL Logical Low Input Voltage DVDD = AVDD = 3.6V 0.8 V (max) IIH Logical High Input Current VIH = DVDD = AVDD = 3.6V 5 µA IIL Logic Low Input Current VIL = 0V, DVDD = AVDD = 3.6V −5 µA CIN Logic Input Capacitance 5 pF Digital Output Characteristics DVDD = 2.7V, IOH = −360µA 2.4 DVDD = 2.7V, IOH = −1.1mA 2.1 1.9 V (min) Low Level Output Voltage DVDD = 2.7V, IOL = 1.6mA 0.32 0.6 V (max) TRI-STATE Leakage Current DVDD = 3.6V, OE = DVDD, VOL = 0V or VOH = DVDD ±20 VOH High Level Output Voltage VOL IOZH, IOZL V (min) µA AC Electrical Characteristics fC1 Maximum Conversion Rate 20 fC2 Minimum Conversion Rate 1 MHz 28 ns tOD Output Delay CLK rise to data rising CLK rise to data falling Pipeline Delay (Latency) MHz (min) 24 ns 2.5 Clock Cycles 3 ns tDS Sampling (Aperture) Delay tAJ Aperture Jitter 30 ps rms tOH Output Hold Time CLK high to data invalid 15 ns tEN OE Low to Data Valid Loaded as in Figure 25 22 ns tDIS OE High to High Z State Loaded as in Figure 25 12 ns ENOB Effective Number of Bits fIN = 1.31 MHz fIN = 3.58 MHz fIN = 7.5 MHz 7.7 7.6 7.4 7.0 Bits (min) SINAD Signal-to- Noise & Distortion fIN = 1.31 MHz fIN = 3.58 MHz fIN = 7.5 MHz 49 47.7 46.5 43 dB (min) SNR Signal-to-Noise Ratio fIN = 1.31 MHz fIN = 3.58 MHz fIN = 7.5 MHz 49 48.7 48.0 44 dB (min) SFDR Spurious Free Dynamic Range fIN = 1.31 MHz fIN = 3.58 MHz fIN = 7.5 MHz 65 55 51 (4) CLK low to acquisition of data 15 dB At least two clock cycles must be presented to the ADC1173 after power up. For details, see THE ADC1173 CLOCK. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 7 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com CONVERTER ELECTRICAL CHARACTERISTICS (continued) The following specifications apply for AVDD = DVDD = +3.0VDC, OE = 0V, VRT = +2.0V, VRB = 0V, CL = 20 pF, fCLK = 15MHz at 50% duty cycle. Boldface limits apply for TA = TMIN to TMAX; all other limits TA = 25°C (1) (2) Symbol THD 8 Parameter Total Harmonic Distortion Conditions fIN = 1.31 MHz fIN = 3.58 MHz fIN = 7.5 MHz Submit Documentation Feedback Typical −62 −54 −51 (3) Limits Units dB Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 TYPICAL PERFORMANCE CHARACTERISTICS INL vs. Temperature DNL vs. Temperature Figure 2. Figure 3. SNR vs. Temperature SNR vs. fIN Figure 4. Figure 5. THD vs. Temperature SINAD vs. Temperature Figure 6. Figure 7. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 9 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) 10 SINAD vs. fIN SFDR vs. Temperature Figure 8. Figure 9. SFDR vs. fIN Differential Gain vs. Temperature Figure 10. Figure 11. Differential Phase vs. Temperature SNR vs. fIN Figure 12. Figure 13. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 TYPICAL PERFORMANCE CHARACTERISTICS (continued) THD vs. fIN SINAD vs. fIN Figure 14. Figure 15. SFDR vs. fIN SNR vs. SUPPLY VOLTAGE Figure 16. Figure 17. THD vs. SUPPLY VOLTAGE SINAD vs. SUPPLY VOLTAGE Figure 18. Figure 19. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 11 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) 12 SFDR vs. SUPPLY VOLTAGE IDDD + IADD vs. fCLK Figure 20. Figure 21. TOD vs. Temperature Spectral Response Figure 22. Figure 23. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 SPECIFICATION DEFINITIONS ANALOG INPUT BANDWIDTH is a measure of the frequency at which the reconstructed output fundamental drops 3 dB below its low frequency value for a full scale input. The test is performed with fIN equal to 100 kHz plus integer multiples of fCLK. The input frequency at which the output is −3 dB relative to the low frequency input signal is the full power bandwidth. APERTURE JITTER is the time uncertainty of the sampling point (tDS), or the range of variation in the sampling delay. BOTTOM OFFSET is the difference between the input voltage that just causes the output code to transition to the first code and the negative reference voltage. Bottom offset is defined as EOB = VZT - VRB, where VZT is the first code transition input voltage. Note that this is different from the normal Zero Scale Error. DIFFERENTIAL GAIN ERROR is the percentage difference between the output amplitudes of a high frequency reconstructed sine wave at two different DC levels. DIFFERENTIAL NON-LINEARITY (DNL) is the measure of the maximum deviation from the ideal step size of 1 LSB. DIFFERENTIAL PHASE ERROR is the difference in the output phase of a reconstructed small signal sine wave at two different DC levels. EFFECTIVE NUMBER OF BITS (ENOB, or EFFECTIVE BITS) is another method of specifying Signal-to-Noise and Distortion Ratio, or SINAD. ENOB is defined as (SINAD - 1.76) / 6.02 and says that the converter is equivalent to a perfect ADC of this (ENOB) number of bits. INTEGRAL NON-LINEARITY (INL) is a measure of the deviation of each individual code from a line drawn from zero scale (½LSB below the first code transition) through positive full scale (½LSB above the last code transition). The deviation of any given code from this straight line is measured from the center of that code value. The end point test method is used. OUTPUT DELAY is the time delay after the rising edge of the input clock before the data update is present at the output pins. OUTPUT HOLD TIME is the length of time that the output data is valid after the rise of the input clock. PIPELINE DELAY (LATENCY) is the number of clock cycles between initiation of conversion and the availability of that conversion result at the output. New data is available at every clock cycle, but the data lags the conversion by the pipeline delay. SAMPLING (APERTURE) DELAY is that time required after the fall of the clock input for the sampling switch to open. The Sample/Hold circuit effectively stops capturing the input signal and goes into the "hold" mode tDS after the clock goes low. SIGNAL TO NOISE RATIO (SNR) is the ratio of the rms value of the input signal to the rms value of the other spectral components below one-half the sampling frequency, not including harmonics or DC. SIGNAL TO NOISE PLUS DISTORTION (S/(N+D) or SINAD) Is the ratio of the rms value of the input signal to the rms value of all of the other spectral components below half the clock frequency, including harmonics but excluding DC. SPURIOUS FREE DYNAMIC RANGE (SFDR) is the difference, expressed in dB, between the rms values of the input signal and the peak spurious signal, where a spurious signal is any signal present in the output spectrum that is not present at the input. TOP OFFSET is the difference between the positive reference voltage and the input voltage that just causes the output code to transition to full scale and is defined as EOT = VFT − VRT. Where VFT is the full scale transition input voltage. Note that this is different from the normal Full Scale Error. TOTAL HARMONIC DISTORTION (THD) is the ratio of the rms total of the first six harmonic components, to the rms value of the input signal. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 13 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com Timing Diagram Figure 24. Timing Diagram Figure 25. tEN , tDIS Test Circuit 14 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 FUNCTIONAL DESCRIPTION The ADC1173 uses a new, unique architecture to achieve 7.4 effective bits at and maintains superior dynamic performance up to ½ the clock frequency. The analog signal at VIN that is within the voltage range set by VRT and VRB are digitized to eight bits at up to 20 MSPS. Input voltages below VRB will cause the output word to consist of all zeroes. Input voltages above VRT will cause the output word to consist of all ones. VRT has a range of 1.0 Volt to the analog supply voltage, AVDD, while VRB has a range of 0 to 2.0 Volts. VRT should always be at least 1.0 Volt more positive than VRB. If VRT and VRTS are connected together and VRB and VRBS are connected together, the nominal values of VRT and VRB are 1.56V and 0.36V, respectively. If VRT and VRTS are connected together and VRB is grounded, the nominal value of VRT is 1.38V. Data is acquired at the falling edge of the clock and the digital equivalent of the data is available at the digital outputs the pipeline delay (2.5 clock cycles) plus tOD later. The ADC1173 will convert as long as the clock signal is present at pin 12. The Output Enable pin OE, when low, enables the output pins. The digital outputs are in the high impedance state when the OE pin is high. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 15 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com APPLICATIONS INFORMATION THE ANALOG INPUT The analog input of the ADC1173 is a switch followed by an integrator. The input capacitance changes with the clock level, appearing as 4 pF when the clock is low, and 11 pF when the clock is high. Since a dynamic capacitance is more difficult to drive than a fixed capacitance, choose an amplifier that can drive this type of load. The LMH6702, LMH6609, LM6152, LM6154, LM6181 and LM6182 have been found to be excellent devices for driving the ADC1173. Do not drive the input beyond the supply rails. shows an example of an input circuit using the LM6181. This circuit has both gain and offset adjustments. If you desire to eliminate these adjustments, you should reduce the signal swing to avoid clipping at the ADC1173 output that can result from normal tolerances of all system components. With no adjustments, the nominal value for the amplifier feedback resistor is 510Ω and the 5.1k resistor at the inverting input should be changed to 860Ω and returned to +3V rather than to the Offset Adjust potentiometer. Driving the analog input with input signals up to 2.8VP-P will result in normal behavior where voltages above VRT will result in a code of FFh and input voltages below VRB will result in an output code of zero. Input signals above 2.8V P-P may result in odd behavior where the output code is not FFh when the input exceeds VRT. REFERENCE INPUTS The reference inputs VRT (Reference Top) and VRB (Reference Bottom) are the top and bottom of the reference ladder. Input signals between these two voltages will be digitized to 8 bits. External voltages applied to the reference input pins should be within the range specified in the Operating Ratings table (1.0V to AVDD for VRT and 0V to (AVDD - 1.0V) for VRB). Any device used to drive the reference pins should be able to source sufficient current into the VRT pin and sink sufficient current from the VRB pin. The reference ladder can be self-biased by connecting VRT to VRTS and connecting VRB to VRBS to provide top and bottom reference voltages of approximately 1.56V and 0.36V, respectively, with VCC = 3.0V. This connection is shown in Figure 26. If VRT and VRTS are tied together, but VRB is tied to analog ground, a top reference voltage of approximately 1.38V is generated. The top and bottom of the ladder should be bypassed with 10µF tantalum capacitors located close to the reference pins. 16 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 Because of resistor tolerances, the reference voltages can vary by as much as 6%. Choose an amplifier that can drive a dynamic capacitance (see text). Figure 26. Simple, Low Component Count, Self -Bias Reference application. The reference self-bias circuit of is very simple and performance is adequate for many applications. Superior performance can generally be achieved by driving the reference pins with a low impedance source. By forcing a little current into or out of the top and bottom of the ladder, as shown in , the top and bottom reference voltages can be trimmed. The resistive divider at the amplifier inputs can be replaced with potentiometers. The LMC662 amplifier shown was chosen for its low offset voltage and low cost. Note that a negative power supply is needed for these amplifiers if their outputs are required to go slightly negative to force the required reference voltages. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 17 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com Self-bias is still used, but the reference voltages are trimmed by providing a small trim current with the operational amplifiers. Figure 27. Better defining the ADC Reference Voltage. 18 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 Driving the reference to force desired values requires driving with a low impedance source, provided by the transistors. Note that pins 16 and 22 are not connected. Figure 28. Driving with a low impedance source If reference voltages are desired that are more than a few tens of millivolts from the self-bias values, the circuit of will allow forcing the reference voltages to whatever levels are desired. This circuit provides the best performance because of the low source impedance of the transistors. Note that the VRTS and VRBS pins are left floating. VRT can be anywhere between VRB + 1.0V and the analog supply voltage, and VRB can be anywhere between ground and 1.0V below VRT. To minimize noise effects and ensure accurate conversions, the total reference voltage range (VRT - VRB) should be a minimum of 1.0V and a maximum of about VA. Best performance can be realized with VRT= 1.56 and VRB= 0.36V. If VRB is not required to be below about +700mV, the -5V points in can be returned to ground and the negative supply eliminated. POWER SUPPLY CONSIDERATIONS Many A/D converters draw sufficient transient current to corrupt their own power supplies if not adequately bypassed. A 10µF tantalum or aluminum electrolytic capacitor should be placed within an inch (2.5 centimeters) of the A/D power pins, with a 0.1 µF ceramic chip capacitor placed as close as possible to the converter's power supply pins. Leadless chip capacitors are preferred because they have low lead inductance. While a single voltage source should be used for the analog and digital supplies of the ADC1173, these supply pins should be well isolated from each other to prevent any digital noise from being coupled to the analog power pins. A 47 Ohm resistor is recommend between the analog and digital supply lines, with a ceramic capacitor close to the analog supply pin. Avoid inductive components in the analog supply line. The converter digital supply should not be the supply that is used for other digital circuitry on the board. It should be the same supply used for the A/D analog supply. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 19 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com As is the case with all high speed converters, the ADC1173 should be assumed to have little a.c. power supply rejection, especially when self-biasing is used by connecting VRT and VRTS together. No pin should ever have a voltage on it that is in excess of the supply voltages or below ground, not even on a transient basis. This can be a problem upon application of power to a circuit. Be sure that the supplies to circuits driving the CLK, OE, analog input and reference pins do not come up any faster than does the voltage at the ADC1173 power pins. Pins 11 and 13 are both labeled DVDD. Pin 11 is the supply point for the digital core of the ADC, where pin 13 is used only to provide power to the ADC output drivers. As such, pin 11 may be connected to a voltage source that is less than the +5V used for AVDD and DVDD to ease interfacing to low voltage devices. Pin 11 should never exceed the pin 13 potential by more than 0.5V. Note that tOD will increase for lower pin 11 voltages. THE ADC1173 CLOCK Although the ADC1173 is tested and its performance is ensured with a 15MHz clock, it typically will function with clock frequencies from 1MHz to 20MHz. If continuous conversions are not required, power consumption can be reduced somewhat by stopping the clock at a logic low when the ADC1173 is not being used. This reduces the current drain in the ADC1173's digital circuitry from a typical value of 2.3mA to about 100µA. Note that powering up the ADC1173 without the clock running may not save power, as it will result in an increased current flow (by as much as 170%) in the reference ladder. In some cases, this may increase the ladder current above the specified limit. Toggling the clock twice at 1MHz or higher and returning it to the low state will eliminate the excess ladder current. An alternative power-saving technique is to power up the ADC1173 with the clock active, then halt the clock in the low state after two or more clock cycles. Stopping the clock in the high state is not recommended as a power-saving technique. LAYOUT AND GROUNDING Proper grounding and proper routing of all signals is essential to ensure accurate conversion. Separate analog and digital ground planes that are connected beneath the ADC1173 may be used, but best EMI practices require a single ground plane. However, it is important to keep analog signal lines away from digital signal lines and away from power supply currents. This latter requirement requires the careful separation and placement of power planes. The use of power traces rather than one or more power planes is not recommended as higher frequencies are not well filtered with lumped capacitances. To filter higher frequency noise, it is necessary to have sufficient capacitance between the power and ground planes. If separate analog and digital ground planes are used, the analog and digital grounds should be in the same layer, but should be separated from each other. If separate analog and digital ground layers are used, they should never overlap each other. Capacitive coupling between a typically noisy digital ground plane and the sensitive analog circuitry can lead to poor performance that may seem impossible to isolate and remedy. The solution is to keep the analog circuity well separated from the digital circuitry. Digital circuits create substantial supply and ground current transients. The logic noise thus generated could have significant impact upon system noise performance. The best logic family to use in systems with A/D converters is one which employs non-saturating transistor designs, or has low noise characteristics, such as the 74HC(T) and 74AC(T)Q families. The worst noise generators are logic families that draw the largest supply current transients during clock or signal edges, like the 74F and the 74AC(T) families. In general, slower logic families, such as 74LS and 74HC(T), will produce less high frequency noise than do high speed logic families. Since digital switching transients are composed largely of high frequency components, total ground plane copper weight will have little effect upon the logic-generated nose. This is because of the skin effect. Total surface area is more important that is total ground plane volume. An effective way to control ground noise is by using a single, solid ground plane, splitting the power plane into analog and digital areas and to have power and ground planes in adjacent board layers. There should be no traces within either the power or the ground layers of the board. The analog and digital power planes should reside in the same board layer so that they can not overlap each other. The analog and digital power planes define the analog and digital areas of the board. 20 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 Generally, analog and digital lines should cross each other at 90 degrees to avoid getting digital noise into the analog path. In high frequency systems, however, avoid crossing analog and digital lines altogether. Clock lines should be isolated from ALL other lines, analog and digital. Even the generally accepted 90 degree crossing should be avoided as even a little coupling can cause problems at high frequencies. Best performance at high frequencies and at high resolution is obtained with a straight signal path. Be especially careful with the layout of inductors. Mutual inductance can change the characteristics of the circuit in which they are used. Inductors should not be placed side by side, not even with just a small part of their bodies being beside each other. The analog input should be isolated from noisy signal traces to avoid coupling of spurious signals into the input. Any external component (e.g., a filter capacitor) connected between the converter's input and ground should be connected to a very clean point in the analog ground return. DYNAMIC PERFORMANCE The ADC1173 is AC tested and its dynamic performance is ensured. To meet the published specifications, the clock source driving the CLK input must be free of jitter. For best a.c. performance, isolating the ADC clock from any digital circuitry should be done with adequate buffers, as with a clock tree. See Figure 29. Figure 29. Isolating the ADC Clock From Digital Circuitry It is good practice to keep the ADC clock line as short as possible and to keep it well away from any other signals. Other signals can introduce jitter into the clock signal. COMMON APPLICATION PITFALLS Driving the inputs (analog or digital) beyond the power supply rails. For proper operation, all inputs should not go more than 50mV below the ground pins or 50mV above the supply pins. Exceeding these limits on even a transient basis can cause faulty or erratic operation. It is not uncommon for high speed digital circuits (e.g., 74F and 74AC devices) to exhibit undershoot that goes more than a volt below ground. A resistor of 50Ω in series with the offending digital input will usually eliminate the problem. Care should be taken not to overdrive the inputs of the ADC1173. Such practice may lead to conversion inaccuracies and even to device damage. Attempting to drive a high capacitance digital data bus. The more capacitance the output drivers must charge for each conversion, the more instantaneous digital current is required from DVDD and DGND. These large charging current spikes can couple into the analog section, degrading dynamic performance. Buffering the digital data outputs (with an 74ACQ541, for example) may be necessary if the data bus to be driven is heavily loaded. Dynamic performance can also be improved by adding 47Ω to 100Ω series resistors at each digital output, reducing the energy coupled back into the converter output pins. Using an inadequate amplifier to drive the analog input. As explained in THE ANALOG INPUT, the capacitance seen at the input alternates between 4 pF and 11 pF with the clock. This dynamic capacitance is more difficult to drive than is a fixed capacitance, and should be considered when choosing a driving device. The LMH6702, LM6152, LM6154, LM6181 and LM6182 have been found to be excellent devices for driving the ADC1173 analog input. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 21 ADC1173 SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 www.ti.com Driving the VRT pin or the VRB pin with devices that can not source or sink the current required by the ladder. As mentioned in REFERENCE INPUTS, care should be taken to see that any driving devices can source sufficient current into the VRT pin and sink sufficient current from the VRB pin. If these pins are not driven with devices than can handle the required current, these reference pins will not be stable, resulting in a reduction of dynamic performance. Using a clock source with excessive jitter, using an excessively long clock signal trace, or having other signals coupled to the clock signal trace. This will cause the sampling interval to vary, causing excessive output noise and a reduction in SNR performance. Simple gates with RC timing is generally inadequate as a clock source. Input test signal contains harmonic distortion that interferes with the measurement of dynamic signal to noise ratio. Harmonic and other interfering signals can be removed by inserting a filter at the signal input. Suitable filters are shown in Figure 30 and Figure 31. The circuit of Figure 30 has cutoff of about 5.5 MHz and is suitable for input frequencies of 1 MHz to 5 MHz. The circuit of Figure 31 has a cutoff of about 11 MHz and is suitable for input frequencies of 5 MHz to 10 MHz. These filters should be driven by a generator of 75 Ohm source impedance and terminated with a 75 ohm resistor. Figure 30. 5.5 MHz Low Pass Filter to Eliminate Harmonics at the Signal Input Use at Input Frequencies of 5 MHz to 10 MHz Figure 31. 11 MHz Low Pass filter to Eliminate Harmonics at the Signal Input. 22 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 ADC1173 www.ti.com SNAS025F – FEBRUARY 1999 – REVISED APRIL 2013 REVISION HISTORY Changes from Revision E (April 2013) to Revision F • Page Changed layout of National Data Sheet to TI format .......................................................................................................... 22 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: ADC1173 23 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) ADC1173CIMTC/NOPB ACTIVE TSSOP PW 24 61 RoHS & Green SN Level-1-260C-UNLIM -40 to 75 ADC1173 CIMTC ADC1173CIMTCX/NOPB ACTIVE TSSOP PW 24 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 75 ADC1173 CIMTC (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of