ADC12DL065
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ADC12DL065 Dual 12-Bit, 65 MSPS, 3.3V, 360mW A/D Converter
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FEATURES
DESCRIPTION
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The ADC12DL065 is a dual, low power monolithic
CMOS analog-to-digital converter capable of
converting analog input signals into 12-bit digital
words at 65 Megasamples per second (MSPS). This
converter uses a differential, pipeline architecture with
digital error correction and an on-chip sample-andhold circuit to minimize power consumption while
providing excellent dynamic performance and a 250
MHz Full Power Bandwidth. Operating on a single
+3.3V power supply, the ADC12DL065 achieves 11.0
effective bits at nyquist and consumes just 360 mW
at 65 MSPS, including the reference current. The
Power Down feature reduces power consumption to
36 mW.
1
2
Single +3.3V Supply operation
Internal Sample-and-Hold
Internal Reference
Outputs 2.4V to 3.6V Compatible
Power Down Mode
Duty Cycle Stabilizer
Multiplexed Output Mode
APPLICATIONS
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Ultrasound and Imaging
Instrumentation
Communications Receivers
Sonar/Radar
xDSL
Cable Modems
DSP Front Ends
KEY SPECIFICATIONS
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Resolution: 12 Bits
DNL: ±0.4 LSB (typ)
SNR (fIN = 10 MHz): 69 dB (typ)
SFDR (fIN = 10 MHz): 86 dB (typ)
Data Latency: 7 Clock Cycles
Power Consumption
– Operating: 360 mW (typ)
– Power Down Mode: 36 mW (typ)
The differential inputs provide a full scale differential
input swing equal to 2 times VREF with the possibility
of a single-ended input. Full use of the differential
input is recommended for optimum performance. The
digital outputs from the two ADC's are available on a
single multiplexed 12-bit bus or on separate buses.
Duty cycle stabilization and output data format are
selectable using a quad state function pin. The output
data can be set for offset binary or two's complement.
To ease interfacing to lower voltage systems, the
digital output driver power pins of the ADC12DL065
can be connected to a separate supply voltage in the
range of 2.4V to the analog supply voltage. This
device is available in the 64-lead TQFP package and
will operate over the industrial temperature range of
−40°C to +85°C. An evaluation board is available to
ease the evaluation process.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005–2013, Texas Instruments Incorporated
ADC12DL065
SNAS249D – MARCH 2005 – REVISED APRIL 2013
www.ti.com
50
51
49
DGND
DR GND
52
DB6
VDR
54
55
56
53
DB7
DB8
DB9
DB10
58
59
60
57
DB11
DR GND
PD
CLK
62
63
61
AGND
VA
AGND
VA
64
Connection Diagram
1
VINB-
VD
48
2
VINB+
DB5
47
3
AGND
DB4
46
VRMB
DB3
45
VRPB
DB2
44
VRNB
DB1
43
VREF
DB0/ABb
42
OEB
41
DR GND
40
4
5
6
7
8
9
AGND
VA
ADC12DL065
DA8
36
VRMA
DA7
35
VINA+
DA6
34
VD
33
32
DGND
DR GND
31
30
29
28
27
26
25
24
23
22
19
18
17
21
VINAAGND
VA
VA
16
AGND
15
20
14
DA5
VDR
37
VRPA
13
DA4
DA9
DA3
38
VRNA
12
DA2
DA10
DA1
MULTIPLEX
DA0
39
11
DR GND
DA11
OEA
AGND
DF/DCS
10
Figure 1. TQFP Package
See Package Number PAG0064A
2
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Block Diagram
VINA+
S/H
Stage 1
Stage 2
Stage 3
Stage 8
Stage 9
Stage 10
VINA3
2
2
2
2
2
2
2
2
2
2
2
Timing
Control
10-Stage Pipeline Converter
MULTIPLEX
21
12
12
Digital
Correction
CLK
MUX
DA0-DA11 or
D0-D11 (MUX)
Output
Buffers
OEA
Duty Cycle
Stabilizer
2
VRPA
VRMA
VRNA
Reference
Select
VREF
Internal
Reference
VRPB
VRMB
VRNB
11
Digital
Correction
2
12
DB1-DB11
DB0/ABb
OEB
Output
Buffers
21
10-Stage Pipeline Converter
Timing
Control
2
2
2
3
2
2
2
2
2
2
2
2
VINB+
S/H
Stage 1
Stage 2
Stage 3
Stage 8
Stage 9
Stage 10
VINB-
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Pin Descriptions and Equivalent Circuits
Pin No.
Symbol
Equivalent Circuit
Description
ANALOG I/O
15
2
VINA+
VINB+
16
1
VINA−
VINB−
VA
Differential analog input pins. With a 1.0V reference voltage the
differential full-scale input signal level is 2.0 VP-P with each input pin
voltage centered on a common mode voltage, VCM. The negative
input pins may be connected to VCM for single-ended operation, but
a differential input signal is required for best performance.
AGND
VA
7
This pin is the reference select pin and the external reference input.
If (VA - 0.3V) < VREF < VA, the internal 1.0V reference is selected.
If AGND < VREF < (AGND + 0.3V), the internal 0.5V reference is
selected.
If a voltage in the range of 0.8V to 1.2V is applied to this pin, that
voltage is used as the reference. VREF should be bypassed to AGND
with a 0.1 µF capacitor when an external reference is used.
VREF
AGND
VA VFloat
21
DF/DCS
13
5
VRPA
VRPB
14
4
VRMA
VRMB
This is a four-state pin.
DF/DCS = VA, output data format is offset binary with duty cycle
stabilization applied to the input clock
DF/DCS = AGND, output data format is 2's complement, with duty
cycle stabilization applied to the input clock.
DF/DCS = VRMA or VRMB , output data is 2's complement without
duty cycle stabilization applied to the input clock
DF/DCS = "float", output data is offset binary without duty cycle
stabilization applied to the input clock.
AGND
VA
VA
VA
12
6
VRNA
VRNB
VA
These pins are high impedance reference bypass pins. All these pins
should each be bypassed to ground with a 0.1 µF capacitor. A 10 µF
capacitor should be placed between the VRPA and VRNA pins and
between the VRPB and VRNB pins.
VRMA and VRMB may be loaded to 1mA for use as a temperature
stable 1.5V reference. The remaining pins should not be loaded.
AGND
AGND
DIGITAL I/O
4
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Pin Descriptions and Equivalent Circuits (continued)
Pin No.
Symbol
Equivalent Circuit
VD
60
22
41
CLK
VA
Description
Digital clock input. The range of frequencies for this input is as
specified in the electrical tables with ensured performance at 65
MHz. The input is sampled on the rising edge.
OEA and OEB are the output enable pins that, when low, holds their
respective data output pins in the active state. When either of these
pins is high, the corresponding outputs are in a high impedance
state.
OEA
OEB
AGND
DGND
VD
59
11
PD
VA
PD is the Power Down input pin. When high, this input puts the
converter into the power down mode. When this pin is low, the
converter is in the active mode.
When low, "A" & "B" data is present on it's respective data output
lines (Parallel Mode).
When high, both "A" and "B" channel data is present on the
"DA0:DA11" digital outputs (Multiplex Mode). The DB0/ABb pin is
used to synchronize the data.
MULTIPLEX
AGND
DGND
24–29
34–39
DA0–DA11
43–47
52–57
DB1–DB11
42
VDR VA
Digital data output pins that make up the 12-bit conversion results of
their respective converters. DA0 and DB0 are the LSBs, while DA11
and DB11 are the MSBs of the output word. Output levels are
TTL/CMOS compatible. Optimum loading is < 10pF.
DB0/ABb
AGND
DR GND
When MULTIPLEX is low, this is DB0.
When MULTIPLEX is high this is the ABb signal, which is used to
synchronize the multiplexed data. ABb changes synchronously with
the Multiplexed "A" and "B" channels. ABb is "high" when "A"
channel data is valid and is "low" when "B" channel data is valid.
ANALOG POWER
9, 18, 19, 62,
63
VA
3, 8, 10, 17,
20, 61, 64
AGND
Positive analog supply pins. These pins should be connected to a
quiet +3.3V source and bypassed to AGND with 0.1 µF capacitors
located within 1 cm of these power pins, and with a 10 µF capacitor.
The ground return for the analog supply.
DIGITAL POWER
33, 48
VD
32, 49
DGND
30, 51
23, 31, 40,
50, 58
Positive digital supply pin. This pin should be connected to the same
quiet +3.3V source as is VA and be bypassed to DGND with a 0.1 µF
capacitor located within 1 cm of the power pin and with a 10 µF
capacitor.
The ground return for the digital supply.
VDR
Positive driver supply pin for the ADC12DL065's output drivers. This
pin should be connected to a voltage source of +2.4V to VD and be
bypassed to DR GND with a 0.1 µF capacitor. If the supply for this
pin is different from the supply used for VA and VD, it should also be
bypassed with a 10 µF capacitor. VDR should never exceed the
voltage on VD. All 0.1 µF bypass capacitors should be located within
1 cm of the supply pin.
DR GND
The ground return for the digital supply for the ADC12DL065's output
drivers. These pins should be connected to the system digital
ground, but not be connected in close proximity to the
ADC12DL065's DGND or AGND pins. See LAYOUT AND
GROUNDING for more details.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2) (3)
VA, VD, VDR
4.2V
≤ 100 mV
|VA–VD|
−0.3V to (VA or VD +0.3V)
Voltage on Any Input or Output Pin
Input Current at Any Pin
(4)
±25 mA
Package Input Current (4)
±50 mA
See (5)
Package Dissipation at TA = 25°C
ESD Susceptibility
Human Body Model
(6)
2500V
Machine Model (6)
250V
Soldering Temperature, Infrared, 10 sec. (7)
235°C
−65°C to +150°C
Storage Temperature
Soldering process must comply with Reflow Temperature Profile specifications. Refer to www.ti.com/packaging. (7)
(1)
(2)
(3)
(4)
(5)
(6)
(7)
All voltages are measured with respect to GND = AGND = DGND = 0V, unless otherwise specified.
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the
Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may
degrade when the device is not operated under the listed test conditions.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
When the input voltage at any pin exceeds the power supplies (that is, VIN < AGND, or VIN > VA), the current at that pin should be
limited to 25 mA. The 50 mA maximum package input current rating limits the number of pins that can safely exceed the power supplies
with an input current of 25 mA to two.
The absolute maximum junction temperature (TJmax) for this device is 150°C. The maximum allowable power dissipation is dictated by
TJmax, the junction-to-ambient thermal resistance (θJA), and the ambient temperature, (TA), and can be calculated using the formula
PDMAX = (TJmax - TA )/θJA. In the 64-pin TQFP, θJA is 50°C/W, so PDMAX = 2 Watts at 25°C and 800 mW at the maximum operating
ambient temperature of 85°C. Note that the power consumption of this device under normal operation will typically be about 390 mW
(360 typical power consumption + 30 mW TTL output loading). The values for maximum power dissipation listed above will be reached
only when the device is operated in a severe fault condition (e.g. when input or output pins are driven beyond the power supply
voltages, or the power supply polarity is reversed). Obviously, such conditions should always be avoided.
Human body model is 100 pF capacitor discharged through a 1.5 kΩ resistor. Machine model is 220 pF discharged through 0Ω.
Reflow temperature profiles are different for lead-free and non-lead-free packages.
Operating Ratings (1) (2)
Operating Temperature
−40°C ≤ TA ≤ +85°C
Supply Voltage (VA, VD)
+3.0V to +3.6V
Output Driver Supply (VDR)
+2.4V to VD
−0.05V to (VD + 0.05V)
CLK, PD, OEA, OEB
Analog Input Pins
0V to 2.6V
VCM
0.5V to 2.0V
≤100mV
|AGND–DGND|
Clock Duty Cycle (DCS On)
20% to 80%
Clock Duty Cycle (DCS Off)
40% to 60%
(1)
(2)
6
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the
Electrical Characteristics. The ensured specifications apply only for the test conditions listed. Some performance characteristics may
degrade when the device is not operated under the listed test conditions.
All voltages are measured with respect to GND = AGND = DGND = 0V, unless otherwise specified.
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Converter Electrical Characteristics
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 65 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On,
parallel output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C (1) (2) (3)
Parameter
Test Conditions
Typical (4)
Limits (4)
Units
(Limits)
STATIC CONVERTER CHARACTERISTICS
12
Bits (min)
INL
Resolution with No Missing Codes
Integral Non Linearity (5)
±0.75
±1.7
LSB (max)
DNL
Differential Non Linearity
±0.4
±1.0
LSB (max)
PGE
Positive Gain Error
±0.2
±2.9
%FS (max)
NGE
Negative Gain Error
±0.2
TC GE
Gain Error Tempco
VOFF
Offset Error (VIN+ = VIN−)
TC VOFF
Offset Error Tempco
−40°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +85°C
±3.2
%FS (max)
10
ppm/°C
0.18
±0.85
%FS (max)
%FS (min)
3.6
ppm/°C
Under Range Output Code
0
0
Over Range Output Code
4095
4095
REFERENCE AND ANALOG INPUT CHARACTERISTICS
VCM
Common Mode Input Voltage
VRMA,
VRMB
Reference Output Voltage
Output load = 1 mA
CIN
VIN Input Capacitance (each pin to GND)
VIN = 2.5 Vdc + 0.7 Vrms
VREF
1.5
(2)
(3)
(4)
(5)
(6)
V (min)
2.0
V (max)
1.5
V
(CLK LOW)
8
pF
(CLK HIGH)
7
External Reference Voltage (6)
1.00
Reference Input Resistance
(1)
0.5
1
pF
0.8
1.2
V (min)
V (max)
MΩ (min)
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings Table. However, errors in the A/D conversion can occur if the input goes
above VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure
accurate conversions. See Figure 2.
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 12-bit LSB is 488 µV.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are ensured to TI's AOQL (Average Outgoing
Quality Level).
Integral Non Linearity is defined as the deviation of the analog value, expressed in LSBs, from the straight line that passes through
positive and negative full-scale.
Optimum performance will be obtained by keeping the reference input in the 0.8V to 1.2V range. The LM4051CIM3-ADJ (SOT-23
package) is recommended for external reference applications.
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Converter Electrical Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 65 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On,
parallel output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C(1)(2)(3)
Parameter
Typical (4)
Test Conditions
Limits (4)
Units
(Limits)
DYNAMIC CONVERTER CHARACTERISTICS
FPBW
SNR
SINAD
Full Power Bandwidth
Signal-to-Noise Ratio
Signal-to-Noise and Distortion
0 dBFS Input, Output at −3 dB
250
fIN = 1 MHz, VIN = −0.5 dBFS
69
fIN = 10 MHz, VIN = −0.5 dBFS
69
67.5
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
68.5
67
dBc (min)
fIN = 1 MHz, VIN = −0.5 dBFS
68.5
fIN = 10 MHz, VIN = −0.5 dBFS
68.5
67.5
dBc (min)
68
67
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
ENOB
THD
H2
H3
SFDR
IMD
Effective Number of Bits
MHz
dBc
dBc
fIN = 1 MHz, VIN = −0.5 dBFS
11.1
fIN = 10 MHz, VIN = −0.5 dBFS
11.1
10.9
Bits (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
11.0
10.8
Bits (min)
Bits
fIN = 1 MHz, VIN = −0.5 dBFS
−82
fIN = 10 MHz, VIN = −0.5 dBFS
−83.5
-75
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
−83
-75
dBc (min)
fIN = 1 MHz, VIN = −0.5 dBFS
−88
fIN = 10 MHz, VIN = −0.5 dBFS
−90
-80
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
−90
-80
dBc (min)
fIN = 1 MHz, VIN = −0.5 dBFS
−88
fIN = 10 MHz, VIN = −0.5 dBFS
−86
-78.5
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
−84
-77
dBc (min)
fIN = 1 MHz, VIN = −0.5 dBFS
86
fIN = 10 MHz, VIN = −0.5 dBFS
86
78.5
dBc (min)
fIN = 32.5 MHz, VIN = −0.5 dBFS
85
77
dBc (min)
fIN = 9.6 MHz and 10.2 MHz, each = −6.0
dBFS
−74
dBFS
Channel—Channel Offset Match
±0.3
%FS
Channel—Channel Gain Match
±4
%FS
10 MHz Tested, Channel;
32.5 MHz Other Channel
90
dBc
10 MHz Tested, Channel;
65 MHz Other Channel
90
dBc
Total Harmonic Distortion
Second Harmonic Distortion
Third Harmonic Distortion
Spurious Free Dynamic Range
Intermodulation Distortion
dBc
dBc
dBc
dBc
INTER-CHANNEL CHARACTERISTICS
Crosstalk
8
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DC and Logic Electrical Characteristics
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 65 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On,
parallel output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C (1) (2) (3)
Parameter
Test Conditions
Typical (4)
Limits (4)
Units
(Limits)
CLK, PD, OEA, OEB DIGITAL INPUT CHARACTERISTICS
VIN(1)
Logical “1” Input Voltage
VD = 3.6V
2.0
V (min)
VIN(0)
Logical “0” Input Voltage
VD = 3.0V
1.0
V (max)
IIN(1)
Logical “1” Input Current
VIN = 3.3V
10
µA
IIN(0)
Logical “0” Input Current
VIN = 0V
−10
µA
CIN
Digital Input Capacitance
5
pF
DA0–DA11, DB0-DB11 DIGITAL OUTPUT CHARACTERISTICS
VOUT(1)
Logical “1” Output Voltage
IOUT = −0.5 mA
VOUT(0)
Logical “0” Output Voltage
IOUT = 1.6 mA, VDR = 3V
IOZ
TRI-STATE Output Current
+ISC
Output Short Circuit Source Current
−ISC
Output Short Circuit Sink Current
COUT
Digital Output Capacitance
VDR = 2.5V
2.3
V (min)
VDR = 3V
2.7
V (min)
0.4
V (max)
VOUT = 2.5V or 3.3V
100
nA
VOUT = 0V
−100
nA
VOUT = 0V
−20
mA
VOUT = VDR
20
mA
5
pF
POWER SUPPLY CHARACTERISTICS
IA
Analog Supply Current
PD Pin = DGND, VREF = VA
PD Pin = VD
90
12
108
mA (max)
mA
ID
Digital Supply Current
PD Pin = DGND
PD Pin = VD , fCLK = 0
19
0
22
mA (max)
mA
IDR
Digital Output Supply Current
PD Pin = DGND, CL = 10 pF (5)
PD Pin = VD, fCLK = 0
15
0
Total Power Consumption
PD Pin = DGND, CL = 10 pF (6)
PD Pin = VD
360
36
Power Supply Rejection Ratio
Rejection of Full-Scale Error with
VA =3.0V vs. 3.6V
62
PSRR1
(1)
(2)
(3)
(4)
(5)
(6)
mA
mA
430
mW (max)
mW
dB
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings Table. However, errors in the A/D conversion can occur if the input goes
above VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure
accurate conversions. See Figure 2.
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 12-bit LSB is 488 µV.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are ensured to TI's AOQL (Average Outgoing
Quality Level).
IDR is the current consumed by the switching of the output drivers and is primarily determined by load capacitance on the output pins,
the supply voltage, VDR, and the rate at which the outputs are switching (which is signal dependent). IDR=VDR(C0 x f0 + C1 x f1 +....C11 x
f11) where VDR is the output driver power supply voltage, Cn is total capacitance on the output pin, and fn is the average frequency at
which that pin is toggling.
Excludes IDR. See DC and Logic Electrical Characteristics, Note 5.
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AC Electrical Characteristics
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, External VREF = +1.0V, fCLK = 65 MHz, fIN = 10 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On,
parallel output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C (1) (2) (3) (4)
Parameter
Typical (5)
Test Conditions
Limits (5)
Units
(Limits)
65
MHz (min)
ns (min)
fCLK1
Maximum Clock Frequency
fCLK
Minimum Clock Frequency
tCH
Clock High Time
Duty Cycle Stabilizer On
7.7
3
tCL
Clock Low Time
Duty Cycle Stabilizer On
7.7
3
ns (min)
tr, tf
Clock Rise and Fall Times
Duty Cycle Stabilizer On
2
4
ns (max)
tCH
Clock High Time
Duty Cycle Stabilizer Off
7.7
6.2
ns (min)
tCL
Clock Low Time
Duty Cycle Stabilizer Off
7.7
6.2
tr, tf
Clock Rise and Fall Times
Duty Cycle Stabilizer Off
2
tCONV
Conversion Latency
Parallel mode
tOD
Data Output Delay after Rising Clock
Edge
Parallel mode
tCONV
Conversion Latency
tCONV
Conversion Latency
2
15
MHz
ns (min)
ns (max)
7
Clock Cycles
3.5
ns (min)
9
ns (max)
Multiplex mode, Channel A
7.5
Clock Cycles
Multiplex mode, Channel B
8
Clock Cycles
6.0
ns (min)
ns (max)
Data Output Delay after Clock Edge
tSKEW
ABb to Data Skew
tAD
Aperture Delay
2
ns
tAJ
Aperture Jitter
1.2
ps rms
tDIS
Data outputs into Hi-Z Mode
10
ns
tEN
Data Outputs Active after Hi-Z Mode
10
ns
1
µs
(1)
(2)
(3)
(4)
(5)
Power Down Mode Exit Cycle
6.0
8
tOD
tPD
Multiplex mode
3.5
±0.5
1.0 µF on pins 4, 14; 0.1 µF on pins
5,6,12,13; 10 µF between pins 5, 6 and
between pins 12, 13
ns (max)
The inputs are protected as shown below. Input voltage magnitudes above VA or below GND will not damage this device, provided
current is limited per Note 4 of the Absolute Maximum Ratings Table. However, errors in the A/D conversion can occur if the input goes
above VA or below GND by more than 100 mV. As an example, if VA is +3.3V, the full-scale input voltage must be ≤+3.4V to ensure
accurate conversions. See Figure 2.
To ensure accuracy, it is required that |VA–VD| ≤ 100 mV and separate bypass capacitors are used at each power supply pin.
With the test condition for VREF = +1.0V (2VP-P differential input), the 12-bit LSB is 488 µV.
Timing specifications are tested at TTL logic levels, VIL = 0.4V for a falling edge and VIH = 2.4V for a rising edge.
Typical figures are at TJ = 25°C, and represent most likely parametric norms. Test limits are ensured to TI's AOQL (Average Outgoing
Quality Level).
VA
I/O
To Internal Circuitry
AGND
Figure 2.
10
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Specification Definitions
APERTURE DELAY is the time after the rising edge of the clock to when the input signal is acquired or held for
conversion.
APERTURE JITTER (APERTURE UNCERTAINTY) is the variation in aperture delay from sample to sample.
Aperture jitter manifests itself as noise in the output.
CLOCK DUTY CYCLE is the ratio of the time during one cycle that a repetitive digital waveform is high to the
total time of one period. The specification here refers to the ADC clock input signal.
COMMON MODE VOLTAGE (VCM) is the common d.c. voltage applied to both input terminals of the ADC.
CONVERSION LATENCY is the number of clock cycles between initiation of conversion and when that data is
presented to the output driver stage. Data for any given sample is available at the output pins the Pipeline Delay
plus the Output Delay after the sample is taken. New data is available at every clock cycle, but the data lags the
conversion by the pipeline delay.
CROSSTALK is coupling of energy from one channel into the other channel.
DIFFERENTIAL NON-LINEARITY (DNL) is the measure of the maximum deviation from the ideal step size of 1
LSB.
EFFECTIVE NUMBER OF BITS (ENOB, or EFFECTIVE BITS) is another method of specifying Signal-to-Noise
and Distortion or SINAD. ENOB is defined as (SINAD - 1.76) / 6.02 and says that the converter is equivalent to a
perfect ADC of this (ENOB) number of bits.
FULL POWER BANDWIDTH is a measure of the frequency at which the reconstructed output fundamental
drops 3 dB below its low frequency value for a full scale input.
GAIN ERROR is the deviation from the ideal slope of the transfer function. It can be calculated as:
Gain Error = Positive Full Scale Error − Offset Error
(1)
A gain of unity occurs when the negative and positive full scale errors are equal to each other, including having
the same sign.
GAIN ERROR MATCHING is the difference in gain errors between the two converters divided by the average
gain of the converters.
INTEGRAL NON LINEARITY (INL) is a measure of the deviation of each individual code from a line drawn from
negative full scale (½ LSB below the first code transition) through positive full scale (½ LSB above the last code
transition). The deviation of any given code from this straight line is measured from the center of that code value.
INTERMODULATION DISTORTION (IMD) is the creation of additional spectral components as a result of two
sinusoidal frequencies being applied to the ADC input at the same time. It is defined as the ratio of the power in
the intermodulation products to the total power in the original frequencies. IMD is usually expressed in dBFS.
LSB (LEAST SIGNIFICANT BIT) is the bit that has the smallest value or weight of all bits. This value is VFS/2n,
where “VFS” is the full scale input voltage and “n” is the ADC resolution in bits.
MISSING CODES are those output codes that will never appear at the ADC outputs. The ADC12DL065 is
ensured not to have any missing codes.
MSB (MOST SIGNIFICANT BIT) is the bit that has the largest value or weight. Its value is one half of full scale.
NEGATIVE FULL SCALE ERROR is the difference between the actual first code transition and its ideal value of
½ LSB above negative full scale.
OFFSET ERROR is the difference between the two input voltages [(VIN+) – (VIN−)] required to cause a transition
from code 2047 to 2048.
OUTPUT DELAY is the time delay after the rising edge of the clock before the data update is presented at the
output pins.
OVER RANGE RECOVERY TIME is the time required after VIN goes from a specified voltage out of the normal
input range to a specified voltage within the normal input range and the converter makes a conversion with its
rated accuracy.
PIPELINE DELAY (LATENCY) See CONVERSION LATENCY.
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POSITIVE FULL SCALE ERROR is the difference between the actual last code transition and its ideal value of
1½ LSB below positive full scale.
POWER SUPPLY REJECTION RATIO (PSRR) is a measure of how well the ADC rejects a change in the power
supply voltage. For the ADC12DL065, PSRR1 is the ratio of the change in Full-Scale Error that results from a
change in the d.c. power supply voltage, expressed in dB. PSRR2 is a measure of how well an a.c. signal riding
upon the power supply is rejected at the output.
SIGNAL TO NOISE RATIO (SNR) is the ratio, expressed in dB, of the rms value of the input signal to the rms
value of the sum of all other spectral components below one-half the sampling frequency, not including
harmonics or d.c.
SIGNAL TO NOISE PLUS DISTORTION (S/N+D or SINAD) Is the ratio, expressed in dB, of the rms value of the
input signal to the rms value of all of the other spectral components below half the clock frequency, including
harmonics but excluding d.c.
SPURIOUS FREE DYNAMIC RANGE (SFDR) is the difference, expressed in dB, between the rms values of the
input signal and the peak spurious signal, where a spurious signal is any signal present in the output spectrum
that is not present at the input.
TOTAL HARMONIC DISTORTION (THD) is the ratio, expressed in dB, of the rms total of the first nine harmonic
levels at the output to the level of the fundamental at the output. THD is calculated as
(2)
where f1 is the RMS power of the fundamental (output) frequency and f2 through f10 are the RMS power of the
first 9 harmonic frequencies in the output spectrum.
SECOND HARMONIC DISTORTION (2ND HARM) is the difference expressed in dB, between the RMS power in
the input frequency at the output and the power in its 2nd harmonic level at the output.
THIRD HARMONIC DISTORTION (3RD HARM) is the difference, expressed in dB, between the RMS power in
the input frequency at the output and the power in its 3rd harmonic level at the output.
12
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Timing Diagram
Sample N + 8
Sample N + 7
Sample N + 9
Sample N + 10
|
Sample N + 6
Sample N
VIN
tAD
Clock N
1
fCLK
Clock N + 7
90%
90%
CLK
|
10%
tCH
10%
tCL
tf
tr
|
OE
(A or B)
tOD
tDIS
tEN
Parallel Output Mode
| |
D0 - D11
(A or B)
Data N - 1
Data N + 2
Data N
Latency
Multiplex Output Mode
|
DB0/ABb
tSKEW
tOD
| |
D0 - D11
DataA
N-1
DataB
N-1
tOD
DataA
N
DataB
N
DataA
N+1
DataB
N+1
DataA
N+2
DataB
N+2
Channel A Latency
Channel B Latency
Figure 3. Output Timing
Transfer Characteristic
Figure 4. Transfer Characteristic
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Typical Performance Characteristics DNL, INL
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 0, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel output
mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
14
DNL
INL
Figure 5.
Figure 6.
DNL vs. fCLK
INL vs. fCLK
Figure 7.
Figure 8.
DNL vs. Clock Duty Cycle
INL vs. Clock Duty Cycle
Figure 9.
Figure 10.
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Typical Performance Characteristics DNL, INL (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 0, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel output
mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
DNL vs. Temperature
INL vs. Temperature
Figure 11.
Figure 12.
DNL vs. VDR, VA = VD = 3.6V
INL vs. VDR, VA = VD = 3.6V
Figure 13.
Figure 14.
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Typical Performance Characteristics
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 32 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel
output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
16
SNR,SINAD,SFDR vs. VA
Distortion vs. VA
Figure 15.
Figure 16.
SNR,SINAD,SFDR vs. VDR, VA = VD = 3.6V
Distortion vs. VDR, VA = VD = 3.6V
Figure 17.
Figure 18.
SNR,SINAD,SFDR vs. VCM
Distortion vs. VCM
Figure 19.
Figure 20.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 32 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel
output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
SNR,SINAD,SFDR vs. fCLK
Distortion vs. fCLK
Figure 21.
Figure 22.
SNR,SINAD,SFDR vs. Clock Duty Cycle
Distortion vs. Clock Duty Cycle
Figure 23.
Figure 24.
SNR,SINAD,SFDR vs. VREF
Distortion vs. VREF
Figure 25.
Figure 26.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 32 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel
output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
18
SNR,SINAD,SFDR vs. fIN
Distortion vs. fIN
Figure 27.
Figure 28.
SNR,SINAD,SFDR vs. Temperature
Distortion vs. Temperature
Figure 29.
Figure 30.
tOD vs. VDR, VA = VD = 3.6V
Parallel Output Mode
tOD vs. VDR, VA = VD = 3.6V
Multiplex Output Mode
Figure 31.
Figure 32.
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Typical Performance Characteristics (continued)
Unless otherwise specified, the following specifications apply for AGND = DGND = DR GND = 0V, VA = VD = +3.3V, VDR =
+2.5V, PD = 0V, VREF = +1.0V, fCLK = 65 MHz, fIN = 32 MHz, tr = tf = 2 ns, CL = 15 pF/pin, Duty Cycle Stabilizer On, parallel
output mode. Boldface limits apply for TJ = TMIN to TMAX: all other limits TJ = 25°C
Spectral Response @ 10 MHz Input
Spectral Response @ 32 MHz Input
Figure 33.
Figure 34.
Intermodulation Distortion, fIN1= 9.6 MHz, fIN2 = 10.2 MHz
Figure 35.
FUNCTIONAL DESCRIPTION
Operating on a single +3.3V supply, the ADC12DL065 uses a pipeline architecture and has error correction
circuitry to help ensure maximum performance. The differential analog input signal is digitized to 12 bits. The
user has the choice of using an internal 1.0 Volt or 0.5 Volt stable reference, or using an external reference. Any
external reference is buffered on-chip to ease the task of driving that pin.
The output word rate is the same as the clock frequency, which can be between 15 MSPS and 65 MSPS
(typical) with fully specified performance at 65 MSPS. The analog input for both channels is acquired at the rising
edge of the clock and the digital data for a given sample is delayed by the pipeline for 7 clock cycles. Duty cycle
stabilization and output data format are selectable using the quad state function DF/DCS pin. The output data
can be set for offset binary or two's complement.
A logic high on the power down (PD) pin reduces the converter power consumption to 36 mW.
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Applications Information
OPERATING CONDITIONS
We recommend that the following conditions be observed for operation of the ADC12DL065:
3.0V ≤ VA ≤ 3.6V
VD = VA
2.4V ≤ VDR ≤ VA
15 MHz ≤ fCLK ≤ 65 MHz
0.8V ≤ VREF ≤ 1.2V (for an external reference)
0.5V ≤ VCM ≤ 2.0V
Analog Inputs
There is one reference input pin, VREF, which is used to select an internal reference, or to supply an external
reference. The ADC12DL065 has two analog signal input pairs, VIN A+ and VIN A- for one converter and VIN B+
and VIN B- for the other converter. Each pair of pins forms a differential input pair.
Reference Pins
The ADC12DL065 is designed to operate with an internal 1.0V or 0.5V reference, or an external 1.0V reference,
but performs well with reference voltages in the range of 0.8V to 1.2V. Lower reference voltages will decrease
the signal-to-noise ratio (SNR) of the ADC12DL065. Increasing the reference voltage (and the input signal swing)
beyond 1.2V may degrade THD for a full-scale input, especially at higher input frequencies.
It is important that all grounds associated with the reference voltage and the analog input signal make connection
to the ground plane at a single, quiet point to minimize the effects of noise currents in the ground path.
The six Reference Bypass Pins (VRPA, VRMA, VRNA, VRPB, VRMB and VRNB) are made available for bypass
purposes. All these pins should each be bypassed to ground with a 0.1 µF capacitor. A 10 µF capacitor should
be placed between the VRPA and VRNA pins and between the VRPB and VRNB pins, as shown in Figure 38. This
configuration is necessary to avoid reference oscillation, which could result in reduced SFDR and/or SNR.
Smaller capacitor values than those specified will allow faster recovery from the power down mode, but may
result in degraded noise performance. Loading any of these pins other than VRMA and VRMB may result in
performance degradation.
The nominal voltages for the reference bypass pins are as follows:
VRM = 1.5 V
VRP = VRM + VREF / 2
VRN = VRM − VREF / 2
User choice of an on-chip or external reference voltage is provided. The internal 1.0 Volt reference is in use
when the the VREF pin is connected to VA. When the VREF pin is connected to AGND, the internal 0.5 Volt
reference is in use. If a voltage in the range of 0.8V to 1.2V is applied to the VREF pin, that is used for the voltage
reference. When an external reference is used, the VREF pin should be bypassed to ground with a 0.1 µF
capacitor close to the reference input pin. There is no need to bypass the VREF pin when the internal reference is
used.
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Signal Inputs
The signal inputs are VIN A+ and VINA− for one ADC and VINB+ and VINB− for the other ADC . The input signal,
VIN, is defined as
VIN A = (VINA+) – (VINA−)
(3)
for the "A" converter and
VIN B = (VINB+) – (VINB−)
(4)
for the "B" converter. Figure 36 shows the expected input signal range. Note that the common mode input
voltage, VCM, should be in the range of 0.5V to 2.0V.
The peaks of the individual input signals should each never exceed 2.6V.
The ADC12DL065 performs best with a differential input signal with each input centered around a common mode
voltage, VCM. The peak-to-peak voltage swing at each analog input pin should not exceed the value of the
reference voltage or the output data will be clipped.
The two input signals should be exactly 180° out of phase from each other and of the same amplitude. For single
frequency inputs, angular errors result in a reduction of the effective full scale input. For complex waveforms,
however, angular errors will result in distortion.
Figure 36. Expected Input Signal Range
For single frequency sine waves the full scale error in LSB can be described as approximately
EFS = 4096 ( 1 - sin (90° + dev))
(5)
Where dev is the angular difference in degrees between the two signals having a 180° relative phase relationship
to each other (see Figure 37). Drive the analog inputs with a source impedance less than 100Ω.
Figure 37. Angular Errors Between the Two Input Signals Will Reduce the Output Level or Cause
Distortion
For differential operation, each analog input pin of the differential pair should have a peak-to-peak voltage equal
to the reference voltage, VREF, be 180 degrees out of phase with each other and be centered around VCM.
Single-Ended Operation
Performance with differential input signals is better than with single-ended signals. For this reason, single-ended
operation is not recommended. However, if single ended-operation is required and the resulting performance
degradation is acceptable, one of the analog inputs should be connected to the d.c. mid point voltage of the
driven input. The peak-to-peak differential input signal at the driven input pin should be twice the reference
voltage to maximize SNR and SINAD performance (Figure 36b). For example, set VREF to 0.5V, bias VIN− to 1.0V
and drive VIN+ with a signal range of 0.5V to 1.5V.
Because very large input signal swings can degrade distortion performance, better performance with a singleended input can be obtained by reducing the reference voltage when maintaining a full-range output. Table 1 and
Table 2 indicate the input to output relationship of the ADC12DL065.
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Table 1. Input to Output Relationship – Differential Input
VIN
VIN−
Binary Output
2’s Complement Output
VCM − VREF/2
VCM + VREF/2
0000 0000 0000
1000 0000 0000
VCM − VREF/4
VCM + VREF/4
0100 0000 0000
1100 0000 0000
VCM
VCM
1000 0000 0000
0000 0000 0000
VCM + VREF/4
VCM − VREF/4
1100 0000 0000
0100 0000 0000
VCM + VREF/2
VCM − VREF/2
1111 1111 1111
0111 1111 1111
+
Table 2. Input to Output Relationship – Single-Ended Input
VIN
VIN−
Binary Output
2’s Complement Output
VCM − VREF
VCM
0000 0000 0000
1000 0000 0000
VCM − VREF/2
VCM
0100 0000 0000
1100 0000 0000
+
VCM
VCM
1000 0000 0000
0000 0000 0000
VCM + VREF/2
VCM
1100 0000 0000
0100 0000 0000
VCM + VREF
VCM
1111 1111 1111
0111 1111 1111
Driving the Analog Inputs
The VIN+ and the VIN− inputs of the ADC12DL065 consist of an analog switch followed by a switched-capacitor
amplifier. The capacitance seen at the analog input pins changes with the clock level, appearing as 8 pF when
the clock is low, and 7 pF when the clock is high.
As the internal sampling switch opens and closes, current pulses occur at the analog input pins, resulting in
voltage spikes at the signal input pins. As a driving amplifier attempts to counteract these voltage spikes, a
damped oscillation may appear at the ADC analog input. Do not attempt to filter out these pulses. Rather, use
amplifiers to drive the ADC12DL065 input pins that are able to react to these pulses and settle before the switch
opens and another sample is taken. The LMH6702 LMH6628, LMH6622 and the LMH6655 are good amplifiers
for driving the ADC12DL065.
To help isolate the pulses at the ADC input from the amplifier output, use RCs at the inputs, as can be seen in
Figure 38 through Figure 40. These components should be placed close to the ADC inputs because the input
pins of the ADC is the most sensitive part of the system and this is the last opportunity to filter that input.
For Nyquist applications the RC pole should be at the ADC sample rate. The ADC input capacitance in the
sample mode should be considered when setting the RC pole. For wideband undersampling applications, the RC
pole should be set at about 1.5 to 2 times the maximum input frequency to maintain a linear delay response.
A single-ended to differential conversion circuit is shown in Figure 40. Table 3 gives resistor values for that circuit
to provide input signals in a range of 1.0V ±0.5V at each of the differential input pins of the ADC12DL065.
Table 3. Resistor Values for Circuit of Figure 40
SIGNAL RANGE
R1
R2
R3
R4
R5, R6
0 - 0.25V
open
0Ω
124Ω
1500Ω
1000Ω
0 - 0.5V
0Ω
openΩ
499Ω
1500Ω
499Ω
±0.25V
100Ω
698Ω
100Ω
698Ω
499Ω
Input Common Mode Voltage
The input common mode voltage, VCM, should be in the range of 0.5V to 2.0V and be a value such that the peak
excursions of the analog signal does not go more negative than ground or more positive than 2.6V. See
Reference Pins.
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DIGITAL INPUTS
Digital TTL/CMOS compatible inputs consist of CLK, OEA, OEB, PD, DF/DCS, and MULTIPLEX.
CLK
The CLK signal controls the timing of the sampling process. Drive the clock input with a stable, low jitter clock
signal in the range of 15 MHz to 65 MHz with rise and fall times of 2 ns or less. The trace carrying the clock
signal should be as short as possible and should not cross any other signal line, analog or digital, not even at
90°.
The CLK signal also drives an internal state machine. If the CLK is interrupted, or its frequency too low, the
charge on internal capacitors can dissipate to the point where the accuracy of the output data will degrade. This
is what limits the lowest sample rate.
The clock line should be terminated at its source in the characteristic impedance of that line. Take care to
maintain a constant clock line impedance throughout the length of the line. Refer to Application Note AN-905
(SNLA035) for information on setting characteristic impedance.
It is highly desirable that the the source driving the ADC CLK pin only drive that pin. However, if that source is
used to drive other things, each driven pin should be a.c. terminated with a series RC to ground, as shown in
Figure 38, such that the resistor value is equal to the characteristic impedance of the clock line and the capacitor
value is
(6)
where tPD is the signal propagation rate down the clock line, "L" is the line length and ZO is the characteristic
impedance of the clock line. This termination should be as close as possible to the ADC clock pin but beyond it
as seen from the clock source. Typical tPD is about 150 ps/inch (60 ps/cm) on FR-4 board material. The units of
"L" and tPD should be the same (inches or centimeters).
The duty cycle of the clock signal can affect the performance of the A/D Converter. Because achieving a precise
duty cycle is difficult, the ADC12DL065 has a Duty Cycle Stabilizer which can be enabled using the DF/DCS pin.
It is designed to maintain performance over a clock duty cycle range of 20% to 80% at 65 MSPS. The Duty Cycle
Stabilizer circuit requires a fast clock edge to produce the internal clock, which is the reason for the rise and fall
time requirement listed in the specifications table.
OEA, OEB
The OEA and OEB pins, when high, put the output pins of their respective converters into a high impedance
state. When either of these pin is low, the corresponding outputs are in the active state. The ADC12DL065 will
continue to convert whether these pins are high or low, but the output can not be read while the pin is high.
Since ADC noise increases with increased output capacitance at the digital output pins, do not use the TRISTATE outputs of the ADC12DL065 to drive a bus. Rather, each output pin should be located close to and drive
a single digital input pin. To further reduce ADC noise, a 100 Ω resistor in series with each ADC digital output
pin, located close to their respective pins, should be added to the circuit.
PD
The PD pin, when high, holds the ADC12DL065 in a power-down mode to conserve power when the converter is
not being used. The power consumption in this state is 36 mW with a 65MHz clock and 40mW if the clock is
stopped when PD is high. The output data pins are undefined and the data in the pipeline is corrupted while in
the power down mode.
The Power Down Mode Exit Cycle time is determined by the value of the components on pins 4, 5, 6, 12, 13 and
14 and is about 500 µs with the recommended components on the VRP, VRM and VRN reference bypass pins.
These capacitors loose their charge in the Power Down mode and must be recharged by on-chip circuitry before
conversions can be accurate. Smaller capacitor values allow slightly faster recovery from the power down mode,
but can result in a reduction in SNR, SINAD and ENOB performance.
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DF/DCS
Duty cycle stabilization and output data format are selectable using this quad state function pin. When enabled,
duty cycle stabilization can compensate for clock inputs with duty cycles ranging from 20% to 80% and generate
a stable internal clock, improving the performance of the part. The Duty Cycle Stabilizer circuit requires a fast
clock edge to produce the internal clock, which is the reason for the rise and fall time requirement listed in the
specifications table. With DF/DCS = VA the output data format is offset binary and duty cycle stabilization is
applied to the clock. With DF/DCS = 0 the output data format is 2's complement and duty cycle stabilization is
applied to the clock. With DF/DCS = VRMA or VRMB the output data format is 2's complement and duty cycle
stabilization is not used. If DF/DCS is floating, the output data format is offset binary and duty cycle stabilization
is not used. While the sense of this pin may be changed "on the fly," doing this is not recommended as the
output data could be erroneous for a few clock cycles after this change is made.
MULTIPLEX
With the MULTIPLEX pin at a logic low, the digital output words from channels A and B are available on separate
digital output buses (Parallel mode). When MULTIPLEX is high, the digital output words are multiplexed on pins
DA0:DA11 (Multiplex Mode). The DB0/ABb pin changes synchronously with the multiplexed outputs, and is high
when channel A data is present on the outputs, and low when channel B data is present.
OUTPUTS
The ADC12DL065 has 12 TTL/CMOS compatible Data Output pins for each output. Valid data is present at
these outputs while the OE and PD pins are low. In the parallel mode, the data should be captured with the CLK
signal. Depending on the setup and hold time requirements of the receiving circuit (ASIC), either the rising edge
or the falling edge of the CLK signal can be used to latch the data. Generally, rising-edge-capture would
maximize setup time with minimal hold time; while falling-edge-capture would maximize hold time with minimal
setup time. However, actual timing for the falling-edge case depends greatly on the CLK frequency and both
cases also depend on the delays inside the ASIC. Refer to the Tod spec in AC Electrical Characteristics.
In Multiplex mode, both channel outputs are available on DA0:DA11. The ABb signal is available to de-multiplex
the output bus. The ABb signal may also be used to latch the data in the ASIC thus avoiding the use of the CLK
signal altogether. However, since the ABb signal edges are provided in-phase with the data transitions, generally
the ASIC circuitry would have to delay the ABb signal with respect to the data in order to use it as the clock for
the capturing latches. It is also possible to use the CLK signal to latch the data in the multiplexed mode as well as described in the previous paragraph.
Be very careful when driving a high capacitance bus. The more capacitance the output drivers must charge for
each conversion, the more instantaneous digital current flows through VDR and DR GND. These large charging
current spikes can cause on-chip ground noise and couple into the analog circuitry, degrading dynamic
performance. Adequate bypassing, limiting output capacitance and careful attention to the ground plane will
reduce this problem. Additionally, bus capacitance beyond the specified 15 pF/pin will cause tOD to increase,
making it difficult to properly latch the ADC output data. The result could be an apparent reduction in dynamic
performance.
To minimize noise due to output switching, minimize the load currents at the digital outputs. This can be done by
connecting buffers (74ACQ541, for example) between the ADC outputs and any other circuitry. Only one driven
input should be connected to each output pin. Additionally, inserting series resistors of about 100Ω at the digital
outputs, close to the ADC pins, will isolate the outputs from trace and other circuit capacitances and limit the
output currents, which could otherwise result in performance degradation. See Figure 38.
24
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+3.0V
3x 0.1 PF
CHOKE
2x 0.1 PF
2
3
4
5
6
7
8
9
2x 0.1 PF
+
1k
7
4
1 PF
5
330
10 PF
6
V DR 30
V DR 51
VA 9
VA 18
VA 19
VA 62
VA 63
VD 33
VD 48
10 PF
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0/ABb
VREF
V RMB
VRP B
V RNB
0.1 PF
0.1 PF
14
1 PF
**
VIN_B
1
0.1 PF
0.1 PF
51
6
13
10 PF 12
2
1
2
51 39 pF
3
1
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
ChB
Output Word
OE
74ACT574
2
3
4
5
6
7
8
9
11
12x100:
0.1 PF
39 pF
0.1 PF
2
VRP A
VRNA
T2
4
V RMA
11
57
56
55
54
53
52
47
46
45
44
43
42
D1
D2
D3
D4
D5
D6
D7
D8
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
OE
74ACT574
VINBVINB+
T4-6T
ADC12DL065
330
**
VIN_A
1
0.1 PF
51
6
T2
39 pF
0.1 PF
2
2
51
3
4
16
15
39 pF
VINA-
DA11
DA10
DA9
DA8
DA7
DA6
DA5
DA4
DA3
DA2
DA1
DA0
VINA+
T4-6T
41
OEB
59
MULTIPLEX
OEA
OEB
3
8
10
17
20
61
64
See
Text
11
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
ChA
Output Word
OE
74ACT574
12x100:
PD
AGND
AGND
AGND
AGND
AGND
AGND
AGND
PD
DF/DCS
DR GND
DR GND
DR GND
DR GND
DR GND
22
OEA
CLK
23
31
40
50
58
11
DGND
DGND
60
21
DF/DCS
32
49
47
Crystal Oscillator
2
3
4
5
6
7
8
9
39
38
37
36
35
34
29
28
27
26
25
24
2
3
4
5
6
7
8
9
11
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
OE
74ACT574
A.
** – May be replaced by Ckt in Figure 40.
Figure 38. Application Circuit using Transformer or Differential Op-Amp Drive Circuit, Parallel mode
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2
3
4
5
6
7
8
9
+3.0V
3x 0.1 PF
CHOKE
2x 0.1 PF
2x 0.1 PF
+
10 PF
7
4
5
0.1 PF
10 PF
0.1 PF
6
1
0.1 PF
6
13
10 PF 12
V RMA
VRP A
2
1
2
51 39 pF
3
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
Channel B
Output Word
OE
74ACT574
57
56
55
54
53
52
47
46
45
44
43
42
2
3
4
5
6
7
8
9
11
***
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
OE
74ACT574
0.1 PF
39 pF
0.1 PF
2
VRP B
1
V RNA
T2
4
V RMB
V RNB
0.1 PF
51
VIN_B
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0/ABb
VREF
0.1 PF
14
0.1 PF
**
V DR 30
V DR 51
VA 9
VA 18
VA 19
VA 62
VA 63
1k
330
VD 33
VD 48
11
D1
D2
D3
D4
D5
D6
D7
D8
VINB-
***
VINB+
T4-6T
ADC12DL065
330
**
VIN_A
1
0.1 PF
51
6
T2
39 pF
0.1 PF
2
2
16
15
39 pF
51
3
4
DA11
DA10
DA9
DA8
DA7
DA6
DA5
DA4
DA3
DA2
DA1
DA0
VINAVINA+
T4-6T
22
OEA
41
OEB
59
OEB
PD
3
8
10
17
20
61
64
See
Text
11
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
Channel A
Output Word
OE
74ACT574
12x100:
OEA
AGND
AGND
AGND
AGND
AGND
AGND
AGND
PD
MULTIPLEX
DR GND
DR GND
DR GND
DR GND
DR GND
11
Multiplex
DF/DCS
23
31
40
50
58
DF/DCS
CLK
DGND
DGND
60
21
32
49
47
Clock In
2
3
4
5
6
7
8
9
39
38
37
36
35
34
29
28
27
26
25
24
2
3
4
5
6
7
8
9
11
1
D1
D2
D3
D4
D5
D6
D7
D8
Q1
Q2
Q3
Q4
Q5
Q6
Q7
Q8
19
18
17
16
15
14
13
12
CLK
OE
74ACT574
A.
** – May be replaced by Ckt in Figure 40.
B.
*** – The delay through the inverters should be adjusted to allow the correct setup and hold time for the latches.
Figure 39. Application Circuit using Transformer or Differential Op-Amp Drive Circuit, Multiplex mode
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2V
R2, 1%
R1, 1%
+
*
2.4k
5k, 1%
SIGNAL
INPUT
-
U1A
-
51
to
V IN+
U2B
+
*
51
39 pF #
5k, 1%
R5, 1%
5k, 1%
5k, 1%
*
R4, 1%
R3, 1%
2.4k
+
*
-
U2A
-
to
V IN-
U1B
+
5k, 1%
39 pF #
51
5k, 1%
*
R6, 1%
5k, 1%
A.
Amplifiers: two LMH6622s or LMH6655s
B.
* – The ground connections indicated with an "*" should be connected to a common point in the analog ground plane.
C.
# – The 39 pF capacitors between the two inputs are for Nyquiest applications and should be removed for
undersampling applications.
Figure 40. Differential Drive Circuit of Figure 38
POWER SUPPLY CONSIDERATIONS
The power supply pins should be bypassed with a 10 µF capacitor and with a 0.1 µF ceramic chip capacitor
within a centimeter of each power pin. Leadless chip capacitors are preferred because they have low series
inductance.
As is the case with all high-speed converters, the ADC12DL065 is sensitive to power supply noise. Accordingly,
the noise on the analog supply pin should be kept below 100 mVP-P.
No pin should ever have a voltage on it that is in excess of the supply voltages, not even on a transient basis. Be
especially careful of this during power turn on and turn off.
The VDR pin provides power for the output drivers and may be operated from a supply in the range of 2.4V to VD.
This can simplify interfacing to lower voltage devices and systems. Note, however, that tOD increases with
reduced VDR. DO NOT operate the VDR pin at a voltage higher than VD.
LAYOUT AND GROUNDING
Proper grounding and proper routing of all signals are essential to ensure accurate conversion. Maintaining
separate analog and digital areas of the board, with the ADC12DL065 between these areas, is required to
achieve specified performance.
The ground return for the data outputs (DR GND) carries the ground current for the output drivers. The output
current can exhibit high transients that could add noise to the conversion process. To prevent this from
happening, the DR GND pins should NOT be connected to system ground in close proximity to any of the
ADC12DL065's other ground pins.
Capacitive coupling between the typically noisy digital circuitry and the sensitive analog circuitry can lead to poor
performance. The solution is to keep the analog circuitry separated from the digital circuitry, and to keep the
clock line as short as possible.
Digital circuits create substantial supply and ground current transients. The logic noise thus generated could
have significant impact upon system noise performance. The best logic family to use in systems with A/D
converters is one which employs non-saturating transistor designs, or has low noise characteristics, such as the
74LS, 74HC(T) and 74AC(T)Q families. The worst noise generators are logic families that draw the largest
supply current transients during clock or signal edges, like the 74F and the 74AC(T) families.
The effects of the noise generated from the ADC output switching can be minimized through the use of 100Ω
resistors in series with each data output line. Locate these resistors as close to the ADC output pins as possible.
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Since digital switching transients are composed largely of high frequency components, total ground plane copper
weight will have little effect upon the logic-generated noise. This is because of the skin effect. Total surface area
is more important than is total ground plane volume.
Generally, analog and digital lines should cross each other at 90° to avoid crosstalk. To maximize accuracy in
high speed, high resolution systems, however, avoid crossing analog and digital lines altogether. It is important to
keep clock lines as short as possible and isolated from ALL other lines, including other digital lines. Even the
generally accepted 90° crossing should be avoided with the clock line as even a little coupling can cause
problems at high frequencies. This is because other lines can introduce jitter into the clock line, which can lead to
degradation of SNR. Also, the high speed clock can introduce noise into the analog chain.
Best performance at high frequencies and at high resolution is obtained with a straight signal path. That is, the
signal path through all components should form a straight line wherever possible.
6 x 100:
COMMON
GROUND
PLANE
Clock line should be short
and cross no other lines.
OSC
LATCH
All Analog Components
mounted over Analog
area of Ground Plane
All Digital
Components
mounted over
Digital area of
Ground Plane
DGND
VDR
DB6
DB7
DB9
DB8
DB11
DB10
PD
DR GND
CLK
AGND
DR GND
DR GND
AGND
DA11
INT/EXT REF
DA10
DA9
VRP A
DA8
VRM A
DA7
VIN A+
DA6
VIN A-
VD
Analog power line should be routed
away from Digital power trace.
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
DGND
VRN A
AGND
16
ADC12DL065
VA
DR GND
15
OEB
VDR
14
AGND
DA5
13
DB0
DA4
12
DB1
VREF
DA3
11
VRN B
DA2
10
DB2
DA1
9
VRP B
DA0
8
DB3
DR GND
7
DB4
OEA
Driving source
located close
to converter. Single Ground entry for all
Reference Components
AGND
VRM B
AGND
5
6
DB5
OF
4
VD
VIN B+
VA
3
VIN B-
VA
2
VA
AGND
1
VA
64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49
Xfmr/Amplifier
48
6 x 100:
47
46
45
44
LATCH
43
42
41
40
39
38
37
36
35
34
33
Digital power line should be routed
away from analog power trace.
Ground entry points
close to ground pins.
Figure 41. Example of a Suitable Layout
Be especially careful with the layout of inductors. Mutual inductance can change the characteristics of the circuit
in which they are used. Inductors should not be placed side by side, even with just a small part of their bodies
beside each other.
The analog input should be isolated from noisy signal traces to avoid coupling of spurious signals into the input.
Any external component (e.g., a filter capacitor) connected between the converter's input pins and ground or to
the reference input pin and ground should be connected to a very clean point in the ground plane.
Figure 41 gives an example of a suitable layout. All analog circuitry (input amplifiers, filters, reference
components, etc.) should be placed in the analog area of the board. All digital circuitry and I/O lines should be
placed in the digital area of the board. The ADC12DL065 should be between these two areas. Furthermore, all
components in the reference circuitry and the input signal chain that are connected to ground should be
connected together with short traces and enter the ground plane at a single, quiet point. All ground connections
should have a low inductance path to ground.
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DYNAMIC PERFORMANCE
To achieve the best dynamic performance, the clock source driving the CLK input must be free of jitter. Isolate
the ADC clock from any digital circuitry with buffers, as with the clock tree shown in Figure 42. The gates used in
the clock tree must be capable of operating at frequencies much higher than those used if added jitter is to be
prevented.
Best performance will be obtained with a differential input drive, compared with a single-ended drive, as
discussed in Single-Ended Operation and Driving the Analog Inputs.
As mentioned in LAYOUT AND GROUNDING, it is good practice to keep the ADC clock line as short as possible
and to keep it well away from any other signals. Other signals can introduce jitter into the clock signal, which can
lead to reduced SNR performance, and the clock can introduce noise into other lines. Even lines with 90°
crossings have capacitive coupling, so try to avoid even these 90° crossings of the clock line.
Figure 42. Isolating the ADC Clock from other Circuitry with a Clock Tree
COMMON APPLICATION PITFALLS
Driving the inputs (analog or digital) beyond the power supply rails. For proper operation, all inputs should
not go more than 100 mV beyond the supply rails (more than 100 mV below the ground pins or 100 mV above
the supply pins). Exceeding these limits on even a transient basis may cause faulty or erratic operation. It is not
uncommon for high speed digital components (e.g., 74F and 74AC devices) to exhibit overshoot or undershoot
that goes above the power supply or below ground. A resistor of about 47Ω to 100Ω in series with any offending
digital input, close to the signal source, will eliminate the problem.
Do not allow input voltages to exceed the supply voltage, even on a transient basis. Not even during power up or
power down.
Be careful not to overdrive the inputs of the ADC12DL065 with a device that is powered from supplies outside
the range of the ADC12DL065 supply. Such practice may lead to conversion inaccuracies and even to device
damage.
Attempting to drive a high capacitance digital data bus. The more capacitance the output drivers must
charge for each conversion, the more instantaneous digital current flows through VDR and DR GND. These large
charging current spikes can couple into the analog circuitry, degrading dynamic performance. Adequate
bypassing and maintaining separate analog and digital areas on the pc board will reduce this problem.
Additionally, bus capacitance beyond the specified 15 pF/pin will cause tOD to increase, making it difficult to
properly latch the ADC output data. The result could, again, be an apparent reduction in dynamic performance.
The digital data outputs should be buffered (with 74ACQ541, for example). Dynamic performance can also be
improved by adding series resistors at each digital output, close to the ADC12DL065, which reduces the energy
coupled back into the converter output pins by limiting the output current. A reasonable value for these resistors
is 100Ω.
Using an inadequate amplifier to drive the analog input. As explained in Signal Inputs, the capacitance seen
at the input alternates between 8 pF and 7 pF, depending upon the phase of the clock. This dynamic load is
more difficult to drive than is a fixed capacitance.
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If the amplifier exhibits overshoot, ringing, or any evidence of instability, even at a very low level, it will degrade
performance. A small series resistor at each amplifier output and a capacitor at the analog inputs (as shown in
Figure 39 and Figure 40) will improve performance. The LMH6702 and the LMH6628 have been successfully
used to drive the analog inputs of the ADC12DL065.
Also, it is important that the signals at the two inputs have exactly the same amplitude and be exactly 180º out of
phase with each other. Board layout, especially equality of the length of the two traces to the input pins, will
affect the effective phase between these two signals. Remember that an operational amplifier operated in the
non-inverting configuration will exhibit more time delay than will the same device operating in the inverting
configuration.
Operating with the reference pins outside of the specified range. As mentioned in Reference Pins, VREF
should be in the range of
0.8V ≤ VREF ≤ 1.2V
(7)
Operating outside of these limits could lead to performance degradation.
Inadequate network on Reference Bypass pins (VRPA, VRNA, VRMA, VRPB, VRNB and VRMB). As mentioned in
Reference Pins, these pins should be bypassed with 0.1 µF capacitors to ground at VRMA and VRMB and with a
series RC of 1.5 Ω and 1.0 µF between pins VRPA and VRNA and between VRPB and VRNB for best performance.
Using a clock source with excessive jitter, using excessively long clock signal trace, or having other
signals coupled to the clock signal trace. This will cause the sampling interval to vary, causing excessive
output noise and a reduction in SNR and SINAD performance.
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REVISION HISTORY
Changes from Revision C (April 2013) to Revision D
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 30
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
(3)
Device Marking
(4/5)
(6)
ADC12DL065CIVS/NOPB
ACTIVE
TQFP
PAG
64
160
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 85
ADC12DL065
CIVS
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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