ADC141S626
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SNAS434B – NOVEMBER 2007 – REVISED MARCH 2013
ADC141S626 14-Bit, 50 kSPS to 250 kSPS, Differential Input, Micro Power A/D Converter
Check for Samples: ADC141S626
FEATURES
1
•
•
2
•
•
•
•
•
True Differential Inputs
Guaranteed Performance from 50 kSPS to 250
kSPS
External Reference
Zero-Power Track Mode
Wide Input Common-Mode Voltage Range
Operating Temperature Range of −40°C to
+85°C
SPI™/QSPI™/MICROWIRE/DSP Compatible
Serial Interface
APPLICATIONS
•
•
•
•
•
•
Automotive Navigation
Portable Systems
Medical Instruments
Instrumentation and Control Systems
Motor Control
Direct Sensor Interface
KEY SPECIFICATIONS
•
•
•
•
•
•
•
Conversion Rate: 50 kSPS to 250 kSPS
INL: ± 0.95 LSB (Max)
DNL: ± 0.95 LSB (Max)
SNR: 82 dBc (Max)
THD: -90 dBc (Typ)
ENOB: 13.3 Bits(Min)
Power Consumption:
– 200 kSPS, 3V: 2.0 mW (Typ)
– 250 kSPS, 5V: 4.8 mW (Typ)
– Power-Down, 3V: 4 µW (Typ)
– Power-Down, 5V: 13 µW (Typ)
DESCRIPTION
The ADC141S626 is a 14-bit, 50 kSPS to 250 kSPS
sampling Analog-to-Digital (A/D) converter. The
converter is based on a successive-approximation
register (SAR) architecture where the differential
nature of the analog inputs is maintained from the
internal sample-and-hold circuits throughout the A/D
converter to provide excellent common-mode signal
rejection. The ADC141S626 features an external
reference that can be varied from 1.0V to VA. It also
features a zero-power track mode where the ADC is
consuming the minimum amount of supply current
while the internal sampling capacitor is tracking the
applied analog input voltage.
The serial data output is binary 2's complement and
is compatible with several standards, such as SPI™,
QSPI™, MICROWIRE, and many common DSP
serial interfaces. The conversion result is clocked out
by the serial clock input and is the result of the
conversion currently in progress; thus, ADC141S626
has no latency.
The ADC141S626 may be operated with independent
analog (VA) and digital input/output (VIO) supplies. VA
and VIO can range from 2.7V to 5.5V and can be set
independent of each other. This allows a user to
maximize performance and minimize power
consumption by operating the analog portion of the
ADC at a VA of 5V while communicating with a 3V
controller on the digital side. With a 3V source, the
power consumption when operating at 200 kSPS is
2.0 mW. With a 5V source, the power consumption
when operating at 250 kSPS is 4.8 mW. The power
consumption drops down to 4 µW and 13 µW
respectively
when
the
ADC141S626
enters
acquisition (power-down) mode. The differential input,
low power consumption, and small size make the
ADC141S626 ideal for direct connection to bridge
sensors and transducers in battery operated systems
or remote data acquisition applications.
Operation is guaranteed over the temperature range
of −40°C to +85°C and clock rates of 0.9 MHz to 4.5
MHz. The ADC141S626 is available in a 10-lead
VSSOP package.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007–2013, Texas Instruments Incorporated
ADC141S626
SNAS434B – NOVEMBER 2007 – REVISED MARCH 2013
www.ti.com
Connection Diagram
VREF
1
10
VA
+IN
2
9
VIO
- IN
3 ADC141S626 8
SCLK
GND
4
7
DOUT
GND
5
6
CS
Block Diagram
SAR
CONTROL
VREF
SERIAL
INTERFACE
+IN
S/H
CDAC
-IN
COMPARATOR
PIN DESCRIPTIONS
Pin No.
Symbol
Description
1
VREF
Voltage Reference Input. A voltage reference between 1V and VA must be applied to this
input. VREF must be decoupled to GND with a minimum ceramic capacitor value of 0.1 µF.
A bulk capacitor value of 1.0 µF to 10 µF in parallel with the 0.1 µFcapacitor is
recommended for enhanced performance.
2
+IN
Non-Inverting Input. +IN is the positive analog input for the differential signal applied to the
ADC141S626.
3
−IN
Inverting Input. −IN is the negative analog input for the differential signal applied to the
ADC141S626.
4
GND
Ground. GND is the ground reference point for all signals applied to the ADC141S626.
5
GND
Ground. GND is the ground reference point for all signals applied to the ADC141S626.
6
CS
7
DOUT
Serial Data Output. The conversion result is provided on DOUT. The serial data output word
is comprised of 2 null bits followed by 14 data bits (MSB first). During a conversion, the
data is output on the falling edges of SCLK and is valid on the subsequent rising edges.
8
SCLK
Serial Clock. SCLK is used to control data transfer and serves as the conversion clock.
9
VIO
Digital Input/Output Power Supply Input. A voltage source between 2.7V and 5.5V must be
applied to this input. VIO must be decoupled to GND with a ceramic capacitor value of 0.1
µF in parallel with a bulk capacitor value of 1.0 µF to 10 µF.
10
VA
Analog Power Supply Input. A voltage source between 2.7V and 5.5V must be applied to
this input. VA must be decoupled to GND with a ceramic capacitor value of 0.1 µF in
parallel with a bulk capacitor value of 1.0 µF to 10 µF.
Chip Select Bar. CS must be active LOW during an SPI conversion, which begins on the
falling edge of CS. The ADC141S626 is in acquisition mode when CS is HIGH.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
2
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Absolute Maximum Ratings (1) (2) (3)
−0.3V to 6.5V
Analog Supply Voltage VA
−0.3V to 6.5V
Digital I/O Supply Voltage VIO
Voltage on Any Analog Input Pin to GND
−0.3V to (VA + 0.3V)
Voltage on Any Digital Input Pin to GND
−0.3V to (VIO + 0.3V)
Input Current at Any Pin
(4)
±10 mA
Package Input Current (4)
±50 mA
See (5)
Power Consumption at TA = 25°C
Human Body Model
ESD Susceptibility (6)
4000V
Machine Model
300V
Charge Device Model
1250V
Junction Temperature
+150°C
Storage Temperature
−65°C to +150°C
(1)
(2)
(3)
(4)
(5)
(6)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see
the Electrical Characteristics. The guaranteed specifications apply only for the test conditions listed. Some performance characteristics
may degrade when the device is not operated under the listed test conditions. Operation of the device beyond the maximum Operating
Ratings is not recommended.
All voltages are measured with respect to GND = 0V, unless otherwise specified.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
When the input voltage at any pin exceeds the power supplies (that is, VIN < GND or VIN > VA), the current at that pin should be limited
to 10 mA. The 50 mA maximum package input current rating limits the number of pins that can safely exceed the power supplies with an
input current of 10 mA to five.
The absolute maximum junction temperature (TJmax) for this device is 150°C. The maximum allowable power dissipation is dictated by
TJmax, the junction-to-ambient thermal resistance (θJA), and the ambient temperature (TA), and can be calculated using the formula
PDMAX = (TJmax − TA)/θJA. The values for maximum power dissipation listed above will be reached only when the ADC141S626 is
operated in a severe fault condition (e.g. when input or output pins are driven beyond the power supply voltages, or the power supply
polarity is reversed). Such conditions should always be avoided.
Human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor. Machine model is a 220 pF capacitor discharged
through 0 Ω. Charge device model simulates a pin slowly acquiring charge (such as from a device sliding down the feeder in an
automated assembler) then rapidly being discharged.
Operating Ratings (1) (2)
−40°C ≤ TA ≤ +85°C
Operating Temperature Range
Supply Voltage, VA
+2.7V to +5.5V
Supply Voltage, VIO
+2.7V to +5.5V
Reference Voltage, VREF
1.0V to VA
Analog Input Pins Voltage Range
0 to VA
−VREF to +VREF
Differential Analog Input Voltage
Input Common-Mode Voltage, VCM
See Figure 41
Digital Input Pins Voltage Range
0 to VIO
Clock Frequency
(1)
(2)
0.9 MHz to 4.5 MHz
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see
the Electrical Characteristics. The guaranteed specifications apply only for the test conditions listed. Some performance characteristics
may degrade when the device is not operated under the listed test conditions. Operation of the device beyond the maximum Operating
Ratings is not recommended.
All voltages are measured with respect to GND = 0V, unless otherwise specified.
Package Thermal Resistance (1) (2)
(1)
(2)
Package
θJA
10-lead VSSOP
240°C / W
Soldering process must comply with TI's Reflow Temperature Profile specifications. Refer to www.ti.com/packaging.
Reflow temperature profiles are different for lead-free packages.
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ADC141S626 Converter Electrical Characteristics (1)
The following specifications apply for VA = VIO = VREF = +2.7V to 5.5V and fSCLK = 0.9 to 3.6 MHz or VA = VIO = VREF = +4.5V
to 5.5V and fSCLK = 3.6 to 4.5 MHz; fIN = 20 kHz and CL = 25 pF, unless otherwise noted. Boldface limits apply for TA = TMIN
to TMAX; all other limits are at TA = 25°C.
Symbol
Parameter
Conditions
Typical
Limits
Units
14
Bits
STATIC CONVERTER CHARACTERISTICS
Resolution with No Missing Codes
INL
Integral Non-Linearity
±0.5
±0.95
LSB (max)
DNL
Differential Non-Linearity
±0.5
±0.95
LSB (max)
OE
Offset Error
−1
±5
LSB (max)
Positive Full-Scale Error
−3
±7
LSB (max)
Negative Full-Scale Error
0.5
±4
LSB (max)
Positive Gain Error
−1.5
±6
LSB (max)
Negative Gain Error
1.5
±6
LSB (max)
VA = VIO = VREF = +3V, −0.1 dBFS
81.9
80.1
dBc (min)
VA = VIO = VREF = +5V, −0.1 dBFS
84.2
82
dBc (min)
VA = VIO = VREF = +3V, −0.1 dBFS
82
80.2
dBc (min)
VA = VIO = VREF = +5V, −0.1 dBFS
84.3
82
dBc (min)
VA = VIO = VREF = +3V, −0.1 dBFS
−102
dBc
VA = VIO = VREF = +5V, −0.1 dBFS
−102
dBc
VA = VIO = VREF = +3V, −0.1 dBFS
97
dBc
VA = VIO = VREF = +5V, −0.1 dBFS
101
VA = VIO = VREF = +3V, −0.1 dBFS
13.3
13.0
bits (min)
VA = VIO = VREF = +5V, −0.1 dBFS
13.7
13.3
bits (min)
FSE
GE
DYNAMIC CONVERTER CHARACTERISTICS
SINAD
Signal-to-Noise Plus Distortion Ratio
SNR
Signal-to-Noise Ratio
THD
SFDR
ENOB
FPBW
Total Harmonic Distortion
Spurious-Free Dynamic Range
Effective Number of Bits
−3 dB Full Power Bandwidth
Output at 70.7%FS
with FS Input
dBc
Differential Input
26
MHz
Single-Ended
Input
22
MHz
ANALOG INPUT CHARACTERISTICS
VIN
Differential Input Range
IDCL
DC Leakage Current
CINA
Input Capacitance
CMRR
Common Mode Rejection Ratio
VIN = VREF or VIN = -VREF
−VREF
V (min)
+VREF
V (max)
±1
µA (max)
In Acquisition Mode
30
pF
In Conversion Mode
3
pF
See the Specification Definitions for the
test condition
76
dB
DIGITAL INPUT CHARACTERISTICS
VIH
Input High Voltage
VIO = +2.7V to 5.5V
1.9
2.3
V (min)
VIL
Input Low Voltage
VIO = +2.7V to 5.5V
1.0
0.7
V (max)
IIN
Input Current
VIN = 0V or VA
CIND
Input Capacitance
±1
µA (max)
2
4
pF (max)
ISOURCE = 200 µA
VA − 0.05
VA − 0.2
V (min)
ISOURCE = 1 mA
VA − 0.16
ISINK = 200 µA
0.01
ISINK = 1 mA
0.05
DIGITAL OUTPUT CHARACTERISTICS
VOH
Output High Voltage
VOL
Output Low Voltage
IOZH, IOZL TRI-STATE Leakage Current
Force 0V or VA
COUT
Force 0V or VA
TRI-STATE Output Capacitance
Output Coding
(1)
4
2
V
0.4
V (max)
±1
µA (max)
4
pF (max)
V
Binary 2'S Complement
Typical values are at TJ = 25°C and represent most likely parametric norms. Test limits are guaranteed to TI's AOQL (Average Outgoing
Quality Level).
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ADC141S626 Converter Electrical Characteristics(1) (continued)
The following specifications apply for VA = VIO = VREF = +2.7V to 5.5V and fSCLK = 0.9 to 3.6 MHz or VA = VIO = VREF = +4.5V
to 5.5V and fSCLK = 3.6 to 4.5 MHz; fIN = 20 kHz and CL = 25 pF, unless otherwise noted. Boldface limits apply for TA = TMIN
to TMAX; all other limits are at TA = 25°C.
Symbol
Parameter
Conditions
Typical
Limits
Units
2.7
V (min)
5.5
V (max)
POWER SUPPLY CHARACTERISTICS
VA
Analog Supply Voltage Range
VIO
Digital Input/Output Supply Voltage Range See (2)
VREF
Reference Voltage Range
IVA
(Conv)
Analog Supply Current, Conversion Mode
IVIO
(Conv)
Digital I/O Supply Current, Conversion
Mode
740
970
µA (max)
fSCLK = 3.6 MHz, VA = 3V, fS = 200
kSPS, fIN = 20 kHz
90
190
µA (max)
fSCLK = 4.5 MHz, VA = 5V, fS = 250
kSPS, fIN = 20 kHz
170
260
µA (max)
fSCLK = 3.6 MHz, VA = 3V, fS = 200
kSPS, fIN = 20 kHz
25
60
µA (max)
fSCLK = 4.5 MHz, VA = 5V, fS = 250
kSPS, fIN = 20 kHz
45
80
µA (max)
8
IVIO (PD)
Digital I/O Supply Current, Power Down
Mode (CS high)
fSCLK = 4.5 MHz, VA = 5V
fSCLK = 0 (3)
0.1
IVREF
(PD)
Reference Current, Power Down Mode
(CS high)
fSCLK = 4.5 MHz, VA = 5V
0.1
fSCLK = 0 (3)
Power Supply Rejection Ratio
V (max)
fSCLK = 4.5 MHz, VA = 5V, fS = 250
kSPS, fIN = 20 kHz
fSCLK = 4.5 MHz, VA = 5V
PSRR
V (min)
VA
µA (max)
Analog Supply Current, Power Down
Mode (CS high)
fSCLK = 0
Power Consumption, Power Down Mode
(CS high)
1.0
760
IVA (PD)
PWR
(PD)
V (max)
540
Reference Current, Conversion Mode
Power Consumption, Conversion Mode
V (min)
5.5
fSCLK = 3.6 MHz, VA = 3V, fS = 200
kSPS, fIN = 20 kHz
IVREF
(Conv)
PWR
(Conv)
2.7
(3)
2
µA
3
µA (max)
0.3
µA (max)
0.1
0.2
µA (max)
fSCLK = 3.6 MHz, fS = 200 kSPS, fIN = 20
kHz, VA = VIO = VREF = 3.0V
2.0
3.0
mW
fSCLK = 4.5 MHz, fS = 250 kSPS, fIN = 20
kHz, VA = VIO = VREF = 5.0V
4.8
6.5
mW
3
4
µW (max)
13
17
µW (max)
3
fSCLK = 0, VA = VIO = VREF = 3.0V (3)
fSCLK = 0, VA = VIO = VREF = 5.0V
(3)
See the Specification Definitions for the
test condition.
−85
VA = VIO = VREF = +2.7V to 5.5V
4.8
µA
µA
dB
AC ELECTRICAL CHARACTERISTICS
fSCLK
Maximum Clock Frequency
4.5
MHz (min)
fSCLK
Minimum Clock Frequency
0.9
MHz (max)
fS
Maximum Sample Rate (4)
250
kSPS (min)
tACQ
Acquisition/Track Time
667
ns (min)
tCONV
Conversion/Hold Time
15
SCLK cycles
tAD
Aperture Delay
(2)
(3)
(4)
See the Specification Definitions.
6
ns
The value of VIO is independent of the value of VA. For example, VIO could be operating at 5V while VA is operating at 3V or VIO could
be operating at 3V while VA is operating at 5V.
This parameter is guaranteed by design and/or characterization and is not tested in production.
While the maximum sample rate is fSCLK/18, the actual sample rate may be lower than this by having the CS rate slower than fSCLK/18.
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ADC141S626
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ADC141S626 Timing Specifications (1)
The following specifications apply for VA = VIO = VREF= +2.7V to 5.5V and fSCLK = 0.9 to 4.5 MHz, CL = 25 pF, Boldface limits
apply for TA = TMIN to TMAX: all other limits TA = 25°C.
Symbol
(1)
(2)
Parameter
Conditions
Typical
Limits
Units
3
6
ns (min)
ns (max)
tCSS
CS Setup Time prior to an SCLK rising edge
1/fSCLK - 3
1/fSCLK - 6
tDH
DOUT Hold Time after an SCLK falling edge
10
6
ns (min)
tDA
DOUT Access Time after an SCLK falling edge
28
40
ns (max)
tDIS
DOUT Disable Time after the rising edge of CS (2)
10
20
ns (max)
tCS
Minimum CS Pulse Width
5
20
ns (min)
tEN
DOUT Enable Time after the falling edge of CS
32
51
ns (max)
tCH
SCLK High Time
67
89
ns (min)
tCL
SCLK Low Time
67
89
ns (min)
tr
DOUT Rise Time
7
ns
tf
DOUT Fall Time
7
ns
Typical values are at TJ = 25°C and represent most likely parametric norms. Test limits are guaranteed to TI's AOQL (Average Outgoing
Quality Level).
tDIS is the time for DOUT to change 10% while being loaded by the Timing Test Circuit.
Timing Diagrams
tACQ (Power-Down)
tCONV (Power-Up)
CS
1
2
3
4
tCH
5
tCS
11
12
13
14
15
17
16
18
1
2
SCLK
tEN
D`
0
tDIS
tCL
0
DB13 DB12
DB5
DB4 DB3
DB2
DB1
DB0
0
0
Figure 1. ADC141S626 Single Conversion Timing Diagram
SCLK
2 mA
1
2
IOL
tCSS
TO OUTPUT
PIN
CS
1.6V
CL
25 pF
2 mA
IOH
Figure 2. Timing Test Circuit
6
Figure 3. Valid CS Assertion Times
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0.9 x VIO
DOUT
CS
0.1 x VIO
tr
VIH
tf
90%
90%
DOUT
10%
tDIS
90%
DOUT
10%
10%
Figure 4. DOUT Rise and Fall Times
Figure 5. Voltage Waveform for tDIS
SCLK
VIL
tDA
2.3V
DOUT
0.7V
tDH
Figure 6. DOUT Hold and Access Times
Specification Definitions
APERTURE DELAY is the time between the first falling edge of SCLK and the time when the input signal is
sampled for conversion.
COMMON MODE REJECTION RATIO (CMRR) is a measure of how well in-phase signals common to both input
pins are rejected.
To calculate CMRR, the change in output offset is measured while the common mode input voltage is changed
from 2V to 3V.
CMRR = 20 LOG ( Δ Common Input / Δ Output Offset)
(1)
CONVERSION TIME is the time required, after the input voltage is acquired, for the ADC to convert the input
voltage to a digital word.
DIFFERENTIAL NON-LINEARITY (DNL) is the measure of the maximum deviation from the ideal step size of 1
LSB.
DUTY CYCLE is the ratio of the time that a repetitive digital waveform is high to the total time of one period. The
specification here refers to the SCLK.
EFFECTIVE NUMBER OF BITS (ENOB, or EFFECTIVE BITS) is another method of specifying Signal-to-Noise
and Distortion or SINAD. ENOB is defined as (SINAD − 1.76) / 6.02 and says that the converter is equivalent to
a perfect ADC of this (ENOB) number of bits.
FULL POWER BANDWIDTH is a measure of the frequency at which the reconstructed output fundamental
drops 3 dB below its low frequency value for a full scale input.
GAIN ERROR is the deviation from the ideal slope of the transfer function. It is the difference between Positive
Full-Scale Error and Negative Full-Scale Error and can be calculated as:
Gain Error = Positive Full-Scale Error − Negative Full-Scale Error
(2)
INTEGRAL NON-LINEARITY (INL) is a measure of the deviation of each individual code from a line drawn from
½ LSB below the first code transition through ½ LSB above the last code transition. The deviation of any given
code from this straight line is measured from the center of that code value.
MISSING CODES are those output codes that will never appear at the ADC outputs. The ADC141S626 is
guaranteed not to have any missing codes.
NEGATIVE FULL-SCALE ERROR is the difference between the differential input voltage at which the output
code transitions from negative full scale to the next code and −VREF + 1 LSB
NEGATIVE GAIN ERROR is the difference between the negative full-scale error and the offset error.
OFFSET ERROR is the difference between the differential input voltage at which the output code transitions from
code 0000h to 0001h and 1 LSB.
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POSITIVE FULL-SCALE ERROR is the difference between the differential input voltage at which the output
code transitions to positive full scale and VREF minus 1 LSB.
POSITIVE GAIN ERROR is the difference between the positive full-scale error and the offset error.
POWER SUPPLY REJECTION RATIO (PSRR) is a measure of how well a change in the analog supply voltage
is rejected. PSRR is calculated from the ratio of the change in offset error for a given change in supply voltage,
expressed in dB. For the ADC141S626, VA is changed from 4.5V to 5.5V.
PSRR = 20 LOG (ΔOutput Offset / ΔVA)
(3)
SIGNAL TO NOISE RATIO (SNR) is the ratio, expressed in dB, of the rms value of the input signal to the rms
value of the sum of all other spectral components below one-half the sampling frequency, not including
harmonics or d.c.
SIGNAL TO NOISE PLUS DISTORTION (S/N+D or SINAD) Is the ratio, expressed in dB, of the rms value of the
input signal to the rms value of all of the other spectral components below one-half the sampling frequency,
including harmonics but excluding d.c.
SPURIOUS FREE DYNAMIC RANGE (SFDR) is the difference, expressed in dB, between the desired signal
amplitude to the amplitude of the peak spurious spectral component below one-half the sampling frequency,
where a spurious spectral component is any signal present in the output spectrum that is not present at the input
and may or may not be a harmonic.
TOTAL HARMONIC DISTORTION (THD) is the ratio of the rms total of the first five harmonic components at the
output to the rms level of the input signal frequency as seen at the output, expressed in dB. THD is calculated as
THD = 20 ‡ log10
A f 22 +
+ A f 62
A f 12
(4)
where Af1 is the RMS power of the input frequency at the output and Af2 through Af6 are the RMS power in the
first 5 harmonic frequencies.
THROUGHPUT TIME is the minimum time required between the start of two successive conversion.
8
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Typical Performance Characteristics
VA = VIO = VREF = +5V, fSCLK = 4.5 MHz, fSAMPLE = 250 kSPS, TA = +25°C, and fIN = 20 kHz unless otherwise stated.
DNL - 250 kSPS
INL - 250 kSPS
Figure 7.
Figure 8.
DNL vs. VA
INL vs. VA
Figure 9.
Figure 10.
DNL vs. VREF
INL vs. VREF
Figure 11.
Figure 12.
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Typical Performance Characteristics (continued)
VA = VIO = VREF = +5V, fSCLK = 4.5 MHz, fSAMPLE = 250 kSPS, TA = +25°C, and fIN = 20 kHz unless otherwise stated.
10
DNL vs. SCLK FREQUENCY
INL vs. SCLK FREQUENCY
Figure 13.
Figure 14.
DNL vs. TEMPERATURE
INL vs. TEMPERATURE
Figure 15.
Figure 16.
SINAD vs. VA
THD vs. VA
Figure 17.
Figure 18.
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Typical Performance Characteristics (continued)
VA = VIO = VREF = +5V, fSCLK = 4.5 MHz, fSAMPLE = 250 kSPS, TA = +25°C, and fIN = 20 kHz unless otherwise stated.
SINAD vs. VREF
THD vs. VREF
Figure 19.
Figure 20.
SINAD vs. SCLK FREQUENCY
THD vs. SCLK FREQUENCY
Figure 21.
Figure 22.
SINAD vs. INPUT FREQUENCY
THD vs. INPUT FREQUENCY
Figure 23.
Figure 24.
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Typical Performance Characteristics (continued)
VA = VIO = VREF = +5V, fSCLK = 4.5 MHz, fSAMPLE = 250 kSPS, TA = +25°C, and fIN = 20 kHz unless otherwise stated.
12
SINAD vs. TEMPERATURE
THD vs. TEMPERATURE
Figure 25.
Figure 26.
VA CURRENT vs. VA
VA CURRENT vs. SCLK FREQUENCY
Figure 27.
Figure 28.
VA CURRENT vs. TEMPERATURE
VREF CURRENT vs. VREF
Figure 29.
Figure 30.
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Typical Performance Characteristics (continued)
VA = VIO = VREF = +5V, fSCLK = 4.5 MHz, fSAMPLE = 250 kSPS, TA = +25°C, and fIN = 20 kHz unless otherwise stated.
VREF CURRENT vs. SCLK FREQUENCY
VREF CURRENT vs. TEMPERATURE
Figure 31.
Figure 32.
VIO CURRENT vs. VIO
VIO CURRENT vs. SCLK FREQUENCY
Figure 33.
Figure 34.
VIO CURRENT vs. TEMPERATURE
SPECTRAL RESPONSE - 250 kSPS
Figure 35.
Figure 36.
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FUNCTIONAL DESCRIPTION
The ADC141S626 is a 14-bit, 50 kSPS to 250 kSPS sampling Analog-to-Digital (A/D) converter. The converter
uses a successive approximation register (SAR) architecture based upon capacitive redistribution containing an
inherent sample-and-hold function. The differential nature of the analog inputs is maintained from the internal
sample-and-hold circuits throughout the A/D converter to provide excellent common-mode signal rejection.
The ADC141S626 operates from independent analog and digital supplies. The analog supply (VA) can range
from 2.7V to 5.5V and the digital input/output supply (VIO) can range from 2.7V to 5.5V. The ADC141S626
utilizes an external reference (VREF), which can be any voltage between 1V and VA. The value of VREF
determines the range of the analog input, while the reference input current (IREF) depends upon the conversion
rate.
The analog input is presented to two input pins: +IN and –IN. Upon initiation of a conversion, the differential input
at these pins is sampled on the internal capacitor array. The inputs are disconnected from the internal circuitry
while a conversion is in progress. The ADC141S626 features a zero-power track mode where the ADC is
consuming the minimum amount of supply current while the internal sampling capacitor is tracking the applied
analog input voltage. Zero-power track mode is exercised by bringing chip select bar (CS) high or low after the
conversion is complete (after the 16th falling edge of the serial clock).
The ADC141S626 communicates with other devices via Serial Peripheral Interface (SPI™) , a synchronous serial
interface that operates using three pins: chip select bar (CS), serial clock (SCLK), and serial data out (DOUT). The
external SCLK controls data transfer and serves as the conversion clock. The duty cycle of SCLK is essentially
unimportant, provided the minimum clock high and low times are met. The minimum SCLK frequency is set by
internal capacitor leakage. Each conversion requires 18 SCLK cycles to complete. If less than 14 bits of
conversion data are required, CS can be brought high at any point during the conversion. This procedure of
terminating a conversion prior to completion is commonly referred to as short cycling.
The digital conversion result is clocked out by the SCLK input and is provided serially, most significant bit (MSB)
first, at the DOUT pin. The digital data that is provided at the DOUT pin is that of the conversion currently in
progress and thus there is no pipe line delay.
REFERENCE INPUT (VREF)
The externally supplied reference voltage (VREF) sets the analog input range. The ADC141S626 will operate with
VREF in the range of 1V to VA.
Operation with VREF below 1V is also possible with slightly diminished performance. As VREF is reduced, the
range of acceptable analog input voltages is reduced. Assuming a proper common-mode input voltage (VCM), the
differential peak-to-peak input range is limited to (2 x VREF). See Input Common Mode Voltage for more details.
Reducing VREF also reduces the size of the least significant bit (LSB). The size of one LSB is equal to [(2 x VREF)
/ 2n], which is 16,384 where n is 14 bits. When the LSB size goes below the noise floor of the ADC141S626, the
noise will span an increasing number of codes and overall performance will suffer. For example, dynamic signals
will have their SNR degrade, while D.C. measurements will have their code uncertainty increase. Since the noise
is Gaussian in nature, the effects of this noise can be reduced by averaging the results of a number of
consecutive conversions.
Additionally, since offset and gain errors are specified in LSB, any offset and/or gain errors inherent in the A/D
converter will increase in terms of LSB size as VREF is reduced.
VREF and analog inputs (+IN and -IN) are connected to the capacitor array through a switch matrix when the input
is sampled. Hence, IREF, I+IN, and I-IN are a series of transient spikes that occur at a frequency dependent on the
operating sample rate of the ADC141S626.
IREF changes only slightly with temperature. See Figure 31 and Figure 32 in Typical Performance Characteristics
for additional details.
14
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ANALOG SIGNAL INPUTS
The ADC141S626 has a differential input where the effective input voltage that is digitized is (+IN) − (−IN). By
using this differential input, small signals common to both inputs are rejected. As shown in Figure 37, noise is
immune at low frequencies where the common-mode rejection ratio (CMRR) is 90 dB. As the frequency
increases to 1 MHz, the CMRR rolls off to 40 dB . In general, operation with a fully differential input signal or
voltage will provide better performance than with a single-ended input. However, if desired, the ADC141S626
can be presented with a single-ended input.
Figure 37. Analog Input CMRR vs. Frequency
The current required to recharge the input sampling capacitor will cause voltage spikes at +IN and −IN. Do not
try to filter out these noise spikes. Rather, ensure that the transient settles out during the acquisition period.
Differential Input Operation
As shown in Figure 38 for a fully differential input signal, a positive full scale output code (01 1111 1111 1111b or
1FFFh or 8191d) will be obtained when (+IN) − (−IN) is greater than or equal to (VREF − 1 LSB). A negative full
scale code (10 0000 0000 0000b or 2000h or -8192d) will be obtained when (+IN) − (−IN) is less than or equal to
(−VREF + 1 LSB). This ignores gain, offset and linearity errors, which will affect the exact differential input voltage
that will determine any given output code. Both inputs should be biased at a common mode voltage (VCM), which
will be thoroughly discussed in Input Common Mode Voltage . Figure 39 shows the ADC141S626 being driven
by a full-scale differential source.
01 1111 1111 1111b
|
00 0000 0000 0000b
+1 LSB
|
- VREF +1LSB
|
-1 LSB
|
ADC Output Code
|
+VREF ±1LSB
|
10 0000 0000 0000b
Analog Input
Figure 38. ADC Output vs. Input for a Differential Input Operation
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VREF
2
VCM
VCM +
RS
VCM
VREF
2
-
VREF
SRC
+
CS
RS
ADC141S626
-
VREF
2
VCM
VCM +
VCM
-
VREF
2
Figure 39. Differential Input
Single-Ended Input Operation
For single-ended operation, the non-inverting input (+IN) of the ADC141S626 can be driven with a signal that has
a peak-to-peak range that is equal to or less than (2 x VREF). The inverting input (−IN) should be biased at a
stable VCM that is halfway between these maximum and minimum values. In order to utilize the entire dynamic
range of the ADC141S626, VREF is limited to VA / 2. This allows +IN a maximum swing range of ground to VA.
Figure 40 shows the ADC141S626 being driven by a full-scale single-ended source.
VCM + VREF
VCM
VCM
- VREF
RS
VREF
SRC
+
CS
ADC141S626
-
VCM
Figure 40. Single-Ended Input
Since the design of the ADC141S626 is optimized for a differential input, the performance degrades slightly when
driven with a single-ended input. Linearity characteristics such as INL and DNL typically degrade by 0.1 LSB and
dynamic characteristics such as SINAD typically degrade by 2 dB. Note that single-ended operation should only
be used if the performance degradation (compared with differential operation) is acceptable.
Input Common Mode Voltage
The allowable input common mode voltage (VCM) range depends upon VA and VREF used for the ADC141S626.
The ranges of VCM are depicted in Figure 41 and Figure 42. Note that these figures only apply to a VA of 5V.
Equations for calculating the minimum and maximum VCM for differential and single-ended operations are shown
in Table 1.
6
6
Single-Ended Input
COMMON-MODE VOLTAGE (V)
COMMON-MODE VOLTAGE (V)
Differential Input
5
VA = 5.0V
3.75
2.5
1.25
0
-1
0.0
1.0
2.0 2.5 3.0
4.0
5
VA = 5.0V
3.75
2.5
1.25
-1
0.0
5.0
VREF (V)
0.75
1.25
1.75
2.5
VREF (V)
Figure 41. VCM range for Differential Input
operation
16
0
Figure 42. VCM range for single-ended operation
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Table 1. Allowable VCM Range
Input Signal
Differential
Single-Ended
Minimum VCM
Maximum VCM
VREF / 2
VA − VREF / 2
VREF
VA − VREF
SERIAL DIGITAL INTERFACE
The ADC141S626 communicates via a synchronous 3-wire serial interface as shown in Figure 1 or re-shown in
Figure 43 for convenience. CS, chip select bar, initiates conversions and frames the serial data transfers. SCLK
(serial clock) controls both the conversion process and the timing of serial data. DOUT is the serial data output
pin, where a conversion result is sent as a serial data stream, MSB first.
A serial frame is initiated on the falling edge of CS and ends on the rising edge of CS. The ADC141S626's DOUT
pin is in a high impedance state when CS is high and is active when CS is low; thus. CS acts as an output
enable.
The ADC141S626 samples the differential input upon the assertion of CS. Assertion is defined as bringing the
CS pin to a logic low state. For the first 15 periods of the SCLK following the assertion of CS, the ADC141S626
is converting the analog input voltage. On the 16th falling edge of SCLK, the ADC141S626 enters acquisition
(tACQ) mode. For the next three periods of SCLK, the ADC141S626 is operating in acquisition mode where the
ADC input is tracking the analog input signal applied across +IN and -IN. During acquisition mode, the
ADC141S626 is consuming a minimal amount of power.
The ADC141S626 can enter conversion mode (tCONV) under three different conditions. The first condition
involves CS going low (asserted) with SCLK high. In this case, the ADC141S626 enters conversion mode on the
first falling edge of SCLK after CS is asserted. In the second condition, CS goes low with SCLK low. Under this
condition, the ADC141S626 automatically enters conversion mode and the falling edge of CS is seen as the first
falling edge of SCLK. In the third condition, CS and SCLK go low simultaneously and the ADC141S626 enters
conversion mode. While there is no timing restriction with respect to the falling edges of CS and SCLK, there is a
minimum and maximum setup time requirements for the falling edge of CS with respect to the rising edge of
SCLK. See Figure 3 in the Timing Diagrams section for more information.
CS Input
The CS (chip select bar) input is active low and is TTL and CMOS compatible. The ADC141S626 enters
conversion mode when CS is asserted and the SCLK pin is in a logic low state. When CS is high, the
ADC141S626 is always in acquisition mode and thus consuming the minimum amount of power. Since CS must
be asserted to begin a conversion, the sample rate of the ADC141S626 is equal to the assertion rate of CS.
Proper operation requires that the fall of CS not occur simultaneously with a rising edge of SCLK. If the fall of CS
occurs during the rising edge of SCLK, the data might be clocked out one bit early. Whether or not the data is
clocked out early depends upon how close the CS transition is to the SCLK transition, the device temperature,
and the characteristics of the individual device. To ensure that the MSB is always clocked out at a given time
(the 3rd falling edge of SCLK), it is essential that the fall of CS always meet the timing requirement specified in
the Timing Specifications.
SCLK Input
The SCLK (serial clock) is used as the conversion clock to shift out the conversion result. SCLK is TTL and
CMOS compatible. Internal settling time requirements limit the maximum clock frequency while internal capacitor
leakage limits the minimum clock frequency. The ADC141S626 offers guaranteed performance with the clock
rates indicated in the Electrical Characteristics.
The ADC141S626 enters acquisition mode on the 16th falling edge of SCLK during a conversion frame.
Assuming that the LSB is clocked into a controller on the 16th rising edge of SCLK, there is a minimum
acquisition time period that must be met before a new conversion frame can begin. Other than the 16th rising
edge of SCLK that was used to latch the LSB into a controller, there is no requirement for the SCLK to transition
during acquisition mode. Therefore, it is acceptable to idle SCLK after the LSB has been latched into the
controller.
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Data Output
The data output format of the ADC141S626 is two’s complement as shown in Figure 38. This figure indicates the
ideal output code for a given input voltage and does not include the effects of offset, gain error, linearity errors, or
noise. Each data output bit is output on the falling edges of SCLK. The 1st and 2nd SCLK falling edges clock out
leading zeros while the 3rd to 16th SCLK falling edges clock out the conversion result, MSB first.
While most receiving systems will capture the digital output bits on the rising edges of SCLK, the falling edges of
SCLK may be used to capture the conversion result if the minimum hold time for DOUT is acceptable. See
Figure 6 for DOUT hold (tDH) and access (tDA) times.
DOUT is enabled on the falling edge of CS and disabled on the rising edge of CS. If CS is raised prior to the 16th
falling edge of SCLK, the current conversion is aborted and DOUT will go into its high impedance state. A new
conversion will begin when CS is driven LOW.
tACQ (Power-Down)
tCONV (Power-Up)
CS
1
2
3
4
tCH
5
tCS
11
12
13
14
15
17
16
18
1
2
SCLK
tEN
D`
0
tDIS
tCL
0
DB13 DB12
DB5
DB4 DB3
DB2
DB1
DB0
0
0
Figure 43. ADC141S626 Single Conversion Timing Diagram
Applications Information
OPERATING CONDITIONS
We recommend that the following conditions be observed for operation of the ADC141S626:
−40°C ≤ TA ≤ +85°C
+2.7V ≤ VA ≤ +5.5V
+2.7V ≤ VIO ≤ +5.5V
1V ≤ VREF ≤ VA
0.9 MHz ≤ fSCLK ≤ 4.5 MHz
VCM: See Input Common Mode Voltage
POWER CONSUMPTION
The architecture, design, and fabrication process allow the ADC141S626 to operate at conversion rates up to
250 kSPS while consuming very little power. The ADC141S626 consumes the least amount of power while
operating in acquisition (power-down) mode. For applications where power consumption is critical, the
ADC141S626 should be operated in acquisition mode as often as the application will tolerate. To further reduce
power consumption, stop the SCLK while CS is high.
Short Cycling
Short cycling refers to the process of halting a conversion after the last needed bit is outputted. Short cycling can
be used to lower the power consumption in those applications that do not need a full 14-bit resolution, or where
an analog signal is being monitored until some condition occurs. In some circumstances, the conversion could be
terminated after the first few bits. This will lower power consumption in the converter since the ADC141S626
spends more time in acquisition mode and less time in conversion mode.
Short cycling is accomplished by pulling CS high after the last required bit is received from the ADC141S626
output. This is possible because the ADC141S626 places the latest converted data bit on DOUT as it is
generated. If only 10-bits of the conversion result are needed, for example, the conversion can be terminated by
pulling CS high after the 10th bit has been clocked out.
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Burst Mode Operation
Normal operation of the ADC141S626 requires the SCLK frequency to be 18 times the sample rate and the CS
rate to be the same as the sample rate. However, in order to minimize power consumption in applications
requiring sample rates below 250 kSPS, the ADC141S626 should be run with an SCLK frequency of 4.5 MHz
and a CS rate as slow as the system requires. When this is accomplished, the ADC141S626 is operating in burst
mode. The ADC141S626 enters into acquisition mode at the end of each conversion, minimizing power
consumption. This causes the converter to spend the longest possible time in acquisition mode. Since power
consumption scales directly with conversion rate, minimizing power consumption requires determining the lowest
conversion rate that will satisfy the requirements of the system.
PCB LAYOUT AND CIRCUIT CONSIDERATIONS
For best performance, care should be taken with the physical layout of the printed circuit board. This is especially
true with a low VREF or when the conversion rate is high. At high clock rates there is less time for settling, so it is
important that any noise settles out before the conversion begins.
Analog and Digital Power Supplies
Any ADC architecture is sensitive to spikes on the power supply, reference, and ground pins. These spikes may
originate from switching power supplies, digital logic, high power devices, and other sources. Power to the
ADC141S626 should be clean and well bypassed. A 0.1 µF ceramic bypass capacitor and a 1 µF to 10 µF
capacitor should be used to bypass the ADC141S626 supply, with the 0.1 µF capacitor placed as close to the
ADC141S626 package as possible.
Since the ADC141S626 has both the VA and VIO pins, the user has three options on how to connect these pins.
The first option is to tie VA and VIO together and power them with the same power supply. This is the most cost
effective way of powering the ADC141S626 but is also the least ideal. As stated previously, noise from VIO can
couple into VA and adversely affect performance. The other two options involve the user powering VA and VIO
with separate supply voltages. These supply voltages can have the same amplitude or they can be different.
They may be set independent of each other to any value between 2.7V and 5.5V.
Best performance will typically be achieved with VA operating at 5V and VIO at 3V. Operating VA at 5V offers the
best linearity and dynamic performance when VREF is also set to 5V; while operating VIO at 3V reduces the power
consumption of the digital logic. Operating the digital interface at 3V also has the added benefit of decreasing the
noise created by charging and discharging the capacitance of the digital interface pins.
Voltage Reference
The reference source must have a low output impedance and needs to be bypassed with a minimum capacitor
value of 0.1 µF. A larger capacitor value of 1 µF to 10 µF placed in parallel with the 0.1 µF is preferred. While the
ADC141S626 draws very little current from the reference on average, there are higher instantaneous current
spikes at the reference.
VREF of the ADC141S626, like all A/D converters, does not reject noise or voltage variations. Keep this in mind if
VREF is derived from the power supply. Any noise and/or ripple from the supply that is not rejected by the external
reference circuitry will appear in the digital results. The use of an active reference source is recommended. The
LM4040 and LM4050 shunt reference families and the LM4132 and LM4140 series reference families are
excellent choices for a reference source.
PCB Layout
Capacitive coupling between the noisy digital circuitry and the sensitive analog circuitry can lead to poor
performance. The solution is to keep the analog circuitry separated from the digital circuitry and the clock line as
short as possible. Digital circuits create substantial supply and ground current transients. The logic noise
generated could have significant impact upon system noise performance. To avoid performance degradation of
the ADC141S626 due to supply noise, avoid using the same supply for the VA and VREF of the ADC141S626 that
is used for digital circuitry on the board.
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Generally, analog and digital lines should cross each other at 90° to avoid crosstalk. However, to maximize
accuracy in high resolution systems, avoid crossing analog and digital lines altogether. It is important to keep
clock lines as short as possible and isolated from ALL other lines, including other digital lines. In addition, the
clock line should also be treated as a transmission line and be properly terminated. The analog input should be
isolated from noisy signal traces to avoid coupling of spurious signals into the input. Any external component
(e.g., a filter capacitor) connected between the converter's input pins and ground or to the reference input pin
and ground should be connected to a very clean point in the ground plane.
A single, uniform ground plane and the use of split power planes are recommended. The power planes should be
located within the same board layer. All analog circuitry (input amplifiers, filters, reference components, etc.)
should be placed over the analog power plane. All digital circuitry should be placed over the digital power plane.
Furthermore, the GND pins on the ADC141S626 and all the components in the reference circuitry and input
signal chain that are connected to ground should be connected to the ground plane at a quiet point. Avoid
connecting these points too close to the ground point of a microprocessor, microcontroller, digital signal
processor, or other high power digital device.
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APPLICATION CIRCUITS
The following figures are examples of the ADC141S626 in typical application circuits. These circuits are basic
and will generally require modification for specific circumstances.
Data Acquisition
Figure 44 shows a typical connection diagram for the ADC141S626 operating at VA of +5V. VREF is connected to
a 4.1V shunt reference, the LM4040-4.1, to define the analog input range of the ADC141S626 independent of
supply variation on the +5V supply line. The VREF pin should be de-coupled to the ground plane by a 0.1 µF
ceramic capacitor and a tantalum capacitor of 10 µF. It is important that the 0.1 µF capacitor be placed as close
as possible to the VREF pin while the placement of the tantalum capacitor is less critical. It is also recommended
that the VA and VIO pins of the ADC141S626 be de-coupled to ground by a 0.1 µF ceramic capacitor in parallel
with a 10 µF tantalum capacitor.
+5V
+
100:
10 PF
ADC141S626
+
LM4040-4.1
VREF
VA
0.1 PF
0.1 PF
10 PF
VIO
+IN
SCLK
- IN
DOUT
GND
CSB
Controller
Figure 44. Low cost, low power Data Acquisition System
Bridge Sensor Application
Figure 45 shows an example of interfacing a bridge sensor to the ADC141S626. The application assumes that
the bridge sensor requires buffering and amplification to fully utilize the dynamic range of the ADC and thus
optimize the performance of the entire signal path. The amplification stage consists of the LMP7702, a dual
precision amplifier, and some gain setting passive components. The amplification stage offers the benefit of high
input impedance and high amplification capability. On the other hand, it offers no common-mode rejection of
common-mode noise or DC-voltage coming from the bridge sensor.
The DAC081S101, a digital-to-analog converter (DAC), is used to bias the bridge sensor. The DAC provides a
mean for dynamically adjusting the gain of the bridge sensor relative to actual maximum and minimum output
conditions. Another option for biasing the bridge sensor would be powering it from the same +5V power supply
voltage as the VA pin on the ADC141S626. This option has the benefit of providing the ideal common-mode input
voltage for the ADC141S626 while keeping design complexity and cost to a minimum. However, any fluctuation
in the +5V supply will still be visible in the converted result. The LM4132-4.1, a 4.1V series reference, is used as
the reference voltage in the application. The ADC141S626, DAC081S101, and the LM4132-4.1 are all powered
from the same +5V voltage source.
+5V
LMP7702
DAC081S101
+
-
470 pF
100 k:
180:
VA VIO
ADC141S626
REF
2 k:
100 k:
+
Bridge
Sensor
+5V +5V
SYNCB
DIN
SCLK
SCLK
DOUT
CSB
180:
LM4132-4.1
AV = 100 V/V
0.1 PF
+
MicroController
4.7 PF
+5V
+
4.7 PF
Figure 45. Interfacing the ADC141S626 to a Bridge Sensor
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Current Sensing Application
Figure 46 shows an example of interfacing a current transducer to the ADC141S626. The current transducer
converts an input current into a voltage that is converted by the ADC. Since the output voltage of the current
transducer is single-ended and centered around a common-mode voltage (VCM) of 2.5V, the ADC141S626 is
configured with the output of the transducer driving the non-inverting input and VCM of the transducer driving the
inverting input. The output of the transducer has an output range of ±2V around VCM of 2.5V. As a result, a series
reference voltage of 2.0V is connected to the ADC141S626. This will allow all of the codes of the ADC141S626
to be available for the application. This configuration of the ADC141S626 is referred to as a single-ended
application of a differential ADC. All of the elements in the application are conveniently powered by the same
+5V power supply, keeping circuit complexity and cost to a minimum.
+5V
+
10 PF
LM4132-2.0
+
VREF
10 PF
0.1 PF
ADC141S626
VA
0.1 PF
VIO
IIN
IOUT
OUT
+5V
IIN
IOUT VCM
GND
LTSR-1513¶V
2.5V + 2.0V
+IN
ADC
2.5V
Serial
Interface
SCLK
DOUT
CSB
-IN
GND
Figure 46. Interfacing the ADC141S626 to a Current Transducer
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ADC141S626
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SNAS434B – NOVEMBER 2007 – REVISED MARCH 2013
REVISION HISTORY
Changes from Revision A (March 2013) to Revision B
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 22
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Copyright © 2007–2013, Texas Instruments Incorporated
Product Folder Links: ADC141S626
23
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADC141S626CIMM/NOPB
ACTIVE
VSSOP
DGS
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
X94C
ADC141S626CIMMX/NOPB
ACTIVE
VSSOP
DGS
10
3500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
X94C
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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