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ADC31JB68
SLASE60B – SEPTEMBER 2015 – REVISED JANUARY 2019
ADC31JB68 Single-channel, 16-bit, 500-MSPS analog-to-digital converter
1 Features
2 Applications
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Single Channel
16-Bit Resolution
Maximum Clock Rate: 500 Msps
Small 40-Pin QFN Package (6 x 6 mm)
Input Buffer Input Bandwidth (3 dB): 1300 MHz
Aperture Jitter: 80 fs
On Chip Clock Divider: /1, /2, /4
On Chip Dither
Consistent Dynamic Performance Using
Foreground and Background Calibration
Input Amplitude and Phase Adjustment
Input Full Scale: 1.7 Vpp
Power Supplies: 1.2/1.8/3 V
JESD204B Interface
– Subclass 1 Compliant
– 2 Lanes at 5 Gbps
Support for Multi-chip Synchronization
Key Specifications
– Power Dissipation: 915 mW at 500 Msps
– Performance at fin = 210 MHz at –1 dBFS
– SNR: 69.3 dBFS
– NSD: –153.3 dBFS/Hz
– SFDR: 80 dBc
– Non-HD2,HD3: –91 dBFS
– Performance at fin = 450 MHz at –1 dBFS
– SNR: 67 dBFS
– NSD: –151 dBFS/Hz
– SFDR: 77 dBc HD2,3
– Non-HD2,HD3: –89 dBFS
High IF Sampling Receivers
Broadband Wireless
Microwave Receivers
Cable CMTS, DOCSIS 3.1 Receivers
Communications Test Equipment
Digitizers
Software Defined Radio (SDR)
Radar and Antenna Arrays
3 Description
The ADC31JB68 is a low-power, wide-bandwidth, 16bit, 500-MSPS analog-to-digital converter (ADC). The
buffered analog input provides uniform input
impedance across a wide frequency range while
minimizing sample-and-hold glitch energy. This
device is designed to sample input signals of up to
1.3 GHz.
The ADC31JB68 provides excellent spurious-free
dynamic range (SFDR) over a large input frequency
range with very-low power consumption. On-chip
dither provides an very-clean noise floor. Embedded
foreground and background calibration provides
consistent performance over the temperature range,
and minimizes part-to-part variation.
This device supports the JESD204B serial interface
with data rates up to 5 Gbps on each of two lanes,
enabling high system integration density.
The ADC31JB68 comes in a 6-mm × 6-mm, 40-pin
QFN package.
Device Information(1)
PART NUMBER
ADC31JB68
PACKAGE
WQFN (40)
BODY SIZE (NOM)
6.00 mm × 6.00 mm
(1) For all available packages, see the package option addendum
at the end of the datasheet.
Transmitted Eye at Output of 18-Inch,
5-mil. FR4 Microstrip Trace at 5 Gb/s
With Optimized De-Emphasis
Spectrum With –1-dBFS, 450-MHz Input
0
-10
-20
Magnitude [dBFS]
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
0
25
50
75
100 125 150 175
Frequency [MHz]
200
225
250
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
ADC31JB68
SLASE60B – SEPTEMBER 2015 – REVISED JANUARY 2019
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
6.9
1
1
1
2
3
6
Absolute Maximum Ratings ...................................... 6
ESD Ratings ............................................................ 6
Recommended Operating Conditions....................... 6
Thermal Information ................................................. 6
Electrical Characteristics: Converter Performance ... 7
Electrical Characteristics: Power Supply .................. 9
Electrical Characteristics: Interface......................... 10
Timing Requirements .............................................. 12
Typical Characteristics ............................................ 16
7
Parameter Measurement Information ................ 20
8
Detailed Description ............................................ 21
7.1 Interface Circuits ..................................................... 20
8.1 Overview ................................................................. 21
8.2 Functional Block Diagram ....................................... 21
8.3 Feature Description................................................. 22
8.4 Device Functional Modes........................................ 30
8.5 Register Map........................................................... 31
9
Application and Implementation ........................ 45
9.1 Application Information............................................ 45
9.2 Typical Application ................................................. 59
10 Power Supply Recommendations ..................... 62
10.1 Power Supply Design............................................ 62
10.2 Decoupling ............................................................ 62
11 Layout................................................................... 63
11.1 Layout Guidelines ................................................. 63
11.2 Layout Example .................................................... 64
12 Device and Documentation Support ................. 65
12.1
12.2
12.3
12.4
12.5
12.6
Related Documentation .......................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
65
65
65
65
65
65
13 Mechanical, Packaging, and Orderable
Information ........................................................... 65
4 Revision History
Changes from Revision A (September 2015) to Revision B
Page
•
Added new note 2 to clarify power sequence in Absolute Maximum Ratings ....................................................................... 6
•
Changed text in last paragraph of Power Supply Design section to clarify power up sequence ......................................... 62
Changes from Original (September 2015) to Revision A
Page
•
Changed From: 1-page Product Preview To: Production datasheet ..................................................................................... 1
•
Changed Features From: Non-HD2,HD3: TBD dBFS To: Non-HD2,HD3: –91 dBFS ........................................................... 1
•
Changed Features From: Non-HD2,HD3: TBD dBFS To: Non-HD2,HD3: –89 dBFS ........................................................... 1
2
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5 Pin Configuration and Functions
The ADC31JB68 is packaged in a 40-pin QFN package (6 x 6 x 0.8, 0.5 mm pin-pitch) with a bottom-side
exposed paddle.
SDO/OVR
34
CSB
VA1.2
35
31
VA1.2
36
SCLK
AGND
37
32
AGND
38
SDI
VA1.8
39
33
VA1.8
40
RTA Package
40 PIN (WQFN)
Top View
VA3.0
1
30
AGND
VA3.0
2
29
VA1.2
AGND
3
28
VA1.8
VIN+
4
27
AGND
VIN-
5
26
SO0-
AGND
6
25
SO0+
VCM
7
24
SO1-
VACLK1.2
8
23
SO1+
VACLK1.2
9
22
AGND
10
21
VA1.2
11
12
13
14
15
16
17
18
19
20
CLKIN+
CLKIN-
AGND
AGND
VA1.8
VA1.8
SYSREF+
SYSREF-
SYNCb+
SYNCb-
AGND
EXPOSED PADDLE ON BOTTOM
OF PACKAGE
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SLASE60B – SEPTEMBER 2015 – REVISED JANUARY 2019
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Pin Functions
PIN
NAME
TYPE or DIAGRAM
NO.
DESCRIPTION
INPUT/REFERENCE
Differential analog input pins.
The differential full-scale signal level is 1.7 Vpp. Each input pin is terminated to the
internal 1.6V common-mode reference with a 100 Ω resistor for a 200 Ω total
differential termination.
VA3.0
VIN+
VIN–
4, 5
VIN+
100:
VA3.0
Input Interface Common mode voltage.
This pin must be bypassed to AGND with a low ESL (equivalent series inductance)
0.1 µF capacitor that is placed as close to the pin as possible to minimize stray
inductance. A 10 µF capacitor should also be placed in parallel. It is recommended to
use VCM to provide the common mode voltage for the differential analog inputs. The
input common-mode bias is provided internally for the ADC input, therefore external
use of VCM is recommended but not strictly required. The recommended decoupling
is always required.
100:
VIN-
VCM
7
VA3.0
VCM
+
-
CLOCK/SYNC
VA1.2
VA3.0
Differential device clock input pins.
AC coupling is recommended for coupling the clock input to these pins. DC biasing of
the clock receiver is provided internally. Each pin is internally terminated to the
500mV DC bias with 50 Ω resistor for a 100 Ω total internal differential termination
resistor. Sampling occurs on the falling edge of the differential signal (CLKIN+) −
(CLKIN-).
CLKIN+
CLKIN+
CLKIN–
50:
11, 12
10k:
+
-
0.5V
50:
AGND
CLKIN-
AGND
VA1.2
VA3.0
3k:
Differential SYSREF signal input pins.
Each pin is internally terminated to the DC bias with a large resistor. An internal 100
Ω differential termination is provided therefore an external termination is not required.
Additional resistive components in the input structure give the SYSREF input a wide
input common mode range.
SYSREF+
1k:
50:
SYSREF+,
SYSREF–
17, 18
50:
SYSREF12k:
+
-
0.5V
1.8V
1.8V
VA3.0
Differential SYNCb signal input pins.
1.5k:
2k:
SYNC+
SYNCb+
SYNCb–
50:
34k:
3pF
19, 20
50:
1.5k:
SYNC2k:
34k:
DC coupling is required for coupling the SYNCb signal to these pins. Each pin is
internally terminated to the DC bias with a large resistor. An internal 100 Ω differential
termination is provided therefore an external termination is not required. Additional
resistive components in the input structure give the SYNCb input a wide input
common mode range. The SYNCb signal is active low and is therefore asserted when
the voltage at SYNCb+ is less than at SYNCb–.
SERIAL INTERFACE (SPI)
SCLK
32
VA3.0
VA1.2
SPI Interface Serial Clock pin.
Serial data is shifted into and out of the device synchronous with this clock signal.
Compatible with 1.2–3.0V CMOS logic levels.
CSB
31
SPI Interface Chip Select pin.
When this signal is asserted, SCLK is used to clock input serial data on the SDI pin or
output serial data on the SDO pin. When this signal is de-asserted, the SDO pin is
high impedance and the input data is ignored. Active low. A 1kΩ pull-up resistor to the
VA1.8 supply is recommended to prevent undesired activation of the SPI bus.
Compatible with 1.2–3.0V CMOS logic levels.
SDI
33
SPI Interface Data Input pin.
Serial data is shifted into the device on this pin while the CSB signal is asserted.
Compatible with 1.2-3.0V CMOS logic levels.
4
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Pin Functions (continued)
PIN
NAME
NO.
TYPE or DIAGRAM
+
-
VA3.0
80:
SDO/OVR
DESCRIPTION
34
SDO/OVR
80:
SPI Data Output and Over-Range pin.
Dual mode pin. When configured as SDO, serial data of the SPI is shifted out of the
device on this pin while CSB is asserted. When configured as OVR, the over-range
signal is output. Pin mode configurable via the SPI. Output voltage is configurable to
1.2V, 1.8V, or 3.0V CMOS logic levels via the SPI. Default configuration outputs the
SDO at a 1.8V logic level.
DIGITAL OUTPUT INTERFACE
VA3.0
SO0+, SO0–,
SO1+, SO1–
25, 26,
23, 24
SOx+
SOx-
Differential High Speed Serial Data Lane pins.
These pins must be AC coupled to the receiving device. The differential trace routing
from these pins must maintain a 100 Ω characteristic impedance.
POWER SUPPLY
3 V Analog Power Supply pin.
VA3.0
1, 2
VA1.8
15, 16,
28, 39,
40
VA1.2
21, 29,
35, 36
Supply Input Pin
This pin must be connected to a quiet source and decoupled to AGND with a 0.1 µF
capacitor located close to each pin and a second 0.1 µF capacitor on bottom layer.
1.8 V Analog Power Supply pins.
Supply Input Pin
These pins must be connected to a quiet source and decoupled to AGND with a
0.1 µF capacitor located close to each pin and a second 0.1 µF capacitor on bottom
layer.
1.2 V Analog Power Supply pins.
Supply Input Pin
These pins must be connected to a quiet source and decoupled to AGND with a 0.1
µF capacitor located close to each pin and a second 0.1 µF capacitor on bottom
layer.
1.2 V Analog Power Supply pins for internal clock path.
VACLK1.2
AGND
8, 9
3, 6, 10,
13, 14,
22, 27,
30, 37,
38
Supply Input Pin
These pins must be connected to a quiet source and decoupled to AGND with a 0.1
µF capacitor located close to each pin and a second 0.1 µF capacitor on bottom
layer.
Analog Ground.
Analog Ground
Exposed
Thermal Pad
Solid ground reference planes under the device are recommended.
Exposed Thermal Pad.
The exposed pad must be connected to the AGND ground plane electrically and with
good thermal dissipation properties to ensure rated performance.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Supply voltage range
(2)
(1)
MIN
MAX
VA3.0
–0.3
4.2
V
VA1.8
–0.3
2.35
V
VA1.2, VACLK1.2
–0.3
1.55
V
VCM – 0.75
VCM + 0.75
V
VCM
–0.3
VA3.0 + 0.3,
not to exceed 4.2 V
V
CLKIN+, CLKIN–, SYSREF+, SYSREF–
–0.3
1.55
V
SYNCb+, SYNCb–
–0.3
VA1.8 + 0.3
V
–0.3
VA3.0 + 0.3,
not to exceed 4.2 V
V
VIN+, VIN–
Voltage applied to input pins
SCLK, SDI, CSb
Operating junction temperature range, TJ (3)
Temperature
(1)
(2)
(3)
125
Storage temperature, Tstg
–65
150
UNIT
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Power sequence must be followed as stated in the Power Supply Design section. Failure to follow the required power sequence may
result in unwanted voltage on the VA1.8 pin that can exceed the absolute maximum voltage of 2.35 VDC.
Prolonged use at this temperature may increase the device failure-in-time (FIT) rate.
6.2 ESD Ratings
VALUE
Electrostatic
discharge
V(ESD)
(1)
(2)
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±1000
Charged device model (CDM), per JEDEC specification JESD22-C101 (2)
±250
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
operating conditions specified over operating free-air temperature range (unless otherwise noted)
Supply voltage range
CLKIN± duty cycle
MIN
MAX
UNIT
VA3.0
2.85
3.15
V
VA1.8
1.7
1.9
V
V
VA1.2, VACLK1.2
1.15
1.25
CLKDIV = 1
30%
70%
–40
85
°C
105
°C
Operating free-air temperature range, TA
Operating junction temperature range, TJ
6.4 Thermal Information
ADC31JB68
THERMAL METRIC (1)
RTA (WQFN)
UNIT
(40) PINS
RθJA
Junction-to-ambient thermal resistance
28.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
13.1
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
1.0
°C/W
RθJB
Junction-to-board thermal resistance
4.5
°C/W
ψJT
Junction-to-top characterization parameter
0.2
°C/W
ψJB
Junction-to-board characterization parameter
4.4
°C/W
(1)
6
For more information about thermal metrics, see the Semiconductor and IC Package Thermal Metrics application report.
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6.5 Electrical Characteristics: Converter Performance
typical values at TA = 25 °C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input; R(term) = 100 Ω (unless
otherwise noted)
PARAMETER
NOTES
MIN
TYP
MAX
UNIT
STATIC CHARACTERISTICS
Resolution
No missing codes
16
GVAR
Gain variation
Part-to-part variation of input voltage to
output code gain between different parts
1
TG
Gain temperature drift
Drift of input voltage to output code gain
across temperature
100
ppm(FSR)/°C
VOFF
Input voltage offset
0.2
%FSR
TVOFF
Input voltage offset temperature
drift
DNL
Differential nonlinearity
Sinusoidal histogram, 10-MHz input
±0.23
LSB
INL
Integral nonlinearity
Sinusoidal histogram, 10-MHz input
±7.1
LSB
See measurement configuration in Figure 64
1300
MHz
fIN = 10 MHz, AIN = –40 dBFS
70.6
fIN = 10 MHz
70.3
fIN = 100 MHz
70.1
9
Bits
%FSR
ppm(FSR)/°C
DYNAMIC AC CHARACTERISTICS
BW3dB
SNR
Analog input bandwidth
Signal to noise ratio
fIN = 210 MHz
68.3
69.3
fIN = 210 MH, TA = 25°C
68.4
69.3
fIN = 350 MHz
68.1
fIN = 450 MHz
NSD
SINAD
Noise spectral density
Signal to noise and distortion
ratio
67.0
fIN = 10 MHz, AIN = –40 dBFS
–154.6
fIN = 10 MHz
–154.3
fIN = 100 MHz
–154.1
fIN = 210 MHz
–152.3
–153.3
fIN = 210 MHz, TA = 25°C
–152.4
–153.3
fIN = 350 MHz
–152.1
fIN = 450 MHz
–151.0
fIN = 10 MHz
69.9
fIN = 100 MHz
69.5
fIN = 210 MHz
67.4
68.9
fIN = 210 MHz, TA = 25ºC
67.8
68.9
fIN = 350 MHz
67.6
fIN = 450 MHz
66.5
fIN = 10 MHz
11.3
fIN = 100 MHz
ENOB
Effective number of bits
10.9
11.2
fIN = 210 MH, TA = 25°C
11.0
11.2
fIN = 350 MHz
10.9
fIN = 450 MHz
10.8
dBFS
dBFS
Bits
83
fIN = 100 MHz
Spurious-free dynamic range
dBFS/Hz
11.3
fIN = 210 MHz
fIN = 10 MHz
SFDR
dBFS
81
fIN = 210 MHz
74
80
fIN = 210 MH, TA = 25°C
75
80
fIN = 350 MHz
79
fIN = 450 MHz
77
dBc
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Electrical Characteristics: Converter Performance (continued)
typical values at TA = 25 °C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input; R(term) = 100 Ω (unless
otherwise noted)
PARAMETER
HD2
HD3
Second-order harmonic
distortion
Third-order harmonic distortion
NOTES
MIN
–83
fIN = 100 MHz
–81
fIN = 210 MHz
–74
–84
fIN = 210 MH, TA = 25°C
–75
–84
fIN = 350 MHz
–84
fIN = 450 MHz
–77
fIN = 10 MHz
–92
fIN = 100 MHz
–83
fIN = 210 MHz
–74
–80
fIN = 210 MHz TA = 25°C
–75
–80
fIN = 350 MHz
–79
fIN = 450 MHz
–87
fIN = 10 MHz
–93
fIN = 100 MHz
NonHD2,
HD3
Spurious free dynamic range
excluding HD2, HD3
IMD3
8
Total harmonic distortion
Two–tone intermodulation
distortion
–83
–91
fIN = 210 MH, TA = 25°C
–83
–91
fIN = 350 MHz
–88
fIN = 450 MHz
–89
fIN = 10 MHz
–81
UNIT
dBc
dBc
dBFS
–78
fIN = 210 MHz
–72
–78
fIN = 210 MH, TA = 25°C
–73
–78
fIN = 350 MHz
–73
fIN = 450 MHz
–72
Worst-case IMD3 spur adjacent to the input
tones, f1 = 200 MHz, f2 = 210 MHz,
AIN = –7-dBFS/tone
–89
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MAX
–89
fIN = 210 MHz
fIN = 100 MHz
THD
TYP
fIN = 10 MHz
dBc
dBc
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6.6 Electrical Characteristics: Power Supply
typical values at TA = 25°C, full temperature range is tMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input, and R(term) = 100 Ω
(unless otherwise noted)
PARAMETER
NOTES
MIN
TYP
MAX
UNIT
2.85
3
3.15
V
1.8-V analog supply (VA1.8)
1.7
1.8
1.9
V
1.2-V analog supply (VA1.2)
1.15
1.2
1.25
V
1.2-V analog supply for clock
signal path (VACLK1.2)
1.15
1.2
1.25
V
3-V analog supply (VA3.0)
I(A3.0)
3-V analog supply current
I(A1.8)
1.8-V analog supply current
61
mA
272
mA
197
mA
915
mW
I(A1.2)
1.2-V analog supply current
Sum of current from VA1.2 and VACLK1.2
supplies
PD
Power dissipation
Total power dissipation;
Fin = 10 MHz
PPD
Power down power dissipation
Power in power down state, no external
clock
17
mW
PSL
Sleep power dissipation
Power in sleep state, no external clock
17
mW
Sensitivity to supply noise
Power of spectral spur resulting from a 100mV sinusoidal signal modulating a supply at
500 kHz. Analog input is a –1-dBFS, 210MHz single tone. In all cases, the spur
appears as part of a pair symmetric about
the fundamental that scales proportionally
with the fundamental amplitude.
dBFS
VA3.0
–81
VA1.8
–55
VA1.2 and VACLK1.2
–35
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6.7 Electrical Characteristics: Interface
typical values at TA = 25°C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input, and R(term) = 100 Ω
(unless otherwise noted) (see the Interface Circuits section)
PARAMETER
NOTES
MIN
TYP
MAX
UNIT
ANALOG INPUTS (VIN+, VIN-)
V(FSR)
Full-scale range voltage
Differential peak-to-peak
1.7
V
VCM
Nominal input common-mode
voltage
1.6
V
ΔVCM
Maximum input common mode
voltage range
VCM ±
0.05
V
RIN
Input termination resistance
Differential resistance at dc
190
Ω
CIN
Input capacitance
Differential
4.6
pF
1.6
V
INPUT COMMON MODE REFERENCE (VCM)
V(VCM)
Common-mode reference voltage
output
I(VCM)
Maximum VCM pin current load
1
mA
CLOCK INPUT (CLKIN+, CLKIN-)
VID-MAX
Maximum input voltage swing (1)
Differential peak voltage
VID-MIN
Minimum input voltage swing (1)
Differential peak voltage
dVSS/dt
Input edge rate at zero
crossing (1)
Recommended minimum
V(IS-
Input common-mode internal bias
voltage (2) (1)
BIAS)
V(IS-IN)
Externally applied common-mode
DC-coupled interface
voltage (1)
(2)
Z(rdiff)
Input termination resistance
Ztt
Common-mode internal bias
source impedance (2)
CT
Input capacitance (2)
Differential resistance at dc
Differential
1000
mV
250
mV
5
V/ns
0.5
V
0.5 ± 0.1
V
100
Ω
10
kΩ
2
pF
SYSREF INPUT (SYSREF+, SYSREF-)
VID-MAX
Maximum input voltage swing (1)
(1)
VID-MIN
Minimum input voltage swing
V(IS-
Input common-mode internal bias
voltage (1)
BIAS)
Differential peak voltage
Differential peak voltage
1000
mV
250
mV
0.5
V
DC-coupled interface,
typical value depends on the configuration of
the SYS_CM parameter
V(IS-IN)
SYS_CM = 00
Externally applied common mode
SYS_CM = 01
voltage (1)
Z(rdiff)
Input termination resistance (2)
Ztt
Common-mode internal bias
source impedance (2)
CT
Input capacitance
(2)
0.5 ± 0.1
0.8 ± 0.2
SYS_CM = 10
1.25 ±
0.25
SYS_CM = 11
1.75 ±
0.25
Differential resistance at dc
Differential
V
100
Ω
14
kΩ
1
pF
350
mV
SYNCb INPUT (SYNCb+, SYNCb-)
VID
Input voltage swing (1)
V(IS-IN)
Externally applied common-mode
DC-coupled interface
voltage (1)
(1)
(2)
10
Differential peak voltage
1.25 ±
0.75
V
Specification applies to the electrical level diagram of Figure 26
Specification applies to the electrical circuit diagram of Figure 27
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Electrical Characteristics: Interface (continued)
typical values at TA = 25°C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input, and R(term) = 100 Ω
(unless otherwise noted) (see the Interface Circuits section)
PARAMETER
NOTES
Z(rdiff)
Input termination resistance (2)
Differential resistance at dc
CT
Input capacitance (2)
Differential
MIN
TYP
MAX
UNIT
100
Ω
1
pF
SERDES OUTPUT (SO0+/-, SO1+/-) Meets JESD204B LV-OIF-11G-SR Standard
Differential peak-peak voltage,
de-emphasis disabled (DEM = 0)
Output differential voltage (3)
VOD
VOD = 0
400
VOD = 1
470
VOD = 2
540
VOD = 3
610
VOD = 4
670
VOD = 5
740
VOD = 6
790
VOD = 7
840
mV
Configurable via SPI, VOD configured to 4
R(deepm) Transmitter de-emphasis range
ISC
Transmitter short circuit current
Z(ddiff)
Differential output impedance
RL(ddiff)
Differential output return loss
magnitude
DEM=0
0.0
DEM=1
–0.8
DEM=2
–2.4
DEM=3
–3.8
DEM=4
–4.9
DEM=5
–6.3
DEM=6
–7.7
DEM=7
–10.3
Transmitter terminals shorted to each other
or ground, power on
(4)
Relative to 100 Ω,
for frequencies from 100 MHz to 0.75 ×
baud rate; default VOD and DEM.
dB
23
mA
100
Ω
–8.5
dB
SCLK, SDI, CSB INPUT
Inputs are compatible with 1.2-V up to 3-V
logic.
VIH
Logical 1 input voltage
0.9
V
VIL
Logical 0 input voltage
IIN0
Logic low input current
4
nA
IIN1
Logic high input current
–8
nA
CIN
Input capacitance
2
pF
VSPI (5)
V
0.3
V
SDO/OVR OUTPUT
VSPI = 1.2 V, 1.8 V, or 3 V, configurable via
SPI
VOH
Logical 1 output voltage (5)
VOL
Logical 0 output voltage
(5)
+ISC
Logic high short circuit current
VSPI = 1.8 V
9
mA
–ISC
Logic low short circuit current
VSPI = 1.8 V
–14
mA
(3)
(4)
(5)
VSPI – 0.2
0
0.3
V
Specification applies to the electrical level diagram of Figure 28
Specification applies to the electrical circuit diagram of Figure 29
The SPI_CFG register must be changed to a supported output logic level after power up and before a SPI read command is executed.
Until that time, the output voltage on SDO/OVR may be as high as the VA3.0 supply during a SPI read command. The SDO/OVR output
is high-Z at all times except during a read command.
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6.8 Timing Requirements
typical values at TA = 25°C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input, and R(term) = 100 Ω
(unless otherwise noted) (see Figure 1 and Figure 2 for timing diagrams)
PARAMETER
NOTES
MIN
NOM
MAX
UNIT
500
MSPS
ADC SAMPLING INSTANT TIMING CHARACTERISTICS
FS
Sampling rate
FCLKIN
Input clock frequency at CLKIN
inputs
Equal to FCLKIN / CLKDIV
100
CLKDIV = 1
100
500
CLKDIV = 2
200
1000
CLKDIV = 4
400
2000
CLKDIV = 1
DC
50% ±
20%
Input clock (CLKIN) duty cycle
CLKDIV = 2 or CLKDIV = 4
tLAT-ADC ADC core latency
tJ
Additive sampling aperture jitter
MHz
50% ± 5%
Delay from a reference sampling instant to
the boundary of the internal LMFC where the
reference sample is the first sample of the
next transmitted multi-frame. In this device,
the frame clock period is equal to the
sampling clock period.
Frame clock
cycles
7
Depends on input CLKIN differential edge
rate at the zero crossing, dVSS/dt, tested
with 5 V/ns edge rate.
fs
CLKDIV = 1
80
CLKDIV = 2, 4
90
OVER-RANGE INTERFACE TIMING CHARACTERISTICS (SDO/OVR (1))
tODH
tODL
OVR assertion delay
OVR de-assertion delay
Functional delay between an overrange
value sampled and OVR asserted
Frame clock
cycles
8
Functional delay between first underrange
value sampled until OVR de-assertion,
configurable via SPI
Configured for minimum delay
tODH
Configured for maximum delay
tODH + 15
Frame clock
cycles
SYSREF TIMING CHARACTERISTICS
SYSREF assertion duration
Required duration of SYSREF assertion
after rising edge event
2
Frame clock
cycles
SYSREF de-assertion duration
Required duration of SYSREF de-assertion
after falling edge event
2
Frame clock
cycles
tS-SYS
SYSREF setup time
Relative to CLKIN rising edge
350
ps
tH-SYS
SYSREF hold time
Relative to CLKIN rising edge
0
ps
tPH-SYS
tPL-SYS
JESD204B INTERFACE LINK TIMING CHARACTERISTICS
Functional delay between SYSREF
assertion latched and LMFC frame
boundary. Depends on CLKDIV setting
4
CLKIN
cycles
4
Frame clock
cycles
10
CLKIN
cycles
5
Frame clock
cycles
18
CLKIN
cycles
4.5
Frame clock
cycles
CLKDIV = 1
tD-LMFC
SYSREF to LMFC delay
CLKDIV = 2
CLKDIV = 4
(1)
12
The SDO/OVR pin is configured in over-range output mode.
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Timing Requirements (continued)
typical values at TA = 25°C, full temperature range is TMIN = –40°C to TMAX = 85°C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3 V; VA1.8 = 1.8 V; VA1.2 = VACLK1.2 = 1.2 V; –1 dBFS differential input, and R(term) = 100 Ω
(unless otherwise noted) (see Figure 1 and Figure 2 for timing diagrams)
PARAMETER
MIN
NOM
MAX
LMFC to K28.5 delay
Functional delay between the start of the
first K28.5 frame during code group
synchronization at the serial output and the
preceding LMFC frame boundary
5.6
6.6
7.6
tD-ILA
LMFC to ILA delay
Functional delay between the start of the
first ILA frame during initial lane
synchronization at the serial output and the
preceding LMFC frame boundary
5.6
6.6
7.6
tD-DATA
LMFC to valid data delay
Functional delay between the start of the
first valid data frame at the serial output and
the preceding LMFC frame boundary
5.6
6.6
7.6
tS-
SYNCb setup time
Required SYNCb setup time-relative to the
internal LMFC boundary (2)
3
SYNCb hold time
Required SYNCb hold time relative to the
internal LMFC boundary (2)
0
tH-SYNCb SYNCb assertion hold time
Required SYNCb hold time after assertion
before de-assertion to initiate a link
resynchronization
4
tILA
Duration of the ILA sequence
4
tD-K28
SYNCb-F
tHSYNCb-F
ILA duration
NOTES
UNIT
Frame clock
cycles
Frame clock
cycles
Multi-frame
clock cycles
SERIAL OUTPUT DATA TIMING CHARACTERISTICS
FSR
Serial bit rate
UI
Unit interval
5.0 Gb/s data rate
1
5.0
Gb/s
tR, tF
Rise/fall times
5.0 Gb/s data rate, default values for VOD
and DEM
p-p UI
Deterministic jitter
Includes periodic jitter (PJ), data dependent
jitter (DDJ), duty cycle distortion (DCD), and
inter-symbol interference (ISI); 5.0 Gb/s data
rate
0.049
DJ
9.82
p-p ps
RJ
Random jitter
Assumes BER of 1e-15 (Q = 15.88); 5.0
Gb/s data rate
0.119
p-p UI
1.50
rms ps
TJ
Total jitter
Sum of DJ and RJ, assumes BER of 1e-15
(Q = 15.88); 5.0 Gb/s data rate
0.169
p-p UI
33.6
p-p ps
200
ps
43
ps
SPI BUS TIMING CHARACTERISTICS (3)
ƒSCLK
Serial clock frequency
tPH
SCLK pulse width – high
6
ns
tPL
SCLK pulse width – low
7
ns
tSSU
SDI input data setup time
3
ns
tSH
SDI input data hold time
1
ns
tODZ
SDO output data driven-to-3state time
10
ns
tOZD
SDO output data 3-state-todriven time
25
ns
tOD
SDO output data delay time
25
ns
tCSS
CSB setup time
tCSH
CSB hold time
tIAG
(2)
(3)
Inter-access gap
fSCLK = 1 / tP
Minimum time CSB must be de-asserted
between accesses
20
MHz
3
ns
1
ns
1
ns
The SYNCb setup and hold times determine the multi-frame after which the ILA is initiated but meeting the setup and hold times are not
required to achieve deterministic latency.
All timing specifications for the SPI given for VSPI = 1.8-V logic levels and a 5-pF capacitive load on the SDO pin. Timing specification
require larger margins for VSPI= 1.2 V. The serial bit rate of the SPI should be limited to 10 Mb/s or lower for VSPI = 1.2-V logic.
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|
Sample N
1
FS
VIN+
tAD
Clock N
|
|
CLKIN+
(CLKDIV=1)
|
Device Latency =
SYSREF+
tH-SYS
tLAT-ADC + tD-DATA
tCLK-DATA
tS-SYS
Digitized Sample N
+
SO0
s
+
SO1
s
1st Octet
of Sample N
2nd Octet
of Sample N
tL-L
Figure 1. Sample Timing Diagram
CLKIN+
< 1/FS
(CLKDIV = 4)
(CLKDIV = 1)
1/FS
Internal
Frame Clock
OVR Output
tODL
Sampling Instant*
(at front-end switch)
*Assumes sampling
phase adjustment is
disabled
tODH
1st Over-range
sample
Over-range
Samples
1st Under-range
sample
Under-range
Samples
Figure 2. Overrange (OVR) Timing Diagram
14
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st
th
1 clock
16
th
clock
24
clock
SCLK
tCSH
tPL
tCSS
tCSS
tPH
tCSH
tP = 1/fSCLK
tIAG
CSB
tSS
tSH
tSS
SDI
D7
tSH
D1
D0
Write Command
COMMAND FIELD
tOD
SDO
Hi-Z
D7
tOZD
D1
Hi-Z
D0
Read Command
tODZ
Figure 3. SPI Timing Diagram
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6.9 Typical Characteristics
Typical values at TA = 25 °C, full temperature range is TMIN = –40 °C to TMAX = 85 °C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3.0 V, VA1.8 = 1.8 V, VA1.2 = VACLK1.2 = 1.2 V, –1 dBFS differential input (unless otherwise noted).
1
8
0.8
6
0.6
4
0.2
INL (LSB)
DNL (LSB)
0.4
0
–0.2
2
0
–2
–0.4
–4
–0.6
–6
–0.8
–1
–8
0
16384
32768
Code
49152
65536
0
16384
Figure 4. DNL vs Output Code
32768
Code
49152
Figure 5. INL vs Output Code
100
10
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
90
5
Magnitude [dBFS]
Max INL Deviation [LSB]
65536
0
85
80
75
70
-5
65
60
-10
1.15
1.175
1.2
VA1.2 Supply [V]
1.225
0
1.25
Figure 6. INL vs Supply
50
100
150 200 250 300 350
Input Frequency [MHz]
400
450
500
Figure 7. SNR, SINAD, SFDR vs Input Frequency
100
110
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
105
100
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
Magnitude [dBFS]
Magnitude [dBFS]
90
95
90
85
80
75
85
80
75
70
70
65
65
60
-50
-45
-40
-35 -30 -25 -20 -15
Input Amplitude [dBFS]
-10
-5
0
60
100
150
200
250
300
350
400
Sampling Rate [MSPS]
450
500
Input Frequency = 210 MHz
Figure 8. SNR, SINAD, SFDR vs Input Amplitude
16
Figure 9. SNR, SINAD, SFDR vs Sampling Rate (FS)
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Typical Characteristics (continued)
Typical values at TA = 25 °C, full temperature range is TMIN = –40 °C to TMAX = 85 °C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3.0 V, VA1.8 = 1.8 V, VA1.2 = VACLK1.2 = 1.2 V, –1 dBFS differential input (unless otherwise noted).
100
100
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
90
Magnitude [dBFS]
Magnitude [dBFS]
90
85
80
75
85
80
75
70
70
65
65
60
2.8
2.85
2.9
2.95
3
3.05
3.1
VA3.0 Supply Voltage [V]
3.15
60
1.7
3.2
Input Frequency = 210 MHz
1.8
1.85
VA1.8 Supply Voltage [V]
1.9
Figure 11. SNR, SINAD, SFDR vs VA1.8 Supply
100
100
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
90
Magnitude [dBFS]
90
Magnitude [dBFS]
1.75
Input Frequency = 210 MHz
Figure 10. SNR, SINAD, SFDR vs VA3.0 Supply
85
80
75
85
80
75
70
70
65
65
60
1.15
1.175
1.2
1.225
VA1.2 Supply Voltage [V]
60
-40 -30 -20 -10
1.25
Input Frequency = 210 MHz
0
10 20 30 40 50 60 70 80 90
Temperature [qC]
Input Frequency = 210 MHz
Figure 12. SNR, SINAD, SFDR vs VA1.2 Supply
Figure 13. SNR, SINAD, SFDR vs Temperature
-60
-60
HD2 [dBFS]
HD3 [dBFS]
Non-HD2,HD3 [dBFS]
THD [dBFS]
-65
-75
-80
-85
HD2 [dBFS]
HD3 [dBFS]
Non-HD2,HD3 [dBFS]
THD [dBFS]
-65
-70
Magnitude [dBFS]
-70
Magnitude [dBFS]
SNR [dBFS]
SINAD [dBFS]
SFDR [dBFS]
95
-75
-80
-85
-90
-95
-90
-100
-95
-105
-100
0
50
100
150 200 250 300 350
Input Frequency [MHz]
400
450
500
-110
-50
-45
-40
-35 -30 -25 -20 -15
Input Amplitude [dBFS]
-10
-5
0
Input Frequency = 210 MHz
Figure 14. HD2, HD3, SPUR, THD vs Input Frequency
Figure 15. HD2, HD3, SPUR, THD vs Input Amplitude
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Typical Characteristics (continued)
Typical values at TA = 25 °C, full temperature range is TMIN = –40 °C to TMAX = 85 °C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3.0 V, VA1.8 = 1.8 V, VA1.2 = VACLK1.2 = 1.2 V, –1 dBFS differential input (unless otherwise noted).
-60
-60
HD2 [dBFS]
HD3 [dBFS]
Non-HD2,HD3 [dBFS]
THD [dBFS]
-65
-70
Magnitude [dBFS]
Magnitude [dBFS]
-70
-75
-80
-85
-75
-80
-85
-90
-90
-95
-95
-100
100
150
200
250
300
350
400
Sampling Rate [MSPS]
450
-100
-40 -30 -20 -10
500
Input Frequency = 210 MHz
0
-10
-20
-20
-30
-30
Magnitude [dBFS]
Magnitude [dBFS]
0
-40
-50
-60
-70
-80
-50
-60
-70
-80
-90
-100
-100
-110
-110
-120
-120
75
100 125 150 175
Frequency [MHz]
200
225
0
250
Figure 18. Output Spectrum, 1-Tone Test at 70 MHz
0
-10
-10
-20
-20
-30
-30
-40
-50
-60
-70
-80
-110
-120
-120
200
225
250
225
250
-80
-110
100 125 150 175
Frequency [MHz]
200
-70
-90
75
100 125 150 175
Frequency [MHz]
-60
-100
50
75
-50
-100
25
50
-40
-90
0
25
Figure 19. Output Spectrum, 1-Tone Test at 210 MHz
0
Magnitude [dBFS]
Magnitude [dBFS]
-40
-90
50
10 20 30 40 50 60 70 80 90
Temperature [qC]
Figure 17. HD2, HD3, SPUR, THD vs Temperature
-10
25
0
Input Frequency = 210 MHz
Figure 16. HD2, HD3, SPUR, THD vs Sampling Rate
0
HD2 [dBFS]
HD3 [dBFS]
Non-HD2,HD3 [dBFS]
THD [dBFS]
-65
0
25
50
75
100 125 150 175
Frequency [MHz]
200
225
250
Input Amplitude = –7 dBFS/Tone
Figure 20. Output Spectrum, 1-Tone Test at 450 MHz
18
Figure 21. Output Spectrum, 2-Tone Test at 200 MHz and
210 MHz
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Typical Characteristics (continued)
Typical values at TA = 25 °C, full temperature range is TMIN = –40 °C to TMAX = 85 °C, ADC sampling rate = 500 MSPS, 50%
clock duty cycle, VA3.0 = 3.0 V, VA1.8 = 1.8 V, VA1.2 = VACLK1.2 = 1.2 V, –1 dBFS differential input (unless otherwise noted).
0.8
0.3
0.7
0.2
0.6
0.1
0
10 20 30 40 50 60 70 80 90
Temperature [qC]
Figure 22. Supply Power and Current vs Temperature
5-mil. FR4 Microstrip Trace at 5 Gb/s with No De-Emphasis
Figure 24. Transmitted Eye at Output of 3-Inch
Power [W]
0.9
0.4
0
0.64
Total Power [W]
IA1.2 [A]
IA1.8 [A]
IA3.0 [A]
1
0.5
0.9
0.5
-40 -30 -20 -10
1.1
Current [A]
Power [W]
1
0.6
Total Power [W]
IA1.2 [A]
IA1.8 [A]
IA3.0 [A]
0.56
0.48
0.8
0.4
0.7
0.32
0.6
0.24
0.5
0.16
0.4
0.08
0.3
100
150
200
250
300
350
400
Sampling Rate [MSPS]
450
Current [A]
1.1
0
500
Figure 23. Supply Power and Current vs Sampling Rate
5-mil. FR4 Microstrip Trace at 5 Gb/s with Optimized De-Emphasis
Figure 25. Transmitted Eye at Output of 18-Inch
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7 Parameter Measurement Information
7.1 Interface Circuits
VID
VSS
VI+
VI-
VIVIS
GND
dVSS/dt
VI+
VI+ and VI- referenced to GND
VID = |VI+ ± VI-|
[mVp]
VIS = |VI+ + VI-| / 2
[V]
VI+ referenced to VIVSS = 2*|VI+ ± VI-| [mVpp]
Figure 26. Electrical Level Diagram for Differential Input Signals
Zrdiff / 2
VI+
Ztt
CT
VI-
+
-
Zrdiff / 2
VIS
Figure 27. Simplified Electrical Circuit Diagram for Differential Input Signals
½ V OD
VO+
VOVOS
GND
VO+ and VO- referenced to GND
VOD = 2*|VO+ ± VO-|
[mVpp]
VOS = |VO+ + VO-| / 2 [V]
Figure 28. Electrical Level Diagram for Differential Output Signals
Zrdiff / 2
VO+
Ztt
+
-
VOS
VOZrdiff / 2
Figure 29. Electrical Circuit Diagram for Differential Output Signals
20
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8 Detailed Description
8.1 Overview
The ADC31JB68 is a low-power, wide-bandwidth, 16-bit, 500-MSPS analog-to-digital converter (ADC). The
buffered analog input provides uniform input impedance across a wide frequency range while minimizing sampleand-hold glitch energy. This device is designed for sampling analog input signals of up to 1300 MHz.
The ADC31JB68 provides excellent spurious-free dynamic range (SFDR) over a large input frequency range with
very low power consumption. On-chip dither provides an exceptionally clean noise floor. Embedded foreground
and background calibration ensures consistent dynamic performance over the entire temperature range and
minimizes part-to-part variation.
The device outputs its digital data from a JESD204B serial interface with two lanes transferring data at up to 5
Gbps/lane. The interface significantly reduces the number of lanes compared to an LVDS interface, allowing high
system integration density. An internal phase locked loop (PLL) transparently generates the necessary clocking
for data serialization.
The ADC31JB68 is offered in a 40- pin QFN (6 x 6mm) package and supports the full industrial temperature
range.
VCM
VIN±
CM
Reference
CLKIN+
CLKIN±
ADC
Buffer
SYNCb+
JESD204B Interface
VIN+
Imbalance
Correction
8.2 Functional Block Diagram
Internal
Reference
CLKIN
Divider
SYNCb±
SO1+
SO1±
SO0+
SO0±
SYSREF+
SYSREF±
SDI
SCLK
CSB
SPI
Interface
Control
Registers
SDO/OVR
Overrange
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8.3 Feature Description
8.3.1 Analog Inputs and Input Buffer
The ADC31JB68 analog signal inputs are designed to be driven differentially. The analog input pins have an
internal analog buffer that drives the sampling circuit. As a result of the analog buffer and internal 200 Ω
termination, the input pins present a time-constant impedance load to the external driving source which enables
great flexibility in the external analog filter design or direct impedance match to the driver. The buffer also helps
to isolate the external driving circuit from the internal switching charge transients of the sampling circuit which
results in a more consistent SFDR performance across input frequencies.
The common-mode voltage of the signal inputs is internally biased to 1.6-V via the internal termination resistors
which allows for AC coupling of the input drive network. Each input pin (VIN+, VIN–) must swing symmetrically
between (VCM + 0.425 V) and (VCM – 0.425 V), resulting in a 1.7 VPP (default) differential input swing.
8.3.2 Amplitude and Phase Imbalance Correction
The ADC performance can be sensitive to amplitude and phase imbalance of the input differential signal. A frontend balance correction circuit is integrated to optimize the second-order distortion (HD2) performance of the ADC
in the presence of an imbalanced input signal. 4-bit control of the phase mismatch and 3-bit control of the
amplitude mismatch corrects the input mismatch before the input buffer. A simplified diagram of the amplitude
and phase correction circuit at the ADC input is shown in Figure 30.
3
V IN+
VIN-
4
INPUT
BUFFER
+
-
V CM
Figure 30. Simplified Input Differential Balance Correction Circuit
Amplitude correction is achieved by varying the single-ended termination resistance of each input while
maintaining constant total differential resistance, thereby adjusting the amplitude at each input but leaving the
differential swing constant. Phase correction, also considered capacitive balance correction, varies the capacitive
load at the ADC input, thereby counter-acting the phase difference between the analog inputs while minimally
affecting amplitude. This function is useful for correcting the balance of transformers or filters that drive the ADC
analog inputs. Figure 31 shows the measured HD2 resulting from an example 300-MHz imbalanced input signal
measured over the available amplitude and phase correction settings. Performance parameters in the Converter
Performance Characteristics are characterized with the amplitude and phase correction settings in the default
condition (no correction).
Figure 31. HD2 Optimization at 300 MHz Using Gain and Phase Imbalance Correction
22
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Feature Description (continued)
8.3.3 Over-Range Detection
Over-range detection is available via the shared SDO/OVR dual-mode pin. Configuration of the SDO/OVR pin
into the over-range mode is done through the SPI. By default, the over-range mode is not selected. The
SDO/OVR pin asserts (logical high) when an over-range signal is detected at the input. The short delay from
when an over-range signal is incident at the input until the SDO/OVR output is asserted allows for almost
immediate detection of over-range signals without delay from the internal ADC pipeline latency or serial link
latency.
The input power threshold to indicate an over-range event is programmable via the SPI in steps of 128 codes
relative to the 16-bit code range of the data at the output of the ADC core.
After an over-range event occurs and the signal at the channel input reduces to a level below full-scale, an
internal counter begins counting to provide a hold function. When the counter reaches the hold counter threshold,
the over-range signal is de-asserted (logical low). The duration of the hold counter is programmable via the SPI
to hold for +3, +7, or +15 frame clock cycles. The counter is disabled (+0 cycles) by default to allow de-assertion
without holding.
8.3.4 Input Clock Divider
An input clock divider allows a high frequency clock signal to be distributed throughout the system and locally
divided down at the ADC device. The frequency at the CLKIN input may be divided down to the sampling rate of
the ADC by factors of 1, 2, or 4. Changing the clock divider setting initiates a JESD204 link re-initialization and
requires re-calibration of the ADC if the sampling rate is changed from the rate during the previous calibration
(see ADC Core Calibration).
8.3.5 SYSREF Detection Gate
When the signal at the SYSREF input is not actively toggling periodically, the SYSREF signal is considered to be
in an idle state. The idle state is recommended at any time the ADC31JB68 spurious performance must be
maximized. The SYSREF detection gate is provided to prevent transitions of the SYSREF signal in and out of the
idle state from impacting the JESD204B core. While the SYSREF signal is In the idle state, the SYSREF
detection gate should be used reject noise that may appear on the SYSREF signal.
The detection gate is the AND gate shown in Figure 72. The gate enables or disables propagation of the
SYSREF signal through to the internal device logic. If the detection gate is disabled and a false edge appears at
the SYSREF input, the signal does not disrupt the internal clock alignment. Note that the SYSREF detection gate
is disabled by default; therefore, the device does not respond to a SYSREF edge until the detection gate is
enabled.
The SYSREF detection gate features is controlled through the SPI.
8.3.6 Serial Differential Output Drivers
The differential drivers that output the serial JESD204B data are voltage mode drivers with amplitude control and
de-emphasis features that may be configured through the SPI for a variety of different channel applications. Eight
amplitude control (VOD) and eight de-emphasis control (DEM) settings are available. Both VOD and DEM
register fields must be configured to optimize the noise performance of the serial link for a particular lossy
channel.
8.3.6.1 De-Emphasis Equalization
De-emphasis of the differential output is provided as a form of continuous-time linear equalization that imposes a
high-pass frequency response onto the output signal to compensate for frequency-dependent attenuation as the
signal propagates through the channel to the receiver. In the time-domain, the de-emphasis appears as the bit
transition transient followed by an immediate reduction in the differential amplitude, as shown in Figure 32. The
characteristic appearance of the waveform changes with differential amplitude and the magnitude of deemphasis applied. The serial lane rate determines the available period of time during which the de-emphasis
transient settles. However, the lane rate does not affect the settling behavior of the applied de-emphasis. The deemphasis value is measured as the ratio (in units of [dB]) between the peak voltage after the signal transition to
the settled voltage value in one bit period. The data rate for this measurement is 1 Gb/s to allow settling of the
de-emphasis transient.
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Feature Description (continued)
Figure 32. De-emphasis of the Differential Output Signal
8.3.6.2 Serial Lane Inversion
The polarity of the individual serial data lanes can be controlled with the serial lane inversion enable function via
the SER_INV register. These controls simplify PCB routing of the serial lanes by allowing the transmitter to be
connected to the receiver with either polarity.
8.3.7 ADC Core Calibration
The ADC core of this device requires foreground calibration to be performed after power-up to achieve full
performance. Immediately after power-up, the ADC31JB68 device detects that the supplies and clock are valid,
waits for a power-up delay, and then performs a foreground calibration of the ADC core automatically. The
power-up delay is 9 × 106 sampling clock cycles or 18 ms at a 500-MSPS sampling rate. The calibration requires
approximately 1.0 × 106 sampling clock cycles.
If the system requires that the ADC31JB68 input clock divider value (CLKDIV) is set to 2 or 4, then ADC
calibration should be performed manually after CLKDIV has been set to the desired value. Manually calibrating
the ADC core is performed by changing to power down mode, returning to normal operation, and monitoring the
CAL_DONE bit in the JESD_STATUS register until calibration is complete. As an alternative to monitoring
CAL_DONE, the system may wait 1.5 × 106 sampling clock cycles until calibration completes.
When the ADC core enters normal conversion, background calibration monitors the performance of the device
and automatically adjusts the core to optimally correct for changes in the operating conditions such as supply
and temperature. The background calibration settling time is less than 375 × 106 sampling clock cycles.
8.3.8 Data Format
Data may be output in the serial stream as 2’s complement format (default) or as offset binary. This selection is
chosen via the SPI. The formatting is performed in the data path prior to JESD204B data framing and 8b/10b
encoding.
8.3.9 JESD204B Supported Features
The ADC31JB68 device supports a specific feature set of the JESD204B standard targeted to its intended
applications but does not implement all the flexibility of the standard. Table 1 summarizes the level of feature
support.
24
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Feature Description (continued)
Table 1. ADC31JB68 Feature Support for the JESD204B Serial Interface
Feature
Supported
Not Supported
Subclass
•
Subclass 1
•
Subclass 0, 2
Device Clock
(CLKIN) and
SYSREF
•
•
•
AC coupled CLKIN
DC coupled CLKIN and SYSREF
Periodic, Pulsed Periodic and One-Shot SYSREF
•
AC coupled SYSREF
Latency
•
Deterministic latency supported for subclass
implementations using standard SYSREF signal
Electrical layer
features
•
•
•
LV-OIF-11G-SR interface and performance
AC coupled serial lanes
TX lane polarity inversion
Transport layer
features and
configuration
•
•
•
L = 2 lanes
K configuration
Scrambling
Data link layer
features
•
•
•
8b/10b encoding
Lane synchronization
•
D21.5, K28.5, ILA, PRBS7, PRBS15, PRBS23, Ramp test
sequences
1 •
Deterministic latency not supported for nonstandard implementations
•
DC coupled serial lanes
•
F, S, L, and HD configuration is not independently
configurable
M, N, N’, CS, CF configuration is not
independently configurable
Idle link mode
Short and Long transport layer test patterns
•
•
•
RPAT/JSPAT test sequences
8.3.10 JESD204B Interface
The JESD204B transmitter block consists of the transport layer, the data scrambler and the link layer. The
transport layer maps the ADC output data into the selected JESD204B frame data format and manages the
transmission of ADC output data or test patterns. The link layer performs the 8b/10b data encoding as well as the
synchronization and initial lane alignment using the SYNCb input signal. Data from the transport layer can be
optionally scrambled.
JESD204B Block
Transport Layer
Frame Data
Mapping
Link Layer
8b/10b
encoding
Scrambler
1+x14+x15
S0/S1
Comma characters
Initial lane alignment
Test Patterns
SYNCb
Figure 33. JESD204B Transmitter Block
8.3.11 Transport Layer Configuration
The transport layer features supported by the ADC31JB68 device are a subset of possible features described in
the JESD204B standard. The configuration options are intentionally simplified to provide the lowest power and
most easy-to-use solution.
8.3.11.1 Lane Configuration
The digital data is output on two serial lanes that support JESD204B. The number of transmission lanes per
channel (L) is only 2. The serial data rate is 10 times the sampling rate. A 500 MSPS sampling rate corresponds
to a 5.0 Gb/s per lane rate.
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8.3.11.2 Frame Format
The lanes per device (L), octets per frame (F), samples per frame (S), and high-density mode (HD) parameters
are not independently configurable. The N, N’, CS, CF, M, and HD parameters are fixed and not configurable.
Table 2 lists the available JESD204B formats and valid ranges for the ADC31JB68. The ranges are limited by
the Serdes line rate and the maximum ADC sample frequency. Figure 34 shows the data format.
Table 2. Available JESD204B Formats and Valid Ranges
L
M
F
S
MAX ADC SAMPLING RATE
(Msps)
MAX fSERDES
(Gbps)
2
1
1
1
500
5.0
D[15:0] = 16-bit Word
Octet 0
(MSB)
(LSB)
Lane 0
D[15] D[14] D[13] D[12] D[11] D[10] D[9]
D[8]
Lane 1
D[7]
D[0]
D[6]
D[5]
D[4]
D[3]
D[2]
D[1]
L=2
S=1
F=1
N=16
CS=0
1¶=16
Figure 34. Transport Layer Definitions for the Supported-Lane Configurations
8.3.11.3 ILA Information
Table 3 summarizes the information transmitted during the initial lane alignment (ILA) sequence. Mapping of
these parameters into the data stream is described in the JESD204B standard.
Table 3. Configuration of the JESD204B Serial-Data Receiver
Logical Value
Encoded Value
ADJCNT
Parameter
DAC LMFC adjustment
0
0
ADJDIR
DAC LMFC adjustment direction
0
0
BID
Bank ID
0
0
CF
Number of control words per frame clock period per link
0
0
CS
Number of control bits per sample
0
0
DID
Device identification number
0
0
F
Number of octets per frame (per lane) (1)
1
0
HD
High-density format
1
1
JESDV
JESD204 version
1
1
Set by register
as 17 to 32
16 to 31
K
Description
Number of frames per multi-frame
(1)
(1)
L
Number of lanes per link
LID
Lane identification number
M
Number of converters per device (1)
N
Converter resolution
N’
2
1
0 (lane 0), 1 (lane 1)
0 or 1
1
0
16
15
Total number of bits per sample (1)
16
15
PHADJ
Phase adjustment request to DAC
0
0
S
Number of samples per converter per frame cycle (1)
1
0
Set by register
as 0 (disabled) or 1
0 or 1
(1)
SCR
Scrambling enabled
SUBCLASSV
Device subclass version
1
1
RES1
Reserved field 1
0
0
RES2
Reserved field 2
0
0
FCHK
Checksum (2)
Computed
Computed
(1)
(2)
26
These parameters have a binary-value-minus-1 encoding applied before being mapped into the link configuration octets. For example, F
= 1 is encoded as 0.
Example: For K=32, lane 0, scrambler disabled, the FCHK value in the ILA will be 0x41 (hex) or 65 (decimal)
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Scrambling of the output serial data is supported and conforms to the JESD204B standard. Scrambling is
disabled by default, but may be enabled via the SPI. When scrambling is enabled, the ADC31JB68 device
supports the early synchronization option by the receiver during the ILA sequence, although the ILA sequence
data is never scrambled.
8.3.12 Test Pattern Sequences
The SPI may enable the following test pattern sequences. Short- and long-transport layer, RPAT, and JSPAT
sequences are not supported.
Table 4. Supported Test Pattern Sequences
Test Pattern
Description
Common Purpose
D21.5
Data is transmitted across a normal link but ADC sampled
data is replaced with D21.5 symbols, resulting in an
alternating 1 and 0 pattern (101010...) on each serial lane.
After enabling this pattern, the JESD204B link must be reinitialized.
Jitter or system debug
K28.5
Continuous K28.5 symbols are output on each serial lane.
Link initialization is not possible nor required.
System debug
Repeated ILA
ILA repeats indefinitely on each serial lane. After enabling this
System debug
pattern, the JESD204B link must be reinitialized.
Ramp
Data is transmitted across a normal link but ADC sampled
data is replaced with a ramp pattern. The ramp ascends
System debug and transport layer verification
through a 16-bit range and the step is programmable. After
enabling this pattern, the JESD204B link must be reinitialized.
PRBS
Standard pseudo-random bit sequences are output on each
serial lane. PRBS 7/15/23 Complies with ITU-T O.150
specification and is compatible with J-BERT equipment. Link
initialization is not possible nor required.
Jitter and bit error rate testing
8.3.13 JESD204B Link Initialization
A JESD204B link is established via link initialization, which involves the following steps: frame alignment, code
group synchronization, and initial lane synchronization. These steps are shown in Figure 35. Link initialization
must occur between the transmitting device (ADC) and receiving device before sampled data may be transmitted
over the link. The link initialization steps described here are specifically for the ADC31JB68 device, supporting
JESD204B subclass 1.
SYSREF assertion
SYNCb assertion
latched
latched
SYNCb de-assertion
latched
tS-SYNCb-F
tS-SYNCb
tS-SYNCb-F
SYNCb
tH-SYNCb-F
tILA
Serial Data
XXX
XXX
tH-SYS
tS-SYS
K28.5
K28.5
ILA
tD-K28
tD-ILA
ILA
Valid Data
tD-DATA
CLKIN
SYSREF
tPL-SYS
tPH-SYS
Tx Frame Clk
Tx LMFC Boundary
tD-LMFC
Frame Clock
Alignment
Code Group
Synchronization
Initial Frame and Lane
Synchronization
Data
Transmission
Figure 35. Link-initialization Timing and Flow Diagram
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8.3.13.1 Frame Alignment
The Frame Alignment step requires alignment of the frame and local multi-frame clocks within the ADC31JB68
device to an external reference. This is accomplished by providing the device clock and SYSREF clock to the
CLKIN and SYSREF inputs, respectively. The ADC31JB68 device aligns its frame clock and LMFC to any
SYSREF rising edge event, offset by a SYSREF-to-LMFC propagation delay.
The SYSREF signal must be source synchronous to the device clock; therefore, the SYSREF rising edge must
meet setup and hold requirements relative to the signal at the CLKIN input. If these requirements cannot be met,
then the alignment of the internal frame and multi-frame clocks cannot be specified. As a result, a link may still
be established, but the latency through the link cannot be deterministic. Frame alignment may occur at any time
but a re-alignment of the internal frame clock and LMFC will break the link. Note that frame alignment is not
required for the ADC31JB68 device to establish a link because the device automatically generates the clocks on
power-up with unknown phase alignment.
8.3.13.2 Code Group Synchronization
Code Group Synchronization is initiated when the receiver sends a synchronization request by asserting the
SYNCb input of the ADC31JB68 device to a logic low state (SYNCb+ < SYNCb–). After the SYNCb assertion is
detected, the ADC31JB68 device outputs K28.5 symbols on all serial lanes. These symbols are used by the
receiver to synchronize and time align its clock and data recovery (CDR) block to the known symbols. The
SYNCb signal must be asserted for at least 4 frame clock cycles otherwise the event is ignored by the
ADC31JB68 device. Code group synchronization is completed when the receiver de-asserts the SYNCb signal to
a logic high state.
After the ADC31JB68 detects a de-assertion of its SYNCb input, the Initial Lane Synchronization step begins on
the following LMFC boundary. The ADC31JB68 device outputs 4 multi-frames of information that compose the
ILA sequence. This sequence contains information about the data transmitted on the link. The initial lane
synchronization step and link initialization conclude when the ILA is finished and immediately transitions into
Data Transmission. During data transmission, valid sampled data is transmitted across the link until the link is
broken.
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ADC Core Calibration Complete
(after power up or Power Down Mode Exit)
Initialize
Default
Frame Clock
and LMFC
Alignment
Clock Alignment and
Synchronization Requests
SYSREF
Assertion
Detected
LMFC
Alignment
Error?
YES
Frame
Alignment
Error?
NO
YES
NO
Re-align
Frame
Clock
& LMFC
SYNCb
Assertion
Detected
Sleep Mode Exit
Serializer
PLL
Calibration
Send K28.5
Characters
Send Encoded
Sampled Data
Valid Data
Transfer
Send ILA
Sequence
Wait for
Next LMFC
Boundary
SYNCb
De-Asserted?
NO
YES
JESD204B Link
Initialization
Figure 36. Device Start-Up and JESD204B Link Synchronization Flow Chart
The flowchart in Figure 36 describes how the ADC31JB68 device initializes the JESD204B link and reacts to
changes in the link. After the ADC core calibration is finished, the ADC31JB68 device begins with PLL calibration
and link initialization using a default frame clock and LMFC alignment by sending K28.5 characters. PLL
calibration requires approximately 153×103 sampling clock cycles. If SYNCb is not asserted, then the device
immediately advances to the ILA sequence at the next LMFC boundary. If SYNCb is asserted, then the device
continues to output K28.5 characters until SYNCb is de-asserted.
When a SYSREF rising edge event is detected, then the ADC31JB68 device compares the SYSREF event to the
current alignment of the LMFC. If the SYSREF event is aligned to the current LMFC alignment, then no action is
taken and the device continues to output data. If misalignment is detected, then the SYSREF event is compared
to the frame clock. If misalignment of the frame clock is also detected, then the clocks are re-aligned and the link
is reinitialized. If the frame clock is not misaligned, then the frame clock alignment is not updated. In the cases
that a SYSREF event causes a link re-initialization, the ADC31JB68 device begins sending K28.5 characters
without a SYNCb assertion and immediately transitions to the ILA sequence on the next LMFC boundary unless
the SYNCb signal is asserted. Anytime the frame clock and LMFC are re-aligned, the serializer PLL must
calibrate before code group synchronization begins. SYSREF events must not occur during ADC31JB68 device
power-up, ADC calibration, or PLL calibration. The JESD_STATUS register is available to check the status of the
ADC31JB68 device and the JESD204B link.
If a SYNCb assertion is detected for at least 4 frame clock cycles, the ADC31JB68 device immediately breaks
the link and sends K28.5 characters until the SYNCb signal is de-asserted.
When exiting sleep mode, the frame clock and LMFC are started with a default (unknown) phase alignment, PLL
calibration is performed, and the device immediately transitions into sending K28.5 characters.
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8.3.14 SPI
The SPI allows access to the internal configuration registers of the ADC through read and write commands to a
specific address. The interface protocol has a 1-bit command, 15-bit address word and 8-bit data word as shown
in Figure 37. A read or write command is 24 bits in total, starting with the read or write command bit where 0
indicates a write command and 1 indicates a read command. The read or write command bit is clocked into the
device on the first rising edge of SCLK after CSb is asserted to 0. During a write command, the 15-bit address
and 8-bit data values follow the read or write bit MSB-first and are latched on the rising edge of SCLK. During a
read command, the SDO output is enabled shortly after the 16th rising edge of SCLK and outputs the read value
MSB first before the SDO output is returned to a high impedance state. The read or write command is completed
on the SCLK rising edge on which the data word’s LSB is latched. CSb may be de-asserted to 1 after the LSB is
latched into the device.
The SPI allows command streaming where multiple commands are made without de-asserting CSb in-between
commands. The commands in the stream must be of similar types, either read or write. Each subsequent
command applies to the register address adjacent to the register accessed in the previous command. The
address order can be configured as either ascending or descending. Command streaming is accomplished by
immediately following a completed command with another set of 8 rising edges of SCLK without de-asserting
CSb. During a write command, an 8-bit data word is input on the SDI input for each subsequent set of SCLK
edges. During a read command, data is output from SDO for each subsequent set of SCLK edges. Each
subsequent command is considered finished after the 8th rising edge of SCLK. De-asserting CSb aborts an
incomplete command.
The SDO output is high impedance at all times other than during the final portion of a read command. During the
time that the SDO output is active, the logic level is determined by a configuration register. The SPI output logic
level must be properly configured after power up and before making a read command to prevent damaging the
receiving device or any other device connected to the SPI bus. Until the SPI_CFG register is properly configured,
voltages on the SDO output may be as high as the VA3.0 supply during a read command. The SDI, SCLK, and
CSB pins are all 1.2-V to 3.0-V compatible.
CSB
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
D2
D1
D0
SCLK
COMMAND FIELD
SDI
R/W
A14
A13
A12
R=1 (MSB)
W=0
A11
A10
DATA FIELD
A9
A8
A7
A6
A5
A4
A3
A2
A1
A0
(LSB)
Address (15-bits)
Hi-Z
D7
(MSB)
D7
SDO
D6
(MSB)
D5
D4
D3
(LSB)
Write DATA (8-bits)
D6
D5
D4
D3
D2
Read DATA (8-bits)
D1
D0
(LSB)
Single Access Cycle
Figure 37. Serial Interface Protocol
8.4 Device Functional Modes
8.4.1 Power-Down and Sleep Modes
Power-down and sleep modes are provided to allow the user to reduce the power consumption of the device
without disabling power supplies. Both modes reduce power consumption by the same amount but they differ in
the amount of time required to return to normal operation. Upon changing from Power Down back to Normal
operation, an ADC calibration routine is performed. Waking from sleep mode does not perform ADC calibration
(see ADC Core Calibration for more details). Neither power-down mode nor sleep mode resets configuration
registers.
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8.5 Register Map
Table 5. ADC31JB68 Register Map
Register
ADDRESS
DFLT
b[7]
b[6]
b[5]
b[4]
CONFIG_A
0x0000
0x3C
SR
Res (0)
ASCEND
Res (1)
b[3]
b[2]
b[1]
b[0]
PAL[3:0]
Address 0x0001 Reserved
DEVICE _CONFIG
0x0002
0x00
CHIP_TYPE
0x0003
0x03
0x0004
0x11
CHIP_ID[7:0]
0x0005
0x00
CHIP_ID[15:8]
0x0006
0x00
CHIP_ID
CHIP _VERSION
Reserved (000000)
PD_MODE[1:0]
Reserved (0000)
CHIP_TYPE[3:0]
CHIP_VERSION[7:0]
Address 0x0007-0x000B Reserved
VENDOR_ID
SPI_CFG
0x000C
0x51
0x000D
0x04
0x0010
0x01
OM1
0x0012
0xC1
OM2
0x0013
0x20
IMB_ADJ
0x0014
0x00
VENDOR_ID[7:0]
VENDOR_ID[15:8]
Reserved (000000)
DF
SYS_CM[1:0]
VSPI[1:0]
SYSG
_EN
Res (00)
Reserved (001000)
Res (0)
Res(01)
CLKDIV [1:0]
AMPADJ[2:0]
PHADJ[3:0]
Address 0x0016-0x003A Reserved
OVR_EN
0x003A
0x00
OVR_HOLD
0x003B
0x00
OVR_TH
0x003C
0x00
DC_MODE
0x003D
0x00
Reserved (0000000)
OVR_EN
Reserved (000000)
OVR_HOLD[1:0]
OVR_TH[7:0]
Reserved (00000)
DC_TC[1:0]
DC_EN
Address 0x003E-0x0046 Reserved
SER_CFG
0x0047
0x00
Res(0)
VOD[2:0]
Res (0)
DEM[2:0]
Address 0x0048-0x005F Reserved
SCR
_EN
JESD_CTRL1
0x0060
0x7F
JESD_CTRL2
0x0061
0x00
0x0062
0x01
JESD_RSTEP[7:0]
0x0063
0x00
JESD_RSTEP[15:8]
0x0064
0x00
JESD_RSTEP
SER_INV
K_M1[4:0]
Res (0)
Reserved (0000)
JESD
_EN
JESD_TEST_MODE[3:0]
Reserved (0000)
SO1_INV
_EN
SO0_IN
V
_EN
ALIGN
PLL
_LOCK
Reserved (00)
Address 0x0065-0x006B Reserved
JESD_STATUS
0x006C
N/A
Res (0)
LINK
SYNC
RE
ALIGN
CAL
_DONE
CLK
_RDY
Address 0x006D- Reserved
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8.5.1 Register Descriptions
8.5.1.1 CONFIG_A (address = 0x0000) [reset = 0x3C]
Figure 38. CONFIG_A
7
SR
R/W
6
Reserved
R/W
5
ASCEND
R/W
4
Reserved
R
3
2
1
0
PAL[3:0]
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 6. CONFIG_A
Bit
(1)
Field
Type
Reset
Description
7
SR
R/W
0
Setting this soft reset bit causes all registers to be reset to their
default state. This bit is self-clearing.
6
Reserved
R/W
0
Reserved and must be written with 0.
5
ASCEND
R/W
1
Order of address change during streaming read or write
commands.
0 : Address is decremented during streaming reads or writes.
1 : Address is incremented during streaming reads or writes
(default).
4
Reserved
R
1
Reserved and must be written with 1.
3:0
PAL[3:0]
R/W
1100
Palindrome bits are bit 3 = bit 4, bit 2 = bit 5, bit 1 = bit 6, and bit
0 = bit 7. (1)
All writes to this register must be a palindrome (for example, bits [3:0] are a mirror image of bits [7:4]). If the data is not a palindrome,
the entire write is ignored.
8.5.1.2 DEVICE CONFIG (address = 0x0002) [reset = 0x00]
Figure 39. DEVICE CONFIG
7
6
5
4
3
Reserved
R/W
2
1
0
PD_MODE [1:0]
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 7. DEVICE CONFIG
Bit
Field
7:2
1:0
32
PD_MODE [1:0]
Type
Reset
Description
R/W
000000
Reserved and must be written with 000000.
R/W
00
Power-down mode
00 : Normal operation (default)
01 : Reserved
10 : Sleep operation (low power, fastest resume)
11 : Power-down (lowest power)
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8.5.1.3 CHIP_TYPE (address = 0x0003 ) [reset = 0x03]
Figure 40. CHIP_TYPE
7
6
5
4
3
2
Reserved
R/W
1
0
CHIP_TYPE
R
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 8. CHIP_TYPE
Bit
Field
7:4
3:0
CHIP_TYPE [3:0]
Type
Reset
Description
R/W
0000
Reserved and must be written with 0000.
R
0011
Chip type that always returns 0x3, indicating that the part is a
high-speed ADC
8.5.1.4 CHIP_ID (address = 0x0005, 0x0004) [reset = 0x00, 0x1B]
Figure 41. CHIP_ID
7
6
5
4
3
2
1
0
1
0
CHIP_ID
R
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 9. CHIP_ID
Bit
Field
Type
Reset
Description
0x0004[7:
0]
CHIP_ID [7:0]
R
0x1B
Chip ID least significant word
0x0005[7:
0]
CHIP_ID [15:8]
R
0x00
Chip ID most significant word
8.5.1.5 CHIP_VERSION (address =0x0006) [reset = 0x00]
Figure 42. CHIP_VERSION
7
6
5
4
3
CHIP_VERSION
R
2
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 10. CHIP_VERSION
Bit
Field
Type
Reset
Description
7:0
CHIP_VERSION [7:0]
R
0x00
Chip version
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8.5.1.6 VENDOR_ID (address = 0x000D, 0x000C) [reset = 0x04, 0x51]
Figure 43. VENDOR_ID
7
6
5
4
3
2
1
0
VENDOR_ID
R
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 11. VENDOR_ID
Bit
Field
Type
Reset
Description
0x000C[7: VENDOR_ID [7:0]
0]
R
0x51
Vendor ID. Texas Instruments vendor ID is 0x0451.
0x000D[7: VENDOR_ID [15:8]
0]
R
0x04
8.5.1.7 SPI_CFG (address = 0x0010 ) [reset = 0x01]
Figure 44. SPI_CFG
7
6
5
4
3
2
1
Reserved
R/W
0
VSP
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 12. SPI_CFG
34
Bit
Field
Type
Reset
Description
7:2
Reserved
R/W
000000
Reserved and must be written with 000000.
1:0
VSPI[1:0]
R/W
01
SPI logic level controls the SDO output logic level.
00 : 1.2 V
01 : 3.0 V (default)
10 : Reserved
11 : 1.8 V
This register must be configured (written) before making a read
command with a SPI that is not a 3-V logic level. The SPI inputs
(SDI, SCLK, and CSb) are compatible with logic levels ranging
from 1.2 to 3 V.
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8.5.1.8 OM1 (Operational Mode 1) (address = 0x0012) [reset = 0xC1]
Figure 45. OM1 (Operational Mode 1)
7
DF
R/W
6
5
4
3
SYS)CM[1:0]
R/W
Reserved
R/W
2
SYSG_EN
R/W
1
0
Reserved
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 13. OM1 (Operational Mode 1)
Bit
Field
Type
Reset
Description
DF
R/W
1
Output data format
0 : Offset binary
1 : Signed 2s complement (default)
6:5
SYS_CM[1:0]
R/W
10
SYSREF Common-Mode Configuration
When the SYSREF signal interface is DC coupled, the
SYSREF+/- input receiver must be configured to appropriately
match the common-mode of the received signal. Set the register
field according to the expected common-mode.
00 : 0.4 V - 0.59 V (RTAIL = Open)
01 : 0.6 V - 0.99 V (RTAIL = 4 kΩ)
10 : 1.0 V - 1.49 V (RTAIL = 1 kΩ, default)
11 : 1.5 V - 2.0 V (RTAIL = 0 Ω)
•
The RTAIL indicates the value of the variable resistor shown
in Figure 72
4:3
Reserved
R/W
00
Reserved and must be written with 00.
2
SYSG_EN
R/W
0
SYSREF detection gate enable
0 : SYSREF gate is disabled; (input is ignored, default)
1 : SYSREF gate is enabled
1:0
Reserved
R/W
01
Reserved. Must be written with 01.
7
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8.5.1.9 OM2 (Operational Mode 2) (address = 0x0013) [reset = 0x20]
Figure 46. OM2 (Operational Mode 2)
7
6
5
4
3
2
1
Reserved
R/W
0
CLKDIV
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 14. OM2 (Operational Mode 2)
Bit
Field
Type
Reset
Description
7:2
Reserved
R/W
001000
Reserved and must be written with 001000.
1:0
CLKDIV[1:0]
R/W
00
Clock divider ratio. Sets the value of the clock divide factor,
CLKDIV
00 : Divide-by-1, CLKDIV = 1 (default)
01 : Divide-by-2, CLKDIV = 2
10 : Divide-by-4, CLKDIV = 4
11 : Reserved
8.5.1.10 IMB_ADJ (Imbalance Adjust) (address = 0x0014) [reset = 0x00]
Figure 47. IMB_ADJ (Imbalance Adjust)
7
Reserved
R/W
6
5
AMPADJ[2:0]
R/W
4
3
2
1
0
PHADJ[3:0]
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 15. IMB_ADJ (Imbalance Adjust)
Bit
7
36
Field
Type
Reset
Description
Reserved
R/W
0
Reserved. Must be written with 0.
6:4
AMPADJ[2:0]
R/W
000
Analog input amplitude imbalance correction
7 = +30 Ω VIN+, –30 Ω VIN–
6 = +20 Ω VIN+, –20 Ω VIN–
5 = +10 Ω VIN+, –10 Ω VIN–
4 = Reserved
3 = –30 Ω VIN+, +30 Ω VIN–
2 = –20 Ω VIN+, +20 Ω VIN–
1 = –10 Ω VIN+, +10 Ω VIN–
0 = +0 Ω VIN+, –0 Ω VIN– (default)
Resistance changes indicate variation of the internal singleended termination
3:0
PHADJ[3:0]
R/W
0000
Analog input phase imbalance correction
15 = +1.68 pF VIN–
...
10 = +0.48 pF VIN–
9 = +0.24 pF VIN–
8 = Reserved
7 = +1.68 pF VIN+
...
2 = +0.48 pF VIN+
1 = +0.24 pF VIN+
0 = +0 pF VIN+, +0 pF VIN– (default)
Capacitance changes indicate the addition of internal capacitive
load on the given pin.
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8.5.1.11 OVR_EN (Over-Range Enable) (address = 0x003A) [reset = 0x00]
Figure 48. OVR_EN (Over-Range Enable)
7
6
5
4
Reserved
R/W
3
2
1
0
OVR_EN
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 16. OVR_EN (Over-Range Enable)
Bit
Field
Type
Reset
Description
7:1
Reserved
R/W
000000
Reserved and must be written as 000000.
0
OVR_EN
R/W
0
Over-range Enable.
0 : Over-Range function is disabled. (default)
1 : Over-Range function is enabled and output via the SDO pin.
8.5.1.12 OVR_HOLD (Over-Range Hold) (address = 0x003B) [reset = 0x00]
Figure 49. OVR_HOLD (Over-Range Hold)
7
6
5
4
3
2
Reserved
R/W
1
0
OVR_HOLD [1:0]
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 17. OVR_HOLD (Over-Range Hold)
Bit
Field
Type
Reset
Description
7:2
Reserved
R/W
000000
Reserved and must be written as 000000.
1:0
OVR_HOLD [1:0]
R/W
00
Over-range hold function. In the event of an input signal larger
than the full-scale range, an over-range event occurs and the
over-range indicators are asserted. OVR_HOLD determines the
amount of time the over-range indicators remain asserted after
the input signal has reduced below full-scale.
00 : OVR indicator extended by +0 clock cycles (default)
01 : OVR indicator extended by +3 clock cycles
10 : OVR indicator extended by +7 clock cycles
11 : OVR indicator extended by +15 clock cycles
Note:
•
The unit of clock cycles corresponds to the period of the
internal sampling clock.
•
The over-range indicators also experience a latency from
when the over-range signal is sampled to when the indicator
is asserted or de-asserted.
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8.5.1.13 OVR_TH (Over-Range Threshold) (address = 0x003C) [reset = 0x00]
Figure 50. OVR_TH (Over-Range Threshold)
7
6
5
4
3
2
1
0
OVR_TH [7:0]
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 18. OVR_TH (Over-Range Threshold)
Bit
Field
Type
Reset
Description
7:0
OVR_TH [7:0]
R/W
00000000
Over-range threshold. This field is an unsigned value from 0 to
255. OVR_TH sets the over-range detection thresholds for the
ADC. If the 16-bit signed data exceeds the thresholds, then
the over-range bit is set. The 16-bit thresholds are ± OVR_TH
× 128 codes from the low and high full-scale codes (32767
and –32768 in signed 2s complement). If OVR_TH is 0, then
the default threshold is used (full scale).
16-bit Threshold
38
OVR_TH
2
Complemen
t
Offset
Binary
Threshold Relative
to Peak Full Scale
[dB]
255 (0xFF)
±32640
65408 / 128
–0.03
254 (0xFE)
±32512
65280 / 256
–0.07
128 (0x80)
±16384
49152 /
16,384
–6.02
2 (0x02)
±256
33024 /
32512
–42.14
1 (0x01)
±128
32896 /
32640
–48.16
0 (0x00)
(default)
+32767 /
–32768
65535 / 0
–0.0
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8.5.1.14 DC_MODE (DC Offset Correction Mode) (address = 0x003D) [reset = 0x00]
Figure 51. DC_MODE (DC Offset Correction Mode)
7
6
5
Reserved
R/W
4
3
2
1
TC_DC[1:0]
R/W
0
DC_EN
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 19. DC_MODE (DC Offset Correction Mode)
Bit
Field
Type
Reset
Description
7:3
Reserved
R/W
00000
Reserved and must be written as 00000.
2:1
TC_DC[1:0]
R/W
00
DC offset filter time constant.
The time constant determines the filter bandwidth of the DC high-pass
filter.
0
DC_EN
R/W
0
TC_TD
Time Constant
(FS = 500 MSPS)
3-dB Bandwidth
(FS = 500 MSPS)
3-dB Bandwidth
(Normalized)
00
8.6 µs
18.5 kHz
37e–6 × Fs
01
65 µs
2.45 kHz
4.9e–6 × Fs
10
526 µs
303 Hz
605e–9 × Fs
11
4.2 ms
38 Hz
76e–9 × Fs
DC offset correction enable
0 : Disable DC offset correction
1 : Enable DC offset correction
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8.5.1.15 SER_CFG (Serial Lane Transmitter Configuration) (address = 0x0047) [reset = 0x00]
Figure 52. SER_CFG (Serial Lane Transmitter Configuration)
7
Reserved
R/W
6
5
VOD[2:0]
R/W
4
3
Reserved
R/W
2
1
DEM[2:0]
R/W
0
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 20. SER_CFG (Serial Lane Transmitter Configuration)
Bit
40
Field
Type
Reset
Description
7
Reserved
R/W
0
Reserved. Must be written as 0.
6:4
VOD[2:0]
R/W
000
Serial-lane transmitter driver output differential peak-to-peak
voltage amplitude.
000 : 0.400 V (default)
001 : 0.470 V
010 : 0.540 V
011 : 0.610 V
100 : 0.670 V
101 : 0.740 V
110 : 0.790 V
111 : 0.840 V
Reported voltage values are nominal values at low-lane rates
with de-emphasis disabled.
3
Reserved
R/W
0
Reserved and must be written as 0.
2:0
DEM[2:0]
R/W
000
Serial lane transmitted de-emphasis.
000 : 0 dB
001 : –0.8 dB
010 : –2.4 dB
011 : –3.8 dB
100 : –4.9 dB
101 : –6.3 dB
110 : –7.7 dB
111 : –10.3 dB
Reported de-emphasis values are nominal values at low-lane
rates with VOD = 4.
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8.5.1.16 JESD_CTRL1 (JESD Configuration Control 1) (address = 0x0060) [reset = 0x7F]
Note: Before altering any parameters in this register, JESD_EN must be set = 0. Changing parameters while
JESD_EN = 1 is not supported.
Figure 53. JESD_CTRL1 (JESD Configuration Control 1)
7
SCR_EN
R/W
6
5
4
K_M1[4:0]
R/W
3
2
1
Reserved
R/W
0
JESD_EN
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 21. JESD_CTRL1 (JESD Configuration Control 1)
Bit
Field
Type
Reset
Description
7
SCR_EN
R/W
0
Scrambler enable.
0 : Disabled (default)
1 : Enabled
Note:
•
JESD_EN must be set to 0 before altering this field.
6:2
K_M1[4:0]
R/W
11111
Number of frames per multi-frame minus 1.
The binary values of K_M1 represent the value (K – 1)
00000 : Reserved
00001 : Reserved
…
01111 : Reserved
10000 : K = 17
…
11111 : K = 32 (default)
Note:
•
K must be in the range 17 to 32. Values outside this range
are either reserved or may produce unexpected results.
•
JESD_EN must be set to 0 before altering this field.
1
Reserved
R/W
1
Reserved and must be written with 1.
0
JESD_EN
R/W
1
JESD204B link enable.
When enabled, the JESD204B link synchronizes and transfers
data normally. When the link is disabled, the serial transmitters
output a repeating, alternating 01010101 stream.
0 : Disabled
1 : Enabled (default)
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8.5.1.17 JESD_CTRL2 (JESD Configuration Control 2) (address = 0x0061) [reset = 0x00]
Figure 54. JESD_CTRL2 (JESD Configuration Control 2)
7
6
5
4
3
Reserved
R/W
2
1
ESD_TEST_MODES[3:0]
R/W
0
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 22. JESD_CTRL2 (JESD Configuration Control 2)
42
Bit
Field
Type
Reset
Description
7:4
Reserved
R/W
0000
Reserved. Must be written as 0000.
3:0
ESD_TEST_MODES[3:0]
R/W
0000
JESD204B test modes.
0000 : Test mode disabled. Normal operation (default)
0001 : PRBS7 test mode
0010 : PRBS15 test mode
0011 : PRBS23 test mode
0100 : Reserved
0101 : ILA test mode
0110 : Ramp test mode
0111 : K28.5 test mode
1000 : D21.5 test mode
1001: Logic low test mode (serial outputs held low)
1010: Logic high test mode (serial outputs held high)
1011 – 1111 : Reserved
Note:
•
JESD_EN must be set to 0 before altering this field.
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8.5.1.18 JESD_RSTEP (JESD Ramp Pattern Step) (address = 0x0063, 0x0062) [reset = 0x00, 0x01]
Figure 55. JESD_RSTEP (JESD Ramp Pattern Step)
7
6
5
4
3
2
1
0
JESD_RSTEP
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 23. JESD_RSTEP (JESD Ramp Pattern Step)
Bit
Field
Type
Reset
Description
0x0062[7:
0]
JESD_RSTEP [7:0]
R/W
0x01
0x0063[7:
0]
JESD_RSTEP [15:8]
R/W
0x00
JESD204B ramp test mode step
The binary value JESD_RSTEP[15:0] corresponds to the step of
the ramp mode step. A value of 0x0000 is not allowed.
Note:
•
JESD_EN must be set to 0 before altering this field.
8.5.1.19 SER_INV (Serial Lane Inversion Control) (address = 0x0064) [reset = 0x00]
Figure 56. SER_INV (Serial Lane Inversion Control)
7
6
5
4
3
SO1_INV_EN
R/W
Reserved
R/W
2
SO0_INV_EN
R/W
1
0
Reserved
R/W
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 24. SER_INV (Serial Lane Inversion Control)
Bit
Field
Type
Reset
Description
7:4
Reserved
R/W
0000
Reserved. Must be written as 0000.
Serial Lane Inversion enable for SO1 and SO0 Output Drivers
0 : Non-Inverted Polarity (default)
1 : Inverted Polarity
Note:
•
JESD_EN must be set to 0 before altering this field.
3
SO1_INV_EN
R/W
0
2:
SO0_INV_EN
R/W
0
1:0
Reserved
R/W
00
Reserved. Must be written as 00.
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8.5.1.20 JESD_STATUS (JESD Link Status) (address = 0x006C) [reset = N/A]
Figure 57. JESD_STATUS (JESD Link Status)
7
Reserved
R
6
LINK
R
5
SYNC
R
4
REALIGN
R/W
3
ALIGN
R/W
2
PLL_LOCK
R
1
CAL_DONE
R
0
CLK_RDY
R
LEGEND: R/W = Read/Write; R = Read only; -n = value after reset
Table 25. JESD_STATUS (JESD Link Status)
Bit
44
Field
Type
Reset
Description
7
Reserved
R
N/A
Reserved.
6
LINK
R
N/A
JESD204B link status
This bit is set when synchronization is finished, transmission of
the ILA sequence is complete, and valid data is being
transmitted.
0 : Link not established
1 : Link established and valid data transmitted
5
SYNC
R
N/A
JESD204B link synchronization request status
This bit is cleared when a synchronization request is received at
the SYNCb input.
0 : Synchronization request received at the SYNCb input and
synchronization is in progress
1 : Synchronization not requested
Note:
•
SYNCb must be asserted for at least four local frame clocks
before synchronization is initiated. The SYNC status bit
reports the status of synchronization, but does not
necessarily report the current status of the signal at the
SYNCb input.
4
REALIGN
R/W
N/A
SYSREF re-alignment status
This bit is set when a SYSREF event causes a shift in the phase
of the internal frame or LMFC clocks.
Note:
•
Write a 1 to REALIGN to clear the bit field to a 0 state.
•
SYSREF events that do not cause a frame or LMFC clock
phase adjustment do not set this register bit.
•
If CLK_RDY becomes low, this bit is cleared.
3
ALIGN
R/W
N/A
SYSREF alignment status
This bit is set when the ADC has processed a SYSREF event
and indicates that the local frame and multi-frame clocks are
now based on a SYSREF event.
Note:
•
Write a 1 to ALIGN to clear the bit field to a 0 state.
•
Rising-edge SYSREF event sets ALIGN bit.
•
If CLK_RDY becomes low, this bit is cleared.
2
PLL_LOCK
R
N/A
PLL lock status. This bit is set when the PLL has achieved lock.
0 : PLL unlocked
1 : PLL locked
1
CAL_DONE
R
N/A
ADC calibration status
This bit is set when the ADC calibration is complete.
0 : Calibration currently in progress or not yet completed
1 : Calibration complete
Note:
•
Calibration must complete before SYSREF detection
(SYS_EN) can be enabled.
0
CLK_RDY
R
N/A
Input clock status
This bit is set when the ADC is powered-up and detects an
active clock signal at the CLKIN input.
0 : CLKIN not detected
1 : CLKIN detected
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
9.1.1 Optimizing Converter Performance
9.1.1.1 Internal Noise Sources
The signal to noise ratio of the ADC is limited by the following noise sources inherent to the device: 1)
Quantization Noise, 2) Thermal Noise, 3) Sampling Instant Noise (Aperture Jitter). These sources combine
together to limit the noise performance of the converter as described by Equation 1. Noise sources external to
the ADC can generally be included in this equation as an RMS summation to determine system performance.
æ SNR ADC [dBc] = -20 ´ log ç 10
ç
è
SNRQuantizatoin Noise
20
2
ö æ ÷ + ç 10
÷ çç
ø è
SNR
2
Thermal Noise
20
ö æ
÷ ç ÷ + ç 10
÷ ç
ø è
SNR
Jitter
20
ö
÷
÷
÷
ø
2
(1)
Quantization noise is independent of the input signal but does not affect the noise performance of the
ADC31JB68 because the SNR limitation of a 16-bit ADC due to quantization noise is 96-dB, well above the SNR
of this device. The thermal noise is input independent, spread evenly across the spectrum, and the main
limitation for signals with lower frequencies or lower amplitudes.
If quantization and thermal noise is ignored, the fundamental SNR limitation due to aperture jitter can be
calculated using Equation 2.
SNRJitter[dBc] = –20 × log(2π × ƒin × Tjitter)
(2)
Signals with larger amplitudes and higher frequencies introduce signal dependent sampling instant noise due to
the Jitter on the sampling clock edge. This noise includes a broadband component that raises the overall noise
spectral density as well as a in-close noise component that is spectrally shaped and concentrates around the
signal in the spectrum. The differential clock receiver of the ADC31JB68 has a very-low noise floor and wide
bandwidth. Minimizing the aperture jitter requires a sampling clock with a steep edge rate at the zero crossing.
Lesser edge rates increase the aperture jitter as demonstrated in Figure 58 which shows the SNR limitation due
to aperture jitter as a function of clock edge range and input signal frequency.
72
71
70
69
SNR [dBc]
68
67
66
65
64
63
10 MHz
70 MHz
170 MHz
350 MHz
62
61
60
0.1
0.2
0.3
0.5 0.7 1
2
3 4 5 6 7 8 10
CLKIN Zero Crossing Edge Rate [V/ns]
Figure 58. Aperture Jitter vs Input Clock Edge Rate, -1 dBFS Input Signal
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Application Information (continued)
9.1.1.2 External Noise Sources
Noise sources external to the ADC device that impact the overall system performance through the ADC include:
1) Analog Input Signal Path noise, 2) Sampling Clock Signal Path Noise, 3) Supply Noise
Analog input signal path noise comes from devices in the signal path prior to the ADC and enters through the
analog input (VIN+/-). An anti-aliasing filter must be included in front of the ADC to limit the noise bandwidth and
prevent the aliasing of noise into any band of interest. Sampling clock signal path noise enters through the
sampling clock input (CLKIN+/-) and adds noise similar to the Aperture Jitter. External clock jitter can be
minimized by using high quality clock sources and jitter cleaners (PLLs) as well as bandpass filters at the clock
input.
The total clock jitter (TJitter), including the aperture jitter of the ADC and external clock noise can be calculated
with Equation 3 and applied to Equation 1 and Equation 2 to determine the system impact. Figure 59 shows the
simulated impact on the SNR of the ADC31JB68 output spectrum for a given total jitter and input signal
frequency.
TJitter =
2
2
(TJitter,Ext.Clock _ Input ) + (TAperture _ ADC )
(3)
74
72
70
SNR [dBc]
68
66
2ps
64
1ps
62
500fs
60
200fs
58
100fs
50fs
56
54
1
10
100
Frequency [MHz]
1000
C001
Figure 59. SNR Limit Due to Total Jitter of Sampling Clock With a -1 dBFS Input Signal
Additional noise may couple to the clock path through power supplies. Take care to provide a very-low noise
power supply and isolated supply return path to minimize noise added to the supply. Spurious noise added to the
clock path results in symmetrical, modulated spurs around large input signals. These spurs have a constant
magnitude in units of dB relative to the input signal amplitude or carrier, [dBc].
9.1.2 Analog Input Considerations
9.1.2.1 Differential Analog Inputs and Full Scale Range
The ADC31JB68 device has one channel with a pair of analog signal input pins: VIN+, VIN−. VIN, the input
differential signal for a channel, is defined as VIN = (VIN+) – (VIN−). Table 26 shows the expected input signal
range when the differential signal swings about the input common mode voltage, VCM. The full-scale differential
peak-to-peak input range is equal to twice the internal reference voltage, VREF. Nominally, the full scale range is
1.7 Vpp-diff, therefore the maximum peak-to-peak single-ended voltage is 0.85 Vpp at each of the VIN+ and
VIN− pins.
The single-ended signals must be opposite in polarity relative to the VCM voltage to provide a purely differential
signal, otherwise the common-mode component may be rejected by the ADC input. Table 26 indicates the input
to output relationship of the ADC31JB68 device where VREF = 0.85 V.
Differential signals with amplitude or phase imbalances result in lower system performance compared to perfectly
balanced signals. Imbalances in signal path circuits lead to differential-to-common-mode signal conversion and
differential signal amplitude loss as shown in Figure 60. This deviation or imbalance directly causes a reduction
in the signal amplitude and may also lead to distortion, particularly even order harmonic distortion, as the signal
propagates through the signal path. The Amplitude and Phase Imbalance Correction feature in the ADC31JB68
helps to correct amplitude or phase errors of the signal.
46
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Application Information (continued)
Table 26. Mapping of the Analog Input Full Scale Range to Digital Codes
VIN+
VIN–
2s Complement Output
Binary Output
Note
VCM + VREF / 2
VCM – VREF / 2
0111 1111 1111 1111
1111 1111 1111 1111
Positive full-scale
VCM + VREF / 4
VCM – VREF / 4
0100 0000 0000 0000
1100 0000 0000 0000
VCM
VCM
0000 0000 0000 0000
1000 0000 0000 0000
VCM – VREF / 4
VCM + VREF / 4
1100 0000 0000 0000
0100 0000 0000 0000
VCM – VREF / 2
VCM + VREF / 2
1000 0000 0000 0000
0000 0000 0000 0000
Single-Ended
Differential Mode
Mid-scale
Negative full-scale
Common Mode
Ideal
VDD
VCM
VCM
0
GND
VIN-DIFF
VIN-
VIN-CM
0
VIN+
Phase
Imbalance
Amplitude
Imbalance
GND
0
Figure 60. Differential Signal Waveform and Signal Imbalance
9.1.2.2 Analog Input Network Model
Matching the impedance of the driving circuit to the input impedance of the ADC can be important for a flat gain
response through the network across frequency. In very broadband applications or lowpass applications, the
ADC driving network must have very low impedance with a small termination resistor at the ADC input to
maximize the bandwidth and minimize the bandwidth limitation posed by the capacitive load of the ADC input. In
bandpass applications, a designer may either design the anti-aliasing filter to match to the complex impedance of
the ADC input at the desired intermediate frequency, or consider the resistive part of the ADC input to be part of
the resistive termination of the filter and the capacitive part of the ADC input to be part of the filter itself. The
capacitive load of the ADC input can be absorbed into most LC bandpass filter designs with a final shunt LC tank
stage.
The analog input circuit of the ADC31JB68 device is a buffered input with an internal differential termination.
Compared to an ADC with a switched-capacitor input sampling network that has a time-varying input impedance,
the ADC31JB68 device provides a time-constant input impedance that simplifies the interface design joining the
ADC and ADC driver. A simplified passive model of the ADC input network is shown in Figure 61 that includes
the termination resistance, input capacitance, parasitic bond-wire inductance, and routing parastics.
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5:
0.960nH
+
²
VIN1.90:
2:
1nH
5:
+
²
5.2pF
1:
100pF
0.1uF
0.5nH
10uF
1.0nH
VCM
5.2pF
2.33pF
4.62pF
4.62pF
50.0:
0.960nH
14.0:
92.0:
Mutual L
K=0.42
50.0:
92.0:
1.90:
0.80pF
VIN+
Estimated External
Components
Figure 61. Simplified Analog Input Network Circuit Model
A more accurate load model is described by the measured differential SDD11 (100-Ω) parameter model. A plot of
the differential impedance derived from the model is shown on the Smith chart of Figure 62.
Figure 62. Differential 1-Port S-Parameter Measurement of the ADC Input (Sdd11, Z0=100 Ω)
9.1.2.3 Input Bandwidth
The input bandwidth response is shown in Figure 63 and depends on the measurement network. The impedance
of the driving source or any series resistance in the driving network greatly impacts the measured bandwidth.
Three separate measurement methodologies are shown here. The first, shown in Figure 64, is the network used
to measure the reported bandwidth in the performance tables. It uses a 50-Ω source, high bandwidth balun, and
custom input network.
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Another measurement using the simplified network of Figure 65 demonstrates the bandwidth of the ADC using a
low impedance source near the ADC pins. The peaking in the frequency response is caused by the resonance
between the package bond wires and input capacitance as well as a parasitic 0.5-nH series trace inductance
leading to the device pins. This peaking is typically made insignificant by the anti-aliasing filter that precedes the
ADC input.
Normalized Magnitude [dB]
The third measurement network of Figure 65 also assumes a low impedance voltage source but shows the
bandwidth flattening impact of adding 10-Ω in series with the VIN+ and VIN– input pins.
10
9
8
7
6
5
4
3
2
1
0
-1
-2
-3
-4
-5
-6
10
Low-: Source
Low-: Source, Series 10: R
50: Source, Custom Network
2.2GHz
1.3GHz
1.7GHz
20 30
50 70 100 200 300 500
Frequency [MHz]
1000 2000
5000
Figure 63. Measured Input Voltage Frequency Response
33:
-6dB, 50:
15:
-3dB, 50:
ADC
50:
P(f)
15:
Picosecond Labs
Matched Balun
5310
-6dB, 50:
33:
Figure 64. 50-Ω Source, Bandwidth Measurement Network
33:
33:
ADC
V(f)
10:
ADC
V(f)
33:
33:
10:
Figure 65. Low-Ω Source Bandwidth Measurement
(Simplified)
Figure 66. Low-Ω Source, Series 10Ω R, Bandwidth
Measurement (Simplified)
9.1.2.4 Driving the Analog Input
The analog input may be driven by a number of network types depending on the end application. The most
important design aspects to consider when designing the ADC voltage driver network are signal coupling,
impedance matching, differential signal balance, anti-alias filtering, and signal level.
An analog signal is AC or DC coupled to the ADC depending on whether signal frequencies near DC must be
sampled without attenuation. DC coupling requires tight control of the output common-mode of the ADC driver to
match the input common-mode of the ADC input. In the case of DC coupling, the bias at the VCM pin may be
used as a reference to set the driver output common-mode, but the load cannot source or sink more current than
the specified value in the electrical parameters. AC coupling does not require strict common-mode control of the
driver and is typically achieved using AC coupling capacitors or a flux-coupled transformer. AC coupling
capacitors should be chosen to have 0.1-Ω impedance or less over the frequency band of interest. LC filter
designs may be customized to achieve either AC or DC coupling.
The internal input network of the ADC31JB68 device has the common-mode voltage bias provided through
internal shunt termination resistors. The recommended bias for the external termination resistors is the commonmode reference voltage from the VCM pin.
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Impedance matching in high speed signal paths using an ADC is dictated by the characteristic impedance of
interconnects and by the design of anti-aliasing filters. Matching the source to the load termination is critical to
ensure maximum power transfer to the load and to maintain gain flatness across the desired frequency band. In
applications with signal transmission lengths greater than 10% of the smallest signal wavelength (0.1 λ),
matching is also desirable to avoid signal reflections and other transmission line effects. Applications that require
high order anti-aliasing filters designs, including LC bandpass filters, require an expected source and load
termination to ensure the passband bandwidth and ripple of the filter design. The recommended range of the
ADC total load termination is from 50- to 200-Ω differential. The ADC31JB68 device has an internal differential
load termination, but additional termination resistance may be added at the ADC input pins to adjust the total
termination. The load termination at the ADC input presents a system-level design tradeoff. Better 2nd order
distortion performance (HD2, IMD2) is achieved by the ADC using a lower load termination resistance, but the
ADC driver must have a higher drive strength and linearity to drive the lower impedance. Choosing a 100-Ω total
load termination is a reasonable balance between these opposing requirements.
Differential signal balance is important to achieve high distortion performance, particularly even order distortion
(HD2, HD4). Circuits such as transformers and filters in the signal path between the signal source and ADC can
disrupt the amplitude and phase balance of the differential signal before reaching the ADC input due to
component tolerances or parasitic mismatches between the two parallel paths of the differential signal. Driving
the ADC31JB68 device with a single-ended signal is not supported due to the tight restriction on the ADC input
common-mode to maintain good distortion performance.
Converting a single-ended signal to a differential signal may be performed by an ADC driver or transformer. The
advantages of the ADC driver over a transformer include configurable gain, isolation from previous stages of
analog signal processing, and superior differential signal balancing. The advantages of using a transformer
include no additional power consumption and little additional noise or distortion.
Z0 = 50 :
MABA007159
VIN+
0.01uF
33 :
ADC
0.01uF
33 :
VINVCM
MABA007159
10 :
0.1uF
0.1uF
10uF
10uF
Figure 67. Transformer Input Network
Figure 67 is an example of driving the ADC input with a cascaded transformer configuration. The cascaded
transformer configuration provides a high degree of differential signal balancing, the series 0.01-µF capacitors
provide AC coupling, and the additional 33-Ω termination resistors provide a total differential load termination of
50 Ω. When additional termination resistors are added to change the ADC load termination, shunt terminations to
the VCM reference are recommended to reduce common-mode fluctuations or sources of common-mode
interference. A differential termination may be used if these sources of common-mode interference are minimal.
In either case, the additional termination components must be placed as close to the ADC pins as possible. The
MABA007159 transmission-line transformer from this example is widely available and results in good differential
balance. Shunt capacitors at the ADC input, used to suppress the charge kickback of an ADC with switchedcapacitor inputs, are not required for this purpose because the buffered input of the ADC31JB68 device does not
kick back a significant amount of charge.
The insertion loss between an ADC driver and the ADC input is important because the driver must overcome the
insertion loss of the connecting network to drive the ADC to full-scale and achieve the best SNR. Minimizing the
loss through the network reduces the output swing and distortion requirements of the driver and usually
translates to a system-level power savings in the driver. This can be accomplished by selecting transformers or
filter designs with low insertion loss. Some filter designs may employ reduced source terminations or impedance
conversions to minimize loss. Many designs require the use of high-Q inductors and capacitors to achieve an
expected passband flatness and profile.
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Sampling theory states that if a signal with frequency ƒIN is sampled at a rate less than 2 × ƒIN, then it
experiences aliasing, causing the signal to fall at a new frequency between 0 and FS / 2 in the sampled spectrum
and become indistinguishable from other signals at that new frequency.
To prevent out-of-band interference from aliasing onto a desired signal at a particular frequency, an anti-aliasing
filter is required at the ADC input to attenuate the interference to a level below the level of the desired signal.
This is accomplished by a lowpass filter in systems with desired signals from DC to FS / 2 or with a bandpass
filter in systems with desired signals greater than FS / 2 (under-sampled signals). If an appropriate anti-aliasing
filter is not included in the system design, the system may suffer from reduced dynamic range due to additional
noise and distortion that aliases into the frequency bandwidth of interest.
LC BPF
Vcc+
Rs
VIN+
0.1uF
RL
ADC
Driver
VIN-
VCM
10 :
0.1uF
0.1uF
10uF
10uF
Figure 68. Bandpass Filter Anti-Aliasing Interface
An anti-aliasing filter is required in front of the ADC input in most applications to attenuate noise and distortion at
frequencies that alias into any important frequency band of interest during the sampling process. An anti-aliasing
filter is typically a LC lowpass or bandpass filter with low insertion loss. The bandwidth of the filter is typically
designed to be less than FS / 2 to allow room for the filter transition bands. Figure 68 is an example architecture
of a 9 pole order LC bandpass anti-aliasing filter with added transmission zeros that can achieve a tight filtering
profile for second Nyquist zone under-sampling applications.
VCC+
LC LPF
RS
VIN+
RL
ADC
Driver
VCMO = VCMA
0.1uF
VCMI
VINVCM
10:
VCC-
0.1uF
0.1uF
10uF
10uF
Figure 69. DC Coupled Interface
DC coupling to the analog input is also possible but the input common-mode must be tightly controlled for
specified performance. The driver device must have an output common-mode that matches the input commonmode of the ADC31JB68 device and the driver must track the VCM output from the ADC31JB68 device, as
shown in the example DC coupled interface of Figure 69, because the input common-mode varies with
temperature. The common-mode path from the VCM output, through the driver device, back to the ADC31JB68
device input, and through a common-mode detector inside the ADC31JB68 device forms a closed tracking loop
that will correct common-mode offset contributed by the driver device but the loop must be stable to ensure
correct performance. The loop requires the large, 10-µF capacitor at the VCM output to establish the dominant
pole for stability and the driver device must reliably track the VCM output voltage bias.
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9.1.2.5 Clipping and Over-Range
The ADC31JB68 device has two regions of signal clipping: code clipping (over-range) and ESD clipping. When
the input signal amplitude exceeds the full-scale reference range, code clipping occurs during which the digital
output codes saturate. If the signal amplitude increases beyond the absolute maximum rating of the analog
inputs, ESD clipping occurs due to the activation of ESD diodes.
Code clipping may be monitored via the SDO/OVR pin, which can be configured to output an quick detect overrange signal. The over-range threshold is programmable via the SPI. An over-range hold feature is also available
to extend the time duration of the indicator longer than the over-range event itself to accommodate the case that
a device monitoring the over-range signal cannot process at the rate of the ADC sampling clock.
ESD clipping and activation of the ESD diodes at the analog input should be avoided to prevent damage or
shortened life of the device. This clipping may be avoided by selecting an ADC driver with an appropriate
saturating output voltage, by placing insertion loss between the driver and ADC, by limiting the maximum
amplitude earlier in the signal path at the system level, or by using a dedicated differential signal limiting device
such as a clamp. Any signal swing limiting device must be chosen carefully to prevent added distortion to the
signal.
9.1.3 CLKIN, SYSREF, and SYNCb Input Considerations
Clocking the ADC31JB68 device shares many common concepts and system design requirements with
previously released ADC products that include a JESD204B interface. A SYSREF signal accompanies the device
clock to provide phase alignment information for the output data serializer (as well as for the sampling instant
when the clock divider is enabled) to ensure that the latency through the JESD204B link is always deterministic,
a concept called deterministic latency. To ensure deterministic latency, the SYSREF signal must meet setup and
hold requirements relative to CLKIN and the design of the clocking interfaces require close attention. As with
other ADCs, the quality of the clock signal also influences the noise and spurious performance of the device.
9.1.3.1 Driving the CLKIN+ and CLKIN– Input
The CLKIN input circuit is composed of a differential receiver and an internal 100-Ω termination to a weakly
driven common-mode of 0.5 V. TI recommends AC coupling to the CLKIN input with 0.1-uF external capacitors to
maintain the optimal common-mode biasing. Figure 70 shows the CLKIN receiver circuit and an example AC
coupled interface.
CLKIN
Receiver
0.1uF
100:-diff
50:
Transmitter
50:
PCB Channel
VIS = 0.5V
10k:
Figure 70. Driving the CLKIN Input With an AC Coupled Interface
DC coupling is allowed as long as the input common-mode range requirements are satisfied. The input commonmode of the CLKIN input is not compatible with many common signaling standards like LVDS and LVPECL.
Therefore, the CLKIN signal driver common-mode must be customized at the transmitter or adjusted along the
interface. Figure 71 shows an example DC coupled interface that uses a resistor divider network to reduce the
common-mode while maintaining a 100-Ω total termination at the load. Design equations are provided with
example values to determine the resistor values.
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VCMO = 1.2V
CLKIN
Receiver
R2
R1
100:-diff
50:
Transmitter
50:
PCB Channel
VIS = 0.5V
10k:
4*R1*R2 + 200*R1 = 1002
R2 / (R1+R2) = 0.5 / VCMO
IDC (Each Side)= VCMO / (R1+R2)
VCMO = 1.2V: R1 = 29.1:, R2 = 20.8: IDC = 24mA
Figure 71. Driving the CLKIN Input With an Example DC Coupled Interface
The CLKIN input supports any type of standard signaling that meets the input signal swing and common-mode
range requirements with an appropriate interface. Generic differential sinusoidal or square-wave clock signals are
also supported. TI does not recommend driving the CLKIN input single-ended. The differential lane trace on the
PCB should be designed to be a controlled 100 Ω and protected from noise sources or other interfering signals.
9.1.3.2 Driving the SYSREF Input
The SYSREF input interface circuit is composed of the differential receiver, internal termination, internal
common-mode bias with common-mode control, and SYSREF detection feature.
A 100-Ω differential termination is provided inside the ADC pins. A high impedance reference biases the input
common-mode through a resistor network that can be configured to support a wide range of input common
modes voltages. Following the receiver, an AND gate provides a method for detecting or ignoring incoming
SYSREF events.
The timing relationship between the CLKIN and SYSREF signal is very stringent in a JESD204B system.
Therefore, the signal path network of the CLKIN and SYSREF signals must be as similar as possible to ensure
that the signal relationship is maintained from the launch of the signal, through their respective channels to the
CLKIN and SYSREF input receivers.
DC coupling of the SYSREF signal to the input pins with the simple interface shown in Figure 72 is required.
Although the input common-mode range of the receiver itself is limited, a wide input common-mode range is
supported using the common-mode control feature which level-shifts the common-mode of the input signal to an
appropriate level for the input receiver. The common-mode control feature is configured via the SPI.
SYSREF
Receiver
CM
Control
100:-diff
Transmitter
PCB Channel
50:
+
-
SYSREF
Detection
Feature
0.5V
Figure 72. SYSREF Input Receiver and DC Coupled Interface
9.1.3.3 SYSREF Signaling
The SYSREF input may be driven by a number of different types of signals. The supported signal types, shown
in Figure 73 (in single-ended form), include periodic, gapped periodic, and one-shot signals. The rising edge of
the SYSREF signal is used as a reference to align the internal frame clock and local multi-frame clock (LMFC).
To ensure proper alignment of these system clocks, the SYSREF signal must be generated along with the CLKIN
signal such that the SYSREF rising edge meets the setup and hold requirements relative to the CLKIN at the
ADC31JB68 device inputs.
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For each rising clock edge that is detected at the SYSREF input, the ADC31JB68 device compares the current
alignment of the internal frame and LMFC with the SYSREF edge and determines if the internal clocks must be
re-aligned. In the case that no alignment is needed, the clocks maintain their current alignment and the
JESD204B data link is not broken. In the case that re-alignment is needed, the JESD204B data link is broken
and the clocks are re-aligned.
Periodic
TSYSREF = n*K*TFRAME
(n=1,2,3...)
Gapped-Periodic
• 2*TFRAME
One-Shot
• 2*TFRAME
Figure 73. SYSREF Signal Types (Single-Ended Representations)
In the case of a periodic SYSREF signal, the frame and LMFC alignment is established at the first rising edge of
SYSREF, and every subsequent rising edge (that properly meets setup and hold requirements) is ignored
because the alignment has already been established. A periodic SYSREF must have a period equal to n × K / FS
where ‘FS’ is the sampling rate, ‘K’ is the JESD204B configuration parameter indicating the number of frames per
multi-frame, and ‘n’ is an integer of one or greater.
Gapped-period signals contain bursts of pulses. The frame and LMFC alignments are established on the first
rising edge of the pulse burst. Any rising edge that does not abide by this rule or does not meet the setup and
hold requirements forces re-alignment of the clocks.
A one-shot signal contains a single rising edge that establishes the frame and LMFC alignment.
For all types of SYSREF signals, the minimum pulse width is 2 × TFRAME.
TI recommends gapped-periodic or one-shot signals for most applications because the SYSREF signal is not
active during normal sampling operation. Periodic signals that toggle constantly introduce spurs into the signal
spectrum that degrade the dynamic range of the system.
9.1.3.4 SYSREF Timing
The SYSREF timing requirements depend on whether deterministic latency of the JESD204B link is required.
If deterministic latency is required, then the SYSREF signal must meet setup and hold requirements relative to
the CLKIN signal. In the case that the internal CLKIN divider is used and a very high-speed signal is provided to
the CLKIN input, the SYSREF signal must meet setup and hold requirements relative to the very high-speed
signal at the CLKIN input.
If deterministic latency is not required, then the SYSREF signal may be supplied as an asynchronous signal
(possibly achieving < ± 2 frame clock cycles latency variation) or not provided at all (resulting in latency variation
as large as the multi-frame period).
9.1.3.5 Effectively Using the Detection Gate Feature
TI recommends the use of the SYSREF detection gate for most applications. The gate is enabled when SYSREF
is being transmitted and the gate is disabled before the SYSREF transmitter is put in the idle state.
Enabling the SYSREF gate immediately sends a logic signal to a logic block responsible for aligning the internal
frame clock and LMFC. If the signal at the SYSREF input is logic high when the gate is enabled, then a "false"
rising edge event causes a re-alignment of the internal clocks, despite the fact that the event is not an actual
SYSREF rising edge. The SYSREF rising edge following the gate enable then causes a subsequent re-alignment
with the desired alignment.
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9.1.3.6 Driving the SYNCb Input
The SYNCb input is part of the JESD204B interface and is used to send synchronization requests from the serial
data receiver to the transmitter. The SYNCb signal, quantified as the (SYNCb+ – SYNCb–), is a differential active
low signal. In the case of the ADC31JB68 device, a JESD204B subclass 1 device, a SYNCb assertion (logic low)
indicates a request for synchronization by the receiver. The SYNCb input is an asynchronous input and does not
have sub-clock-cycle setup and hold requirements relative to the CLKIN or any other input to the ADC31JB68
device.
The SYNCb input is a differential receiver as shown in Figure 74. Resistors provide an internal 100-Ω differential
termination as well as a voltage divider circuit that gives the SYNCb receiver a wide input common-mode range.
The SYNCb signal must be DC coupled from the driver to the SYNCb inputs; therefore, the wide common-mode
range allows the use of many different logic standards including LVDS and LVPECL. No additional external
components are needed for the SYNCb signal path as shown in the interface circuit of Figure 74, but providing
an electrical probing site is recommended for system debug.
1.8V
1.5k:
1.5k:
SYNCb
Receiver
2k:
100:-diff
2k:
50:
Transmitter
PCB Channel
50:
34k:
34k:
3pF
Figure 74. SYNCb Input Receiver and Interface
9.1.4 Output Serial Interface Considerations
9.1.4.1 Output Serial-Lane Interface
The output high speed serial lanes must be AC coupled to the receiving device with 0.01-µF capacitors as shown
in Figure 75. DC coupling to the receiving device is not supported. The lane channel on the PCB must be a
100-Ω differential transmission line with dominant coupling recommended between the differential traces instead
of to adjacent layers. The lane must terminate at a 100-Ω termination inside the receiving device. Avoid changing
the direction of the channel traces abruptly at angles larger than 45°.
0.01uF
100:-diff
Serial Lane
Transmitter
PCB Channel
100:
Serial Lane
Receiver
Figure 75. High-Speed Serial-Lane Interface
The recommended spacing between serial lanes is 3× the differential line spacing or greater. High speed serial
lanes should be routed on top of or below adjacent, quiet ground planes to provide shielding. TI recommends
that other high speed signal traces do not cross the serial lanes on adjacent PCB layers. If absolutely necessary,
crossing should occur at a 90° angle with the trajectory of the serial lane to minimize coupling.
The integrity of the data transfer from the transmitter to receiver is limited by the accuracy of the lane impedance
and the attenuation as the signal travels down the lane. Inaccurate or varying impedance and frequency
dependent attenuation results in increased ISI (part of deterministic jitter) and reduced signal-to-noise ratio,
which limits the ability of the receiver to accurately recover the data.
Two features are provided in the ADC31JB68 device serial transmitters to compensate attenuation and ISI
caused by the serial lane: voltage swing control (VOD) and de-emphasis (DEM).
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9.1.4.2 Voltage Swing and De-Emphasis Optimization
Voltage swing control (VOD) compensates for attenuation across all frequencies through the channel at the
expense of power consumption. Increasing the voltage swing increases the power consumption. De-emphasis
(DEM) compensates for the frequency dependent attenuation of the channel but results in attenuation at lower
frequencies. The voltage swing control and de-emphasis feature may be used together to optimally compensate
for attenuation effects of the channel.
The frequency response of the PCB channel is typically lowpass with more attenuation occurring at higher
frequencies. The de-emphasis implemented in the ADC31JB68 device is a form of linear, continuous-time
equalization that shapes the signal at the transmitter into a high-pass response to counteract the low-pass
response of the channel. The de-emphasis setting should be selected such that the equalizer’s frequency
response is the inverse of the channel’s response. Therefore, transferring data at the highest speeds over long
channel lengths requires knowledge of the channel characteristics.
Optimization of the de-emphasis and voltage swing settings is only necessary if the ISI and losses caused by the
channel are too great for reception at the desired bit rate. Many applications will perform with an adequate BER
using the default settings.
9.1.4.3 Minimizing EMI
High data-transfer rates have the potential to emit radiation. EMI may be minimized using the following
techniques:
• Use differential stripline channels on inner layer sandwiched between ground layers instead of routing
microstrip pairs on the top layer.
• Avoid routing lanes near the edges of boards.
• Enable data scrambling to spread the frequency content of the transmitted data.
• If the serial lane must travel through an interconnect, choose a connector with good differential pair channels
and shielding.
• Ensure lanes are designed with an accurate, 100-Ω characteristic impedance and provide accurate 100-Ω
terminations inside the receiving device.
9.1.5 JESD204B System Considerations
9.1.5.1 Frame and LMFC Clock Alignment Procedure
Frame and LMFC clocks are generated inside the ADC31JB68 device and are used to properly align the phase
of the serial data leaving the device. The phases of the frame and multi-frame clocks are determined by the
frame alignment step for JESD204B link initialization as shown in Figure 35. These clocks are not accessible
outside the device. The frequencies of the frame and LMFC must be equal to the frame and LMFC of the device
receiving the serial data.
When the ADC31JB68 device is powered-up, the internal frame and local multi-frame clocks initially assume a
default phase alignment. To ensure determinist latency through the JESD204B link, the frame and LMFC clocks
of the ADC31JB68 device must be aligned in the system. Perform the following steps to align the device clocks:
1. Enable the SYSREF signal driver. See SYSREF Signaling for more information.
2. Enable detection of the SYSREF signal at the ADC31JB68 device by enabling the SYSREF detection gate.
See Effectively Using the Detection Gate Feature for more information.
3. Apply the desired SYSREF signal at the ADC31JB68 device SYSREF input.
4. Disable detection of the SYSREF signal by disabling the SYSREF gate.
5. Configure the SYSREF driver into its idle state.
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9.1.5.2 Link Interruption
The internal frame and multi-frame clocks must be stable to maintain the JESD204B link. The ADC31JB68 is
designed to maintain the JESD204B link in most conditions but some features interrupt the internal clocks and
break the link.
The following actions cause a break in the JESD204B link:
• The ADC31JB68 device is configured into power-down mode or sleep mode
• The ADC31JB68 device CLKIN clock divider setting is changed
• The serial data receiver performs a synchronization request
• The ADC31JB68 device detects a SYSREF assertion that is not aligned with the internal frame or multi-frame
clocks
• The CLKIN input is interrupted
• Power to the device is interrupted
The following actions do not cause a change in clock alignment nor break the JESD204B link:
• The ambient temperature or operating voltages are varied across the ranges specified in the normal operating
conditions.
• The ADC31JB68 device detects a SYSREF assertion that is aligned with the internal frame and multi-frame
clocks.
9.1.5.3 Clock Configuration Examples
The features provided in the ADC31JB68 device allow for a number of clock and JESD204B link configurations.
These examples in Table 27 show some common implementations and may be used as a starting point for a
more customized implementation.
Table 27. Example ADC31JB68 Clock Configurations
Parameter
Example 1
Example 2
Example 3
CLKIN frequency
500 MHz
1000 MHz
1966.08 MHz
CLKIN divider
1
2
4
Sampling rate
500 MSPS
500 MSPS
491.52 MSPS
K (Frames per multiframe)
20
32
32
LMFC Frequency
25 MHz
15.625 MHz
15.36 MHz
SYSREF
Frequency (1)
25 MHz
11.5625 MHz
7.68 MHz (1)
Serial bit rate for each
lane
5.0 Gb/s
5.0 Gb/s
4.9152 Gb/s
(1)
The SYSREF frequency for a continuous SYSREF signal can be the indicated frequency ƒLMFC or integer sub-harmonic such as ƒLMFC /
2, ƒLMFC / 3, and so forth. Gapped-periodic SYSREF signals should have pulses spaced by the associated periods 1 / ƒLMFC, 2 / ƒLMFC,
3 / ƒLMFC, and so forth.
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9.1.6 SPI
Figure 76 demonstrates a typical circuit to interface the ADC31JB68 device to a SPI master using a shared SPI
bus. The 4-wire interface (SCLK, SDI, SDO, CSb) is compatible with 1.2-, 1.8-, or 3.0-V logic. The input pins
(SCLK, SDI, CSb) use thick-oxide devices to tolerate 3.0-V logic although the input threshold levels are relative
to 1.2-V logic. A low-capacitance protection diode may also be added with the anode connected to the SDO
output and the cathode connected to the desired voltage supply to prevent an accidental pre-configured read
command from causing damage.
1.2V 1.8V 3.0V
Configurable bus access logic level
(Must configure BEFORE read command)
Hi-Z idle state
SDO
SPI Slave 1
(ADC31JB68)
22 :
3.0V tolerant inputs
1.2V logic thresholds
1.2V
SCLK
22 :
VMASTER
SPI
Master
10k:
SDI
CSB1
SDO
22 :
SCLK
SPI Slave 2
SDI
CSB2
Figure 76. Typical SPI Application
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9.2 Typical Application
DOWN-CONVERTING
MIXER
LNA
BPF
BPF
DRIVER
ADC31JB68
(ADC)
BPF
SUPERHETERODYNE
RECEIVER
RF PLL
FPGA /
ASIC
JESD204
SYSTEM
CLOCK
GENERATOR
SYNTH
Figure 77. Typical Application Circuit
VDDA1.8
10 k:
VDDA1.8
VDDA1.2
0.1, 0.1 uF
SPI
Master
VA3.0
VDDA3.0
VA3.0
0.1, 0.1 uF
50 :
0.01 uF
0.01 uF
50 :
AGND
VIN+
AntiAliasing
Filter
VINAGND
100 :
Driver
10 :
0.1 uF
10 uF
VCM
10 uF
0.1 uF
VACLK1.2
VACLK1.2
VDDA1.2
AGND
CSB
22 :
31
32
SCLK
SDI
33
34
VA1.2
35
SDO/OVR
VA1.2
36
AGND
37
VA1.8
AGND
38
39
40
VA1.8
0.1, 0.1 uF
1
2
30
AGND
29
VA1.2
3
28
4
27
5
26
ADC31JB68
6
25
7
24
8
23
9
22
10
21
0.1, 10 uF
VDDA1.2
VA1.8
VDDA1.8
0.1, 10 uF
AGND
100 : Differential
SO0SO0+
0.01 uF
100 :
0.01 uF
100 :
SO1SO1+
AGND
VDDA1.2
VA1.2
JESD204
Clock
Generator
19
20
SYNCb-
SYNCb+
18
SYSREF-
17
0.1, 0.1 uF
100 : Differential
0.1 uF
VDDA1.8
JESD204
Receiver
100 :
16
SYSREF+
14
15
VA1.8
VA1.8
AGND
13
AGND
12
CLKIN0.1 uF
CLKIN+
11
0.1, 0.1 uF
0.1, 0.1 uF
100 : Differential
Trace Matched
100 :
100 :
Figure 78. Typical Circuit Implementation
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Typical Application (continued)
9.2.1 Design Requirements
The following are example design requirements expected of the ADC in a typical high-IF, 200-MHz bandwidth
receiver, and are met by the ADC31JB68 device:
Table 28. Example Design Requirements for a High-IF Application
Example Design Requirement (1)
Specification
ADC31JB68 Capability
Sampling rate
> 450 MSPS to allow 200 MHz unaliased
bandwidth
Up to 500 MSPS
Input bandwidth
> 500-MHz, 1 dB flatness
1000 MHz, 1 dB Bandwidth
Full-scale range
< 2 Vpp-diff
1.7 Vpp-diff
Small signal noise spectral density
< –152 dBFS/Hz
–154.5 dBFS/Hz
Large-signal SNR
> 69 dBFS for a –1 dBFS, 210 MHz Input
69.7 dBFS for a –1 dBFS, 210 MHz Input
SFDR
> 75 dBFS for a –1 dBFS, 210 MHz input
79 dBFS for a –1 dBFS, 210 MHz input
HD2, HD3
< –75 dBFS for a –1 dBFS, 210 MHz input
–79 dBFS for a –1 dBFS, 210 MHz input
Next largest SPUR
< –88 dBFS for a –1 dBFS, 210 MHz input
–90 dBFS for a –1 dBFS, 210 MHz input
Over-range detection
Included
Fast over-range detection on SDO/OVR pin
Digital interface
JESD204B interface, 2 lane/channel
JESD204B subclass 1 interface, 2 lane/channel,
5.0 Gb/s bit rate
Configuration interface
SPI configuration, 4-wire, 1.8 V logic, SCLK >
1 MHz
SPI configuration, 4-Wire, 1.8 V Logic, SCLK up to
20 MHz
Package size
< 10 × 10 × 1 mm
6 × 6 × 0.8 mm
(1)
These example design requirements do not represent the capabilities of the ADC31JB68, rather the requirements are satisfied by the
ADC31JB68.
9.2.2 Design Procedure
The following procedure can be followed to design the ADC31JB68 device into most applications:
• Choose an appropriate ADC driver and analog input interface.
– Optimize the signal chain gain leading up to the ADC to make use of the full ADC dynamic range.
– Identify whether DC or AC coupling is required.
– Determine the desired analog input interface, such as a bandpass filter or a transformer.
– Use the provided input network models to design and verify the interface.
– Refer to the interface recommendations in Analog Input Considerations.
• Determine the core sampling rate of the ADC.
– Must satisfy the bandwidth requirements of the application .
– Must also provide enough margin to prevent aliasing or to accommodate the transitions bands of an antialiasing filter.
– Ensure the application initialization sequence properly handles ADC core calibration as described in ADC
Core Calibration.
• Determine the system latency requirements.
– Total allowable latency through the ADC and JESD204B link.
– Is the system tolerant of latency variation over time or conditions or between power cycles?
• Determine the desired JESD204B link configuration as discussed in JESD204B Supported Features.
– Based on the system latency requirements, determine whether deterministic latency is required across the
JESD204B link.
– Choose the number of frames per multi-frame, K.
– Choose whether scrambling is desired.
• Choose an appropriate clock generator, CLKIN interface, and SYSREF interface.
– Determine the system clock distribution scheme and the clock frequencies for the CLKIN and SYSREF
inputs.
– Determine the allowable amount of sampling clock phase noise in the system and then select a CLKIN
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•
•
•
•
•
•
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edge rate that satisfies this requirement as discussed in Internal Noise Sources.
– Choose an appropriate CLKIN interface as discussed in Driving the CLKIN+ and CLKIN– Input.
– Based on the latency requirements, determine whether SYSREF must meet setup and hold requirements
relative to CLKIN.
– Choose the SYSREF signal type as discussed in SYSREF Signaling.
– Choose an appropriate SYSREF interface as discussed in Driving the SYSREF Input.
– Choose a CLKIN and SYSREF clock generator based on the above requirements. The signals must come
from the same generator in many cases.
Design the SYNCb interface as discussed in Driving the SYNCb Input.
Choose appropriate configurations for the output serial data interface.
– Design the serial lane interface according to Output Serial-Lane Interface.
– Choose the required PCB materials, keeping in mind the desired rate of the serial lanes.
– Characterize the signal lane channels that connect the ADC serial output transmitters to the receiving
device either through simulation or bench characterization.
– Optimize the VOD and DEM parameters to achieve the required signal integrity according to Voltage
Swing and De-Emphasis Optimization.
Design the SPI bus interface.
– Verify the electrical and functional compatibility of the ADC SPI with the SPI controller.
– Interface the ADC to the SPI bus according to SPI.
– Ensure that the application initialization sequence properly configures the output SDO voltage before the
first read command.
Design the power supply architecture and de-coupling.
– Choose appropriate power supply and supply filtering devices to provide stable, low-noise supplies as
described in Power Supply Recommendations.
– Design the capacitive de-coupling around the ADC, also described in Power Supply Recommendations,
while paying close attention to placing the capacitors as close to the device as possible.
– Time the power architecture to satisfy the power sequence requirements described in Power Supply
Design.
Ensure that the application initialization sequence satisfies the JESD204B link initialization requirements
described in JESD204B Link Initialization.
Refer to Figure 78 for an example hardware design.
9.2.3 Application Curve
0
-10
-20
Magnitude [dBFS]
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
0
25
50
75
100 125 150 175
Frequency [MHz]
200
225
250
f1 = 200 MHz; f2 = 210 MHz, Input Amplitude = –7 dBFS/Tone
Figure 79. Output Spectrum, 2-Tone Test
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10 Power Supply Recommendations
10.1 Power Supply Design
The ADC31JB68 device is a very-high dynamic range device, and therefore requires very-low noise power
supplies. LDO-type regulators, capacitive decoupling, and series isolation devices like ferrite beads are all
recommended.
LDO-type, low-noise regulators should be used to generate the 1.2-V, 1.8-V, and 3.0-V supplies used by the
device. To improve power efficiency, a switching-type regulator may precede the LDO to efficiently drop a supply
to an intermediate voltage that satisfies the drop-out requirements of the LDO. The recommended supply path
includes a switching-type regulator followed by an LDO to provide an efficient and low noise source. Additional
ferrite beads and LC filters may be used to further suppress noise. Supplying power to multiple devices in a
system from one regulator may result in noise coupling between the multiple devices; therefore, series isolation
devices and additional capacitive decoupling is recommended to improve the isolation. VACLK1.2, a sensitive
supply that powers the internal clock path, can be sourced by the same regulator as the VA1.2 supply, but the
VACLK1.2 supply should be isolated using a ferrite device.
The power supplies must be applied to the ADC31JB68 device in this specific sequence:
1. VA3.0
2. VA1.8 and VA1.2 and VACLK1.2
First, apply the VA3.0 (+3.0 V) to provide the bias for the ESD diodes. Next, within 1 ms after enabling the VA3.0
supply, apply the VA1.8 (+1.8-V) and VA1.2 (+1.2-V) and VACLK1.2 (+1.2-V) supplies. Use the reverse order to
turn off the power supplies off.
10.2 Decoupling
Decoupling capacitors must be used at each supply pin to prevent supply or ground noise from degrading the
dynamic performance of the ADC and to provide the ADC with a well of charge to minimize voltage ripple caused
by current transients. The recommended supply decoupling scheme is to have a ceramic X7R 0201 0.1-μF and a
X7R 0402 0.1-μF capacitor at each supply pin. The 0201 capacitor must be placed on the same layer as the
device as close to the pin as possible to minimize the AC decoupling path length from the supply pin, through the
capacitor, to the nearest adjacent ground pin. The 0402 capacitor should also be close to the pins. TI does not
recommend placing all capacitors on the opposite board side. Each voltage supply should also have a single 10μF decoupling capacitor near the device but the proximity to the supply pins is less critical.
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11 Layout
11.1 Layout Guidelines
The design of the PCB is critical to achieve the full performance of the ADC31JB68 device. Defining the PCB
stackup should be the first step in the board design. Experience has shown that at least 6 layers are required to
adequately route all required signals to and from the device. Each signal routing layer must have an adjacent
solid ground plane to control signal return paths to have minimal loop areas and to achieve controlled
impedances for microstrip and stripline routing. Power planes must also have adjacent solid ground planes to
control supply return paths. Minimizing the spacing between supply and ground planes improves performance by
increasing the distributed decoupling. A recommended stack-up for a 6-layer board design is shown in Figure 80.
Although the ADC31JB68 device consists of both analog and digital circuitry, TI highly recommends solid ground
planes that encompass the device and its input and output signal paths. TI does not recommend split ground
planes that divide the analog and digital portions of the device. Split ground planes may improve performance if a
nearby, noisy, digital device is corrupting the ground reference of the analog signal path. When split ground
planes are employed, one must carefully control the supply return paths and keep the paths on top of their
respective reference planes.
Quality analog input signal and clock signal path layout is required for full dynamic performance. Symmetry of the
differential signal paths and discrete components in the path is mandatory and symmetrical shunt-oriented
components should have a common grounding via. The high frequency requirements of the input and clock
signal paths necessitate using differential routing with controlled impedances and minimizing signal path stubs
(including vias) when possible.
Coupling onto or between the clock and input signal paths must be avoided using any isolation techniques
available including distance isolation, orientation planning to prevent field coupling of components like inductors
and transformers, and providing well coupled reference planes. Via stitching around the clock signal path and the
input analog signal path provides a quiet ground reference for the critical signal paths and reduces noise
coupling onto these paths. Sensitive signal traces must not cross other signal traces or power routing on
adjacent PCB layers, rather a ground plane should separate the traces. If necessary, the traces should cross at
90° angles to minimize crosstalk.
The substrate dielectric materials of the PCB are largely influenced by the speed and length of the high speed
serial lanes. The affordable and common FR4 variety may not offer the consistency or low loss to support the
highest speed transmission (5 Gb/s) and long lengths (> 8 inch). Although the VOD and DEM features are
available to improve the signal integrity of the serial lanes, some of the highest performing applications may still
require special dielectric materials.
Coupling of ambient signals into the signal path is reduced by providing quiet, close reference planes and by
maintaining signal path symmetry to ensure the coupled noise is common-mode. Faraday caging may be used in
very noisy environments and high dynamic range applications to isolate the signal path.
L1 ± SIG
0.007''
L2 ± GND
0.004''
L3 ± PWR/SIG
0.0625''
L4 ± PWR
0.004''
L5 ± GND
0.007''
L6 ± SIG
1 oz. Copper on L2-5, 2 oz. Copper on L1, L6
100 Differential Signaling on SIG Layers
Low loss dielectric adjacent very high speed trace layers
Figure 80. Recommended PCB Layer Stack-Up for a Six-Layer Board
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ADC31JB68
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11.2 Layout Example
Figure 81. Layout
64
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ADC31JB68
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12 Device and Documentation Support
12.1 Related Documentation
For related documentation see the following: ADC31JB68EVM User's Guide
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADC31JB68RTAT
ACTIVE
WQFN
RTA
40
250
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 85
31JB68
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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