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ADS1282
SBAS418I – SEPTEMBER 2007 – REVISED MARCH 2015
ADS1282 High-Resolution Analog-To-Digital Converter
1 Features
3 Description
•
The ADS1282 is an extremely high-performance,
single-chip analog-to-digital converter (ADC) with an
integrated, low-noise programmable gain amplifier
(PGA) and two-channel input multiplexer (mux). The
ADS1282 is suitable for the demanding needs of
energy
exploration
and
seismic
monitoring
environments.
1
•
•
•
•
•
•
•
•
•
•
•
High Resolution:
130-dB SNR (250 SPS)
High Accuracy:
THD: –122 dB
INL: 0.5 ppm
Low-Noise PGA
Two-Channel Input Mux
Inherently-Stable Modulator with Fast Responding
Overrange Detection
Flexible Digital Filter:
Sinc + FIR + IIR (Selectable)
Linear or Minimum Phase Response
Programmable High-Pass Filter
Selectable FIR Data Rates: 250 SPS to 4 kSPS
Filter Bypass Option
Low Power Consumption: 25 mW
Shutdown: 10 μW
Offset and Gain Calibration Engine
SYNC Input
Analog Supply:
Unipolar (+5 V) or Bipolar (±2.5 V)
Digital Supply: 1.8 V to 3.3 V
2 Applications
•
•
•
Energy Exploration
Seismic Monitoring
High-Accuracy Instrumentation
The converter uses a fourth-order, inherently stable,
delta-sigma (ΔΣ) modulator that provides outstanding
noise and linearity performance. The modulator is
used either in conjunction with the on-chip digital
filter, or can be bypassed for use with post
processing filters.
The flexible input MUX provides an additional
external input for measurement, as well as internal
self-test connections. The PGA features outstanding
low noise (5 nV/√Hz) and high input impedance,
allowing easy interfacing to geophones and
hydrophones over a wide range of gains.
The digital filter provides selectable data rates from
250 to 4000 samples per second (SPS). The highpass filter (HPF) features an adjustable corner
frequency. On-chip gain and offset scaling registers
support system calibration.
The synchronization input (SYNC) can be used to
synchronize the conversions of multiple ADS1282s.
The SYNC input also accepts a clock input for
continuous alignment of conversions from an external
source.
Together, the amplifier, modulator, and filter dissipate
25 mW. The ADS1282 is available in a compact
TSSOP-28 package and is fully specified from –40°C
to +85°C, with a maximum operating range to
+125°C.
AVDD
VREFN
VREFP
DVDD
CLK
ADS1282
MUX
Input 1
Input 2
PGA
4th-Order
DS
Modulator
Programmable
Digital Filter
Calibration
SPI
Interface
SCLK
DOUT
DIN
DRDY
Control
SYNC
RESET
PWDN
VCOM
Over-Range
Modulator Output
3
AVSS
DGND
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
ADS1282
SBAS418I – SEPTEMBER 2007 – REVISED MARCH 2015
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
4 Ordering Information
For the most current package and ordering information, see the Package Option Addendum at the end of this
document, or visit the device product folder at ti.com.
5 Specifications
5.1
Absolute Maximum Ratings (1)
Over operating free-air temperature range (unless otherwise noted).
ADS1282, ADS1282H
UNIT
AVDD to AVSS
–0.3 to +5.5
V
AVSS to DGND
–2.8 to +0.3
V
DVDD to DGND
–0.3 to +3.9
V
Input current
100, momentary
mA
Input current
10, continuous
mA
AVSS – 0.3 to AVDD + 0.3
V
Analog input voltage
Digital input voltage to DGND
–0.3 to DVDD + 0.3
V
+150
°C
Operating temperature range
–40 to +125
°C
Storage temperature range
–60 to +150
°C
Maximum junction temperature
(1)
2
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not implied.
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5.2 Electrical Characteristics
Limit specifications at –40°C to +85°C. Typical specifications at +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK (1) = 4.096MHz,
VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, and fDATA = 1000SPS, unless otherwise
noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ANALOG INPUTS
Full-scale input voltage
Absolute input range
VIN = (AINP – AINN)
AINP or AINN
±VREF/(2 × PGA)
AVSS + 0.7
PGA input voltage noise density
Differential input impedance
V
5
nV/√Hz
Chop on
1
GΩ
Chop off
100
GΩ
100
MΩ
Common-mode input impedance
Input bias current
Crosstalk
V
AVDD – 1.25
f = 31.25Hz
MUX on-resistance
1
nA
–135
dB
30
Ω
PGA OUTPUT (CAPP, CAPN)
Absolute output range
AVSS + 0.4
PGA differential output impedance
AVDD – 0.4
V
Ω
600
Output impedance tolerance
±10%
External bypass capacitance
10
Modulator differential input impedance
55
100
nF
kΩ
AC PERFORMANCE
Signal-to-noise ratio (2)
Total harmonic distortion (3)
Spurious-free dynamic
range
SNR
THD
120
124
dB
PGA = 1...16
–122
–114
PGA = 32
–117
–110
PGA = 64
–115
SFDR
dB
dB
123
DC PERFORMANCE
Resolution
Data rate
fDATA
Integral nonlinearity (INL) (4)
No missing codes
31
FIR filter mode
250
Sinc filter mode
8000
Differential input
Offset error
Offset error after calibration (6)
Shorted input
Offset drift
Gain error (7)
–1.5%
Gain error after calibration (6)
128,000
SPS
0.00005
0.0004
% FSR (5)
50
200
μV
1
μV
0.02
μV/°C
–1.0%
PGA = 1
2
PGA = 16
9
Gain matching (8)
0.3%
fCM = 60Hz (9)
Common-mode rejection
(1)
(2)
(3)
(4)
(5)
(6)
(7)
(8)
(9)
SPS
–0.5%
0.0002%
Gain drift
Power-supply rejection
Bits
4000
AVDD, AVSS
DVDD
fPS = 60Hz (9)
95
110
80
90
90
115
ppm/°C
ppm/°C
0.8%
dB
dB
fCLK = system clock.
VIN = 20mVDC/PGA; see Table 1.
VIN = 31.25Hz, –0.5dBFS.
Best-fit method.
FSR: Full-scale range = ±VREF/(2 × PGA).
Calibration accuracy is on the level of noise reduced by 4 (calibration averages 16 readings).
The PGA output impedance and the modulator input impedance results in –1% systematic gain error.
Gain match relative to PGA = 1.
fCM is the input common-mode frequency. fPS is the power-supply frequency.
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SBAS418I – SEPTEMBER 2007 – REVISED MARCH 2015
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Electrical Characteristics (continued)
Limit specifications at –40°C to +85°C. Typical specifications at +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK (1) = 4.096MHz,
VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, and fDATA = 1000SPS, unless otherwise
noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
(VREF = VREFP – VREFN)
0.5
5
(AVDD – AVSS)
+ 0.2
V
V
VOLTAGE REFERENCE INPUTS
Reference input voltage
Negative reference input
VREFN
AVSS – 0.1
VREFP – 0.5
Positive reference input
VREFP
VREFN + 0.5
AVDD + 0.1
Reference input impedance
85
V
kΩ
DIGITAL FILTER RESPONSE
Passband ripple
±0.003
Passband (–0.01dB)
0.375 × fDATA
Bandwidth (–3dB)
0.413 × fDATA
High-pass filter corner
0.1
Stop band attenuation (10)
135
Hz
Hz
10
0.500 × fDATA
Settling time (latency)
Hz
dB
Stop band
Group delay
dB
Minimum phase filter (11)
5/fDATA
Linear phase filter
31/fDATA
Minimum phase filter
62/fDATA
Linear phase filter
62/fDATA
Hz
s
s
DIGITAL INPUT/OUTPUT
VIH
VIL
0.8 × DVDD
DVDD
V
DGND
0.2 × DVDD
V
VOH
IOH = 1mA
VOL
IOL = 1mA
0.2 × DVDD
V
0 < VDIGITAL IN < DVDD
±10
μA
4.096
MHz
fCLK/2
MHz
Input leakage
Clock input
Serial clock rate
0.8 × DVDD
fCLK
V
1
fSCLK
POWER SUPPLY
AVSS
–2.6
0
V
AVDD
AVSS + 4.75
AVSS + 5.25
V
DVDD
1.65
3.6
V
AVDD, AVSS current
DVDD current
Power dissipation
Normal operation
4.5
6.5
|mA|
Standby mode
1
15
|μA|
Power-down mode
1
15
|μA|
Normal operation
0.6
0.8
mA
Modulator mode
0.1
Standby mode
25
50
Power-down mode (12)
1
15
μA
Normal operation
25
35
mW
Standby mode
90
250
μW
Power-down mode
10
125
μW
mA
μA
(10) Input frequencies in the range of NfCLK/512 ± fDATA/2 (N = 1, 2, 3...) can mix with the modulator chopping clock. In these frequency
ranges intermodulation = 120dB, typ.
(11) At dc; see Figure 44.
(12) CLK input stopped.
4
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SBAS418I – SEPTEMBER 2007 – REVISED MARCH 2015
6 Timing Diagram
tSCDL
tSCLK
tSPWH
SCLK
tDIST
tSPWL
tSCDL
DIN
tDIHD
tDOHD
DOUT
tDOPD
6.1
Timing Requirements
At TA = –40°C to +85°C and DVDD = 1.65V to 3.6V, unless otherwise noted.
PARAMETER DESCRIPTION
tSCLK
(1)
(2)
MAX
UNITS
2
16
1/fCLK
SCLK pulse width, high and low (1)
0.8
10
1/fCLK
tDIST
DIN valid to SCLK rising edge: setup time
50
50
tSPWH, L
SCLK period
MIN
ns
tDIHD
Valid DIN to SCLK rising edge: hold time
tDOPD
SCLK falling edge to valid new DOUT: propagation delay (2)
ns
tDOHD
SCLK falling edge to DOUT invalid: hold time
0
ns
tSCDL
Final SCLK rising edge of command to first SCLK rising edge for register read/write
data. (Also between consecutive commands.)
24
1/fCLK
100
ns
Holding SCLK low for 64 DRDY falling edges resets the serial interface.
Load on DOUT = 20pF || 100kΩ.
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7 Pin Configuration and Functions
PW Package
TSSOP-28
(Top View)
CLK
1
28
BYPAS
SCLK
2
27
DGND
DRDY
3
26
DVDD
DOUT
4
25
DGND
RESET
DIN
5
24
DGND
6
23
PWDN
MCLK
7
22
VREFP
M1
8
21
VREFN
ADS1282
M0
9
20
AVSS
SYNC
10
19
AVDD
MFLAG
11
18
AINN1
DGND
12
17
AINP1
CAPN
13
16
AINN2
CAPP
14
15
AINP2
Pin Functions
6
NAME
NO.
FUNCTION
CLK
1
Digital input
Master clock input
DESCRIPTION
SCLK
2
Digital input
Serial clock input
DRDY
3
Digital output
Data ready output: read data on falling edge
DOUT
4
Digital output
Serial data output
DIN
5
Digital input
Serial data input
MCLK
7
Digital I/O
Modulator clock output; if in modulator mode:
MCLK: Modulator clock output
Otherwise, the pin is an unused input (must be tied).
M1
8
Digital I/O
Modulator data output 1; if in modulator mode:
M1: Modulator data output 1
Otherwise, the pin is an unused input (must be tied).
M0
9
Digital I/O
Modulator data output 0; if in modulator mode:
M0: Modulator data output 0
Otherwise, the pin is an unused input (must be tied).
SYNC
10
Digital input
Synchronize input
MFLAG
11
Digital output
Modulator Over-Range flag: 0 = normal, 1 = modulator over-range
DGND
6, 12, 25, 27
Digital ground
Digital ground, pin 12 is the key ground point
CAPN
13
Analog
PGA outputs: Connect 10nF capacitor from CAPP to CAPN
CAPP
14
Analog
PGA outputs: Connect 10nF capacitor from CAPP to CAPN
AINP2
15
Analog input
Positive analog input 2
AINN2
16
Analog input
Negative analog input 2
AINP1
17
Analog input
Positive analog input 1
AINN1
18
Analog input
Negative analog input 1
AVDD
19
Analog supply
Positive analog power supply
AVSS
20
Analog supply
Negative analog power supply
VREFN
21
Analog input
Negative reference input
VREFP
22
Analog input
Positive reference input
PWDN
23
Digital input
Power-down input, active low
RESET
24
Digital input
Reset input, active low
DVDD
26
Digital supply
BYPAS
28
Analog
Digital power supply: +1.8V to +3.3V
Sub-regulator output: Connect 1μF capacitor to DGND
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8 Typical Characteristics
At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, PGA = 1,
CAPN – CAPP = 10nF, and fDATA = 1000SPS, unless otherwise noted.
0
-60
-80
-100
-120
8192-Point FFT
VIN = -20dBFS, 31.25Hz
PGA = 1
THD = -120.1dB
-20
-40
Amplitude (dB)
-40
Amplitude (dB)
0
8192-Point FFT
VIN = -0.5dBFS, 31.25Hz
PGA = 1
THD = -124.0dB
-20
-60
-80
-100
-120
-140
-140
-160
-160
-180
-180
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
500
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
Figure 1. Output Spectrum
0
-60
-80
-100
-120
8192-Point FFT
Shorted Input
SNR = 124.0dB
-20
-40
Amplitude (dB)
-40
Amplitude (dB)
Figure 2. Output Spectrum
0
8192-Point FFT
VIN = -0.5dBFS, 31.25Hz
PGA = 16
THD = -122.4dB
-20
-60
-80
-100
-120
-140
-140
-160
-160
-180
-180
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
500
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
Figure 3. Output Spectrum
8192-Point FFT
20mVDC
SNR = 124.2dB
-20
500
Figure 4. Output Spectrum
-40
-60
-80
-100
-120
-140
-100
Total Harmonic Distortion (dB)
0
Amplitude (dB)
500
THD Limited by
Signal Generator
PGA = 1
PGA = 8
-105
-110
-115
-120
-125
-160
VIN = -0.5dBFS
-130
-180
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
500
10
Figure 5. Output Spectrum
20
30
40
50
60
70
Input Frequency (Hz)
80
90
100
Figure 6. THD vs Input Frequency
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Typical Characteristics (continued)
At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, PGA = 1,
CAPN – CAPP = 10nF, and fDATA = 1000SPS, unless otherwise noted.
-110
VIN = 20mVDC
Total Harmonic Distortion (dB)
Signal-to-Noise Ratio (dB)
126
125
124
123
122
121
120
-55
-35
-15
5
25
45
65
Temperature (°C)
85
105
PGA = 8
VIN = 31.25Hz, -0.5dBFS
-115
-120
-125
-130
-55
125
-35
-15
Figure 7. SNR vs Temperature
25
45
65
Temperature (°C)
85
105
125
Figure 8. THD vs Temperature
130
-110
PGA = 1
PGA = 8
Total Harmonic Distortion (dB)
Signal-to-Noise Ratio (dB)
5
125
PGA = 8
120
115
110
105
-115
-120
-125
PGA = 1
-130
100
0
1
2
3
VREF (V)
4
5
0
5.5
Figure 9. SNR vs Reference Voltage
-110
124
123
122
121
120
VIN = 20mVDC
Data Rate = fCLK/4096
119
0.5
1.0
1.5
2.0
2.5
3.0
fCLK (MHz)
3.5
4.0
4.5
Total Harmonic Distortion (dB)
Signal-to-Noise Ratio (dB)
2
3
VREF (V)
4
5
6
Figure 10. THD vs Reference Voltage
125
PGA = 8
VIN = 31.25Hz, -0.5dBFS
Data Rate = fCLK/4096
-115
-120
-125
-130
0.5
Figure 11. SNR vs Clock Frequency
8
1
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1.0
1.5
2.0
2.5
3.0
fCLK (MHz)
3.5
4.0
4.5
Figure 12. THD vs Clock Frequency
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Typical Characteristics (continued)
At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, PGA = 1,
CAPN – CAPP = 10nF, and fDATA = 1000SPS, unless otherwise noted.
130
140
120
120
Power-Supply Rejection (dB)
Common-Mode Rejection (dB)
DVDD
110
100
90
80
70
AVDD
100
80
AVSS
60
40
20
0
10
100
1k
10k
Input Frequency (Hz)
100k
1M
10
Figure 13. CMR vs Input Frequency
100
1k
10k
100k
Power-Supply Frequency (Hz)
1M
Figure 14. Power-Supply Rejection vs Frequency
4
4
Integral Nonlinearity (ppm)
Integral Nonlinearity (ppm)
3
2
PGA = 2
1
0
-1
PGA = 8
PGA = 32
-2
3
2
1
-3
-4
-100
-75
-50
-25
0
25
50
Input Amplitude (% Full-Scale)
75
0
-55
100
-35
Figure 15. INL vs Input Amplitude
5
25
45
65
Temperature (°C)
85
105
125
105
125
Figure 16. INL vs Temperature
0
30
Shorted Input
8192-Point FFT
Adjacent Channel VIN = -0.5dBFS, 31.25Hz
-20
-40
-60
25
Power (mW)
Amplitude (dB)
-15
-80
-100
-120
20
15
-140
-160
-180
0
50
100 150 200 250 300 350 400 450
Frequency (Hz)
500
10
-55
Figure 17. Crosstalk Output Spectrum
-35
-15
5
25
45
65
Temperature (°C)
85
Figure 18. Power vs Temperature
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Typical Characteristics (continued)
At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, PGA = 1,
CAPN – CAPP = 10nF, and fDATA = 1000SPS, unless otherwise noted.
30
30
25
25
20
20
25 Units
Occurrences
Power (mW)
PGA = 8
15
10
15
10
PGA = 1
5
5
0
0
1.0
1.5
2.0
2.5
3.0
fCLK (MHz)
3.5
4.0
4.5
-100 -80 -60 -40 -20
0
20
Offset (mV)
Figure 19. Power vs Clock Frequency
60
80
100
Figure 20. Offset Histogram
90
10
25 Units
25 Units Based on
+20°C Intervals
Over the Range of
-40°C to +85°C
80
8
70
Occurrences
Occurrences
40
6
4
60
50
40
30
20
2
PGA = 8
PGA =1
10
0.08
0.10
0.06
0.02
0.06
0
0.04
-0.3
0.02
-0.4
-0.02
-0.5
-0.04
-0.8 -0.7 -0.6
Gain Error (%)
-0.06
-0.9
-0.08
0
-1.1 -1.0
-0.10
0
-1.2
Offset Drift (mV/°C)
Figure 21. Gain Error Histogram
Figure 22. Offset Drift Histogram
90
80
25 Units Based on +20°C Intervals
Over the Range of -40°C to +85°C
8
70
6
PGA = 16
Occurrences
Occurrences
PGA = 32
60
50
Worst-Case Gain Match Relative PGA = 1 (25 Units)
PGA = 1, 2, 4
PGA = 8, 64
40
30
4
2
20
10
0.10
0.14
0.18
0.22
0.26
0.30
0.34
0.38
0.42
0.46
-15
-14
-13
-12
-11
-10
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
10
0.50
0
0
Gain Drift (ppm/°C)
Gain Error (%)
Figure 23. Gain Drift Histogram
Figure 24. Gain Match Histogram
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9 Overview
The ADS1282 is a high-performance analog-to-digital
converter (ADC) intended for energy exploration,
seismic monitoring, chomatography, and other
exacting applications. The converter provides 24- or
32-bit output data in data rates from 250SPS to
4000SPS. Figure 25 shows the block diagram of the
ADS1282.
The
two-channel
input
MUX
allows
five
configurations: Input 1; Input 2; Input 1 and Input 2
shorted together; shorted with 400Ω test; and
common-mode test. The input MUX is followed by a
continuous time PGA, featuring very low noise of
5nV/√Hz. The PGA is controlled by register settings,
allowing gains of 1 to 64.
The inherently-stable, fourth-order, delta-sigma
modulator measures the differential input signal
VIN = (AINP – AINN) PGA against the differential
reference VREF = (VREFP – VREFN). A digital output
(MFLAG) indicates that the modulator is in overload
as a result of an overdrive condition. The modulator
output is available directly on the MCLK, M0, and M1
output pins. The modulator connects to an on-chip
digital filter that provides the output code readings.
The SYNC input resets the operation of both the
digital
filter
and
the
modulator,
allowing
synchronization conversions of multiple ADS1282
devices to an external event. The SYNC input
supports a continuously-toggled input mode that
accepts an external data frame clock locked to the
conversion rate.
The RESET input resets the register settings and
also restarts the conversion process. The PWDN
input sets the device into a micro-power state. Note
that register settings are not retained in PWDN mode.
Use the STANDBY command in its place if it is
desired to retain register settings (the quiescent
current in the Standby mode is slightly higher).
Noise-immune Schmitt-trigger and clock-qualified
inputs (RESET and SYNC) provide increased
reliability in high-noise environments. The serial
interface is used to read conversion data, in addition
to reading from and writing to the configuration
registers.
VREFP
VREFN
AVDD
CAPN
CAPP
The digital filter consists of a variable decimation rate,
fifth-order sinc filter followed by a variable phase,
decimate-by-32, finite-impulse response (FIR) lowpass filter with programmable phase, and then by an
adjustable high-pass filter for dc removal of the output
reading. The output of the digital filter can be taken
from the sinc, the FIR low-pass, or the infinite impulse
response (IIR) high-pass sections.
Gain and offset registers scale the digital filter output
to produce the final code value. The scaling feature
can be used for calibration and sensor gain matching.
The output data word is provided as either a 24-bit
word or a full 32-bit word, allowing complete
utilization of the inherently high resolution.
ADS1282
AINP2
AINN2
AINP1
AINN1
BYPAS
CLK
+1.8V
(Digital core)
DVDD
LDO
MUX
300W
400W
PGA
4th-Order
DS
Modulator
300W
400W
Programmable
Digital Filter
Calibration
Serial
Interface
Over-Range
Detection
SYNC
Control
RESET
PWDN
AVDD + AVSS
2
AVSS
DRDY
SCLK
DIN
DOUT
MFLAG
MCLK
M0
M1
DGND
Figure 25. ADS1282 Block Diagram
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9.2 Noise Performance
The device features unipolar and bipolar analog
power supplies (AVDD and AVSS, respectively) for
input range flexibility and a digital supply accepting
1.8V to 3.3V. The analog supplies may be set to +5V
to accept unipolar signals (with input offset) or set
lower in the range of ±2.5V to accept true bipolar
input signals (ground referenced).
The ADS1282 offers outstanding noise performance
(SNR). SNR depends on the data rate, the PGA
setting. As the bandwidth is reduced by decreasing
the data rate, the SNR improves correspondingly.
Similarly, as the PGA gain is increased, the SNR
decreases. Table 1 summarizes the noise
performance versus data rate and PGA setting.
An internal sub-regulator is used to supply the digital
core from DVDD. The BYPAS pin (pin 28) is the subregulator output and requires a 1μF capacitor for
noise reduction. BYPAS should not be used to drive
external circuitry.
9.3 Input-Referred Noise
The input-referred noise is related to SNR by
Equation 1:
FSRRMS
SNR = 20log
NRMS
9.1 ADS1282H
The H version of the ADS1282 has an improved input
stage compared to the base version ADS1282. The
ADS1282H design is optimized for use with high
impedance sensors, such as hydrophones. The
ADS1282H is recommended when interfacing to
hydrophone sensors and can also be used for lowimpedance, geophone sensors as well. The base
version ADS1282 should only be used with lowimpedance geophone sensors, where the associated
external terminating resistance is < 50kΩ (per
resistor).
where:
FSRRMS = Full-scale range RMS = VREF/(2 × √2 ×
PGA)
NRMS = Noise RMS (input-referred)
(1)
9.4 Idle Tones
The ADS1282 modulator incorporates an internal
dither signal that randomizes the idle tone energy.
Low-level idle tones may still be present, typically
–137dB below full-scale. The low-level idle tones can
be shifted out of the passband with an external offset
= 20mV/PGA. See the Application Information section
for the recommended offset circuit.
Table 1. Signal-to-Noise Ratio (dB) (1)
(1)
12
PGA
DATA RATE
(SPS)
1
2
4
8
16
32
64
250
130
130
129
128
125
119
114
500
127
127
126
125
122
116
111
1000
124
124
123
122
119
113
108
2000
121
121
120
119
116
111
106
4000
118
118
117
116
113
108
103
VIN = 20mVDC/PGA.
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9.5 Analog Inputs and Multiplexer
Analog Inputs and Multiplexer (continued)
A diagram of the input multiplexer is shown in
Figure 26.
Table 2. Multiplexer Modes
MUX[2:0]
ESD diodes protect the multiplexer inputs. If either
input is taken below AVSS – 0.3V or above AVDD +
0.3V, the ESD protection diodes may turn on. If these
conditions are possible, external Schottky clamp
diodes and/or series resistors may be required to limit
the input current to safe values (see the Absolute
Maximum Ratings table).
Also, overdriving one unused input may affect the
conversions of the other input. If overdriven inputs
are possible, it is recommended to clamp the signal
with external Schottky diodes.
AVDD
S1
AINP1
ESD Diodes
S2
AINP2
400W
(+)
S3
To PGA
400W
S4
S5
AINN1
ESD Diodes
AINN2
(-)
S6
Total Harmonic Distortion (dB)
2
000
S1, S5
001
S2, S6
AINP2 and AINN2 connected to
preamplifier
010
S3, S4
Preamplifier inputs shorted together
through 400Ω internal resistors
011
S1, S5, S2, S6
AINP1, AINN1 and AINP2, AINN2
connected together and to the preamplifier
100
S6, S7
External short, preamplifier inputs shorted
to AINN2 (common-mode test)
0
AVDD + AVSS
AVDD
DESCRIPTION
AINP1 and AINN1 connected to
preamplifier
The typical on-resistance (RON) of the multiplexer
switch is 30Ω. When the multiplexer is used to drive
an external load on one input by a signal generator
on the other input, on-resistance and on-resistance
amplitude dependency can lead to measurement
errors. Figure 27 shows THD versus load resistance
and amplitude. Note that THD improves with highimpedance loads and with lower amplitude drive
signals. The data are measured with the circuit from
Figure 28 with MUX[2:0] = 011.
S7
AVSS
SWITCHES
-40
-60
-80
-100
-120
-140
0.1k
AVSS
PGA = 1
PGA = 2
PGA = 4
PGA = 8
PGA = 16
PGA = 32
PGA = 64
-20
1k
10k
100k
1M
10M
RLOAD (W)
Figure 26. Analog Inputs and Multiplexer
The specified input operating range of the PGA is
shown in Equation 2:
AVSS + 0.7V < (AINN or AINP) < AVDD - 1.25V (2)
Figure 27. THD Versus External Load and Signal
Magnitude (PGA) (see Figure 28)
500W
Absolute input levels (input signal level and commonmode level) should be maintained within these limits
for best operation.
500W
The multiplexer connects one of the two external
differential inputs to the preamplifier inputs, in
addition to internal connections for various self-test
modes. Table 2 summarizes the multiplexer
configurations for Figure 26.
RLOAD
ADS1282
Input 1
Input 2
Figure 28. Driving an External Load Through the
Mux
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The PGA of the ADS1282 is a low-noise, continuoustime, differential-in/differential-out CMOS amplifier.
The gain is programmable from 1 to 64, set by
register bits, PGA[2:0]. The PGA differentially drives
the modulator through 300Ω internal resistors. A C0G
capacitor (10nF typical) must be connected to CAPP
and CAPN to filter modulator sampling glitches. The
external capacitor also serves as an anti-alias filter.
The corner frequency is given in Equation 3:
1
fP =
6.3 ´ 600 ´ C
(3)
Referring to Figure 29, amplifiers A1 and A2 are
chopped to remove the offset, offset drift, and the 1/f
noise. Chopping moves the effects to fCLK/128 (8kHz),
which is safely out of the passband. Chopping can be
disabled by setting the CHOP register bit = 0. When
chopping is disabled, the input impedance of the PGA
increases substantially (100GΩ). As shown in
Figure 30, chopping maintains flat noise density; if
chopping is disabled, however, it results in a rising 1/f
noise profile.
PGA (Programmable Gain
Amplifier) (continued)
100
PGA Noise (nV/ÖHz)
9.6 PGA (Programmable Gain Amplifier)
PGA CHOP Off
10
PGA CHOP On
1
10
The PGA has programmable gains from 1 to 64.
Table 3 shows the register bit setting for the PGA and
resulting full-scale differential range.
Table 3. PGA Gain Settings
PGA[2:0]
GAIN
DIFFERENTIAL
INPUT RANGE
(V)(1)
000
1
±2.5
001
2
±1.25
010
4
±0.625
011
8
±0.312
100
16
±0.156
101
32
±0.078
110
64
±0.039
300W
A1
CAPP
CHOP
Gain Control
PGA[2:0] Bits
10nF
300W
CAPN
A2
(55kW, typ)
Modulator
Effective
Impedance
(1) VREF = 5V.
MUX (-)
Chopping Control CHOP Bit
AVSS
Figure 29. PGA Block Diagram
14
1k
Figure 30. PGA Noise
AVDD
MUX (+)
100
Frequency (Hz)
The specified output operating range of the PGA is
shown in Equation 4:
AVSS + 0.4V < (CAPN or CAPP) < AVDD - 0.4V (4)
PGA output levels (signal plus common-mode) should
be maintained within these limits for best operation.
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9.7 ADC
Modulator (continued)
The ADC block of the ADS1282 is composed of two
sections: a high-accuracy modulator and a
programmable digital filter.
appears at the chopping frequency (fCLK/512 = 8kHz).
The component at 5.8kHz is the tone frequency,
shifted out of band by an external 20mV/PGA offset.
The frequency of the tone is proportional to the
applied dc input and is given by PGA × VIN/0.003 (in
kHz).
The high-performance modulator is an inherentlystable, fourth-order, ΔΣ, 2 + 2 pipelined structure, as
Figure 31 shows. It shifts the quantization noise to a
higher frequency (out of the passband) where digital
filtering can easily remove it. The modulator can be
filtered either by the on-chip digital filter or by use of
post-processing filters.
fCLK/4
Analog Input (VIN)
MCLK
2nd-Order
DS
1st-Stage
0
1Hz Resolution
VIN = 20mVDC
-20
-40
Magnitude (dB)
9.8 Modulator
-60
-80
-100
-120
-140
M0
-160
-180
1
2nd-Order
DS
2nd-Stage
10
100
1k
10k
100k
Frequency (Hz)
M1
Figure 32. Modulator Output Spectrum
4th-Order Modulator
Figure 31. Fourth-Order Modulator
The modulator first stage converts the analog input
voltage into a pulse-code modulated (PCM) stream.
When the level of differential analog input (AINP –
AINN) is near one-half the level of the reference
voltage 1/2 × (VREFP – VREFN), the ‘1’ density of
the PCM data stream is at its highest. When the level
of the differential analog input is near zero, the PCM
‘0’ and ‘1’ densities are nearly equal. At the two
extremes of the analog input levels (+FS and –FS),
the ‘1’ density of the PCM streams is approximately
+90% and +10%, respectively.
The modulator second stage produces a '1' density
data stream designed to cancel the quantization
noise of the first stage. The data streams of the two
stages are then combined before the digital filter
stage, as shown in Equation 5.
Y[n] = 3M0[n - 2] - 6M0[n - 3] + 4M0[n - 4]
+ 9(M1[n] - 2M1[n - 1] + M1[n - 2])
(5)
M0[n] represents the most recent first-stage output
while M0[n – 1] is the previous first-stage output.
When the modulator output is enabled, the digital
filter shuts down to save power.
The modulator is optimized for input signals within a
4kHz passband. As Figure 32 shows, the noise
shaping of the modulator results in a sharp increase
in noise above 6kHz. The modulator has a chopped
input structure that further reduces noise within the
passband. The noise moves out of the passband and
9.9 Modulator Over-Range
The ADS1282 modulator is inherently stable, and
therefore, has predictable recovery behavior resulting
from an input overdrive condition. The modulator
does not exhibit self-resetting behavior, which often
results in an unstable output data stream.
The ADS1282 modulator outputs a 1s density data
stream at 90% duty cycle with the positive full-scale
input signal applied (10% duty cycle with the negative
full-scale signal). If the input is overdriven past 90%
modulation, but below 100% modulation (10% and
0% for negative overdrive, respectively), the
modulator remains stable and continues to output the
1s density data stream. The digital filter may or may
not clip the output codes to +FS or –FS, depending
on the duration of the overdrive. When the input
returns to the normal range from a long duration
overdrive (worst case), the modulator returns
immediately to the normal range, but the group delay
of the digital filter delays the return of the conversion
result to within the linear range (31 readings for linear
phase FIR). 31 additional readings (62 total) are
required for completely settled data.
If the inputs are sufficiently overdriven to drive the
modulator to full duty cycle, all 1s or all 0s, the
modulator enters a stable saturated state. The digital
output code may clip to +FS or –FS, again depending
on the duration. A small duration overdrive may not
always clip the output code. When the input returns to
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Modulator Over-Range (continued)
the normal range, the modulator requires up to 12
modulator clock cycles (fMOD) to exit saturation and
return to the linear region. The digital filter requires an
additional 62 conversions for fully settled data (linear
phase FIR).
In the extreme case of over-range, either input is
overdriven, exceeding the voltage of either analog
supply voltage plus an internal ESD diode drop. The
internal diodes begin to conduct and the signal on the
input is clipped. When the input overdrive is removed,
the diodes recover quickly. Keep in mind that the
input current must be limited to 100mA peak or 10mA
continuous if an overvoltage condition is possible.
9.11 Modulator Over-Range Detection
(MFLAG)
The ADS1282 has a fast-responding over-range
detection that indicates when the differential input
exceeds 100% or –100% full-scale. The threshold
tolerance is ±2.5%.The MFLAG output asserts high
when in an over-range condition. As Figure 33 and
Figure 34 illustrate, the absolute differential input is
compared to 100% of range. The output of the
comparator is sampled at the rate of fMOD/2, yielding
the MFLAG output. The minimum MFLAG pulse width
is fMOD/2.
AINP
å
9.10 Modulator Input Impedance
Where:
fMOD = Modulator sample frequency (CLK / 4)
CS = Input sampling capacitor (17pF, typ)
The resulting modulator input impedance for CLK =
4.096MHz is 55kΩ. The modulator input impedance
and the PGA output resistors result in a systematic
gain error of –1%. CS can vary ±20% over production
lots, affecting the gain error.
P
100% FS
AINN
Q
MFLAG
Pin
fMOD/2
Figure 33. Modulator Over-Range Block Diagram
VIN (% of Full-Scale)
The modulator samples the buffered input voltage
with an internal capacitor to perform conversions. The
charging of the input sampling capacitor draws a
transient current from the PGA output. The average
value of the current can be used to calculate an
effective input impedance of REFF = 1/(fMOD × CS).
IABSI
+100
(AINP - AINN)
0
Time
-100
MFLAG
Pin
Figure 34. Modulator Over-Range Flag Operation
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9.12 Modulator Output Mode
The modulator digital stream output is accessible
directly, bypassing and disabling the internal digital
filter. The modulator output mode is activated by
setting the CONFIG0 register bits FILTR[1:0] = 00.
Pins M0 and M1 then become the modulator data
outputs and the MCLK becomes the modulator clock
output. When not in the modulator mode, these pins
are inputs and must be tied.
The modulator output is composed of three signals:
one output for the modulator clock (MCLK) and two
outputs for the modulator data (M0 and M1). The
modulator clock output rate is fMOD (fCLK / 4).
Synchronization resets the MCLK phase, as shown in
Figure 35. The SYNC input is latched on the rising
edge of CLK. The MCLK resets and the next rising
edge of MCLK occurs three or five CLK periods later,
as shown in Figure 35.
9.13 Voltage Reference Inputs
(VREFP, VREFN)
The voltage reference for the ADS1282 is the
differential voltage between VREFP and VREFN:
VREF = VREFP – VREFN. The reference inputs use a
structure similar to that of the analog inputs with the
circuitry of the reference inputs shown in Figure 36.
The average load presented by the switched
capacitor reference input can be modeled with an
effective differential impedance of REFF = tSAMPLE/CIN
(tSAMPLE = 1/fMOD). Note that the effective impedance
of the reference inputs loads the external reference.
The modulator output data are two bits wide, which
must be merged together before being filtered. Use
the time domain equation of Equation 5 to merge the
data outputs.
AVDD
ESD
Diodes
VREFP
REFF = 85kW
(fMOD = 1.024MHz)
11.5pF
VREFN
tCSHD
ESD
Diodes
CLK
REFF =
tCMD
tSCSU
SYNC
tSYMD
MCLK
1
fMOD ´ CX
AVSS
(1)
Figure 36. Simplified Reference Input Circuit
tMCD0, 1
M0
M1
(1) MCLK = fCLK / 4.
Figure 35. Modulator Mode Timing
Table 4. Modulator Output Timing For Figure 35
PARAMETER
tMCD0,
(1)
1
DESCRIPTION
MIN
TYP
MCLK rising edge to M0, M1 valid propagation delay (1)
tCMD
CLK rising edge to MCLK rising edge reset time (after
synchronization)
tCSHD
CLK to SYNC hold time to not latch on CLK edge
10
tSCSU
SYNC to CLK setup time to latch on CLK edge
10
tSYMD
SYNC to stable bit stream
MAX
UNIT
100
ns
3
1/fCLK
ns
ns
16
1/fMOD
Load on M0 and M1 = 20pF || 100kΩ.
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The ADS1282 reference inputs are protected by ESD
diodes. In order to prevent these diodes from turning
on, the voltage on either input must stay within the
range shown in Equation 6:
Digital Filter (continued)
Table 5. Digital Filter Selection
FILTR[1:0] BITS
DIGITAL FILTERS SELECTED
00
Bypass; modulator output mode
AVSS - 300mV < (VREFP or VREFN) < AVDD + 300mV
01
Sinc
(6)
10
Sinc + FIR
Note that the minimum valid input for VREFN is
AVSS – 0.1V and maximum valid input for VREFP is
AVDD + 0.1V.
11
Sinc + FIR + HPF
(low-pass and high-pass)
A high-quality +5V reference voltage is necessary for
achieving the best performance from the ADS1282.
Noise and drift on the reference degrade overall
system performance, and it is critical that special care
be given to the circuitry generating the reference
voltages in order to achieve full performance. See the
Application Information section for reference
recommendations.
9.14
Digital Filter
The digital filter receives the modulator output and
decimates the data stream. By adjusting the amount
of filtering, tradeoffs can be made between resolution
and data rate: filter more for higher resolution, filter
less for higher data rate.
The digital filter is comprised of three cascaded filter
stages: a variable-decimation, fifth-order sinc filter; a
fixed-decimation FIR, low-pass filter (LPF) with
selectable phase; and a programmable, first-order,
high-pass filter (HPF), as shown in Figure 37.
9.14.1 Sinc Filter Stage (Sinx/X)
The sinc filter is a variable decimation rate, fifth-order,
low-pass filter. Data are supplied to this section of the
filter from the modulator at the rate of fMOD (fCLK/4).
The sinc filter attenuates the high-frequency noise of
the modulator, then decimates the data stream into
parallel data. The decimation rate affects the overall
data rate of the converter; it is set by the DR[2:0]
register bits, as shown in Table 6.
Equation 7 shows the scaled Z-domain transfer
function of the sinc filter.
5
1 - Z-N
H(Z) =
-1
N(1 - Z )
Where:
N = decimation ratio
The output can be taken from one of the three filter
blocks, as Figure 37 shows. To implement the digital
filter completely off-chip, select the filter bypass
setting (modulator output). For partial filtering by the
ADS1282, select the sinc filter output. For complete
on-chip filtering, activate both the sinc and FIR
stages. The HPF can then be included to remove dc
and low frequencies from the data. Table 5 shows the
filter options.
(7)
Table 6. Sinc Filter Data Rates (CLK = 4.096MHz)
DR[2:0] REGISTER
DECIMATION
RATIO (N)
SINC DATA RATE
(SPS)
000
128
8,000
001
64
16,000
010
32
32,000
011
16
64,000
100
8
128,000
3
Direct Modulator
Bit Stream
Filter Mode
(Register Select)
30
Filter
MUX
From Modulator
Sinc Filter
(Decimate by
8 to 128)
Coefficient Filter
(FIR)
(Decimate by 32)
High-Pass Filter
(IIR)
CAL
Block
Code
Clip
To Output Register
31
Figure 37. Digital Filter and Output Code Processing
18
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Equation 8 shows the frequency domain transfer
function of the sinc filter.
0
-0.5
5
pN ´ f
fMOD
½H(f)½ =
-1.0
Gain (dB)
sin
p´f
fMOD
N sin
-1.5
-2.0
(8)
-2.5
where:
N = decimation ratio (see Table 6)
-3.0
0
The sinc filter has notches (or zeroes) that occur at
the output data rate and multiples thereof. At these
frequencies, the filter has zero gain. Figure 38 shows
the frequency response of the sinc filter and
Figure 39 shows the roll-off of the sinc filter.
0
-20
Gain (dB)
-40
-60
-80
-100
-120
-140
0
1
2
3
4
Normalized Frequency (fIN/fDATA)
5
0.05
0.10
0.15
0.20
Normalized Frequency (fIN/fDATA)
Figure 39. Sinc Filter Roll-Off
9.14.2 FIR Stage
The second stage of the ADS1282 digital filter is an
FIR low-pass filter. Data are supplied to this stage
from the sinc filter. The FIR stage is segmented into
four sub-stages, as shown in Figure 40. The first two
sub-stages are half-band filters with decimation ratios
of 2. The third sub-stage decimates by 4 and the
fourth sub-stage decimates by 2. The overall
decimation of the FIR stage is 32. Note that two
coefficient sets are used for the third and fourth
sections, depending on the phase selection. Table 34
(in the Appendix section at the end of this document)
lists the FIR stage coefficients. Table 7 lists the data
rates and overall decimation ratio of the FIR stage.
Table 7. FIR Filter Data Rates
Figure 38. Sinc Filter Frequency Response
(N = 32)
Sinc
Filter
FIR Stage 2
Decimate by 2
FIR Stage 1
Decimate by 2
DR[2:0] REGISTER
DECIMATION
RATIO (N)
FIR DATA RATE
(SPS)
000
4096
250
001
2048
500
010
1024
1000
011
512
2000
100
256
4000
FIR Stage 3
Decimate by 4
FIR Stage 4
Decimate by 2
Output
Coefficients
Linear
Minimum
PHASE Select
Figure 40. FIR Filter Sub-Stages
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As shown in Figure 41, the FIR frequency response
provides a flat passband to 0.375 of the data rate
(±0.003dB passband ripple). Figure 42 shows the
transition from passband to stop band.
2.0
1.5
9.15 Group Delay and Step Response
1.0
Magnitude (mdB)
back (or alias) into the passband and cause errors. A
low-pass signal filter reduces the effect of aliasing.
Often, the RC low-pass filter provided by the PGA
output resistors and the external capacitor connected
to CAPP and CAPN provides sufficient signal
attenuation.
The FIR block is implemented as a multi-stage FIR
structure with selectable linear or minimum phase
response. The passband, transition band, and stop
band responses of the filters are nearly identical but
differ in the respective phase responses.
0.5
0
-0.5
-1.0
9.15.1 Linear Phase Response
-1.5
-2.0
0
0.05
0.10
0.15
0.20
0.25
0.30
0.35
0.40
Normalized Input Frequency (fIN/fDATA)
Figure 41. FIR Passband Magnitude Response
(fDATA = 500Hz)
20
0
Linear phase filters exhibit constant delay time versus
input frequency (that is, constant group delay). Linear
phase filters have the property that the time delay
from any instant of the input signal to the same
instant of the output data is constant and is
independent of the signal nature. This filter behavior
results in essentially zero phase error when analyzing
multi-tone signals. However, the group delay and
settling time of the linear phase filter are somewhat
larger than the minimum phase filter, as shown in
Figure 43.
-40
1.4
-60
1.2
-80
1.0
Minimum Phase Filter
Amplitude (dB)
Magnitude (dB)
-20
-100
-120
-140
-160
0
0.1
0.2 0.3 0.4 0.5 0.6 0.7 0.8
Normalized Input Frequency (fIN/fDATA)
0.9
1.0
Figure 42. FIR Transition Band Magnitude
Response
0.8
0.6
0.4
0.2
Linear Phase Filter
0
-0.2
0
Although not shown in Figure 42, the passband
response repeats at multiples of the modulator
frequency (NfMOD – f0 and NfMOD + f0, where N = 1, 2,
etc. and f0 = passband). These image frequencies, if
present in the signal and not externally filtered, fold
20
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5
10 15 20 25 30 35 40 45 50 55 60 65
Time Index (1/fDATA)
Figure 43. FIR Step Response
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Group Delay and Step Response (continued)
Group Delay and Step Response (continued)
9.15.2 Minimum Phase Response
9.15.3 HPF Stage
The minimum phase filter provides a short delay from
the arrival of an input signal to the output, but the
relationship (phase) is not constant versus frequency,
as shown in Figure 44. The filter phase is selected by
the PHS bit, as Table 8 shows.
The last stage of the ADS1282 filter block is a firstorder HPF implemented as an IIR structure. This filter
stage blocks dc signals and rolls off low-frequency
components below the cut-off frequency. The transfer
function for the filter is shown in Equation 14 of the
Appendix.
35
25
The high-pass corner frequency is programmed by
registers HPF[1:0], in hexadecimal. Equation 9 is
used to set the high-pass corner frequency. Table 9
lists example values for the high-pass filter.
20
HPF[1:0] = 65,536 1 -
Linear Phase Filter
Group Delay (1/fDATA)
30
1-2
cos wN + sin wN - 1
cos wN
15
(9)
10
Minimum Phase Filter
5
0
20
40
60
80 100 120
Frequency (Hz)
140 160 180
200
Figure 44. FIR Group Delay (fDATA = 500Hz)
Table 8. FIR Phase Selection
Where:
HPF = High-pass filter register value (converted
to hexadecimal)
ωN = 2πfHP/fDATA (normalized frequency,
radians)
fHP = High-pass corner frequency (Hz)
fDATA = Data rate (Hz)
Table 9. High-Pass Filter Value Examples
PHS BIT
FILTER PHASE
0
Linear
1
Minimum
fHP (Hz)
DATA RATE (SPS)
HPF[1:0]
0.5
250
0337h
1.0
500
0337h
1.0
1000
019Ah
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Group Delay and Step Response (continued)
9.16 Master Clock Input (CLK)
The HPF causes a small gain error, in which case the
magnitude of the error depends on the ratio of
fHP/fDATA. For many common values of (fHP/fDATA), the
gain error is negligible. Figure 45 shows the gain
error of the HPF. The gain error factor is illustrated in
Equation 13 (see the Appendix at the end of this
document).
The ADS1282 requires a clock input for operation.
The clock is applied to the CLK pin. The data
conversion rate scales directly with the CLK
frequency. Power consumption versus CLK frequency
is relatively constant (see the Typical Characteristics).
0
Gain Error (dB)
-0.10
9.17 Synchronization
(Sync Pin and Sync Command)
-0.20
The ADS1282 can be synchronized to an external
event, as well as synchronized to other ADS1282
devices if the sync event is applied simultaneously.
-0.30
-0.40
-0.50
0.0001
0.001
0.01
0.1
Frequency Ratio (fHP/fDATA)
Figure 45. HPF Gain Error
0
90
-7.5
75
60
Amplitude
45
-22.5
Phase
-30.0
30
-37.5
15
-45.0
0.01
The ADS1282 has two sources for synchronization:
the SYNC input pin and the SYNC command. The
ADS1282 also has two synchronizing modes: Pulsesync and Continuous-sync. In Pulse-sync mode, the
ADS1282 synchronizes to a single sync event. In
Continuous-sync mode, either a single SYNC event is
used to synchronize conversions or a continuous
clock is applied to the pin with a period equal to
integer multiples of the data rate. When the periods of
the sync input and the DRDY output do not match,
the ADS1282 re-synchronizes and conversions are
restarted.
Phase (°)
Amplitude (dB)
Figure 46 shows the first-order amplitude and phase
response of the HPF. Note that in the case of
applying step inputs or synchronizing, the settling
time of the filter should be taken into account.
-15.0
As with any high-speed data converter, a high-quality,
low-jitter clock is essential for optimum performance.
Crystal clock oscillators are the recommended clock
source. Make sure to avoid excess ringing on the
clock input; keep the clock trace as short as possible
and use a 50Ω series resistor close to the source.
0
0.1
1
10
Normalized Frequency (f/fC)
100
Figure 46. HPF Amplitude and Phase Response
22
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9.18 Pulse-Sync Mode
Continuous-Sync Mode (continued)
In Pulse-sync mode, the ADS1282 stops and restarts
the conversion process when a sync event occurs (by
pin or command). When the sync event occurs, the
device resets the internal memory; DRDY goes high
(pulse SYNC mode) otherwise in Continuous SYNC
mode, DRDY continues to toggle, and after the digital
filter has settled, new conversion data are available,
as shown in Figure 47 and Table 10.
Note that a SYNC clock input should be applied after
the Continuous-Sync mode is set. The first rising
edge of SYNC then causes a synchronization.
tCSHD
System Clock
(fCLK)
tSCSU
SYNC Command
Note that resynchronization occurs on the next rising
CLK edge after the rising edge of the SYNC pin or
after the eighth rising SCLK edge for opcode SYNC
commands. To be effective, the SYNC opcode should
be broadcast to all devices simultaneously.
tSPWH
SYNC Pin
New Data
Ready
tSPWL
tDR
DRDY
(Pulse-Sync)
9.19 Continuous-Sync Mode
New Data
Ready
tDR
In Continuous-sync mode, either a single sync pulse
or a continuous clock may be applied. When a single
sync pulse is applied (rising edge), the device
behaves similar to the Pulse-sync mode. However, in
this mode, DRDY continues to toggle unaffected but
the DOUT output is held low until data are ready, 63
DRDY periods later. When the conversion data are
non-zero, new conversion data are ready (as shown
in Figure 47).
DRDY
(Continuous-Sync)
DOUT
Figure 47. Pulse-Sync Timing, Continuous-Sync
Timing with Single Sync
When a continuous clock is applied to the SYNC pin,
the period must be an integral multiple of the output
data rate or the device re-synchronizes. Note that
synchronization results in the restarting of the digital
filter and an interruption of 63 readings (refer to
Table 10).
When the sync input is first applied, the device resynchronizes (under the condition tSYNC ≠ N/fDATA).
DRDY continues to output but DOUT is held low until
the new data are ready. Then, if SYNC is applied
again and the period matches an integral multiple of
the output data rate, the device freely runs without resynchronization. Note that the phase of the applied
clock and output data rate (DRDY) are not matched
because of the initial delay of DRDY after SYNC is
first applied. Figure 48 shows the timing for
Continuous-Sync mode.
tSCSU
tCSHD
System Clock
(fCLK)
tSPWL
tSPWH
SYNC
tSYNC
DRDY
1/fDATA
Figure 48. Continuous-Sync Timing with Sync
Clock
Table 10. Pulse-Sync Timing for Figure 47 and Figure 48
PARAMETER
DESCRIPTION
MIN
MAX
UNITS
tSYNC
SYNC period (1)
1
Infinite
n/fDATA
tCSHD
CLK to SYNC hold time to not latch on CLK edge
10
tSCSU
SYNC to CLK setup time to latch on CLK edge
10
ns
SYNC pulse width, high or low
2
1/fCLK
tSPWH,
tDR
(1)
L
Time for data ready (SINC filter)
Time for data ready (FIR filter)
ns
See Appendix, Table 35
62.98046875/fDATA + 468/fCLK
Continuous-Sync mode; a free-running SYNC clock input without causing re-synchronization.
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9.20 Reset (RESET Pin and Reset
Command)
The ADS1282 may be reset in two ways: toggle the
RESET pin low or send a Reset command. When
using the RESET pin, take it low and hold for at least
2/fCLK to force a reset. The ADS1282 is held in reset
until the pin is released. By command, RESET takes
effect on the next rising edge of fCLK after the eighth
rising edge of SCLK of the command. Note that in
order to ensure the Reset command can function, the
SPI interface may require resetting itself; see the
Serial Interface section.
In reset, registers are set to default and the
conversions are synchronized on the next rising edge
of CLK. New conversion data are available, as shown
in Figure 49 and Table 11.
Settled
Data
DRDY
tDR
tCRHD
System Clock
(fCLK)
tRST
Power-Down
(PWDN Pin and Standby
Command) (continued)
In power-down, note that the device outputs remain
active and the device inputs must not float. When the
Standby command is sent, the SPI port and the
configuration registers are kept active. Figure 50 and
Table 12 show the timing.
PWDN Pin
Wakeup
Command
DRDY
tDR
Figure 50. PWDN Pin and Wake-Up Command
Timing
(Table 12 Shows tDR)
tRCSU
9.22 Power-On Sequence
RESET Pin
or
RESET Command
Figure 49. Reset Timing
Table 11. Reset Timing for Figure 49
PARAMETER DESCRIPTION
MIN
UNITS
tCRHD
CLK to RESET hold time
10
ns
tRCSU
RESET to CLK setup time
10
ns
tRST
RESET low
2
1/fCLK
tDR
Time for data ready
62.98046875/
fDATA + 468/fCLK
The ADS1282 has three power supplies: AVDD,
AVSS, and DVDD. Figure 51 shows the power-on
sequence of the ADS1282. The power supplies can
be sequenced in any order. The supplies [the
difference of (AVDD – AVSS) and DVDD] generate
an internal reset whose outputs are summed to
generate a global internal reset. After the supplies
have crossed the minimum thresholds, 216 fCLK cycles
are counted before releasing the internal reset. After
the internal reset is released, new conversion data
are available, as shown in Figure 51 and Table 12.
AVDD - AVSS
DVDD
9.21 Power-Down
(PWDN Pin and Standby Command)
There are two ways to power-down the ADS1282:
take the PWDN pin low or send a Standby command.
When the PWDN pin is pulled low, the internal
circuitry is disabled to minimize power and the
contents of the register settings are reset.
3.5V nom
1V nom
CLK
16
Internal Reset
2
fCLK
DRDY
tDR
Figure 51. Power-On Sequence
Table 12. Power-On, PWDN Pin, and Wake-Up Command Timing for New Data
PARAMETER
DESCRIPTION
FILTER MODE
16
tDR
(1)
(2)
24
Time for data ready 2 CLK cycles after power-on;
and new data ready after PWDN pin or Wake-Up command
SINC (1)
See Appendix, Table 35
62.98046875/fDATA + 468/fCLK
(2)
FIR
Supply power-on and PWDN pin default is 1000SPS FIR.
Subtract two CLK cycles for the Wake-Up command. The Wake-Up command is timed from the next rising edge of CLK to after the
eighth rising edge of SCLK during command to DRDY falling.
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9.23 DVDD Power Supply
Serial Interface (continued)
The DVDD supply operates over the range of +1.65V
to +3.6V. If DVDD is operated at less than 2.25V,
connect the DVDD pin to the BYPAS pin. If DVDD is
greater than or equal to 2.25V, do not connect DVDD
to the BYPAS pin. Figure 52 shows this connection.
9.24.1 Serial Clock (SCLK)
1.65V to 3.6V
DVDD
Data are shifted into DIN on the rising edge of SCLK
and data are shifted out of DOUT on the falling edge
of SCLK. If SCLK is held low for 64 DRDY cycles,
data transfer or commands in progress terminate and
the SPI interface resets. The next SCLK pulse starts
a new communication cycle. This timeout feature can
be used to recover the interface when a transmission
is interrupted or SCLK inadvertently glitches. SCLK
should remain low when not active.
ADS1282
Tie DVDD to BYPAS if
DVDD power is < 2.25V.
Otherwise float BYPAS.
The serial clock (SCLK) is an input that is used to
clock data into (DIN) and out of (DOUT) the
ADS1282. This input is a Schmitt-trigger input that
has a high degree of noise immunity. However, it is
recommended to keep SCLK as clean as possible to
prevent possible glitches from inadvertently shifting
the data.
BYPAS
1 mF
Figure 52. DVDD Power
9.24 Serial Interface
9.24.2 Data Input (DIN)
A serial interface is used to read the conversion data
and access the configuration registers. The interface
consists of three basic signals: SCLK, DIN, and
DOUT. An additional output, DRDY, transitions low in
Read Data Continuous mode when data are ready for
retrieval. Figure 53 shows the connection when
multiple converters are used.
The data input pin (DIN) is used to input register data
and commands to the ADS1282. Keep DIN low when
reading conversion data in the Read Data Continuous
mode (except when issuing a STOP Read Data
Continuous command). Data on DIN are shifted into
the converter on the rising edge of SCLK.
9.24.3 Data Output (DOUT)
FPGA or Processor
SCLK
DOUT1
ADS1282
DIN2
DRDY1
SCLK
DOUT2
ADS1282
DIN2
DRDY2
SCLK
The data output pin (DOUT) is used to output data
from the ADS1282. Data are shifted out on DOUT on
the falling edge of SCLK.
DOUT1
DIN1
IRQ
SCLK (optional)
DOUT2
DIN2
IRQ (optional)
Figure 53. Interface for Multiple Devices
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Serial Interface (continued)
9.25 Data Format
9.24.4 Data Ready (DRDY)
The ADS1282 provides 32 bits of conversion data in
binary twos complement format, as shown in
Table 13. The LSB of the data is a redundant sign bit:
'0' for positive numbers and '1' for negative numbers.
However, when the output is clipped to +FS, the
LSB = 1; when the output is clipped to –FS, the
LSB = 0. If desired, the data readback may be
stopped at 24 bits. Note that in sinc filter mode, the
output data are scaled by 1/2.
DRDY is an output; when it transitions low, this
transition indicates new conversion data are ready, as
shown in Figure 54. When reading data by the
continuous mode, the data must be read within four
CLK periods before DRDY goes low again or the data
are overwritten with new conversion data. When
reading data by the command mode, the read
operation can overlap the occurrence of the next
DRDY without data corruption.
Table 13. Ideal Output Code Versus Input Signal
INPUT SIGNAL VIN
(AINP – AINN)
DRDY
32-BIT IDEAL OUTPUT
CODE(1)
FIR FILTER
DOUT
Bit 31
Bit 30
Bit 29
VREF
>
2 x PGA
SINC
FILTER(2)
7FFFFFFFh
(3)
7FFFFFFEh
3FFFFFFFh
00000002h
00000001h
00000000h
00000000h
FFFFFFFFh
FFFFFFFFh
80000001h
C0000000h
80000000h
(3)
SCLK
VREF
2 x PGA
Figure 54. DRDY with Data Retrieval
VREF
DRDY resets high on the first falling edge of SCLK.
Figure 54 and Figure 55 show the function of DRDY
with and without data readback, respectively.
2PGA ´ (230 - 1)
If data are not retrieved (no SCLK provided), DRDY
pulses high for four fCLK periods during the update
time, as shown in Figure 55.
-VREF
4/fCLK
0
2PGA ´ (230 - 1)
230
-VREF
´
2PGA
Data Updating
<
DRDY
26
230
-VREF
2PGA
Figure 55. DRDY with No Data Retrieval
230 - 1
´
230 - 1
(1) Excludes effects of noise, linearity, offset, and gain errors.
(2) Due to the reduction in oversampling ratio (OSR) related to
the sinc filter high data rates, full resolution may not be
available.
(3) In sinc filter mode, the output does not clip at half-scale code
when the full-scale range is exceeded.
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9.26 Reading Data
Reading Data (continued)
The ADS1282 has two ways to read conversion data:
Read Data Continuous and Read Data By Command.
When a Stop Read Data Continuous command is
issued, the DRDY output is blocked but the ADS1282
continues conversions. In stop continuous mode, the
data can only be read by command.
9.26.1 Read Data Continuous
In the Read Data Continuous mode, the conversion
data are shifted out directly from the device without
the need for sending a read command. This mode is
the default mode at power-on. This mode is also
enabled by the RDATAC command. When DRDY
goes low, indicating that new data are available, the
MSB of data appears on DOUT, as shown in
Figure 56. The data are normally read on the rising
edge of SCLK, at the occurrence of the first falling
edge of SCLK, DRDY returns high. After 32 bits of
data have been shifted out, further SCLK transitions
cause DOUT to go low. If desired, the read operation
may be stopped at 24 bits. The data shift operation
must be completed within four CLK periods before
DRDY falls again or the data may be corrupted.
9.26.2 Read Data by Command
The Read Data Continuous mode is stopped by the
SDATAC command. In this mode, conversion data
are read by command. In the Read Data By
Command mode, a read data command must be sent
to the device for each data conversion (as shown in
Figure 57). When the read data command is received
(on the eighth SCLK rising edge), data are available
to read only when DRDY goes low (tDR). When DRDY
goes low, conversion data appear on DOUT. The
data may be read on the rising edge of SCLK.
DRDY
1
2
3
4
5
6
7
8
9
10 11 12 13 14 15 16
25 26 27 28 29 30
31 32
SCLK
DOUT
Data Byte 1 (MSB)
Data Byte 2 (MSB - 1)
Data Byte 4 (LSB)
tDDPD
DIN
Figure 56. Read Data Continuous
Table 14. Timing Data for Figure 56
PARAMETER
DESCRIPTION
(1)
MIN
TYP
DRDY to valid MSB on DOUT propagation delay (1)
tDDPD
MAX
UNITS
100
ns
Load on DOUT = 20pF || 100kΩ.
DRDY
tDR
1
2
3
4
5
6
7
8
9
10 11 12 13 14 15 16
33 34 35 36 37 38
39 40
SCLK
DOUT
Don't Care
DIN
Data Byte 1 (MSB)
Date Byte 4 (LSB)
tDDPD
Command Byte (0001 0010)
Figure 57. Read Data by Command, RDATA (tDDPD Timing is Given in Table 14)
Table 15. Read Data Timing for Figure 57
PARAMETER
tDR
DESCRIPTION
MIN
Time for new data after data read command
0
TYP
MAX
UNITS
1
fDATA
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9.27 One-Shot Operation
The ADS1282 can perform very power-efficient, oneshot conversions using the STANDBY command
while under software control. Figure 58 shows this
sequence. First, issue the STANDBY command to set
the Standby mode.
When ready to make a measurement, issue the
WAKEUP command. Monitor DRDY; when it goes
low, the fully settled conversion data are ready and
may be read directly in Read Data Continuous mode.
Afterwards, issue another STANDBY command.
When ready for the next measurement, repeat the
cycle starting with another WAKEUP command.
9.28 Offset and Full-Scale Calibration
Registers
The conversion data can be scaled for offset and gain
before yielding the final output code. As shown in
Figure 59, the output of the digital filter is first
subtracted by the offset register (OFC) and then
multiplied by the full-scale register (FSC).
Equation 10 shows the scaling:
FSC[2:0]
Final Output Data = (Input - OFC[2:0]) ´
400000h
(10)
The values of the offset and full-scale registers are
set by writing to them directly, or they are set
automatically by calibration commands.
Note that the offset and full-scale calibrations apply to
specific PGA settings. When the PGA is changed,
these registers generally require recalculation.
Calibration is bypassed in the sinc filter mode.
Standby
ADS1282 Status
Performing One-Shot Conversion
Standby
DRDY
DIN
(1)
STANDBY
STANDBY
WAKEUP
Settled
Data
DOUT
(1) See Figure 50 and Table 12 for time to new data.
Figure 58. One-Shot Conversions Using the Standby Command
AINP
Modulator
AINN
Digital
Filter
+
S
´
OFC
Register
FSC Register
400000h
-
Output Data
Clipped to 32 Bits
Final Output
Figure 59. Calibration Block Diagram
28
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9.28.1 Ofc[2:0] Registers
9.28.2 FSC[2:0] Registers
The offset calibration is a 24-bit word, composed of
three 8-bit registers, as shown in Table 18. The offset
register is left-justified to align with the 32-bits of
conversion data. The offset is in twos complement
format with a maximum positive value of 7FFFFFh
and a maximum negative value of 800000h. This
value is subtracted from the conversion data. A
register value of 00000h has no offset correction
(default value). Note that while the offset calibration
register value can correct offsets ranging from –FS to
+FS (as shown in Table 16), to avoid input overload,
the analog inputs cannot exceed the full-scale range.
The full-scale calibration is a 24-bit word, composed
of three 8-bit registers, as shown in Table 19. The
full-scale calibration value is 24-bit, straight offset
binary, normalized to 1.0 at code 400000h. Table 17
summarizes the scaling of the full-scale register. A
register value of 400000h (default value) has no gain
correction (gain = 1). Note that while the full-scale
calibration register value corrects gain errors above 1
(gain correction < 1), the full-scale range of the
analog inputs should not exceed 103% to avoid input
overload.
Table 17. Full-Scale Calibration Register Values
Table 16. Offset Calibration Values
FSC REGISTER
GAIN CORRECTION
OFC REGISTER
FINAL OUTPUT CODE(1)
800000h
2.0
7FFFFFh
80000000h
400000h
1.0
000001h
FFFFFF00h
200000h
0.5
000000h
00000000h
000000h
0
FFFFFFh
00000100h
800000h
7FFFFF00h
(1) Full 32-bit final output code with zero code input.
Table 18. Offset Calibration Word
REGISTER
BYTE
OFC0
LSB
B7
B6
B5
B4
BIT ORDER
B3
B2
B1
OFC1
MID
B15
B14
B13
B12
B11
B10
B9
B8
OFC2
MSB
B23 (MSB)
B22
B21
B20
B19
B18
B17
B16
B0 (LSB)
B0 (LSB)
Table 19. Full-Scale Calibration Word
REGISTER
BYTE
FSC0
LSB
B7
B6
B5
B4
BIT ORDER
B3
B2
B1
FSC1
MID
B15
B14
B13
B12
B11
B10
B9
B8
FSC2
MSB
B23 (MSB)
B22
B21
B20
B19
B18
B17
B16
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9.29 Calibration Commands
Calibration Commands (continued)
Calibration commands may be sent to the ADS1282
to calibrate the conversion data. The values of the
offset and gain calibration registers are internally
written to perform calibration. The appropriate input
signals must be applied to the ADS1282 inputs
before sending the commands. Use slower data rates
to achieve more consistent calibration results; this
effect is a byproduct of the lower noise that these
data rates provide. Also, if calibrating at power-on, be
sure the reference voltage is fully settled.
9.29.1 OFSCAL Command
Figure 60 shows the calibration command sequence.
After the analog input voltage (and reference) have
stabilized, send the Stop Data Continuous command
followed by the SYNC and Read Data Continuous
commands. 64 data periods later, DRDY goes low.
After DRDY goes low, send the Stop Data
Continuous, then the Calibrate command followed by
the Read Data Continuous command. After 16 data
periods, calibration is complete and conversion data
may be read at this time. The SYNC input must
remain high during the calibration sequence.
Note that the calibration commands apply to specific
PGA settings. If the PGA is changed, recalibration is
necessary. Calibration is bypassed in the sinc filter
mode.
The OFSCAL command performs an offset
calibration. Before sending the offset calibration
command sequence (Figure 60), a zero input signal
must be applied to the ADS1282 and the inputs
allowed to stabilize. When the command sequence
(Figure 60) is sent, the ADS1282 averages 16
readings and then writes this value to the OFC
register. The contents of the OFC register may be
subsequently read or written. During offset
calibration, the full-scale correction is bypassed.
9.29.2 GANCAL Command
The GANCAL command performs a gain calibration.
Before sending the GANCAL command sequence
(Figure 60), a dc input must be applied (typically fullscale input, but not to exceed 103% full-scale). After
the signal has stabilized, the command sequence can
be sent. The ADS1282 averages 16 readings, then
computes a gain value that makes the applied input
the new full-scale. The gain value is written to the
FSC register, whose contents may be subsequently
read or written.
VIN
Fully stable input and reference voltage.
Commands
SDATAC
DRDY
SYNC
RDATAC
SDATAC
OFSCAL or
GANCAL
RDATAC
16 Data
Periods
64 Data Periods
Calibration
Complete
SYNC
Figure 60. Offset and Gain Calibration Timing
30
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9.30 User Calibration
User Calibration (continued)
System calibration of the ADS1282 can be performed
without using the calibration commands. This
procedure requires the calibration values to be
externally calculated and then written to the
calibration registers. The steps for this procedure are:
1. Set the OFSCAL[2:0] register = 0h and
GANCAL[2:0] = 400000h. These values set the
offset and gain registers to 0 and 1, respectively.
2. Apply a zero differential input to the input of the
system. Wait for the system to settle and then
average n output readings. Higher numbers of
averaged readings result in more consistent
calibration. Write the averaged value to the OFC
register.
3. Apply a differential dc signal, or an ac signal
(typically full-scale, but not to exceed 103% fullscale). Wait for the system to settle and then
average the n output readings.
The value written to the FSC registers is calculated
by Equation 11.
DC signal calibration is shown in Equation 11. The
expected output code is based on 31-bit output data.
FSC[2:0] = 400000h ´
Expected Output Code
Actual Output Code
(11)
For ac signal calibration, use an RMS value of
collected data (as shown in Equation 12).
Expected RMS Value
FSC[2:0] = 400000h ´
Actual RMS Value
(12)
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10 Commands
The commands listed in Table 20 control the
operation of the ADS1282. Most commands are
stand-alone (that is, 1 byte in length); the register
reads and writes require a second command byte in
addition to the actual data bytes.
In Read Data Continuous mode, the ADS1282 places
conversion data on the DOUT pin as SCLK is
applied. As a consequence of the potential conflict of
conversion data on DOUT and data placed on DOUT
resulting from a register or Read Data By Command
operation, it is necessary to send a STOP Read Data
Continuous command before Register or Data Read
By Command. The STOP Read Data Continuous
command disables the direct output of conversion
data on the DOUT pin.
A delay of 24 fCLK cycles between commands and
between bytes within a command is required, starting
from the last SCLK rising edge of one command to
the first SCLK rising edge of the following command.
This delay is shown in Figure 61.
DIN
Command
Byte
Command
Byte
SCLK
(1)
tSCLKDLY
(1)
tSCLKDLY = 24/fCLK (min).
Figure 61. Consecutive Commands
Table 20. Command Descriptions
COMMAND
TYPE
DESCRIPTION
1st COMMAND BYTE (1) (2)
WAKEUP
Control
Wake-up from Standby mode
0000 000X (00h or 01h)
STANDBY
Control
Enter Standby mode
0000 001X (02h or 03h)
SYNC
Control
Synchronize the A/D conversion
0000 010X (04h or 5h)
RESET
Control
Reset registers to default values
0000 011X (06h or 07h)
RDATAC
Control
Read data continuous
0001 0000 (10h)
SDATAC
Control
Stop read data continuous
0001 0001 (11h)
RDATA
Data
Read data by command (4)
0001 0010 (12h)
RREG
Register
Read nnnnn register(s) at address rrrrr (4)
001r rrrr (20h + 000r rrrr)
000n nnnn (00h + n nnnn)
000n nnnn (00h + n nnnn)
WREG
Register
Write nnnnn register(s) at address rrrrr
010r rrrr (40h + 000r rrrr)
OFSCAL
Calibration
Offset calibration
0110 0000 (60h)
GANCAL
Calibration
Gain calibration
0110 0001 (61h)
(1)
(2)
(3)
(4)
32
2nd COMMAND BYTE (3)
X = don't care.
rrrrr = starting address for register read and write commands.
nnnnn = number of registers to be read/written – 1. For example, to read/write three registers, set nnnnn = 2 (00010).
Required to cancel Read Data Continuous mode before sending a command.
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WAKEUP: Wake-Up from Standby Mode
SDATAC: Stop Read Data Continuous
Description: This command is used to exit the
standby mode. Upon sending the command, the time
for the first data to be ready is illustrated in Figure 50
and Table 13. Sending this command during normal
operation has no effect; for example, reading data by
the Read Data Continuous method with DIN held low.
Description: This command stops the Read Data
Continuous mode. Exiting the Read Data Continuous
mode is required before sending Register and Data
read commands. This command suppresses the
DRDY output, but the ADS1282 continues
conversions.
STANDBY: Standby Mode
RDATA: Read Data by Command
Description: This command places the ADS1282
into Standby mode. In Standby, the device enters a
reduced power state where a low quiescent current
remains to keep the register settings and SPI
interface active. For complete device shutdown, take
the PWDN pin low (register settings are not saved).
To exit Standby mode, issue the WAKEUP command.
The operation of Standby mode is shown in
Figure 62.
Description: This command reads the conversion
data. See the Read Data By Command section for
more details.
DIN
0000 001X
(STANDBY)
0000 000X
(WAKEUP)
Standby Mode
Description: This command is used to read single or
multiple register data. The command consists of a
two-byte op-code argument followed by the output of
register data. The first byte of the op-code includes
the starting address, and the second byte specifies
the number of registers to read – 1.
First command byte: 001r rrrr, where rrrrr is the
starting address of the first register.
SCLK
Operating
RREG: Read Register Data
Operating
Figure 62. Standby Command Sequence
SYNC: Synchronize the A/D Conversion
Description: This command synchronizes the
analog-to-digital (A/D) conversion. Upon receipt of the
command, the reading in progress is cancelled and
the conversion process is re-started. In order to
synchronize multiple ADS1282s, the command must
be sent simultaneously to all devices. Note that the
SYNC pin must be high for this command.
RESET: Reset the Device
Description: The RESET command resets the
registers to default values, enables the Read Data
Continuous mode, and restarts the conversion
process; the RESET command is functionally the
same as the RESET pin. See Figure 49 for the
RESET command timing.
RDATAC: Read Data Continuous
Description: This command enables the Read Data
Continuous mode (default mode). In this mode,
conversion data can be read from the device directly
without the need to supply a data read command.
Each time DRDY falls low, new data are available to
read. See the Read Data Continuous section for
more details.
Second command byte: 000n nnnn, where nnnnn is
the number of registers – 1 to read.
Starting with the 16th falling edge of SCLK, the
register data appear on DOUT.
The RREG command is illustrated in Figure 63. Note
that a delay of 24 fCLK cycles is required between
each byte transaction.
WREG: Write to Register
Description: This command writes single or multiple
register data. The command consists of a two-byte
op-code argument followed by the input of register
data. The first byte of the op-code contains the
starting address and the second byte specifies the
number of registers to write – 1.
First command byte: 001r rrrr, where rrrrr is the
starting address of the first register.
Second command byte: 000n nnnn, where nnnnn is
the number of registers – 1 to write.
Data byte(s): one or more register data bytes,
depending on the number of registers specified.
Figure 64 illustrates the WREG command.
Note that a delay of 24 fCLK cycles is required
between each byte transaction.
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OFSCAL: Offset Calibration
GANCAL: Gain Calibration
Description: This command performs an offset
calibration. The inputs to the converter (or the inputs
to the external pre-amplifier) should be zeroed and
allowed to stabilize before sending this command.
The offset calibration register updates after this
operation. See the Calibration Commands section for
more details.
Description: This command performs a gain
calibration. The inputs to the converter should have a
stable dc input (typically full-scale, but not to exceed
103% full-scale). The gain calibration register updates
after this operation. See the Calibration Commands
section for more details.
tDLY
1
2
3
4
5
6
7
8
9
tDLY
10 11 12 13 14 15 16
tDLY
17 18 19 20 21 22 23 24
25 26
SCLK
DIN
Command Byte 1
DOUT
Command Byte 2
Don't Care
Register Data 5
Register Data 6
Example: Read six registers, starting at register 05h (OFC0)
Command Byte 1 = 0010 0101
Command Byte 2 = 0000 0101
Figure 63. Read Register Data (Table 21 Shows tDLY)
tDLY
1
2
3
4
5
6
7
8
9
10 11 12 13 14 15 16
tDLY
tDLY
17 18 19 20 21 22 23 24
25 26
SCLK
DIN
Command Byte 1
Command Byte 2
Register Data 5
Register Data 6
Example: Write six registers, starting at register 05h (OFC0)
Command Byte 1 = 0100 0101
Command Byte 2 = 0000 0101
Figure 64. Write Register Data (Table 21 Shows tDLY)
Table 21. tDRY Value
34
PARAMETER
MIN
tDLY
24/fCLK
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11 Register Map
Collectively, the registers contain all the information needed to configure the part, such as data rate, filter
selection, calibration, etc. The registers are accessed by the RREG and WREG commands. The registers can be
accessed individually or as a block of registers by sending or receiving consecutive bytes. Note that after a
register write operation the ADC resets, resulting in an interruption of 63 readings.
Table 22. Register Map
ADDRESS
REGISTER
RESET
VALUE
BIT 7
BIT 6
BIT 5
BIT 4
BIT 3
BIT 2
BIT 1
00h
ID
X0h
ID3
ID2
ID1
ID0
0
0
0
0
01h
CONFIG0
52h
SYNC
1
DR2
DR1
DR0
PHS
FILTR1
FILTR0
02h
CONFIG1
08h
0
MUX2
MUX1
MUX0
CHOP
PGA2
PGA1
PGA0
03h
HPF0
32h
HPF07
HPF06
HPF05
HPF04
HPF03
HPF02
HPF01
HPF00
04h
HPF1
03h
HPF15
HPF14
HPF13
HPF12
HPF11
HPF10
HPF09
HPF08
05h
OFC0
00h
OFC07
OFC06
OFC05
OFC04
OFC03
OFC02
OFC01
OFC00
06h
OFC1
00h
OFC15
OFC14
OFC13
OFC12
OFC11
OFC10
OFC09
OFC08
07h
OFC2
00h
OFC23
OFC22
OFC21
OFC20
OFC19
OFC18
OFC17
OFC16
08h
FSC0
00h
FSC07
FSC06
FSC05
FSC04
FSC03
FSC02
FSC01
FSC00
09h
FSC1
00h
FSC15
FSC14
FSC13
FSC12
FSC11
FSC10
FSC09
FSC08
0Ah
FSC2
40h
FSC23
FSC22
FSC21
FSC20
FSC19
FSC18
FSC17
FSC16
BIT 0
Table 23. ID : ID Register (Address 00h)
7
6
5
4
3
2
1
0
ID3
ID2
ID1
ID0
0
0
0
0
Reset value = X0h.
Bit[7:4]
ID[3:0]
Factory-programmed identification bits (read-only)
Bit[3:0]
Reserved
Always write '0'
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Table 24. CONFIG0 : Configuration Register 0 (Address 01h)
7
6
5
4
3
2
1
0
SYNC
1
DR2
DR1
DR0
PHASE
FILTR1
FILTR0
Reset value = 52h.
Bit[7]
SYNC
Synchronization mode
0: Pulse SYNC mode (default)
1: Continuous SYNC mode
Bit[6]
Reserved
Always write '1'
Bit[5:3]
Data Rate Select
DR[2:0]
000: 250SPS
001: 500SPS
010: 1000SPS (default)
011: 2000SPS
100: 4000SPS
Bit[2]
FIR Phase Response
PHASE
0: Linear phase (default)
1: Minimum phase
Bit[1:0]
Digital Filter Select
FILTR[1:0]
Digital filter configuration
00: On-chip filter bypassed, modulator output mode
01: Sinc filter block only
10: Sinc + LPF filter blocks (default)
11: Sinc + LPF + HPF filter blocks
36
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Table 25. CONFIG1 : Configuration Register 1 (Address 02h)
7
6
5
4
3
2
1
0
0
MUX2
MUX1
MUX0
CHOP
PGA2
PGA1
PGA0
Reset value = 08h.
Bit[7]
Reserved
Always write '0'
Bit[6:4]
MUX Select
MUX[2:0]
000: AINP1 and AINN1 (default)
001: AINP2 and AINN2
010: Internal short via 400Ω
011:AINP1 and AINN1 connected to AINP2 and AINN2
100: External short to AINN2
Bit[3]
PGA Chopping Enable
CHOP
0: PGA chopping disabled
1: PGA chopping enabled (default)
Bit[2:0]
PGA Gain Select
PGA[2:0]
000: G
001: G
010: G
011: G
100: G
101: G
110: G
=
=
=
=
=
=
=
1 (default)
2
4
8
16
32
64
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HPF1 and HPF0
These two bytes (high-byte and low-byte, respectively) set the corner frequency of the high-pass filter.
Table 26. HPF0: High-Pass Filter Corner Frequency, Low Byte (Address 03h)
7
6
5
4
3
2
1
0
HP07
HP06
HP05
HP04
HP03
HP02
HP01
HP00
Reset value = 32h.
Table 27. HPF1: High-Pass Filter Corner Frequency, High Byte (Address 04h)
7
6
5
4
3
2
1
0
HP15
HP14
HP13
HP12
HP11
HP10
HP09
HP08
Reset value = 03h.
OFC2, OFC1, OFC0
These three bytes set the offset calibration value.
Table 28. OFC0: Offset Calibration, Low Byte (Address 05h)
7
6
5
4
3
2
1
0
OC07
OC06
OC05
OC04
OC03
OC02
OC01
OC00
Reset value = 00h.
Table 29. OFC1: Offset Calibration, Mid Byte (Address 06h)
7
6
5
4
3
2
1
0
OC15
OC14
OC13
OC12
OC11
OC10
OC09
OC08
Reset value = 00h.
Table 30. OFC2: Offset Calibration, High Byte (Address 07h)
7
6
5
4
3
2
1
0
OC23
OC22
OC21
OC20
OC19
OC18
OC17
OC16
Reset value = 00h.
FSC2, FSC1, FSC0
These three bytes set the full-scale calibration value.
Table 31. FSC0: Full-Scale Calibration, Low Byte (Address 08h)
7
6
5
4
3
2
1
0
FSC07
FSC06
FSC05
FSC04
FSC03
FSC02
FSC01
FSC00
Reset value = 00h.
Table 32. FSC1: Full-Scale Calibration, Mid Byte (Address 09h)
7
6
5
4
3
2
1
0
FSC15
FSC14
FSC13
FSC12
FSC11
FSC10
FSC09
FSC08
Reset value = 00h.
Table 33. FSC2: Full-Scale Calibration, High Byte (Address 0ah)
7
6
5
4
3
2
1
0
FSC23
FSC22
FSC21
FSC20
FSC19
FSC18
FSC17
FSC16
Reset value = 40h.
38
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12 Configuration Guide
After RESET or power-on, the registers can be
configured using the following procedure:
1. Reset the serial interface. Before using the
serial interface, it may be necessary to recover
the serial interface (undefined I/O power-up
sequencing may cause false SCLK detection). To
reset the SPI interface, toggle the RESET pin or,
when in Read Data Continuous mode, hold SCLK
low for 64 DRDY periods.
2. Configure the registers. The registers are
configured by either writing to them individually or
as a group. Software may be configured in either
mode. The SDATAC command must be sent
before register read/write operations to cancel the
Read Data Continuous mode.
3. Verify register data. The register may be read
back for verification of device communications.
4. Set the data mode. After register configuration,
the device may be configured for Read Data
Continuous mode, either by the Read Data
Continuous command or configured in Read Data
By Register mode using SDATAC command.
5. Synchronize readings. Whenever SYNC is high,
the ADS1282 freely runs the data conversions.
To stop and re-sync the conversions, take SYNC
low and then high.
6. Read data. If the Read Data Continuous mode is
active, the data are read directly after DRDY falls
by applying SCLK pulses. If the Read Data
Continuous mode is inactive, the data can only
be read by Read Data By Command. The Read
Data opcode command must be sent in this mode
to read each conversion result (note that DRDY
only asserts after each read data command is
sent).
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13 Application Information
The ADS1282 is a very high-resolution ADC. Optimal
performance requires giving special attention to the
support circuitry and printed circuit board (PCB)
design. Locate noisy digital components, such as
microcontrollers, oscillators, etc, in an area of the
PCB away from the converter or front-end
components. Locating the digital components close to
the power-entry point keeps the digital current path
short and separate from sensitive analog
components.
A typical geophone front-end application is shown in
Figure 65. The application shows the ADS1282
operation with dual ±2.5V analog supplies. The
ADS1282 can also operate with a single +5V analog
supply.
The geophone input signal is filtered both
differentially, by components C4 and R1 to R4 and
filtered independently by components C2, C3 and R1,
R2. The differential filter removes high-frequency
normal mode components from the input signal. The
independent
filters
remove
high-frequency
components that are common to both input signals
leads (common-mode filter). The recommended input
filters may not be required for all applications
depending on the system requirements.
Resistors R5 and R6 bias the signals inputs to
midsupply (ground), and also provide the bias current
return path for the ADS1282 inputs. For single-supply
operation, set the bias to a low impedance +2.5V
(AVDD/2).
40
Optional diode clamps protect the ADS1282 inputs
from voltage transients and overloads. The diodes
provide input protection when possible high-level
transients may exceed the internal ESD diode rating.
The REF02 +5V reference provides the reference to
the ADS1282. The reference output is filtered by the
optional R7 and C5 filter network. The filter requires
several seconds to settle after power-on. Capacitor
C7 provides high-frequency bypassing of the
reference inputs and should be placed close to the
ADS1282 pins. Note that R7 (1kΩ) results in a
systematic gain error of 1.2%.
Alternatively, the REF5050 (5V) or REF5045 (4.5V)
reference can be used. The REF5045 reference has
the advantage of operating from the +5V power
supply. The REF5050 requires +5.2V minimum power
supply.
Optional components R8, and R9 provides a 20mV
offset to the ADS1282. The internal 300Ω resistors
form a voltage divider with the external resistors to
provide the offset. The offset moves the low level idle
tones out of the passband. Note that the offset is
independent of the PGA setting. The offset resistors
also result in a small additional gain error. To
maintain good CMR performance, R10 and R11 should
be matched to 0.1%, and the traces routed back
directly to the reference.
Capacitor C6 (10nF) filters the PGA output glitches
caused by sampling of the modulator. The capacitor
also forms a low-pass filter on the input signal with a
cut-off frequency ≉25kHz.
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+2.5V
15
+2.5V
Test
Source
-2.5V
19
20
AVDD
AVSS
AINP2
16
AINN2
R1
100W
Geophone
R3
100W
17
C2
1nF, C0G
R5
20kW
R6
20kW
(2)
R2
100W
C3
1nF, C0G
AINP1
C4
10nF
C0G
R4
100W
18
AINN1
ADS1282
-2.5V
R8
(1)
75kW
14
C6
10nF
C0G 13
+6.5V
(+2.8V REF5050)
1 mF
CAPN
R9
R7
1kW
REF02
(REF5050)
CAPP
(1)
75kW
22
VREFP
1mF
+
C5
100mF
-2.5V
C7
0.1mF
21
VREFN
DGND
6, 12, 25, 27
(1) Optional 20mV offset. Match to 0.1% to maintain CMR.
(2) Optional external diode clamps.
Figure 65. Geophone Interface Application
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Figure 66 shows the digital connection to a field
programmable gate array (FPGA) device. In this
example, two ADS1282s are shown connected. The
DRDY output from each ADS1282 can be used;
however, when the devices are synchronized, the
DRDY output from only one device is sufficient. A
shared SCLK line between the devices is optional.
The modulator over-range flag (MFLAG) from each
device ties to the FPGA. For synchronization, one
SYNC control line connects all ADS1282 devices.
The RESET line also connects to all ADS1282
devices.
For best performance, the FPGA and the ADS1282s
should operate from the same clock. Avoid ringing on
the digital inputs. 47Ω resistors in series with the
digital traces can help to reduce ringing by controlling
impedances. Place the resistors at the source (driver)
end of the trace. Unused digital inputs should not
float; tie them to DVDD or GND. This includes the
modulator data pins, M0, M1, and MCLK.
4.096MHz Clock
47W
+3.3V
26
(1)
DVDD
1mF
28
CLK
1
ADS1282
24
RESET
BYPAS
DOUT
DIN
1mF
SCLK
SYNC
DGND
MFLAG
4
47W
RESET
47W
DOUT1
47W
5
CLK Input
DIN1
47W
2
SCLK1
47W
10
SYNC
47W
11
MFLAG1
4, 12, 23
+3.3V
26
(1)
1mF
DVDD
CLK
ADS1282
RESET
28
BYPAS
DOUT
DIN
1mF
SCLK
SYNC
MFLAG
DGND
DRDY
1
FPGA
24
4
47W
DOUT2
5
DIN2
2
SCLK2
47W
10
47W
11
3
47W
MFLAG2
DRDY
6, 12, 25
NOTE: Dashed line is optional.
(1) For DVDD < 2.25V, see the DVDD Power Supply section.
Figure 66. Microcontroller Interface with Dual ADS1282s
42
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SBAS418I – SEPTEMBER 2007 – REVISED MARCH 2015
14 Appendix
Table 34. FIR Stage Coefficients
SESSION 1
SESSION 2
SESSION 3
SESSION 4
Scaling = 134217728
Scaling = 134217728
COEFFICIENT
LINEAR PHASE
SCALING = 1/512
LINEAR PHASE
SCALING =
1/8388608
LINEAR
PHASE
MINIMUM PHASE
LINEAR
PHASE
b0
3
–10944
0
819
–132
11767
b1
0
0
0
8211
–432
133882
b2
–25
103807
–73
44880
–75
769961
b3
0
0
–874
174712
2481
2940447
b4
150
–507903
–4648
536821
6692
8262605
b5
256
0
–16147
1372637
7419
17902757
b6
150
2512192
–41280
3012996
–266
30428735
b7
0
4194304
–80934
5788605
–10663
40215494
b8
–25
2512192
–120064
9852286
–8280
39260213
b9
0
0
–118690
14957445
10620
23325925
b10
3
–507903
–18203
20301435
22008
–1757787
b11
0
224751
24569234
348
–21028126
b12
103807
580196
26260385
–34123
–21293602
b13
0
893263
24247577
–25549
–3886901
b14
–10944
891396
18356231
33460
14396783
b15
293598
9668991
61387
16314388
b16
–987253
327749
–7546
1518875
b17
–2635779
–7171917
–94192
–12979500
b18
–3860322
–10926627
–50629
–11506007
b19
–3572512
–10379094
101135
2769794
b20
–822573
–6505618
134826
12195551
b21
4669054
–1333678
–56626
6103823
b22
12153698
2972773
–220104
–6709466
b23
19911100
5006366
–56082
–9882714
b24
25779390
4566808
263758
–353347
b25
27966862
2505652
231231
8629331
b26
25779390
126331
–215231
5597927
b27
19911100
–1496514
–430178
–4389168
b28
12153698
–1933830
34715
–7594158
b29
4669054
–1410695
580424
–428064
b30
–822573
–502731
283878
6566217
b31
–3572512
245330
–588382
4024593
b32
–3860322
565174
–693209
–3679749
b33
–2635779
492084
366118
–5572954
b34
–987253
231656
1084786
332589
b35
293598
–9196
132893
5136333
b36
891396
–125456
–1300087
2351253
b37
893263
–122207
–878642
–3357202
b38
580196
–61813
1162189
–3767666
b39
224751
–4445
1741565
1087392
b40
–18203
22484
–522533
3847821
b41
–118690
22245
–2490395
919792
b42
–120064
10775
–688945
–2918303
MINIMUM PHASE
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Table 34. FIR Stage Coefficients (continued)
SESSION 1
44
LINEAR PHASE
SCALING =
1/8388608
SESSION 3
SESSION 4
Scaling = 134217728
Scaling = 134217728
LINEAR
PHASE
MINIMUM PHASE
LINEAR
PHASE
MINIMUM PHASE
b43
–80934
940
2811738
–2193542
b44
–41280
–2953
2425494
1493873
b45
–16147
–2599
–2338095
2595051
b46
–4648
–1052
–4511116
–79991
b47
–874
–43
641555
–2260106
b48
–73
214
6661730
–963855
b49
0
132
2950811
1482337
b50
0
33
–8538057
1480417
b51
0
0
COEFFICIENT
LINEAR PHASE
SCALING = 1/512
SESSION 2
–10537298
–586408
b52
9818477
–1497356
b53
41426374
–168417
b54
56835776
1166800
b55
41426374
644405
b56
9818477
–675082
b57
–10537298
–806095
b58
–8538057
211391
b59
2950811
740896
b60
6661730
141976
b61
641555
–527673
b62
–4511116
–327618
b63
–2338095
278227
b64
2425494
363809
b65
2811738
–70646
b66
–688945
–304819
b67
–2490395
–63159
b68
–522533
205798
b69
1741565
124363
b70
1162189
–107173
b71
–878642
–131357
b72
–1300087
31104
b73
132893
107182
b74
1084786
15644
b75
366118
–71728
b76
–693209
–36319
b77
–588382
38331
b78
283878
38783
b79
580424
–13557
b80
34715
–31453
b81
–430178
–1230
b82
–215231
20983
b83
231231
7729
b84
263758
–11463
b85
–56082
–8791
b86
–220104
4659
b87
–56626
7126
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Table 34. FIR Stage Coefficients (continued)
SESSION 1
LINEAR PHASE
SCALING =
1/8388608
SESSION 3
SESSION 4
Scaling = 134217728
Scaling = 134217728
LINEAR
PHASE
MINIMUM PHASE
b88
134826
–732
b89
101135
–4687
b90
–50629
–976
b91
–94192
2551
b92
–7546
1339
b93
61387
–1103
b94
33460
–1085
b95
–25549
314
b96
–34123
681
b97
348
16
b98
22008
–349
COEFFICIENT
LINEAR PHASE
SCALING = 1/512
SESSION 2
LINEAR
PHASE
MINIMUM PHASE
b99
10620
–96
b100
–8280
144
b101
–10663
78
b102
–266
–46
b103
7419
–42
b104
6692
9
b105
2481
16
b106
–75
0
b107
–432
–4
b108
–132
0
b109
0
0
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ADS1282
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1+
www.ti.com
cos wN + sin wN - 1
1-2
cos wN
HPF Gain =
2-
cos wN + sin wN - 1
cos wN
(13)
See the HPF Stage section for an example of how to use this equation.
HPF Transfer Function
-1
2-a
1-Z
´
HPF(Z) =
-1
1 - bZ
2
(14)
where b is calculated as shown in Equation 15:
1 + (1 - a)2
b=
2
(15)
Table 35. tDR Time for Data Ready (Sinc Filter)
fDATA
(1)
fCLK
(1)
128k
440
64k
616
32k
968
16k
1672
8k
2824
For SYNC and Wake-Up commands, fCLK = number of CLK cycles from next rising CLK edge directly after eighth rising SCLK edge to
DRDY falling edge. For Wake-Up command only, subtract two fCLK cycles.
Table 35 is referenced by Table 10 and Table 12.
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15 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (August 2013) to Revision I
Page
•
Deleted low-power mode from data sheet, and made high-resolution mode default............................................................. 1
•
Changed MODE bit to 1 in register map after removing mode option ................................................................................. 35
•
Deleted MODE bit from CONFIG0; changed to 1 ............................................................................................................... 36
Changes from Revision G (May 2010) to Revision H
Page
•
Added ADS1282H device to data sheet................................................................................................................................. 1
•
Added Chop off test condition to Differential Input Impedance parameter............................................................................. 3
•
Added ADS1282H description .............................................................................................................................................. 12
Changes from Revision F (March 2009) to Revision G
Page
•
Corrected typical specification in Digital Filter Response, Minimum phase filter settling time in Electrical
Characteristics table ............................................................................................................................................................... 4
•
Corrected units typo of Figure 41 ......................................................................................................................................... 20
•
Moved Equation 14 and Equation 15 to the Appendix from the HPF Stage section ........................................................... 21
•
Added footnote 2 to Table 13, Ideal Output Code................................................................................................................ 26
•
Corrected sign typo in Equation 13 ...................................................................................................................................... 46
Changes from Revision E (October 2008) to Revision F
Page
•
Added tCMD specification for low-power mode in Modulator Output Timing table................................................................. 17
•
Updated Equation 7 ............................................................................................................................................................. 18
•
Updated Equation 8 ............................................................................................................................................................. 19
•
Updated Figure 41 ............................................................................................................................................................... 20
•
Minor graphical edits to Figure 47 ....................................................................................................................................... 23
•
Minor graphical edits to Figure 48 ....................................................................................................................................... 23
•
Changed 466/fCLK to 468/fCLK in tDR row of Table 10 ........................................................................................................... 23
•
Updated Figure 65, showing alternate bias resistor location................................................................................................ 41
•
Corrected Table 34 (Appendix, FIR Stage Coefficients) ...................................................................................................... 43
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47
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADS1282HIPW
ACTIVE
TSSOP
PW
28
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ADS1282HI
ADS1282HIPWR
ACTIVE
TSSOP
PW
28
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ADS1282HI
ADS1282IPW
ACTIVE
TSSOP
PW
28
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ADS1282
ADS1282IPWR
ACTIVE
TSSOP
PW
28
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ADS1282
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of