ADS7806UG4

ADS7806 ADS 780 6 ADS 780 6 SBAS021B – NOVEMBER 1992 – REVISED SEPTEMBER 2003 Low-Power 12-Bit Sampling CMOS ANALOG-to-DIGITAL CONVERTER FEATURES DESCRIPTION ● ● ● ● ● ● ● ● ● ● ● The ADS7806 is a low-power, 12-bit, sampling Analog-toDigital (A/D) converter using state of the art CMOS structures. It contains a complete 12-bit, capacitor-based, Successive Approximation Register (SAR) A/D converter with sampleand-hold, clock, reference, and a microprocessor interface with parallel and serial output drivers. POWER DISSIPATION: 35mW max POWER-DOWN MODE: 50µW ACQUISITION AND CONVERSION: 25µs max ±1/2 LSB MAX INL AND DNL 72dB MIN SINAD WITH 1kHz INPUT INPUT RANGES: ±10V, 0V to +5V, and 0V TO +4V SINGLE +5V SUPPLY OPERATION PARALLEL AND SERIAL DATA OUTPUT PIN-COMPATIBLE WITH THE 16-BIT ADS7807 USES INTERNAL OR EXTERNAL REFERENCE 0.3" DIP-28 AND SO-28 Clock The ADS7806 can acquire and convert to full 12-bit accuracy in 25µs max, while consuming only 35mW max. Laser trimmed scaling resistors provide standard industrial input ranges of ±10V and 0V to +5V. In addition, a 0V to +4V range allows development of complete single-supply systems. The ADS7806 is available in a 0.3" DIP-28 and SO-28, both fully specified for operation over the industrial –40°C to +85°C temperature range. R/C CS BYTE Power Down Successive Approximation Register and Control Logic 40kΩ CDAC R1IN BUSY Parallel 20kΩ 40kΩ Serial Data Clock and 10kΩ Comparator R2IN Serial Serial Data Data CAP Out Parallel Data Buffer 6kΩ REF 8 Internal +2.5V Ref Reference Power-Down Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright © 1992-2003, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com ELECTROSTATIC DISCHARGE SENSITIVITY ABSOLUTE MAXIMUM RATINGS(1) Analog Inputs: R1IN ............................................................................ ±12V R2IN ........................................................................... ±5.5V CAP ................................... VANA + 0.3V to AGND2 – 0.3V REF .......................................... Indefinite Short to AGND2, Momentary Short to VANA Ground Voltage Differences: DGND, AGND1, and AGND2 ............. ±0.3V VANA ....................................................................................................... 7V VDIG to VANA ...................................................................................... +0.3V VDIG ........................................................................................................ 7V Digital Inputs ............................................................. –0.3V to VDIG + 0.3V Maximum Junction Temperature ................................................... +165°C Internal Power Dissipation ............................................................. 825mW Lead Temperature (soldering, 10s) ............................................... +300°C This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. Exposure to absolute maximum conditions for extended periods may affect device reliability. PACKAGE/ORDERING INFORMATION PRODUCT MAXIMUM INTEGRAL LINEARITY ERROR (LSB) MINIMUM SIGNAL-TOSPECIFIED (NOISE + DISTORTION) PACKAGE TEMPERATURE RATIO (DB) PACKAGE-LEAD DESIGNATOR(1) RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY ADS7806P ADS7806PB ±0.9 ±0.45 70 72 DIP-28 NT –40°C to +85°C " " " ADS7806P ADS7806PB ADS7806P ADS7806PB Tubes, 13 Tubes, 13 ADS7806U " ±0.9 " 70 " SO-28 " DW " –40°C to +85°C " ADS7806U ADS7806U ADS7806U ADS7806U/1K Tubes, 28 Tape and Reel, 1000 ADS7806UB " ±0.45 " 72 " " " " " " " ADS7806UB ADS7806UB ADS7806UB Tubes, 28 ADS7806UB/1K Tape and Reel, 1000 NOTE: (1) For the most current specifications and package information, refer to our web site at www.ti.com. ELECTRICAL CHARACTERISTICS At TA = –40°C to +85°C, fS = 40kHz, VDIG = VANA = +5V, and using internal reference and fixed resistors (see Figure 7b), unless otherwise specified. ADS7806P, U PARAMETER CONDITIONS MIN TYP RESOLUTION DC ACCURACY Integral Linearity Error Differential Linearity Error No Missing Codes Transition Noise(2) Gain Error Full-Scale Error(3,4) Full-Scale Error Drift Full-Scale Error(3,4) Full-Scale Error Drift Bipolar Zero Error(3) Bipolar Zero Error Drift Unipolar Zero Error(3) Unipolar Zero Error Drift Recovery Time to Rated Accuracy from Power-Down(5) Power-Supply Sensitivity (VDIG = VANA = VS) 2 MAX MIN TYP 12 ANALOG INPUT Voltage Ranges Impedance Capacitance THROUGHPUT SPEED Conversion Time Complete Cycle Throughput Rate ADS7806PB, UB MAX UNITS ✻ Bits ±10, 0 to +5, 0 to +4 (See Table I) V ✻ 35 20 25 Acquire and Convert ±0.15 ±0.15 Tested 0.1 ±0.2 ±7 ±0.5 ±0.5 ±0.5 1 ±0.9 ±0.9 ±0.5 ±0.5 µs µs kHz ✻ ✻ ✻ ✻ ±0.1 ±0.45 ±0.45 LSB(1) LSB Bits LSB % % ppm/°C % ppm/°C mV ppm/°C mV ppm/°C ms ±5 ±0.25 ±0.25 ✻ ±10 ✻ ✻ ±3 ✻ ✻ ✻ ±0.5 +4.75V < VS < +5.25V ✻ ✻ ✻ 40 Ext. 2.5000V Ref Ext. 2.5000V Ref ±10V Range ±10V Range 0V to 5V, 0V to 4V Ranges 0V to 5V, 0V to 4V Ranges 2.2µF Capacitor to CAP pF ✻ LSB ADS7806 www.ti.com SBAS021B ELECTRICAL CHARACTERISTICS (Cont.) At TA = –40°C to +85°C, fS = 40kHz, VDIG = VANA = +5V, and using internal reference and fixed resistors (see Figure 7b), unless otherwise specified. ADS7806P, U PARAMETER AC ACCURACY Spurious-Free Dynamic Range Total Harmonic Distortion Signal-to-(Noise + Distortion) Signal-to-Noise Usable Bandwidth(7) Full-Power Bandwidth (–3dB) SAMPLING DYNAMICS Aperture Delay Aperture Jitter Transient Response Over-Voltage Recovery(8) REFERENCE Internal Reference Voltage Internal Reference Source Current (Must use external buffer.) Internal Reference Drift External Reference Voltage Range for Specified Linearity External Reference Current Drain DIGITAL INPUTS Logic Levels VIL VIH(9) IIL IIH DIGITAL OUTPUTS Data Format Data Coding VOL VOH Leakage Current CONDITIONS fIN = fIN = fIN = fIN = 1kHz, 1kHz, 1kHz, 1kHz, ±10V ±10V ±10V ±10V 80 90 –90 73 73 130 600 70 70 FS Step MAX MIN ✻ –80 72 72 No Load 2.48 2.5 1 8 2.5 External 2.5000V Ref VIL = 0V VIH = 5V Must be ≤ VANA ✻ 2.7 ✻ ✻ ✻ ✻ ✻ Parallel 12-bits in 2-bytes; Serial Binary Two’s Complement or Straight Binary +0.8 ✻ VD +0.3V ✻ ±10 ±10 +0.4 VANA = VDIG = 5V, fS = 40kHz REFD HIGH PWRD and REFD HIGH ✻ ✻ 2.52 ✻ +5 +5 0.6 5.0 28 23 50 –40 –55 –65 dB(6) dB dB dB kHz kHz ns ps µs ns V µA ppm/°C V ✻ µA ✻ ✻ ✻ ✻ V V µA µA ✻ ±5 ✻ V V µA 15 ✻ pF 83 83 ✻ ✻ ns ns ✻ ✻ V V mA mA mW mW µW +5.25 +5.25 ✻ ✻ 35 +85 +125 +150 75 75 UNITS ✻ ✻ +4 +4.75 +4.75 MAX ✻ 100 –0.3 +2.0 RL = 3.3kΩ, CL = 50pF RL = 3.3kΩ, CL = 10pF ✻ ✻ ✻ ✻ ✻ ✻ 5 2.3 DIGITAL TIMING Bus Access Time Bus Relinquish Time TYP ✻ ✻ 750 Output Capacitance TEMPERATURE RANGE Specified Performance Derated Performance Storage Thermal Resistance (θJA) DIP SO TYP 40 20 ISINK = 1.6mA ISOURCE = 500µA High-Z State, VOUT = 0V to VDIG High-Z State POWER SUPPLIES Specified Performance VDIG VANA IDIG IANA Power Dissipation MIN ADS7806PB, UB ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ °C °C °C °C/W °C/W ✻ Specifications same as ADS7806 P, U. NOTES: (1) LSB means Least Significant Bit. One LSB for the ±10V input range is 4.88mV. (2) Typical rms noise at worst-case transition. (3) As measured with fixed resistors, see Figure 7b. Adjustable to zero with external potentiometer. (4) Full-scale error is the worst case of –Full-Scale or +Full-Scale untrimmed deviation from ideal first and last code transitions, divided by the transition voltage (not divided by the full-scale range) and includes the effect of offset error. (5) This is the time delay after the ADS7806 is brought out of Power-Down mode until all internal settling occurs and the analog input is acquired to rated accuracy. A Convert command after this delay will yield accurate results. (6) All specifications in dB are referred to a full-scale input. (7) Usable bandwidth defined as full-scale input frequency at which Signal-to-(Noise + Distortion) degrades to 60dB. (8) Recovers to specified performance after 2 • FS input overvoltage. (9) The minimum VIH level for the DATACLK signal is 3V. ADS7806 SBAS021B www.ti.com 3 PIN DESCRIPTIONS DIGITAL I/O PIN # NAME 1 2 3 4 5 6 7 8 9 R1IN AGND1 R2IN CAP REF AGND2 SB/BTC EXT/INT D7 10 11 12 13 14 15 16 17 18 19 20 21 22 D6 D5 D4 D3 DGND D2 D1 D0 DATACLK SDATA TAG BYTE R/C O O O I/O O I I I 23 CS I 24 BUSY O 25 26 27 28 PWRD REFD VANA VDIG I I DESCRIPTION Analog Input. See Figure 7. Analog Sense Ground Analog Input. See Figure 7. Reference Buffer Output. 2.2µF tantalum capacitor to ground. Reference Input/Output. 2.2µF tantalum capacitor to ground. Analog Ground Selects Straight Binary or Binary Two’s Complement for Output Data Format. External/Internal data clock select. Data Bit 3 if BYTE is HIGH. Data bit 11 (MSB) if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. Leave unconnected when using serial output. Data Bit 2 if BYTE is HIGH. Data bit 10 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. Data Bit 1 if BYTE is HIGH. Data bit 9 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. Data Bit 0 (LSB) if BYTE is HIGH. Data bit 8 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. LOW if BYTE is HIGH. Data bit 7 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. Digital Ground LOW if BYTE is HIGH. Data bit 6 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. LOW if BYTE is HIGH. Data bit 5 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. LOW if BYTE is HIGH. Data bit 4 if BYTE is LOW. Hi-Z when CS is HIGH and/or R/C is LOW. Data Clock Output when EXT/INT is LOW. Data clock input when EXT/INT is HIGH. Serial Output Synchronized to DATACLK Serial Input When Using an External Data Clock Selects 8 most significant bits (LOW) or 4 least significant bits (HIGH) on parallel output pins. With CS LOW and BUSY HIGH, a Falling Edge on R/C Initiates a New Conversion. With CS LOW, a rising edge on R/C enables the parallel output. Internally OR’ed with R/C. If R/C is LOW, a falling edge on CS initiates a new conversion. If EXT/INT is LOW, this same falling edge will start the transmission of serial data results from the previous conversion. At the start of a conversion, BUSY goes LOW and stays LOW until the conversion is completed and the digital outputs have been updated. PWRD HIGH shuts down all analog circuitry except the reference. Digital circuitry remains active. REFD HIGH shuts down the internal reference. External reference will be required for conversions. Analog Supply. Nominally +5V. Decouple with 0.1µF ceramic and 10µF tantalum capacitors. Digital Supply. Nominally +5V. Connect directly to pin 27. Must be ≤ VANA. I I O O O O O PIN CONFIGURATION Top View DIP, SO R1IN 1 28 VDIG AGND1 2 27 VANA R2IN 3 26 REFD CAP 4 25 PWRD REF 5 24 BUSY AGND2 6 23 CS SB/BTC 7 ANALOG INPUT RANGE CONNECT R1IN VIA 200Ω TO CONNECT R2IN VIA 100Ω TO IMPEDANCE ±10V 0V to 5V 0V to 4V VIN AGND VIN CAP VIN VIN 45.7kΩ 20.0kΩ 21.4kΩ TABLE I. Input Range Connections. See Figure 7. 22 R/C ADS7806 4 EXT/INT 8 21 BYTE D7 9 20 TAG D6 10 19 SDATA D5 11 18 DATACLK D4 12 17 D0 D3 13 16 D1 DGND 14 15 D2 ADS7806 www.ti.com SBAS021B TYPICAL CHARACTERISTICS TA = +25°C, fS = 40kHz, VDIG = VANA = +5V, and using internal reference and fixed resistors (see Figure 7b), unless otherwise specified. FREQUENCY SPECTRUM (8192 Point FFT; fIN = 15kHz, 0dB) 0 0 –20 –20 –40 –40 Amplitude (dB) –60 –80 –60 –80 –100 –100 –120 –120 0 5 10 15 0 20 5 10 Frequency (kHz) 90 90 80 80 0dB 70 70 60 SINAD (dB) SINAD (dB) 20 SIGNAL-TO-(NOISE + DISTORTION) vs INPUT FREQUENCY AND INPUT AMPLITUDE SIGNAL-TO-(NOISE + DISTORTION) vs INPUT FREQUENCY (fIN = 0dB) 60 50 40 –20dB 50 40 30 30 20 20 10 –60dB 0 10 100 1k 10k 100k 0 1M 2 4 6 8 10 12 14 16 Input Signal Frequency (Hz) Input Signal Frequency (kHz) SIGNAL-TO-(NOISE + DISTORTION) vs TEMPERATURE (fIN = 1kHz, 0dB; fS = 10kHz to 40kHz) AC PARAMETERS vs TEMPERATURE (fIN = 1kHz, 0dB) 74.0 110 73.9 10kHz 73.8 40kHz 73.7 73.6 –75 SFDR, SNR, and SINAD (dB) 20kHz 20 –60 –65 100 –70 95 –75 90 –80 85 –85 80 –90 SNR and SINAD 75 –95 70 –100 THD 65 –105 60 –50 –25 0 25 50 75 Temperature (°C) 100 125 150 –110 –75 –50 –25 0 25 50 75 100 125 150 Temperature (°C) ADS7806 SBAS021B 18 SFDR 105 30kHz SINAD (dB) 15 Frequency (kHz) THD (dB) Amplitude (dB) FREQUENCY SPECTRUM (8192 Point FFT; fIN = 1kHz, 0dB) www.ti.com 5 TYPICAL CHARACTERISTICS (Cont.) TA = +25°C, fS = 40kHz, VDIG = VANA = +5V, and using internal reference and fixed resistors (see Figure 7b), unless otherwise specified. POWER-SUPPLY RIPPLE SENSITIVITY INL/DNL DEGRADATION PER LSB OF P-P RIPPLE 1 Linearity Degradation (LSB/LSB) 12-Bit (LSBs) 0.10 0 All Codes INL –0.10 12-Bit (LSBs) 0.10 0 –0.10 All Codes DNL 0 512 10–1 10–2 INL 10–3 10–4 DNL 10–5 1024 1536 2048 2560 3072 3584 4095 101 102 mV From Ideal Percent From Ideal BPZ Error 0 +FS Error –FS Error 0 –0.20 –75 –50 –25 0 25 50 75 100 125 3 2 1 0 –1 –2 0.40 107 0.20 0 0.40 +FS Error (4V Range) –FS Error (5V Range) 0.20 0 150 –75 –50 –25 0 25 50 75 100 125 150 125 150 Temperature (°C) INTERNAL REFERENCE VOLTAGE vs TEMPERATURE CONVERSION TIME vs TEMPERATURE 2.520 15.10 2.515 15.00 2.510 Conversion Time (µs) Internal Reference (V) 106 UPO Error Temperature (°C) 2.505 2.500 2.495 2.490 2.485 14.90 14.80 14.70 14.60 14.50 14.40 14.30 14.20 2.480 –75 –50 –25 0 25 50 75 100 125 150 –75 –50 –25 0 25 50 75 100 Temperature (°C) Temperature (°C) 6 105 ENDPOINT ERRORS (Unipolar Ranges) Percent From Ideal Percent From Ideal Percent From Ideal mV From Ideal ENDPOINT ERRORS (20V Bipolar Range) –0.20 0.20 104 Power-Supply Ripple Frequency (Hz) Decimal Code 3 2 1 0 –1 –2 0.20 103 ADS7806 www.ti.com SBAS021B BASIC OPERATION PARALLEL OUTPUT Figure 1a shows a basic circuit to operate the ADS7806 with a ±10V input range and parallel output. Taking R/C (pin 22) LOW for 40ns (12µs max) will initiate a conversion. BUSY (pin 24) will go LOW and stay LOW until the conversion is completed and the output register is updated. If BYTE (pin 21) is LOW, the eight Most Significant Bits (MSBs) will be valid when BUSY rises; if BYTE is HIGH, the four Least Significant Bits (LSBs) will be valid when BUSY rises. Data will be output in Binary Two’s Complement (BTC) format. BUSY going HIGH can be used to latch the data. After the first byte has been read, BYTE can be toggled allowing the remaining byte to be read. All convert commands will be ignored while BUSY is LOW. The ADS7806 will begin tracking the input signal at the end of the conversion. Allowing 25µs between convert commands assures accurate acquisition of a new signal. The offset and gain are adjusted internally to allow external trimming with a single supply. The external resistors compensate for this adjustment and can be left out if the offset and gain will be corrected in software (refer to the Calibration section). SERIAL OUTPUT Figure 1b shows a basic circuit to operate the ADS7806 with a ±10V input range and serial output. Taking R/C (pin 22) LOW for 40ns (12µs max) will initiate a conversion and output valid data from the previous conversion on SDATA (pin 19) synchronized to 12 clock pulses output on DATACLK (pin 18). BUSY (pin 24) will go LOW and stay LOW until the conversion is completed and the serial data has been transmitted. Data will be output in BTC format, MSB first, and will be valid on both the rising and falling edges of the data clock. BUSY going HIGH can be used to latch the data. All convert commands will be ignored while BUSY is LOW. The ADS7806 will begin tracking the input signal at the end of the conversion. Allowing 25µs between convert commands assures accurate acquisition of a new signal. The offset and gain are adjusted internally to allow external trimming with a single supply. The external resistors compensate for this adjustment and can be left out if the offset and gain will be corrected in software (refer to the Calibration section). STARTING A CONVERSION The combination of CS (pin 23) and R/C (pin 22) LOW for a minimum of 40ns immediately puts the sample-and-hold of the ADS7806 in the hold state and starts conversion ‘n’. BUSY (pin 24) will go LOW and stay LOW until conversion ‘n’ is completed and the internal output register has been updated. All new convert commands during BUSY LOW will be ignored. CS and/or R/C must go HIGH before BUSY goes HIGH, or a new conversion will be initiated without sufficient time to acquire a new signal. Serial Output Parallel Output 200Ω ±10V 1 66.5kΩ 2 27 3 26 2.2µF +5V 0.1µF 10µF + + ±10V 4 2.2µF + 5 25 6 23 66.5kΩ 100Ω BUSY 24 Convert Pulse 2.2µF R/C 7 Pin 21 B11 B10 LOW (MSB) B9 Pin 21 HIGH B1 B3 B2 B8 B7 B0 LOW (LSB) 2.2µF + BYTE 21 9 20 10 19 11 18 12 17 28 2 27 3 26 4 25 5 24 6 23 7 40ns min 0.1µF 10µF + + 22 NC(1) 8 21 9 20 NC(1) 10 19 NC(1) 13 16 NC(1) 14 15 NC(1) 12 17 NC(1) NC(1) 13 16 NC(1) 14 15 NC(1) B6 B5 +5V BUSY Convert Pulse R/C ADS7806 B4 11 18 40ns min SDATA DATACLK LOW LOW LOW NOTE: (1) These pins should be left unconnected. They will be active when R/C is HIGH. NOTE: (1) SDATA (pin 19) is always active. FIGURE 1a. Basic ±10V Operation, both Parallel and Serial Output. FIGURE 1b. Basic ±10V Operation with Serial Output. ADS7806 SBAS021B + 22 ADS7806 8 1 +5V +5V 100Ω + 200Ω 28 www.ti.com 7 The ADS7806 will begin tracking the input signal at the end of the conversion. Allowing 25µs between convert commands assures accurate acquisition of a new signal. Refer to Tables II and III for a summary of CS , R/C, and BUSY states, and Figures 2 through 6 for timing diagrams. CS R/C BUSY 1 X X OPERATION None. Databus is in Hi-Z state. ↓ 0 1 Initiates conversion ‘n’. Databus remains in Hi-Z state. 0 ↓ 1 Initiates conversion ‘n’. Databus enters Hi-Z state. 0 1 ↑ Conversion ‘n’ completed. Valid data from conversion ‘n’ on the databus. ↓ 1 1 Enables databus with valid data from conversion ‘n’. ↓ 1 0 Enables databus with valid data from conversion ‘n – 1’(1). Conversion ‘n’ in progress. 0 ↑ 0 Enables databus with valid data from conversion ‘n – 1’(1). Conversion ‘n’ in progress. CS and R/C are internally OR’ed and level triggered. There is no requirement which input goes LOW first when initiating a conversion. If, however, it is critical that CS or R/C initiates conversion ‘n’, be sure the less critical input is LOW at least 10ns prior to the initiating input. If EXT/INT (pin 8) is LOW when initiating conversion ‘n’, serial data from conversion ‘n – 1’ will be output on SDATA (pin 19) following the start of conversion ‘n’. See Internal Data Clock in the Reading Data section. To reduce the number of control pins, CS can be tied LOW using R/C to control the read and convert modes. This will have no effect when using the internal data clock in the serial output mode. The parallel output and the serial output (only when using an external data clock), however, will be affected whenever R/C goes HIGH. Refer to the Reading Data section. 0 0 ↑ New conversion initiated without acquisition of a new signal. Data will be invalid. CS and/or R/C must be HIGH when BUSY goes HIGH. X X 0 New convert commands ignored. Conversion ‘n’ in progress. NOTE: (1) See Figures 2 and 3 for constraints on data valid from conversion ‘n – 1’. TABLE II. Control Functions When Using Parallel Output (DATACLK tied LOW, EXT/INT tied HIGH). READING DATA The ADS7806 outputs serial or parallel data in Straight Binary (SB) or Binary Two’s Complement data output format. If SB/BTC (pin 7) is HIGH, the output will be in SB format, and if LOW, the output will be in BTC format. Refer to Table IV for ideal output codes. The parallel output can be read without affecting the internal output registers; however, reading the data through the serial port will shift the internal output registers one bit per data clock pulse. As a result, data can be read on the parallel port CS R/C BUSY EXT/INT DATACLK ↓ 0 1 0 Output OPERATION Initiates conversion ‘n’. Valid data from conversion ‘n – 1’ clocked out on SDATA. 0 ↓ 1 0 Output Initiates conversion ‘n’. Valid data from conversion ‘n – 1’ clocked out on SDATA. ↓ 0 1 1 Input Initiates conversion ‘n’. Internal clock still runs conversion process. 0 ↓ 1 1 Input Initiates conversion “n”. Internal clock still runs conversion process. ↓ 1 1 1 Input Conversion ‘n’ completed. Valid data from conversion ‘n’ clocked out on SDATA synchronized to external data clock. ↓ 1 0 1 Input Valid data from conversion ‘n – 1’ output on SDATA synchronized to external data clock. Conversion ‘n’ in progress. 0 ↑ 0 1 Input Valid data from conversion ‘n – 1’ output on SDATA synchronized to external data clock. Conversion ‘n’ in progress. 0 0 ↑ X X New conversion initiated without acquisition of a new signal. Data will be invalid. CS and/or R/C must be HIGH when BUSY goes HIGH. X X 0 X X New convert commands ignored. Conversion “n” in progress. NOTE: (1) See Figures 4, 5, and 6 for constraints on data valid from conversion ‘n – 1’. TABLE III. Control Functions When Using Serial Output. DESCRIPTION Full-Scale Range Least Significant Bit (LSB) ANALOG INPUT ±10 4.88mV 0V to 5V 1.22mV 0V to 4V 976µV DIGITAL OUTPUT BINARY TWO’S COMPLEMENT STRAIGHT BINARY (SB/BTC LOW) (SB/BTC HIGH) HEX +Full-Scale (FS – 1LSB) Midscale One LSB Below Midscale –Full-Scale HEX BINARY CODE CODE BINARY CODE 9.99512V 4.99878V 3.999024V 0111 1111 1111 1111 7FF 1111 1111 1111 1111 CODE FFF 0V 2.5V 2V 0000 0000 0000 0000 000 1000 0000 0000 0000 800 –4.88mV 2.49878V 1.999024V 1111 1111 1111 1111 FFF 0111 1111 1111 1111 7FF –10V 0V 0V 1000 0000 0000 0000 800 0000 0000 0000 0000 000 TABLE IV. Output Codes and Ideal Input Voltages. 8 ADS7806 www.ti.com SBAS021B prior to reading the same data on the serial port, but data cannot be read through the serial port prior to reading the same data on the parallel port. PARALLEL OUTPUT To use the parallel output, tie EXT/INT (pin 8) HIGH and DATACLK (pin 18) LOW. SDATA (pin 19) should be left unconnected. The parallel output will be active when R/C (pin 22) is HIGH and CS (pin 23) is LOW. Any other combination of CS and R/C will tri-state the parallel output. Valid conversion data can be read in two 8-bit bytes on D7D0 (pins 9-13 and 15-17). When BYTE (pin 21) is LOW, the eight most significant bits will be valid with the MSB on D7. When BYTE is HIGH, the four least significant bits will be valid with the LSB on D4. BYTE can be toggled to read both bytes within one conversion cycle. PARALLEL OUTPUT (AFTER A CONVERSION) After conversion ‘n’ is completed and the output registers have been updated, BUSY (pin 24) will go HIGH. Valid data from conversion ‘n’ will be available on D7-D0 (pins 9-13 and 15-17). BUSY going high can be used to latch the data. Refer to Table V and Figures 2 and 3 for timing constraints. PARALLEL OUTPUT (DURING A CONVERSION) After conversion ‘n’ has been initiated, valid data from conversion ‘n – 1’ can be read and will be valid up to 12µs after the start of conversion ‘n’. Do not attempt to read data beyond 12µs after the start of conversion ‘n’ until BUSY (pin 24) goes HIGH; this may result in reading invalid data. Refer to Table V and Figures 2 and 3 for timing constraints. Upon initial power up, the parallel output will contain indeterminate data. t1 t1 R/C t3 t3 t4 BUSY t5 t6 t6 t7 Convert Acquire MODE t8 Acquire t12 t11 Parallel Data Bus Previous High Byte Valid Previous High Byte Valid Hi-Z Convert t12 t10 Previous Low Byte Valid Not Valid Low Byte Valid High Byte Valid Hi-Z t9 t2 t12 t12 t9 High Byte Valid t12 t12 BYTE FIGURE 2. Conversion Timing with Parallel Output (CS and DATACLK tied LOW, EXT/INT tied HIGH). t21 t21 t21 t21 t21 t21 t21 t21 t21 t21 R/C t1 CS t3 BUSY t4 BYTE DATA BUS Hi-Z State High Byte t12 Hi-Z State t9 Low Byte t12 Hi-Z State t9 FIGURE 3. Using CS to Control Conversion and Read Timing with Parallel Outputs. ADS7806 SBAS021B www.ti.com 9 SYMBOL DESCRIPTION t1 Convert Pulse Width t2 Data Valid Delay after R/C LOW t3 BUSY Delay from INTERNAL DATA CLOCK MIN TYP MAX UNITS 0.04 14.7 12 µs 20 µs 85 ns 20 µs (During A Conversion) To use the internal data clock, tie EXT/INT (pin 8) LOW. The combination of R/C (pin 22) and CS (pin 23) LOW will initiate conversion ‘n’ and activate the internal data clock (typically a 900kHz clock rate). The ADS7806 will output 12 bits of valid data, MSB first, from conversion ‘n – 1’ on SDATA (pin 19), synchronized to 12 clock pulses output on DATACLK (pin 18). The data will be valid on both the rising and falling edges of the internal data clock. The rising edge of BUSY (pin 24) can be used to latch the data. After the 12th clock pulse, DATACLK will remain LOW until the next conversion is initiated, while SDATA will go to whatever logic level was input on TAG (pin 20) during the first clock pulse. Refer to Table VI and Figure 4. Start of Conversion t4 BUSY LOW 14.7 t5 BUSY Delay after 90 ns End of Conversion t6 Aperture Delay 40 t7 Conversion Time 14.7 ns 20 µs 5 µs 83 ns t8 Acquisition Time t9 Bus Relinquish Time 10 t10 BUSY Delay after Data Valid 20 60 ns t11 Previous Data Valid 12 14.7 µs t12 Bus Access Time and BUSY Delay t13 Start of Conversion after Start of Conversion 83 ns 1.4 µs to DATACLK Delay t14 DATACLK Period t15 Data Valid to DATACLK 1.1 µs EXTERNAL DATA CLOCK 20 75 ns 400 600 ns To use an external data clock, tie EXT/INT (pin 8) HIGH. The external data clock is not a conversion clock; it can only be used as a data clock. To enable the output mode of the ADS7806, CS (pin 23) must be LOW and R/C (pin 22) must be HIGH. DATACLK must be HIGH for 20% to 70% of the total data clock period; the clock rate can be between DC and 10MHz. Serial data from conversion ‘n’ can be output on SDATA (pin 19) after conversion ‘n’ is completed or during conversion ‘n + 1’. HIGH Delay t16 Data Valid after DATACLK LOW Delay t17 External DATACLK Period 100 ns t18 External DATACLK LOW 40 ns t19 External DATACLK HIGH 50 ns t20 CS and R/C to External 25 ns ns DATACLK Setup Time t21 R/C to CS Setup Time 10 t22 Valid Data after DATACLK HIGH 25 t7 + t8 Throughput Time ns 25 An obvious way to simplify control of the converter is to tie CS LOW and use R/C to initiate conversions. While this is perfectly acceptable, there is a possible problem when using an external data clock. At an indeterminate point from 12µs after the start of conversion ‘n’ until BUSY rises, the internal logic will shift the results of conversion ‘n’ into the output register. If CS is LOW, R/C is HIGH, and the external clock is HIGH at this point, data will be lost. So, with CS LOW, either R/C and/or DATACLK must be LOW during this period to avoid losing valid data. µs TABLE V. Conversion and Data Timing. TA = –40°C to +85°C. SERIAL OUTPUT Data can be clocked out with the internal data clock or an external data clock. When using serial output, be careful with the parallel outputs, D7-D0 (pins 9-13 and 15-17), as these pins will come out of Hi-Z state whenever CS (pin 23) is LOW and R/C (pin 22) is HIGH. The serial output can not be tristated and is always active. t 7 + t8 CS or R/C(1) t14 DATACLK t13 1 2 3 11 12 1 2 Bit 9 Valid Bit 1 Valid LSB Valid MSB Valid Bit 10 Valid t16 t15 MSB Valid Bit 10 Valid SDATA (Results from previous conversion.) BUSY NOTE: (1) If controlling with CS , tie R/C LOW. Data bus pins will remain Hi-Z at all times. If controlling with R/C, tie CS LOW. Data bus pins will be active when R/C is HIGH, and should be left unconnected. FIGURE 4. Serial Data Timing Using Internal Data Clock (TAG tied LOW). 10 ADS7806 www.ti.com SBAS021B FIGURE 5. Conversion and Read Timing with External Clock (EXT/INT Tied HIGH) Read after Conversion. ADS7806 SBAS021B www.ti.com 11 TAG SDATA BUSY R/C CS EXTERNAL DATACLK t21 t3 t1 0 t18 t17 Tag 0 t19 t21 1 t22 Tag 1 Bit 11 (MSB) t20 2 Tag 2 Bit 10 3 11 Tag 11 Bit 1 12 Tag 12 Bit 0 (LSB) 13 Tag 13 Tag 0 14 Tag 14 Tag 1 t20 EXTERNAL DATA CLOCK TAG FEATURE (After a Conversion) TAG (pin 20) inputs serial data synchronized to the external or internal data clock. After conversion ‘n’ is completed and the output registers have been updated, BUSY (pin 24) will go HIGH. With CS LOW and R/C HIGH, valid data from conversion ‘n’ will be output on SDATA (pin 19) synchronized to the external data clock input on DATACLK (pin 18). The MSB will be valid on the first falling edge and the second rising edge of the external data clock. The LSB will be valid on the 12th falling edge and 13th rising edge of the data clock. TAG (pin 20) will input a bit of data for every external clock pulse. The first bit input on TAG will be valid on SDATA on the 13th falling edge and the 14th rising edge of DATACLK; the second input bit will be valid on the 14th falling edge and the 15th rising edge, etc. With a continuous data clock, TAG data will be output on SDATA until the internal output registers are updated with the results from the next conversion. Refer to Table V and Figure 5. EXTERNAL DATA CLOCK (During a Conversion) After conversion ‘n’ has been initiated, valid data from conversion ‘n – 1’ can be read and will be valid up to 12µs after the start of conversion ‘n’. Do not attempt to clock out data from 12µs after the start of conversion ‘n’ until BUSY (pin 24) rises; this will result in data loss. NOTE: For the best possible performance when using an external data clock, data should not be clocked out during a conversion. The switching noise of the asynchronous data clock can cause digital feedthrough degrading the converter’s performance. Refer to Table VI and Figure 6. When using an external data clock, the serial bit stream input on TAG will follow the LSB output on SDATA until the internal output register is updated with new conversion results. See Table VI and Figures 5 and 6. The logic level input on TAG for the first rising edge of the internal data clock will be valid on SDATA after all 12 bits of valid data have been output. INPUT RANGES The ADS7806 offers three input ranges: standard ±10V, 0V-5V, and a 0V-4V range for complete, single-supply systems. See Figures 7a and 7b for the necessary circuit connections for implementing each input range and optional offset and gain adjust circuitry. Offset and full-scale error(1) specifications are tested with the fixed resistors, see Figure 7b. Adjustments for offset and gain are described in the Calibration section of this data sheet. The offset and gain are adjusted internally to allow external trimming with a single supply. The external resistors compensate for this adjustment and can be left out if the offset and gain will be corrected in software (refer to the Calibration section). The input impedance, summarized in Table II, results from the combination of the internal resistor network (see the front page of this data sheet) and the external resistors used for NOTE: (1) Full-scale error includes offset and gain errors measured at both +FS and –FS. t17 t18 t19 EXTERNAL DATACLK t20 t22 CS t21 t20 R/C t1 t11 BUSY t3 DATA TAG Tag 0 Bit 11 (MSB) Bit 0 (LSB) Tag 0 Tag 1 Tag 1 Tag 12 Tag 13 Tag 14 FIGURE 6. Conversion and Read Timing with External Clock (EXT/INT tied HIGH) Read During a Conversion. 12 ADS7806 www.ti.com SBAS021B each input range (see Figure 8). The input resistor divider network provides inherent over-voltage protection to at least ±12V on R1IN and ±5.5V on R2IN. Analog inputs above or below the expected range will yield either positive full-scale or negative full-scale digital outputs, respectively. There will be no wrapping or folding over for analog inputs outside the nominal range. GAIN ADJUST RANGE (mV) ±10V ±15 ±60 0 to 5V ±4 ±30 0 to 4V ±3 ±30 TABLE VI. Offset and Gain Adjust Ranges for Hardware Calibration (see Figure 7a). CALIBRATION are necessary. See the No Calibration section for more details on the external resistors. Refer to Table VII for the range of offset and gain errors with and without the external resistors. HARDWARE CALIBRATION To calibrate the offset and gain of the ADS7806 in hardware, install the resistors shown in Figure 7a. Table VI lists the hardware trim ranges relative to the input for each input range. NO CALIBRATION Figure 7b shows circuit connections. Note that the actual voltage dropped across the external resistors is at least two orders of magnitude lower than the voltage dropped across the internal resistor divider network. This should be consid- SOFTWARE CALIBRATION To calibrate the offset and gain in software, no external resistors are required. However, to get the data sheet specifications for offset and gain, the resistors shown in Figure 7b ±10V OFFSET ADJUST RANGE (mV) INPUT RANGE 0V-5V 0V-4V 33.2kΩ 200Ω 200Ω 1 VIN 2 R1IN 1 2 AGND1 100Ω 3 R1IN 200Ω 1 33.2kΩ 3 VIN R2IN 100Ω 2 VIN R2IN 100Ω 3 +5V 4 33.2kΩ 2.2µF +5V + 1MΩ 50kΩ 2.2µF + 50kΩ 50kΩ 6 + 2.2µF 50kΩ REF + 1MΩ 2.2µF AGND2 AGND1 R2IN +5V CAP 4 5 REF + 6 +5V CAP 2.2µF 5 50kΩ 4 R1IN AGND1 1MΩ 5 50kΩ REF + 2.2µF AGND2 CAP 6 AGND2 FIGURE 7a. Circuit Diagrams (With Hardware Trim). ±10V 0V-5V 0V-4V 33.2kΩ 200Ω 200Ω 1 VIN 2 R1IN 1 2 AGND1 100Ω 66.5kΩ 3 +5V R2IN R1IN 200Ω 1 33.2kΩ 3 VIN 2 VIN R2IN 100Ω 3 100Ω 4 2.2µF + CAP 2.2µF 5 2.2µF 4 5 REF + 6 + 2.2µF CAP 4 2.2µF + REF 5 + AGND2 6 R1IN AGND1 AGND2 2.2µF AGND1 R2IN CAP REF + 6 AGND2 FIGURE 7b. Circuit Diagrams (Without Hardware Trim). ADS7806 SBAS021B www.ti.com 13 ered when choosing the accuracy and drift specifications of the external resistors. In most applications, 1% metal-film resistors will be sufficient. The external resistors, see Figure 7b, may not be necessary in some applications. These resistors provide compensation for an internal adjustment of the offset and gain which allows calibration with a single supply. Not using the external resistors will result in offset and gain errors in addition to those listed in the electrical characteristics section. Offset refers to the equivalent voltage of the digital output when converting with the input grounded. A positive gain error occurs when the equivalent output voltage of the digital output is larger than the analog input. Refer to Table VII for nominal ranges of gain and offset errors with and without the external resistors. Refer to Figure 8 for typical shifts in the transfer functions which occur when the external resistors are removed. To further analyze the effects of removing any combination of the external resistors, consider Figure 9. The combination of the external and the internal resistors form a voltage divider which reduces the input signal to a 0.3125V to 2.8125V input range at the Capacitor Digital-to-Analog Converter (CDAC). The internal resistors are laser trimmed to high relative accuracy to meet full specifications. The actual input impedance of the internal resistor network looking into pin 1 or pin 3, however, is only accurate to ±20% due to process variations. This should be taken into account when determining the effects of removing the external resistors. REFERENCE The ADS7806 can operate with its internal 2.5V reference or an external reference. By applying an external reference to pin 5, the internal reference can be bypassed; REFD (pin 26) OFFSET ERROR WITH RESISTORS INPUT RANGE (V) GAIN ERROR WITHOUT RESISTORS WITH RESISTORS WITHOUT RESISTORS RANGE (mV) RANGE (mV) TYP (mV) RANGE (% FS) RANGE (% FS) TYP –10 ≤ BPZ ≤ 10 0 ≤ BPZ ≤ 35 +15 –0.4 ≤ G ≤ 0.4 0.15 ≤ G(1) ≤ 0.15 –0.3 ≤ G ≤ 0.5 –0.1 ≤ G(1) ≤ 0.2 +0.05 +0.05 0 to 5 –3 ≤ UPO ≤ 3 –12 ≤ UPO ≤ –3 –7.5 –0.4 ≤ G ≤ 0.4 0.15 ≤ G(1) ≤ 0.15 –1.0 ≤ G ≤ 0.1 –0.55 ≤ G(1) ≤ –0.05 –0.2 –0.2 0 to 4 –3 ≤ UPO ≤ 3 –10.5 ≤ UPO ≤ –1.5 –6 –0.4 ≤ G ≤ 0.4 –0.15 ≤ G(1) ≤ 0.15 –1.0 ≤ G ≤ 0.1 –0.55 ≤ G(1) ≤ –0.05 –0.2 –0.2 ±10 NOTE: (1) High Grade. TABLE VII. Range of Offset and Gain Errors With and Without External Resistors (a) Bipolar (b) Unipolar Digital Output Digital Output +Full-Scale +Full-Scale Analog Input –Full-Scale Analog Input –Full-Scale Typical Transfer Functions With External Resistors Typical Transfer Functions Without External Resistors FIGURE 8. Typical Transfer Functions With and Without External Resistors. 14 ADS7806 www.ti.com SBAS021B 200Ω 39.8kΩ VIN CDAC (High Impedance) (0.3125V to 2.8125V) 66.5kΩ 40kΩ 20kΩ 9.9kΩ +5V 100Ω +2.5V +2.5V 200Ω 39.8kΩ CDAC (High Impedance) (0.3125V to 2.8125V) 33.2kΩ 100Ω 20kΩ 9.9kΩ 40kΩ VIN +2.5V 200Ω +2.5V 39.8kΩ VIN CDAC (High Impedance) (0.3125V to 2.8125V) 33.2kΩ 100Ω 40kΩ 20kΩ 9.9kΩ +2.5V +2.5V FIGURE 9. Circuit Diagrams Showing External and Internal Resistors. tied HIGH will power-down the internal reference reducing the overall power consumption of the ADS7806 by approximately 5mW. ZCAP The internal reference has approximately an 8 ppm/°C drift (typical) and accounts for approximately 20% of the full-scale error (FSE = ±0.5% for low grade, ±0.25% for high grade). CAP (Pin 4) The ADS7806 also has an internal buffer for the reference voltage. Figure 10 shows characteristic impedances at the input and output of the buffer with all combinations of powerdown and reference down. CDAC Buffer Internal Reference REF (Pin 5) ZREF REF REF (pin 5) is an input for an external reference or the output for the internal 2.5V reference. A 2.2µF tantalum capacitor should be connected as close as possible to the REF pin from ground. This capacitor and the output resistance of REF create a low-pass filter to bandlimit noise on the reference. Using a smaller value capacitor will introduce more noise to the reference, degrading the SNR and SINAD. The REF pin should not be used to drive external AC or DC loads, as shown in Figure 10. The range for the external reference is 2.3V to 2.7V and determines the actual LSB size. Increasing the reference voltage will increase the full-scale range and the LSB size of the converter which can improve the SNR. PWRD 0 REFD 1 PWRD 1 REFD 0 PWRD 1 REFD 1 ZCAP (Ω) 1 1 200 200 ZREF (Ω) 6k 100M 6k 100M FIGURE 10. Characteristic Impedances of Internal Buffer. CAP CAP (pin 4) is the output of the internal reference buffer. A 2.2µF tantalum capacitor should be placed as close as possible to the CAP pin from ground to provide optimum switching currents for the CDAC throughout the conversion ADS7806 SBAS021B PWRD 0 REFD 0 www.ti.com 15 cycle. This capacitor also provides compensation for the output of the buffer. Using a capacitor any smaller than 1µF can cause the output buffer to oscillate and may not have sufficient charge for the CDAC. Capacitor values larger than 2.2µF will have little affect on improving performance. See Figures 10 and 11. 7000 6000 5000 REFD HIGH will power-down the internal 2.5V reference. All other analog circuitry, including the reference buffer, will be active. REFD should be HIGH when using an external reference to minimize power consumption and the loading effects on the external reference. See Figure 10 for the characteristic impedance of the reference buffer’s input for both REFD HIGH and LOW. The internal reference consumes approximately 5mW. LAYOUT 4000 µs POWER 3000 For optimum performance, tie the analog and digital power pins to the same +5V power supply and tie the analog and digital grounds together. As noted in the electrical characteristics, the ADS7806 uses 90% of its power for the analog circuitry. The ADS7806 should be considered as an analog component. 2000 1000 0 0.1 1 10 100 “CAP” Pin Value (µF) FIGURE 11. Power-Down to Power-Up Time vs Capacitor Value on CAP. The output of the buffer is capable of driving up to 1mA of current to a DC load. Using an external buffer will allow the internal reference to be used for larger DC loads and AC loads. Do not attempt to directly drive an AC load with the output voltage on CAP. This will cause performance degradation of the converter. REFERENCE AND POWER-DOWN The +5V power for the A/D converter should be separate from the +5V used for the system’s digital logic. Connecting VDIG (pin 28) directly to a digital supply can reduce converter performance due to switching noise from the digital logic. For best performance, the +5V supply can be produced from whatever analog supply is used for the rest of the analog signal conditioning. If +12V or +15V supplies are present, a simple +5V regulator can be used. Although it is not suggested, if the digital supply must be used to power the converter, be sure to properly filter the supply. When using either a filtered digital supply or a regulated analog supply, both VDIG and VANA should be tied to the same +5V source. GROUNDING The ADS7806 has analog power-down and reference power down capabilities via PWRD (pin 25) and REFD (pin 26), respectively. PWRD and REFD HIGH will power-down all analog circuitry maintaining data from the previous conversion in the internal registers, provided that the data has not already been shifted out through the serial port. Typical power consumption in this mode is 50µW. Power recovery is typically 1ms, using a 2.2µF capacitor connected to CAP. Figure 11 shows power-down to power-up recovery time relative to the capacitor value on CAP. With +5V applied to VDIG, the digital circuitry of the ADS7806 remains active at all times, regardless of PWRD and REFD states. PWRD PWRD HIGH will power-down all of the analog circuitry except for the reference. Data from the previous conversion will be maintained in the internal registers and can still be read. With PWRD HIGH, a convert command yields meaningless data. 16 REFD Three ground pins are present on the ADS7806. DGND is the digital supply ground. AGND2 is the analog supply ground. AGND1 is the ground to which all analog signals internal to the A/D converter are referenced. AGND1 is more susceptible to current induced voltage drops and must have the path of least resistance back to the power supply. All the ground pins of the A/D converter should be tied to an analog ground plane, separated from the system’s digital logic ground, to achieve optimum performance. Both analog and digital ground planes should be tied to the “system” ground as near to the power supplies as possible. This helps to prevent dynamic digital ground currents from modulating the analog ground through a common impedance to power ground. SIGNAL CONDITIONING The FET switches used for the sample hold on many CMOS A/D converters release a significant amount of charge injection which can cause the driving op amp to oscillate. The amount of charge injection due to the sampling FET switch ADS7806 www.ti.com SBAS021B on the ADS7806 is approximately 5% to 10% of the amount on similar A/D converters with the charge redistribution Digital-to-Analog Converter (DAC) CDAC architecture. There is also a resistive front end which attenuates any charge which is released. The end result is a minimal requirement for the drive capability on the signal conditioning preceding the A/D converter. Any op amp sufficient for the signal in an application will be sufficient to drive the ADS7806. The resistive front end of the ADS7806 also provides a specified ±25V over-voltage protection. In most cases, this eliminates the need for external over-voltage protection circuitry. INTERMEDIATE LATCHES The ADS7806 does have tri-state outputs for the parallel port, but intermediate latches should be used if the bus will be active during conversions. If the bus is not active during conversion, the tri-state outputs can be used to isolate the A/D converter from other peripherals on the same bus. Before enabling the QSPI interface, the microcontroller must be configured to monitor the slave-select line. When a transition from LOW to HIGH occurs on Slave Select (SS) from BUSY (indicating the end of the current conversion), the port can be enabled. If this is not done, the microcontroller and the A/D converter may be “out-of-sync.” Figure 13 shows another interface between the ADS7806 and a QSPI equipped microcontroller. The interface allows the microcontroller to give the convert pulses while also allowing multiple peripherals to be connected to the serial bus. This interface and the following discussion assume a master clock for the QSPI interface of 16.78MHz. Notice that the serial data input of the microcontroller is tied to the MSB (D7) of the ADS7806 instead of the serial output (SDATA). Using D7 instead of the serial port offers tri-state capability which allows other peripherals to be connected to the MISO pin. When communication is desired with those peripherals, PCS0 and PCS1 should be left HIGH; that will keep D7 tristated and prevent a conversion from taking place. Intermediate latches are beneficial on any monolithic A/D converter. The ADS7806 has an internal LSB size of 610µV. Transients from fast switching signals on the parallel port, even when the A/D converter is tri-stated, can be coupled through the substrate to the analog circuitry causing degradation of converter performance. The effects of this phenomenon will be more obvious when using the pin-compatible ADS7807 or any of the other 16-bit converters in the ADS Family. This is due to the smaller internal LSB size of 38µV. QSPI™ APPLICATIONS INFORMATION Figure 12 shows a simple interface between the ADS7806 and any QSPI equipped microcontroller. This interface assumes that the convert pulse does not originate from the microcontroller and that the ADS7806 is the only serial peripheral. Convert Pulse QSPI™ ADS7806 R/C PCS0/SS MOSI SCK BUSY SDATA DATACLK CS EXT/INT CPOL = 0 (Inactive State is LOW) CPHA = 1 (Data valid on falling edge) QSPI port is in slave mode. BYTE FIGURE 12. QSPI Interface to the ADS7806. R/C PCS1 CS +5V EXT/INT SCK DATACLK MISO D7 (MSB) BYTE FIGURE 13. QSPI Interface to the ADS7806. Processor Initiates Conversions. In this configuration, the QSPI interface is actually set to do two different serial transfers. The first, an 8-bit transfer, causes PCS0 (R/C) and PCS1 (CS ) to go LOW, starting a conversion. The second, a 12-bit transfer, causes only PCS1 (CS ) to go LOW. This is when the valid data will be transferred. For both transfers, the DT register (delay after transfer) is used to cause a 19µs delay. The interface is also set up to wrap to the beginning of the queue. In this manner, the QSPI is a state machine which generates the appropriate timing for the ADS7806. This timing is thus locked to the crystal-based timing of the microcontroller and not interrupt driven. So, this interface is appropriate for both AC and DC measurements. For the fastest conversion rate, the baud rate should be set to 2 (4.19MHz SCK), DT set to 10, the first serial transfer set to eight bits, the second set to 12 bits, and DSCK disabled (in the command control byte). This will allow for a 23kHz maximum conversion rate. For slower rates, DT should be increased. Do not slow SCK as this may increase the chance of affecting the conversion results or accidently initiating a second conversion during the first 8-bit transfer. ADS7806 SBAS021B PCS0 CPOL = 0 CPHA = 0 QSPI™ INTERFACING ADS7806 www.ti.com 17 In addition, CPOL and CPHA should be set to zero (SCK normally LOW and data captured on the rising edge). The command control byte for the 8-bit transfer should be set to 20H and for the 12-bit transfer to 61H. Convert Pulse DSP56000 ADS7806 SPI™ INTERFACE R/C The SPI interface is generally only capable of 8-bit data transfers. For some microcontrollers with SPI interfaces, it might be possible to receive data in a similar manner as shown for the QSPI interface in Figure 12. The microcontroller will need to fetch the eight most significant bits before the contents are overwritten by the least significant bits. SC1 BUSY SRD SDATA SCO DATACLK CS EXT/INT A modified version of the QSPI interface, see Figure 13, might be possible. For most microcontrollers with SPI interface, the automatic generation of the start-of-conversion pulse will be impossible and will have to be done with software. This will limit the interface to ‘DC’ applications due to the insufficient jitter performance of the convert pulse tself. SYN = 0 (Asychronous) BYTE GCK = 1 (Gated clock) SCD1 = 0 (SC1 is an input) SHFD = 0 (Shift MSB first) WL1 = 0 WL0 = 1 (Word length = 12 bits) FIGURE 14. DSP56000 Interface to the ADS7806. DSP56000 INTERFACING The DSP56000 serial interface has an SPI compatibility mode with some enhancements. Figure 14 shows an interface between the ADS7806 and the DSP56000 which is very similar to the QSPI interface seen in Figure 12. As mentioned in the QSPI section, the DSP56000 must be programmed to enable the interface when a LOW-to-HIGH transition on SC1 is observed (BUSY going HIGH at the end of conversion). The DSP56000 can also provide the convert pulse by including a monostable multi-vibrator as seen in Figure 15. The receive and transmit sections of the interface are decoupled (asynchronous mode) and the transmit section is set to generate a word length frame sync every other transmit frame (frame rate divider set to two). The prescale modulus should be set to five. The monostable multi-vibrator in this circuit will provide varying pulse widths for the convert pulse. The pulse width will be determined by the external R and C values used with the multi-vibrator. The 74HCT123N data sheet shows that the pulse width is (0.7)RC. Choosing a pulse width as close to the minimum value specified in this data sheet will offer the best performance. See the Starting A Conversion section of this data sheet for details on the conversion pulse width. The maximum conversion rate for a 20.48MHz DSP56000 is 35.6kHz. If a slower oscillator can be tolerated on the DSP56000, a conversion rate of 40kHz can be achieved by using a 19.2MHz clock and a prescale modulus of four. SPI is a registered trademark of Motorola. DSP56000 74HCT123N +5V +5V R B1 REXT1 C SC2 CLR1 ADS7806 CEXT1 Q1 A1 R/C SC0 DATACLK SRD SDATA CS SYN = 0 (Asychronous) GCK = 1 (Gated clock) SCD2 = 1 (SC2 is an output) SHFD = 0 (Shift MSB first) WL1 = 0 WL0 = 1 (Word length = 16 bits) EXT/INT BYTE FIGURE 15. DSP56000 Interface to the ADS7806. Processor initiates conversions. 18 ADS7806 www.ti.com SBAS021B PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) ADS7806UB ACTIVE SOIC DW 28 20 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7806U B (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of