ADS8321E

® ADS8321 For most current data sheet and other product information, visit www.burr-brown.com 16-Bit, High Speed, MicroPower Sampling ANALOG-TO-DIGITAL CONVERTER FEATURES q BIPOLAR INPUT RANGE q 100kHz SAMPLING RATE q MICRO POWER: 4.5mW at 100kHz 1mW at 10kHz q POWER DOWN: 3µA max q 8-LEAD MSOP PACKAGE q PIN-COMPATIBLE TO ADS7816 AND ADS7822 q SERIAL (SPI/SSI) INTERFACE DESCRIPTION The ADS8321 is a 16-bit sampling analog-to-digital converter with guaranteed specifications over a 4.75V to 5.25V supply range. It requires very little power even when operating at the full 100kHz data rate. At lower data rates, the high speed of the device enables it to spend most of its time in the power-down mode— the average power dissipation is less than 1mW at 10kHz data rate. The ADS8321 also features a synchronous serial (SPI/SSI compatible) interface, and a differential input. The reference voltage can be set to any level within the range of 500mV to 2.5V. Ultra-low power and small size make the ADS8321 ideal for portable and battery-operated systems. It is also a perfect fit for remote data acquisition modules, simultaneous multi-channel systems, and isolated data acquisition. The ADS8321 is available in an 8-lead MSOP package. APPLICATIONS q q q q BATTERY OPERATED SYSTEMS REMOTE DATA ACQUISITION ISOLATED DATA ACQUISITION SIMULTANEOUS SAMPLING, MULTI-CHANNEL SYSTEMS q INDUSTRIAL CONTROLS q ROBOTICS q VIBRATION ANALYSIS SAR VREF ADS8321 DOUT +In CDAC –In S/H Amp Comparator CS/SHDN Serial Interface DCLOCK International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 © 1999 Burr-Brown Corporation PDS-1537A Printed in U.S.A. Setpember, 1999 SPECIFICATIONS: +VCC = +5V At –40°C to +85°C, VREF = +2.5V, –In = 2.5V, fSAMPLE = 100kHz, and fCLK = 24 • fSAMPLE, unless otherwise specified. ADS8321E PARAMETER RESOLUTION ANALOG INPUT Full-Scale Input Span Absolute Input Range Capacitance Leakage Current SYSTEM PERFORMANCE No Missing Codes Integral Linearity Error Offset Error Offset Temperature Drift Gain Error, Positive Negative Gain Temperature Drift Noise Common-Mode Rejection Ratio Power Supply Rejection Ratio SAMPLING DYNAMICS Conversion Time Acquisition Time Throughput Rate Clock Frequency Range DYNAMIC CHARACTERISTICS Total Harmonic Distortion SINAD Spurious Free Dynamic Range SNR REFERENCE INPUT Voltage Range Resistance Current Drain fSAMPLE = 10kHz CS = VCC DIGITAL INPUT/OUTPUT Logic Family Logic Levels: VIH VIL VOH VOL Data Format POWER SUPPLY REQUIREMENTS VCC VCC Range(2) Quiescent Current Power Dissipation Power Down TEMPERATURE RANGE Specified Performance T Specifications same as grade to the left. NOTES: (1) LSB means Least Significant Bit. (2) See Typical Performance Curves for more information. (3) fCLK = 2.4MHz, CS = VCC for 216 clock cycles out of every 240. (4) See the Power Dissipation section for more information regarding lower sample rates. VIN = 5Vp-p at 10kHz VIN = 5Vp-p at 10kHz VIN = 5Vp-p at 10kHz 14 ±0.008 ±0.4 ±1 ±0.018 ±2 ±0.05 ±0.05 ±0.3 60 80 3 16 4.5 0.024 –84 82 84 85 0.5 CS = GND, fSAMPLE = 0Hz CS = VCC 5 5 40 0.8 0.1 CMOS IIH = +5µA IIL = +5µA IOH = –250µA IOL = 250µA 3.0 –0.3 4.0 VCC + 0.3 0.8 T T T T T T T T T T T T T T T T T V V µA µA mW µA °C VCC/2 T T T T T T T T T T V V V V 100 2.9 T T –86 84 86 87 T T T T T T T T +In – (–In) +In –In –VREF –0.1 –0.1 25 1 15 ±0.006 ±0.2 T ±0.012 ±1 ±0.024 ±0.024 CONDITIONS MIN TYP MAX 16 +VREF VCC + 0.1 +4.0 T T T T T MIN ADS8321EB TYP MAX T T T T UNITS Bits V V V pF nA Bits % of FSR mV µV/°C % % ppm/°C µVrms dB LSB(1) Clk Cycles Clk Cycles kHz MHz dB dB dB dB V GΩ GΩ µA µA µA +4.7V < VCC < 5.25V 80 3 T 0.4 Binary Two’s Complement Specified Performance 4.75 2.7 1100 250 5.5 0.3 –40 5.25 5.25 1700 8.5 3 +85 fSAMPLE = 10kHz(3, 4) CS = VCC The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® ADS8321 2 PIN CONFIGURATION Top View MSOP ABSOLUTE MAXIMUM RATINGS(1) VCC ....................................................................................................... +6V Analog Input .............................................................. –0.3V to (VCC + 0.3V) Logic Input ............................................................................... –0.3V to 6V Case Temperature ......................................................................... +100°C Junction Temperature .................................................................... +150°C Storage Temperature ..................................................................... +125°C External Reference Voltage .............................................................. +5.5V NOTE: (1) Stresses above these ratings may permanently damage the device. VREF +In –In GND 1 2 ADS8321 3 4 8 7 6 5 +VCC DCLOCK DOUT CS/SHDN PIN ASSIGNMENTS PIN 1 2 3 4 5 6 NAME VREF +In –In GND CS/SHDN DOUT DESCRIPTION Reference Input Non Inverting Input Inverting Input Ground Chip Select when LOW, Shutdown Mode when HIGH. The serial output data word is comprised of 16 bits of data. In operation the data is valid on the falling edge of DCLOCK. The second clock pulse after the falling edge of CS enables the serial output. After one null bit, data is valid for the next 16 edges. Data Clock synchronizes the serial data transfer and determines conversion speed. Power Supply. ELECTROSTATIC DISCHARGE SENSITIVITY Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. BurrBrown Corporation recommends that all integrated circuits be handled and stored using appropriate ESD protection methods. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. 7 8 DCLOCK +VCC PACKAGE/ORDERING INFORMATION MAXIMUM INTEGRAL LINEARITY ERROR (LSB) 0.018% NO MISSING CODES ERROR (LSB) 14 PACKAGE DRAWING NUMBER(1) 337 SPECIFICATION TEMPERATURE RANGE –40°C to +85°C PRODUCT ADS8321E PACKAGE MSOP ORDERING NUMBER ADS8321E/250 ADS8321E/2K5 ADS8321EB/250 ADS8321EB/2K5 TRANSPORT MEDIA Tape and Reel Tape and Reel Tape and Reel Tape and Reel " ADS8321EB " 0.012% " 15 " MSOP " 337 " –40°C to +85°C " " " " " " NOTES: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of “ADS8321EB/2K5” will get a single 2500-piece Tape and Reel. For detailed Tape and Reel mechanical information, refer to Appendix B of Burr-Brown IC Data Book. ® 3 ADS8321 TYPICAL PERFORMANCE CURVES At TA = +25°C, VCC = +5V, VREF = +2.5V, fSAMPLE = 100kHz, fCLK = 24 • fSAMPLE, unless otherwise specified. FREQUENCY SPECTRUM (8192 Point FFT, fIN = 10.03kHz, –0.3dB) 0 –20 INTEGRAL LINEARITY ERROR vs CODE (+25°C) 3 2 Integral Linearity Error (LSB) –40 1 0 –1 –2 –3 –4 –5 –6 Amplitude (dB) –60 –80 –100 –120 –140 –160 –180 0 10 20 30 40 50 Frequency (kHz) 0000H 4000H 7FF9H Hex Code C000H FFFDH DIFFERENTIAL LINEARITY ERROR vs CODE (+25°C) 3.0 SUPPLY CURRENT vs TEMPERATURE 1400 1200 Supply Current (µA) Differential Linearity Error (LSB) 2.5 2.0 1.5 1.0 0.5 0 –0.5 –1.0 –1.5 0000H 3FFFH 7FFCH Hex Code C000H FFFDH 1000 800 600 400 200 0 –50 0 50 Temperature (°C) 100 POWER-DOWN SUPPLY CURRENT vs TEMPERATURE 600 500 400 5V 300 200 100 0 –50 –25 0 25 50 75 100 Temperature (°C) Quiescent Current (mA) Supply Current (nA) QUIESCENT CURRENT vs VCC 1.20 1.10 1.00 0.90 0.80 0.70 0.60 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 VCC (V) ® ADS8321 4 TYPICAL PERFORMANCE CURVES (Cont.) At TA = +25°C, VCC = +5V, VREF = +2.5V, fSAMPLE = 100kHz, fCLK = 24 • fSAMPLE, unless otherwise specified. MAXIMUM SAMPLE RATE vs VCC 1000 90 SIGNAL-TO-NOISE AND SIGNAL-TO-(NOISE + DISTORTION) vs INPUT FREQUENCY SNR 85 100 SNR and SINAD (dB) Sample Rate (kHz) 80 SINAD 75 10 70 1 1 2 3 VCC (V) 4 5 65 0.1 1 10 Input Frequency (kHz) 100 SPURIOUS FREE DYNAMIC RANGE AND TOTAL HARMONIC DISTORTION vs INPUT FREQUENCY 90 SFDR 30 Reference Current (µA) REFERENCE CURRENT vs SAMPLE RATE 35 2.5V 85 SNR and SINAD (dB) 25 20 15 10 5 0 1.25V THD 80 75 70 65 0.1 1 10 Input Frequency (kHz) 100 0 20 40 60 80 100 Sample Rate (kHz) 120 140 NOISE vs REFERENCE VOLTAGE 18 16 Peak-to-Peak Noise (LSB) 10 15 CHANGE IN GAIN vs REFERENCE VOLTAGE Change in Gain (LSB) 14 12 10 8 6 4 2 0 0.1 1 Reference Voltage (V) 10 5 0 –5 –10 –15 0 0.5 1.0 1.5 2.0 Reference Voltage (V) 2.5 3.0 ® 5 ADS8321 TYPICAL PERFORMANCE CURVES (Cont.) At TA = +25°C, VCC = +5V, VREF = +2.5V, fSAMPLE = 100kHz, fCLK = 24 • fSAMPLE, unless otherwise specified. CHANGE IN BIPOLAR ZERO vs REFERENCE VOLTAGE 6.0 5.0 CHANGE IN OFFSET vs TEMPERATURE 5.0 4.0 Change from +25°C (LSB) Change in BPZ (LSB) 4.0 3.0 2.0 1.0 0 –1.0 –2.0 –3.0 0 0.5 1.0 1.5 2.0 2.5 3.0 Reference Voltage (V) 3.0 2.0 1.0 0 –1.0 –2.0 –3.0 –4.0 –5.0 –50 0 50 Temperature (°C) 100 CHANGE IN GAIN vs TEMPERATURE 5.0 4.0 Change from +25°C (LSB) 90 80 70 CMRR (dB) 60 50 40 30 20 10 0 COMMON-MODE REJECTION RATIO vs FREQUENCY 3.0 2.0 1.0 0 –1.0 –2.0 –3.0 –4.0 –5.0 –50 0 50 Temperature (°C) 100 VCM = 1Vp-p Sinewave 1 10 100 1k Frequency (Hz) 10k 100k 1M REFERENCE CURRENT vs TEMPERATURE 70 60 Reference Current (µA) 50 5V 40 30 20 10 –50 –25 0 25 Temperature (°C) 50 75 100 ® ADS8321 6 THEORY OF OPERATION The ADS8321 is a classic Successive Approximation Register (SAR) analog-to-digital (A/D) converter. The architecture is based on capacitive redistribution which inherently includes a sample/hold function. The converter is fabricated on a 0.6µ CMOS process. The architecture and process allow the ADS8321 to acquire and convert an analog signal at up to 100,000 conversions per second while consuming less than 5.5mW from +VCC. The ADS8321 requires an external reference, an external clock, and a single power source (VCC). The external reference can be any voltage between 500mV and 2.5V. The value of the reference voltage directly sets the range of the analog input. The reference input current depends on the conversion rate of the ADS8321. The external clock can vary between 24kHz (1kHz throughput) and 2.4MHz (100kHz throughput). The duty cycle of the clock is essentially unimportant as long as the minimum high and low times are at least 200ns (4.75V or greater). The minimum clock frequency is set by the leakage on the capacitors internal to the ADS8321. The analog input is provided to two input pins: +In and –In. When a conversion is initiated, the differential input on these pins is sampled on the internal capacitor array. While a conversion is in progress, both inputs are disconnected from any internal function. The digital result of the conversion is clocked out by the DCLOCK input and is provided serially, most significant bit first, on the DOUT pin. The digital data that is provided on the DOUT pin is for the conversion currently in progress—there is no pipeline delay. It is possible to continue to clock the ADS8321 after the conversion is complete and to obtain the serial data least significant bit first. See the digital timing section for more information. 2 • VREF peak-to-peak Common Voltage Single-Ended Input ADS8321 VREF peak-to-peak Common Voltage ADS821 VREF peak-to-peak Differential Input FIGURE 1. Methods of Driving the ADS8321—Single-Ended or Differential. 5 VCC = 5V 4.0 4 Common Voltage Range (V) 3 Single-Ended Input 2.8 2.2 2 1 0 –0.3 –1 0.0 0.5 1.0 VREF (V) 1.5 2.0 2.5 ANALOG INPUT The analog input is bipolar and fully differential. There are two general methods of driving the analog input of the ADS8321: single-ended or differential (see Figure 1). When the input is single-ended, the –In input is held at a fixed voltage. The +In input swings around the same voltage and the peak-to-peak amplitude is 2 • VREF. The value of VREF determines the range over which the common voltage may vary (see Figure 2). When the input is differential, the amplitude of the input is the difference between the +In and –In input, or; +In – (–In). A voltage or signal is common to both of these inputs. The peak-to-peak amplitude of each input is VREF about this common voltage. However, since the input are 180°C outof-phase, the peak-to-peak amplitude of the difference voltage is 2 • VREF. The value of VREF also determines the range of the voltage that may be common to both inputs (see Figure 3). In each case, care should be taken to ensure that the output impedance of the sources driving the +In and –In inputs are matched. If this is not observed, the two inputs could have 7 FIGURE 2. Single-Ended Input—Common Voltage Range vs VREF. 5 4.0 4 VCC = 5V Common Voltage Range (V) 3 Differential Input 2.75 2 1 1.95 0 –0.3 –1 0.0 0.5 1.0 VREF (V) 1.5 2.0 2.5 FIGURE 3. Differential Input—Common Voltage Range vs VREF. ® ADS8321 different settling times. This may result in offset error, gain error, and linearity error which change with both temperature and input voltage. If the impedance cannot be matched, the errors can be lessened by giving the ADS8321 additional acquisition time. The input current on the analog inputs depends on a number of factors: sample rate, input voltage, and source impedance. Essentially, the current into the ADS8321 charges the internal capacitor array during the sample period. After this capacitance has been fully charged, there is no further input current. The source of the analog input voltage must be able to charge the input capacitance (25pF) to 16-bit settling level within 4.5 clock cycles. When the converter goes into the hold mode or while it is in the power-down mode, the input impedance is greater than 1GΩ. Care must be taken regarding the absolute analog input voltage. The +In input should always remain within the range of GND – 300mV to VCC + 300mW. The –In input should always remain within the range of GND – 300mV to 4V. Outside of these ranges, the converter’s linearity may not meet specifications. NOISE The noise floor of the ADS8321 itself is extremely low, as can be seen from Figures 4 and 5, and is much lower than competing A/D converters. It was tested by applying a low noise DC input and a 2.5V reference to the ADS8321 and initiating 5,000 conversions. The digital output of the A/D converter will vary in output code due to the internal noise of the ADS8321. This is true for all 16-bit SAR-type A/D converters. Using a histogram to plot the output codes, the distribution should appear bell-shaped with the peak of the bell curve representing the nominal code for the input value. The ±1σ, ±2σ, and ±3σ distributions will represent the 68.3%, 95.5%, and 99.7%, respectively, of all codes. The transition noise can be calculated by dividing the number of codes measured by 6 and this will yield the ±3σ distribution or 99.7% of all codes. Statistically, up to 3 codes could fall outside the distribution when executing 1000 conversions. The ADS8321, with five output codes for the ±3σ distribution, will yield a ±0.8LSB transition noise. Remember, to achieve this low noise performance, the peak-to-peak noise of the input signal and reference must be < 50µV. REFERENCE INPUT The external reference sets the analog input range. The ADS8321 will operate with a reference in the range of 500mV to 2.5V. There are several important implications of this. As the reference voltage is reduced, the analog voltage weight of each digital output code is reduced. This is often referred to as the Least Significant Bit (LSB) size and is equal to 2 • VREF divided by 65,535. This means that any offset or gain error inherent in the A/D converter will appear to increase, in terms of LSB size, as the reference voltage is reduced. The noise inherent in the converter will also appear to increase with lower LSB size. With a +2.5V reference, the internal noise of the converter typically contributes only 5 LSB peak-to-peak of potential error to the output code. When the external reference is 500mV, the potential error contribution from the internal noise will be 10 times larger—15 LSBs. The errors due to the internal noise are gaussian in nature and can be reduced by averaging consecutive conversion results. For more information regarding noise, consult the typical performance curve “Noise vs Reference Voltage.” Note that the Effective Number of Bits (ENOB) figure is calculated based on the converter’s signal-to-(noise + distortion) ratio with a 1kHz, 0dB input signal. SINAD is related to ENOB as follows: SINAD = 6.02 • ENOB + 1.76 With lower reference voltages, extra care should be taken to provide a clean layout including adequate bypassing, a clean power supply, a low-noise reference, and a low-noise input signal. Because the LSB size is lower, the converter will also be more sensitive to external sources of error such as nearby digital signals and electromagnetic interference. 1639 1260 981 0 12 192 13 14 15 Code 16 24 17 0 18 FIGURE 4. Histogram of 5,000 Conversions of a DC Input at the Code Transition. 2318 836 696 244 0 12 13 14 15 Code 16 2 17 0 18 FIGURE 5. Histogram of 5,000 Conversions of a DC Input at the Code Center. ® ADS8321 8 AVERAGING The noise of the A/D converter can be compensated by averaging the digital codes. By averaging conversion results, transition noise will be reduced by a factor of 1/√n, where n is the number of averages. For example, averaging 4 conversion results will reduce the transition noise by 1/2 to ±0.25 LSBs. Averaging should only be used for input signals with frequencies near DC. For AC signals, a digital filter can be used to low pass filter and decimate the output codes. This works in a similar manner to averaging; for every decimation by 2, the signalto-noise ratio will improve 3dB. SYMBOL tSMPL tCONV tCYC tCSD tSUCS thDO tdDO tdis ten DESCRIPTION Analog Input Sample Time Conversion Time Throughput Rate CS Falling to DCLOCK LOW CS Falling to DCLOCK Rising DCLOCK Falling to Current DOUT Not Valid DCLOCK Falling to Next DOUT Valid CS Rising to DOUT Tri-State DCLOCK Falling to DOUT Enabled DOUT Fall Time DOUT Rise Time MIN 4.5 TYP MAX 5.0 UNITS Clk Cycles Clk Cycles 16 100 0 kHz ns 20 ns 5 15 30 70 20 5 7 50 100 50 25 25 ns ns ns ns ns ns DIGITAL INTERFACE SIGNAL LEVELS The digital inputs of the ADS8321 can accommodate logic levels up to 5.5V regardless of the value of VCC. The CMOS digital output (DOUT) will swing 0V to VCC. If VCC is 3V and this output is connected to a 5V CMOS logic input, then that IC may require more supply current than normal and may have a slightly longer propagation delay. SERIAL INTERFACE The ADS8321 communicates with microprocessors and other digital systems via a synchronous 3-wire serial interface as shown in Figure 6 and Table I. The DCLOCK signal synchronizes the data transfer with each bit being transmitted on the falling edge of DCLOCK. Most receiving systems will capture the bitstream on the rising edge of DCLOCK. However, if the minimum hold time for DOUT is acceptable, the system can use the falling edge of DCLOCK to capture each bit. tf tr TABLE I. Timing Specifications (VCC = 5V) –40°C to +85°C. A falling CS signal initiates the conversion and data transfer. The first 4.5 to 5.0 clock periods of the conversion cycle are used to sample the input signal. After the fifth falling DCLOCK edge, DOUT is enabled and will output a LOW value for one clock period. For the next 16 DCLOCK periods, DOUT will output the conversion result, most significant bit first. After the least significant bit (B0) has been output, subsequent clocks will repeat the output data but in a least significant bit first format. After the most significant bit (B15) has been repeated, DOUT will tri-state. Subsequent clocks will have no effect on the converter. A new conversion is initiated only when CS has been taken HIGH and returned LOW. Complete Cycle CS/SHDN tSUCS Sample DCLOCK tCSD DOUT Hi-Z 0 tSMPL B15 B14 B13 B12 B11 B10 B9 B8 (MSB) tCONV B7 B6 B5 B4 B3 B2 B1 B0 (LSB) Use positive clock edge for data transfer Hi-Z Conversion Power Down NOTE: Minimum 22 clock cycles required for 16-bit conversion. Shown are 24 clock cycles. If CS remains LOW at the end of conversion, a new datastream with LSB-first is shifted out again. FIGURE 6. ADS8321 Basic Timing Diagrams. ® 9 ADS8321 DATA FORMAT The output data from the ADS8321 is in Binary Two’s Complement format as shown in Table II. This table represents the ideal output code for the given input voltage and does not include the effects of offset, gain error, or noise. DESCRIPTION Full-Scale Range Least Significant Bit (LSB) +Full Scale Midscale Midscale – 1LSB –Full Scale ANALOG VALUE 2 • VREF 2 • VREF/65536 BINARY CODE +VREF – 1 LSB 0V 0V – 1 LSB –VREF 0111 1111 1111 1111 0000 0000 0000 0000 1111 1111 1111 1111 1000 0000 0000 0000 HEX CODE 7FFF 0000 FFFF 8000 DIGITAL OUTPUT BINARY TWO’S COMPLEMENT POWER DISSIPATION The architecture of the converter, the semiconductor fabrication process, and a careful design allow the ADS8321 to convert at up to a 100kHz rate while requiring very little power. Still, for the absolute lowest power dissipation, there are several things to keep in mind. The power dissipation of the ADS8321 scales directly with conversion rate. Therefore, the first step to achieving the lowest power dissipation is to find the lowest conversion rate that will satisfy the requirements of the system. In addition, the ADS8321 is in power-down mode under two conditions: when the conversion is complete and whenever CS is HIGH (see Figure 6). Ideally, each conversion should occur as quickly as possible, preferably at a 2.4MHz clock rate. This way, the converter spends the longest possible time in the power-down mode. This is very important as the converter not only uses power on each DCLOCK transition (as is typical for digital CMOS components) but also uses some current for the analog circuitry, such as the comparator. The analog section dissipates power continuously, until the power down mode is entered. TABLE II. Ideal Input Voltages and Output Codes. 1.4V 3kΩ DOUT 100pF CLOAD Test Point DOUT tr tf VOH VOL Voltage Waveforms for DOUT Rise and Fall Times, tr, tf Load Circuit for tdDO, tr, and tf Test Point DCLOCK VIL tdDO DOUT thDO Voltage Waveforms for DOUT Delay Times, tdDO VOH VOL DOUT 3kΩ 100pF CLOAD Load Circuit for tdis and ten VCC tdis Waveform 2, ten tdis Waveform 1 CS/SHDN VIH CS/SHDN DCLOCK 1 2 DOUT Waveform 1(1) tdis DOUT Waveform 2(2) Voltage Waveforms for tdis 90% 10% DOUT ten Voltage Waveforms for ten VOL B11 NOTES: (1) Waveform 1 is for an output with internal conditions such that the output is HIGH unless disabled by the output control. (2) Waveform 2 is for an output with internal conditions such that the output is LOW unless disabled by the output control. FIGURE 7. Timing Diagrams and Test Circuits for the Parameters in Table I. ® ADS8321 10 1000 Supply Current (µA) TA = 25°C VCC = 5.0V VREF = 2.5V fCLK = 2.4MHz 100 10 1 0.1 1 10 100 Sample Rate (kHz) FIGURE 8. Maintaining fCLK at the Highest Possible Rate Allows Supply Current to Drop Linearly with Sample Rate. Figure 8 shows the current consumption of the ADS8321 versus sample rate. For this graph, the converter is clocked at 2.4MHz regardless of the sample rate—CS is HIGH for the remaining sample period. Figure 9 also shows current consumption versus sample rate. However, in this case, the DCLOCK period is 1/24th of the sample period—CS is HIGH for one DCLOCK cycle out of every 16. There is an important distinction between the power-down mode that is entered after a conversion is complete and the full power-down mode which is enabled when CS is HIGH. CS LOW will shut down only the analog section. The digital section is completely shutdown only when CS is HIGH. Thus, if CS is left LOW at the end of a conversion and the converter is continually clocked, the power consumption will not be as low as when CS is HIGH. See Figure 10 for more information. SHORT CYCLING Another way of saving power is to utilize the CS signal to short cycle the conversion. Because the ADS8321 places the latest data bit on the DOUT line as it is generated, the converter can easily be short cycled. This term means that the conversion can be terminated at any time. For example, if only 14 bits of the conversion result are needed, then the conversion can be terminated (by pulling CS HIGH) after the 14th bit has been clocked out. This technique can be used to lower the power dissipation (or to increase the conversion rate) in those applications where an analog signal is being monitored until some condition becomes true. For example, if the signal is outside a predetermined range, the full 16-bit conversion result may not be needed. If so, the conversion can be terminated after the first n bits, where n might be as low as 3 or 4. This results in lower power dissipation in both the converter and the rest of the system, as they spend more time in the power-down mode. 1000 Supply Current (µA) 100 10 TA = 25°C VCC = 5.0V VREF = 2.5V fCLK = 24 • fSAMPLE 0.1 1 10 100 1 Sample Rate (kHz) FIGURE 9. Scaling fCLK Reduces Supply Current Only Slightly with Sample Rate. LAYOUT 1000 800 Supply Current (µA) 600 400 200 0.250 TA = 25°C VCC = 5.0V VREF = 2.5V fCLK = 24 • fSAMPLE CS LOW (GND) CS HIGH (VCC) 0.00 0.1 1 10 100 Sample Rate (kHz) FIGURE 10. Shutdown Current with CS HIGH is 50nA Typically, Regardless of the Clock. Shutdown Current with CS LOW Varies with Sample Rate. For optimum performance, care should be taken with the physical layout of the ADS8321 circuitry. This will be particularly true if the reference voltage is low and/or the conversion rate is high. At a 100kHz conversion rate, the ADS8321 makes a bit decision every 416ns. That is, for each subsequent bit decision, the digital output must be updated with the results of the last bit decision, the capacitor array appropriately switched and charged, and the input to the comparator settled to a 16-bit level all within one clock cycle. The basic SAR architecture is sensitive to spikes on the power supply, reference, and ground connections that occur just prior to latching the comparator output. Thus, during any single conversion for an n-bit SAR converter, there are n “windows” in which large external transient voltages can easily affect the conversion result. Such spikes might originate from switching power supplies, digital logic, and high ® 11 ADS8321 power devices, to name a few. This particular source of error can be very difficult to track down if the glitch is almost synchronous to the converter’s DCLOCK signal—as the phase difference between the two changes with time and temperature, causing sporadic misoperation. With this in mind, power to the ADS8321 should be clean and well bypassed. A 0.1µF ceramic bypass capacitor should be placed as close to the ADS8321 package as possible. In addition, a 1µF to 10µF capacitor and a 5Ω or 10Ω series resistor may be used to lowpass filter a noisy supply. The reference should be similarly bypassed with a 0.1µF capacitor. Again, a series resistor and large capacitor can be used to lowpass filter the reference voltage. If the reference voltage originates from an op amp, be careful that the op amp can drive the bypass capacitor without oscillation (the series resistor can help in this case). Keep in mind that while the ADS8321 draws very little current from the reference on average, there are still instantaneous current demands placed on the external input and reference circuitry. Burr-Brown’s OPA627 op amp provides optimum performance for buffering both the signal and reference inputs. For low cost, low voltage, single-supply applications, the OPA2350 or OPA2340 dual op amps are recommended. Also, keep in mind that the ADS8321 offers no inherent rejection of noise or voltage variation in regards to the reference input. This is of particular concern when the reference input is tied to the power supply. Any noise and ripple from the supply will appear directly in the digital results. While high frequency noise can be filtered out as described in the previous paragraph, voltage variation due to the line frequency (50Hz or 60Hz), can be difficult to remove. The GND pin on the ADS8321 should be placed on a clean ground point. In many cases, this will be the “analog” ground. Avoid connecting the GND pin too close to the grounding point for a microprocessor, microcontroller, or digital signal processor. If needed, run a ground trace directly from the converter to the power supply connection point. The ideal layout will include an analog ground plane for the converter and associated analog circuitry. APPLICATION CIRCUITS Figure 11 shows a basic data acquisition system. The ADS8321 input range is 0V to VCC, as the reference input is connected directly to the power supply. The 5Ω resistor and 1µF to 10µF capacitor filter the microcontroller “noise” on the supply, as well as any high-frequency noise from the supply itself. The exact values should be picked such that the filter provides adequate rejection of the noise. 5V 5Ω + ADS8321 +2.5V Reference 0.1µF 0V to 5V +In –In GND CS DOUT DCLOCK VREF VCC + 1µF to 10µF Microcontroller 1µF to 10µF FIGURE 11. Basic Data Acquisition System. ® ADS8321 12