ADS8324
SBAS172A – AUGUST 2001 – REVISED MARCH 2004
14-Bit, High Speed, 1.8V MicroPower Sampling
ANALOG-TO-DIGITAL CONVERTER
FEATURES
DESCRIPTION
●
●
●
●
The ADS8324 is a 14-bit, sampling Analog-to-Digital (A/D)
converter with tested specifications using a 1.8V supply
voltage. It requires very little power, even when operating at
the full 50kHz data rate. At lower data rates, the high speed
of the device enables it to spend most of its time in the
power-down mode—the average power dissipation is less
than 1mW at 10kHz data rate.
The ADS8324 also features a synchronous serial (SPI/SSI
compatible) interface, and a differential input. The reference voltage can be set to any level within the range of
500mV to VCC/2.
Ultra-low power and small size make the ADS8324 ideal
for portable and battery-operated systems. It is also a
perfect fit for remote data acquisition modules, simultaneous multi-channel systems, and isolated data acquisition. The ADS8324 is available in an MSOP-8 package.
BIPOLAR INPUT RANGE
1.8V OPERATION
50kHz SAMPLING RATE
MICRO POWER:
5.0mW at 2.7V
2.5mW at 1.8V
● POWER DOWN: 3μA max
● MSOP-8 PACKAGE
● PIN-COMPATIBLE TO 12-BIT ADS7817
● SERIAL (SPI/SSI) INTERFACE
APPLICATIONS
●
●
●
●
BATTERY OPERATED SYSTEMS
REMOTE DATA ACQUISITION
ISOLATED DATA ACQUISITION
SIMULTANEOUS SAMPLING,
MULTI-CHANNEL SYSTEMS
● INDUSTRIAL CONTROLS
● ROBOTICS
● VIBRATION ANALYSIS
SAR
VREF
ADS8324
DOUT
+In
Serial
Interface
CDAC
–In
DCLOCK
S/H Amp
Comparator
CS/SHDN
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2001-2004, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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ABSOLUTE MAXIMUM RATINGS(1)
PIN CONFIGURATION
VCC ....................................................................................................... +6V
Analog Input ............................................................. –0.3V to (VCC + 0.3V)
Logic Input .............................................................................. –0.3V to 6V
Case Temperature ......................................................................... +100°C
Junction Temperature .................................................................... +150°C
Storage Temperature ..................................................................... +125°C
External Reference Voltage .............................................................. +5.5V
Top View
MSOP
NOTE: (1) Stresses above these ratings may permanently damage the device.
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
8
+VCC
7
DCLOCK
3
6
DOUT
4
5
CS/SHDN
VREF
1
+In
2
–In
GND
ADS8324
PIN ASSIGNMENTS
PIN
NAME
DESCRIPTION
1
VREF
Reference Input
2
+In
Non Inverting Input
3
–In
Inverting Input
ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric
changes could cause the device not to meet its published
specifications.
4
GND
5
CS/SHDN
Ground
6
DOUT
The serial output data word is comprised of 16
bits of data. In operation, the data is valid on the
rising edge of DCLOCK. The fifth falling edge of
DCLOCK after the falling edge of CS enables
the serial output. After one null bit, data is valid
for the next 16 edges.
7
DCLOCK
Data Clock synchronizes the serial data transfer
and determines conversion speed.
8
+VCC
Chip Select when LOW, Shutdown Mode when
HIGH.
Power Supply
PACKAGE/ORDERING INFORMATION
PRODUCT
MAXIMUM
INTEGRAL
LINEARITY
ERROR (LSB)
NO
MISSING
CODES
ERROR (LSB)
ADS8324E
±3
ADS8324EB
±2
"
"
"
"
PACKAGE
PACKAGE
DRAWING
NUMBER(1)
SPECIFICATION
TEMPERATURE
RANGE
PACKAGE
MARKING(2)
ORDERING
NUMBER(3)
TRANSPORT
MEDIA
14
MSOP
337
–40°C to +85°C
A24
14
MSOP
337
–40°C to +85°C
A24
"
"
"
"
"
ADS8324E/250
ADS8324E/2K5
ADS8324EB/250
ADS8324EB/2K5
Tape and Reel
Tape and Reel
Tape and Reel
Tape and Reel
"
"
"
"
"
NOTES: (1) For detail drawing and dimension table, please see end of data sheet or package drawing file on web. (2) Performance grade information is marked
on the reel. (3) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500
pieces of “ADS8324EB/2K5” will get a single 2500-piece Tape and Reel.
2
ADS8324
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SBAS172A
ELECTRICAL CHARACTERISTICS: +VCC = +1.8V
At –40°C to +85°C, VREF = 0.9V, –In = 0.9V, fSAMPLE = 50kHz, and fCLK = 24 • fSAMPLE, unless otherwise specified.
ADS8324E
PARAMETER
CONDITIONS
MIN
TYP
RESOLUTION
ANALOG INPUT
Full-Scale Input Span
Absolute Input Range
+In – (–In)
+In
–In
–VREF
–0.1
0.8
REFERENCE INPUT
Voltage Range
Resistance
Power Dissipation
Power Down
✻
✻
✻
at DCC
+1.8V < VCC < +3.6V
±3
±8
±2
✻
✻
✻
✻
✻
✻
±8
50
1.8
✻
1.3
–0.3
1.4
80
3
✻
✻
✻
✻
✻
✻
✻
1.8
1400
250
2.5
0.3
Clk Cycles
Clk Cycles
kHz
MHz
dB
dB
dB
dB
✻
✻
0.4
Binary Two’s Complement
–40
✻
Bits
LSB
LSB
μV/°C
LSB
ppm/°C
μVrms
dB
LSB(1)
V
GΩ
GΩ
μA
μA
μA
✻
VCC + 0.3
0.5
1.8
TEMPERATURE RANGE
Specified Performance
±2
±4
✻
✻
✻
✻
✻
✻
CMOS
fSAMPLE = 10kHz(3, 4)
VCC = 1.8V
CS = VCC
V
V
V
pF
nA
–86
78
86
✻
VCC/2
5
5
40
0.8
0.1
Specified Performance
✻
✻
✻
✻
✻
✻
–84
77
85
78
0.5
IIH = +5μA
IIL = +5μA
IOH = –250μA
IOL = 250μA
Bits
✻
0.024
CS = GND, fSAMPLE = 0Hz
CS = VCC
UNITS
✻
✻
16
4.5
VIN = 5Vp-p at 10kHz
VIN = 5Vp-p at 10kHz
VIN = 5Vp-p at 10kHz
MAX
14
±4
±0.1
±4
±0.4
60
74
3
fSAMPLE = 10kHz
CS = VCC
POWER SUPPLY REQUIREMENTS
VCC
VCC Range(2)
Quiescent Current
TYP
✻
✻
14
Current Drain
DIGITAL INPUT/OUTPUT
Logic Family
Logic Levels:
VIH
VIL
VOH
VOL
Data Format
+VREF
VCC + 0.1
+1.0
25
1
SAMPLING DYNAMICS
Conversion Time
Acquisition Time
Throughput Rate
Clock Frequency Range
DYNAMIC CHARACTERISTICS
Total Harmonic Distortion
SINAD
Spurious Free Dynamic Range
SNR
MIN
14
Capacitance
Leakage Current
SYSTEM PERFORMANCE
No Missing Codes
Integral Linearity Error
Bipolar Zero Error
Bipolar Zero Error Drift
Gain Error
Gain Temperature Drift
Noise
Common-Mode Rejection Ratio
Power Supply Rejection Ratio
ADS8324EB
MAX
3.6
1700
✻
✻
✻
✻
✻
3.0
3.0
+85
✻
V
V
V
V
✻
✻
V
V
μA
μA
mW
μA
✻
°C
✻
✻
✻ Specifications same as ADS8324E.
NOTES: (1) LSB means Least Significant Bit. (2) See Typical Performance Curves for more information. (3) fCLK = 1.2MHz, CS = VCC for 216 clock cycles out
of every 240. (4) See the Power Dissipation section for more information regarding lower sample rates.
ADS8324
SBAS172A
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3
TYPICAL CHARACTERISTICS
At TA = +25°C, VCC = 1.8V, VREF = 0.9V, fSAMPLE = 50kHz, fCLK = 24 • fSAMPLE, unless otherwise specified.
FREQUENCY SPECTRUM
(4096 point FFT, fIN = 0.989kHz, –0.2dB)
–20
–20
–40
–40
–60
–80
–60
–80
–100
–100
–120
–120
–140
–140
0
0
5
10
15
20
25
0
5
10
FREQUENCY SPECTRUM
(4096 point FFT, fIN = 20.001kHz, –0.2dB)
SIGNAL-TO-NOISE RATIO AND
SIGNAL-TO-(NOISE + DISTORTION)
vs INPUT FREQUENCY
90
SNR and SINAD (dB)
85
–40
–60
–80
25
SNR
80
75
SINAD
70
65
–120
–140
60
0
5
10
15
20
1
25
10
SIGNAL-TO-NOISE RATIO AND
TOTAL HARMONIC DISTORTION
vs INPUT FREQUENCY
COMMON-MODE REJECTION vs FREQUENCY
95
–95
75
90
–90
70
SFDR
85
–85
80
–80
75
–75
THD(1)
70
NOTE: (1) First nine harmonics
of the input frequency.
65
60
1
10
CMRR (dB)
80
THD (dB)
–100
100
100
Frequency (kHz)
Frequency (kHz)
SFDR (dB)
20
Frequency (kHz)
–100
65
60
55
–70
50
–65
45
–60
100
Frequency (kHz)
4
15
Frequency (kHz)
–20
Amplitude (dB)
FREQUENCY SPECTRUM
(4096 point FFT, fIN = 9.998kHz, –0.2dB)
0
Amplitude (dB)
Amplitude (dB)
0
VCM = 400mVp-p sinewave centered around VREF
40
100
1k
10k
100k
1M
Frequency (kHz)
ADS8324
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SBAS172A
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VCC = 1.8V, VREF = 0.9V, fSAMPLE = 50kHz, fCLK = 24 • fSAMPLE, unless otherwise specified.
DIFFERENTIAL LINEARITY ERROR vs CODE
INTEGRAL LINEARITY ERROR vs CODE
2
2
ILE (LSBs)
DLE (LSBs)
1
1
0
0
–1
–1
2000H
3000H
0000H
1000H
–2
2000H
1FFFH
3000H
Output Code
QUIESCENT CURRENT vs VCC
1FFFH
REFERENCE CURRENT vs TEMPERATURE
2
1.5
1
0.5
0
1.5
2
2.5
3
3.5
9
8
7
6
5
4
3
2
1
0
–50
4
–30
–10
VCC (V)
1.8
9
1.6
8
Reference Current (μA)
10
1.4
1.2
1
0.8
0.6
0.4
0.2
50
70
90
7
6
5
4
3
2
1
0
–30
–10
10
30
50
70
90
0
Temperature (°C)
20
40
60
80
Sample Rate (kHz)
ADS8324
SBAS172A
30
REFERENCE CURRENT vs SAMPLE RATE
2
0
–50
10
Temperature (°C)
SUPPLY CURRENT vs TEMPERATURE
Supply Current (μA)
1000H
10
Average Reference Current (μA)
Quiescent Current (mA)
2.5
0000H
Output Code
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5
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VCC = 1.8V, VREF = 0.9V, fSAMPLE = 50kHz, fCLK = 24 • fSAMPLE, unless otherwise specified.
CHANGE IN GAIN vs TEMPERATURE
1
0.8
0.8
Change from +25°C (LSB)
Change from +25°C (LSB)
CHANGE IN BIPOLAR OFFSET vs TEMPERATURE
1
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
–1
–50
–30
–10
10
30
50
70
90
–1
–50
110
–30
–10
Temperature (°C)
CHANGE IN BPZ vs REFERENCE VOLTAGE
30
50
70
90
110
CHANGE IN GAIN vs REFERENCE VOLTAGE
10
10
8
8
6
6
Change in GAIN (LSB)
Change in BPZ (LSB)
10
Temperature (°C)
4
2
0
–2
–4
–6
4
2
0
–2
–4
–6
–8
–8
–10
–10
0.4
0.5
0.6
0.7
0.8
0.9
1
1.1
0.4
0.5
0.6
Reference Voltage (V)
0.7
0.8
0.9
1
1.1
Reference Voltage (V)
MAXIMUM SAMPLING RATE vs SUPPLY VOLTAGE
NOISE vs REFERENCE VOLTAGE
1000
10
8
Sampling Rate (kHz)
Peak-to-Peak Noise (LSB)
9
7
6
5
4
3
100
10
2
1
1
0
0.4
0.5
0.6
0.7
0.8
0.9
1
1.5
1.1
6
2
2.5
3
3.5
4
Supply (V)
Reference Voltage (V)
ADS8324
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SBAS172A
THEORY OF OPERATION
2 • VREF
peak-to-peak
Single-Ended Input
VREF
peak-to-peak
Common
Voltage
ADS8324
VREF
peak-to-peak
Differential Input
FIGURE 1. Methods of Driving the ADS8324—Single-Ended
or Differential.
2
1.8
1.6
1.4
1.2
1
0.8
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
–1
VCC = 1.8V
Single-Ended Input
0.5
0.6
0.7
0.8
0.9
1
VREF (V)
The analog input is bipolar and fully differential. There are
two general methods of driving the analog input of the
ADS8324: single-ended or differential, as shown in Figure
1. When the input is single-ended, the –In input is held at a
fixed voltage. The +In input swings around the same voltage
and the peak-to-peak amplitude is 2 • VREF. The value of
VREF determines the range over which the common voltage
may vary, as shown in Figure 2.
When the input is differential, the amplitude of the input is
the difference between the +In and –In input, or, +In – (–In).
A voltage or signal is common to both of these inputs. The
peak-to-peak amplitude of each input is VREF about this
common voltage. However, since the inputs are 180° out-ofphase, the peak-to-peak amplitude of the difference voltage
is 2 • VREF. The value of VREF also determines the range of
the voltage that may be common to both inputs, as shown in
Figure 3.
In each case, care should be taken to ensure that the output
impedance of the sources driving the +In and –In inputs are
matched. If this is not observed, the two inputs could have
Common Voltage Range (V)
FIGURE 2. Single-Ended Input—Common Voltage Range
vs VREF.
ANALOG INPUT
2
1.8
1.6
1.4
1.2
1
0.8
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
–1
Differential Input
VCC = 1.8V
0.5
0.6
0.7
0.8
0.9
1
VREF (V)
FIGURE 3. Differential Input—Common Voltage Range vs
VREF.
ADS8324
SBAS172A
ADS8324
Common
Voltage
Common Voltage Range (V)
The ADS8324 is a classic Successive Approximation Register (SAR) A/D converter. The architecture is based on
capacitive redistribution that inherently includes a sampleand-hold function. The converter is fabricated on a 0.6μ
CMOS process. The architecture and process allow the
ADS8324 to acquire and convert an analog signal at up to
50,000 conversions per second while consuming less than
3.0mW from +VCC.
The ADS8324 requires an external reference, an external
clock, and a single power source (VCC). The external reference can be any voltage between 500mV and VCC /2. The
value of the reference voltage directly sets the range of the
analog input. The reference input current depends on the
conversion rate of the ADS8324.
The external clock can vary between 24kHz (1kHz throughput) and 1.2MHz (50kHz throughput). The duty cycle of the
clock is essentially unimportant as long as the minimum
high and low times are at least 200ns. The minimum clock
frequency is set by the leakage on the capacitors internal to
the ADS8324.
The analog input is provided to two input pins: +In and –In.
When a conversion is initiated, the differential input on these
pins is sampled on the internal capacitor array. While a
conversion is in progress, both inputs are disconnected from
any internal function.
The digital result of the conversion is clocked out by the
DCLOCK input and is provided serially, most significant bit
first, on the DOUT pin. The digital data that is provided on the
DOUT pin is for the conversion currently in progress—there
is no pipeline delay. It is possible to continue to clock the
ADS8324 after the conversion is complete and to obtain the
serial data least significant bit first. See the digital timing
section for more information.
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7
different settling times. This may result in offset error, gain
error, and linearity error that changes with both temperature
and input voltage. If the impedance cannot be matched, the
errors can be lessened by giving the ADS8324 additional
acquisition time.
The input current on the analog inputs depends on a number
of factors: sample rate, input voltage, and source impedance.
Essentially, the current into the ADS8324 charges the internal capacitor array during the sample period. After this
capacitance has been fully charged, there is no further input
current. The source of the analog input voltage must be able
to charge the input capacitance (25pF) to the 14-bit settling
level within 4.5 clock cycles. When the converter goes into
the hold mode, or while it is in the power-down mode, the
input impedance is greater than 1GΩ.
Care must be taken regarding the absolute analog input
voltage. The +In input should always remain within the
range of GND – 100mV to VCC + 100mV. The –In input
should always remain within the range of GND – 100mV to
2V. Outside of these ranges, the converter’s linearity may
not meet specifications.
NOISE
The noise floor of the ADS8324 itself is extremely low, as
can be seen from Figure 4, and is much lower than competing A/D converters. It was tested by applying a low noise
DC input and a 0.9V reference to the ADS8324 and initiating 5,000 conversions. The digital output of the A/D converter will vary in output code due to the internal noise of
the ADS8324. This is true for all 14-bit SAR-type A/D
converters. Using a histogram to plot the output codes, the
distribution should appear bell-shaped, with the peak of the
bell curve representing the nominal code for the input value.
The ±1σ, ±2σ, and ±3σ distributions will represent the
68.3%, 95.5%, and 99.7%, respectively, of all codes. The
transition noise can be calculated by dividing the number of
codes measured by 6 and this will yield the ±3σ distribution
or 99.7% of all codes. Statistically, up to 3 codes could fall
outside the distribution when executing 1000 conversions.
The ADS8324, with five output codes for the ±3σ distribution, will yield a ±0.8LSB transition noise. Remember, to
achieve this low-noise performance, the peak-to-peak noise
of the input signal and reference must be < 50μV.
REFERENCE INPUT
3857
The external reference sets the analog input range. The
ADS8324 will operate with a reference in the range of
500mV to VCC /2. There are several important implications
of this. As the reference voltage is reduced, the analog
voltage weight of each digital output code is reduced. This
is often referred to as the Least Significant Bit (LSB) size
and is equal to 2 • VREF divided by 16,384. This means that
any offset or gain error inherent in the A/D converter will
appear to increase, in terms of LSB size, as the reference
voltage is reduced.
The noise inherent in the converter will also appear to increase
with lower LSB size. With a 0.9V reference, the internal noise
of the converter typically contributes only 5LSB peak-to-peak
of potential error to the output code. When the external
reference is 500mV, the potential error contribution from the
internal noise will be 7LSBs. The errors due to the internal
noise are gaussian in nature and can be reduced by averaging
consecutive conversion results.
For more information regarding noise, consult the typical
performance curve “Noise vs Reference Voltage.” Note that
the Effective Number of Bits (ENOB) figure is calculated
based on the converter’s signal-to-(noise + distortion) ratio
with a 1kHz, 0dB input signal. SINAD is related to ENOB
as follows:
SINAD = 6.02 • ENOB + 1.76
With lower reference voltages, extra care should be taken to
provide a clean layout including adequate bypassing, a clean
power supply, a low-noise reference, and a low-noise input
signal. Because the LSB size is lower, the converter will also
be more sensitive to external sources of error such as nearby
digital signals and electromagnetic interference.
8
583
560
0
3FFEH
0
3FFFH
0000H
0001H
0002H
Code
FIGURE 4. Histogram of 5,000 Conversions of a DC Input
at the Code Transition.
AVERAGING
The noise of the A/D converter can be compensated by
averaging the digital codes. By averaging conversion results, transition noise will be reduced by a factor of 1/√n,
where n is the number of averages. For example, averaging
4 conversion results will reduce the transition noise by 1/2
to ±0.25LSBs. Averaging should only be used for input
signals with frequencies near DC.
For AC signals, a digital filter can be used to low-pass filter
and decimate the output codes. This works in a similar
manner to averaging; for every decimation by 2, the signalto-noise ratio will improve 3dB.
ADS8324
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SBAS172A
DIGITAL INTERFACE
SYMBOL
SIGNAL LEVELS
The CMOS digital output (DOUT) will swing from 0V to
VCC. If VCC is 3V, and this output is connected to a 5V
CMOS logic input, then that IC may require more supply
current than normal and may have a slightly longer propagation delay.
DESCRIPTION
MIN
tSMPL
Analog Input Sample Time
4.5
TYP
MAX
UNITS
5.0
Clk Cycles
tCONV
Conversion Time
tCYC
Throughput Rate
50
kHz
tCSD
CS Falling to
0
ns
16
Clk Cycles
DCLOCK LOW
CS Falling to
tSUCS
50
ns
DCLOCK Rising
SERIAL INTERFACE
The ADS8324 communicates with microprocessors and
other digital systems via a synchronous 3-wire serial interface, as shown in Figure 5 and Table I. The DCLOCK signal
synchronizes the data transfer with each bit being transmitted on the falling edge of DCLOCK. Most receiving systems
will capture the bitstream on the rising edge of DCLOCK.
However, if the minimum hold time for DOUT is acceptable,
the system can use the falling edge of DCLOCK to
capture each bit.
A falling CS signal initiates the conversion and data transfer.
The first 4.5 to 5.0 clock periods of the conversion cycle are
used to sample the input signal. After the fifth falling
DCLOCK edge, DOUT is enabled and will output a LOW
value for one clock period. For the next 16 DCLOCK
periods, DOUT will output the conversion result, most significant bit first followed by two zeros on clock cycles 15
and 16. After the two zero “dummy bits” have been output,
subsequent clocks will repeat the output data but in a least
significant bit first format starting with a zero.
CS must be taken HIGH following a conversion in order to
place DOUT in tri-state. Subsequent clocks will have no
effect on the converter. A new conversion is initiated only
when CS has been taken HIGH and returned LOW.
thDO
DCLOCK Falling to
Current DOUT Not Valid
tdDO
DCLOCK Falling to Next
DOUT Valid
tdis
ten
5
20
ns
100
250
ns
CS Rising to DOUT Tri-State
50
100
ns
DCLOCK Falling to DOUT
Enabled
100
200
ns
tf
DOUT Fall Time
50
150
ns
tr
DOUT Rise Time
75
200
ns
TABLE I. Timing Specifications (VCC = 1.8V) –40°C to
+85°C.
See Figure 6 for test conditions.
DATA FORMAT
The output data from the ADS8324 is in Binary Two’s
Complement format, as shown in Table II. This table represents the ideal output code for the given input voltage and
does not include the effects of offset, gain error, or noise.
DESCRIPTION
ANALOG VALUE
DIGITAL OUTPUT
Full-Scale Range
2 • VREF
Least Significant
Bit (LSB)
2 • VREF/16384
BINARY CODE
HEX CODE
+Full Scale
+VREF – 1 LSB
0111 1111 1111 1100
7FFC
0V
0000 0000 0000 0000
0000
0V – 1 LSB
1111 1111 1111 1100
FFFC
–VREF
1000 0000 0000 0000
8000
Midscale
Midscale – 1LSB
–Full Scale
BINARY TWO’S COMPLEMENT
TABLE II. Ideal Input Voltages and Output Codes.
Complete Cycle
CS/SHDN
tSUCS
Sample
Power Down
Conversion
DCLOCK
tCSD
DOUT
Use positive clock edge for data transfer
Hi-Z
0
B13 B12 B11 B10 B9
(MSB)
tSMPL
B8
B7
B6
B5
B4
B3 B2 B1
B0 0
(LSB)
0
Hi-Z
tCONV
NOTE: Minimum 22 clock cycles required for 14-bit conversion. Shown are 24 clock cycles.
If CS remains LOW at the end of conversion, a new datastream with LSB-first is shifted out again.
FIGURE 5. ADS8324 Basic Timing Diagrams.
ADS8324
SBAS172A
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9
POWER DISSIPATION
The architecture of the converter, the semiconductor fabrication process, and a careful design allow the ADS8324 to
convert at up to a 50kHz rate while requiring very little
power. Still, for the absolute lowest power dissipation, there
are several things to keep in mind.
The power dissipation of the ADS8324 scales directly with
the conversion rate. Therefore, the first step to achieving the
lowest power dissipation is to find the lowest conversion rate
that will satisfy the requirements of the system.
In addition, the ADS8324 is in power-down mode under two
conditions: when the conversion is complete and whenever
CS is HIGH (see Figure 5). Ideally, each conversion should
occur as quickly as possible, preferably at a 1.2MHz clock
rate. This way, the converter spends the longest possible
time in the power-down mode. This is very important as the
converter not only uses power on each DCLOCK transition
(as is typical for digital CMOS components) but also uses
some current for the analog circuitry, such as the comparator. The analog section dissipates power continuously, until
the power-down mode is entered.
0.9V
3kΩ
DOUT
VOH
DOUT
VOL
Test Point
tr
30pF
CLOAD
tf
Voltage Waveforms for DOUT Rise and Fall Times, tr, tf
Load Circuit for tdDO, tr, and tf
Test Point
DCLOCK
VIL
VCC
DOUT
tdDO
VOH
DOUT
tdis Waveform 2, ten
3kΩ
tdis Waveform 1
30pF
CLOAD
VOL
thDO
Load Circuit for tdis and ten
Voltage Waveforms for DOUT Delay Times, tdDO
VIH
CS/SHDN
DOUT
Waveform 1(1)
CS/SHDN
90%
DCLOCK
5
6
tdis
DOUT
Waveform 2(2)
VOL
DOUT
10%
B11
ten
Voltage Waveforms for tdis
Voltage Waveforms for ten
NOTES: (1) Waveform 1 is for an output with internal conditions such that the output
is HIGH unless disabled by the output control. (2) Waveform 2 is for an output with
internal conditions such that the output is LOW unless disabled by the output control.
FIGURE 6. Timing Diagrams and Test Circuits for the Parameters in Table I.
10
ADS8324
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SBAS172A
10000
TA = 25°C
VCC = 1.8V
VREF = 0.9V
fCLK = 24 • fSAMPLE
1000
800
Supply Current (μA)
Figure 7 shows the current consumption of the ADS8324
versus sample rate. For this graph, the converter is clocked
at 1.2MHz regardless of the sample rate—CS is HIGH for
the remaining sample period. Figure 8 also shows current
consumption versus sample rate. However, in this case, the
DCLOCK period is 1/24th of the sample period—CS is
HIGH for one DCLOCK cycle out of every 16.
There is an important distinction between the power-down
mode that is entered after a conversion is complete and the
full power-down mode that is enabled when CS is HIGH. CS
LOW will shut down only the analog section. The digital
section is completely shutdown only when CS is HIGH.
Thus, if CS is left LOW at the end of a conversion and the
converter is continually clocked, the power consumption
will not be as low as when CS is HIGH, shown in Figure 9.
600
400
CS LOW (GND)
200
0.250
CS HIGH (VCC)
0.00
0.1
1
10
100
Sample Rate (kHz)
Supply Current (μA)
10000
FIGURE 9. Shutdown Current with CS HIGH is 50nA
Typically, Regardless of the Clock. Shutdown
Current with CS LOW Varies with Sample
Rate.
TA = 25°C
VCC = 1.8V
VREF = 0.9V
fCLK = 2.4MHz
1000
LAYOUT
100
10
0.1
1
10
100
Sample Rate (kHz)
FIGURE 7. Maintaining fCLK at the Highest Possible Rate
Allows Supply Current to Drop Linearly with
Sample Rate.
Supply Current (μA)
10000
1000
100
TA = 25°C
VCC = 1.8V
VREF = 0.9V
fCLK = 24 • fSAMPLE
10
0.1
1
10
100
Sample Rate (kHz)
FIGURE 8. Scaling fCLK Reduces Supply Current Only
Slightly with Sample Rate.
For optimum performance, care should be taken with the
physical layout of the ADS8324 circuitry. This will be
particularly true if the reference voltage is low and/or the
conversion rate is high. At a 50kHz conversion rate, the
ADS8324 makes a bit decision every 213ns. That is, for each
subsequent bit decision, the digital output must be updated
with the results of the last bit decision, the capacitor array
appropriately switched and charged, and the input to the
comparator settled to a 14-bit level all within one clock
cycle.
The basic SAR architecture is sensitive to spikes on the
power supply, reference, and ground connections that occur
just prior to latching the comparator output. Thus, during
any single conversion for an n-bit SAR converter, there are
n “windows” in which large external transient voltages can
easily affect the conversion result. Such spikes might originate from switching power supplies, digital logic, and high
power devices, to name a few. This particular source of error
can be very difficult to track down if the glitch is almost
synchronous to the converter’s DCLOCK signal—as the
phase difference between the two changes with time and
temperature, causing sporadic misoperation.
With this in mind, power to the ADS8324 should be clean
and well bypassed. A 0.1μF ceramic bypass capacitor should
be placed as close to the ADS8324 package as possible. In
addition, a 1μF to 10μF capacitor and a 5Ω or 10Ω series
resistor may be used to low-pass filter a noisy supply.
The reference should be similarly bypassed with a 0.1μF
capacitor. Again, a series resistor and large capacitor can be
used to low-pass filter the reference voltage. If the reference
voltage originates from an op amp, be careful that the op
ADS8324
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11
amp can drive the bypass capacitor without oscillation (the
series resistor can help in this case). Keep in mind that while
the ADS8324 draws very little current from the reference on
average, there are still instantaneous current demands placed
on the external input and reference circuitry.
Texas Instruments OPA627 op amp provides optimum performance for buffering both the signal and reference inputs.
For low-cost, low-voltage, single-supply applications, the
OPA2350 or OPA2340 dual op amps are recommended.
Also, keep in mind that the ADS8324 offers no inherent
rejection of noise or voltage variation in regards to the
reference input. This is of particular concern when the
reference input is tied to the power supply. Any noise and
ripple from the supply will appear directly in the digital
results. While high frequency noise can be filtered out as
described in the previous paragraph, voltage variation due to
the line frequency (50Hz or 60Hz), can be difficult to
remove.
The GND pin on the ADS8324 should be placed on a clean
ground point. In many cases, this will be the “analog”
ground. Avoid connecting the GND pin too close to the
grounding point for a microprocessor, microcontroller, or
digital signal processor. If needed, run a ground trace directly from the converter to the power supply connection
point. The ideal layout will include an analog ground plane
for the converter and associated analog circuitry.
APPLICATION CIRCUITS
Figure 10 shows a basic data acquisition system. The
ADS8324 input range is 0V to VCC, as the reference input is
connected directly to the power supply. The 5Ω resistor and
1μF to 10μF capacitor filter the microcontroller “noise” on
the supply, as well as any high-frequency noise from the
supply itself. The exact values should be picked such that the
filter provides adequate rejection of the noise.
1.8V
5Ω
+
1μF to 10μF
ADS8324
0.9V
Reference
VREF
VCC
0.1μF
0V to 1.8V
+In
CS
–In
DOUT
GND
+ 1μF to
10μF
Microcontroller
DCLOCK
FIGURE 10. Basic Data Acquisition System.
12
ADS8324
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SBAS172A
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
ADS8324E/250
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
A24
Samples
ADS8324E/2K5
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
A24
Samples
ADS8324EB/2K5
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
A24
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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