ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
16-Bit, High-Speed, 2.7V to 5.5V microPower Sampling
ANALOG-TO-DIGITAL CONVERTER
Check for Samples: ADS8326
FEATURES
APPLICATIONS
•
•
•
•
•
1
23
•
•
•
•
•
•
•
16 Bits No Missing Codes (Full-Supply Range,
High or Low Grade)
Very Low Noise: 3LSBPP
Excellent Linearity:
±1LSB typ, ±1.5LSB max INL
±0.6LSB typ, ±1LSB max DNL
±1mV max Offset
±12LSB typ Gain Error
microPower:
10mW at 5V, 250kHz
4mW at 2.7V, 200kHz
2mW at 2.7V, 100kHz
0.2mW at 2.7V, 10kHz
MSOP-8 and SON-8 Packages
(SON-8 package same as 3x3 QFN)
16-Bit Upgrade to the 12-Bit ADS7816 and
ADS7822
Pin-Compatible with the ADS7816, ADS7822,
ADS7826, ADS7827, ADS7829, ADS8320, and
ADS8325
Serial ( SPI™/SSI) Interface
•
•
•
Battery-Operated Systems
Remote Data Acquisition
Isolated Data Acquisition
Simultaneous Sampling, Multichannel
Systems
Industrial Controls
Robotics
Vibration Analysis
DESCRIPTION
The ADS8326 is a 16-bit, sampling, analog-to-digital
(A/D) converter specified for a supply voltage range
from 2.7V to 5.5V. It requires very little power, even
when operating at the full data rate. At lower data
rates, the high speed of the device enables it to
spend most of its time in the power-down mode. For
example, the average power dissipation is less than
0.2mW at a 10kHz data rate.
The ADS8326 offers excellent linearity and very low
noise and distortion. It also features a synchronous
serial (SPI/SSI-compatible) interface and a differential
input. The reference voltage can be set to any level
within the range of 0.1V to VDD.
Low power and small size make the ADS8326 ideal
for portable and battery-operated systems. It is also a
perfect fit for remote data-acquisition modules,
simultaneous multichannel systems, and isolated
data acquisition. The ADS8326 is available in either
an MSOP-8 and an SON-8 package.
SAR
REF
ADS8326
DOUT
+IN
CDAC
Serial
Interface
-IN
DCLOCK
S/H Amp
Comparator
CS/SHDN
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SPI is a trademark of Motorola, Inc.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007–2009, Texas Instruments Incorporated
ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
PRODUCT
MAXIMUM
INTEGRAL
LINEARITY
ERROR
(LSB) (2)
NO
MISSING
CODES
ERROR
(LSB)
PACKAGELEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ADS8326I
±3
16
MSOP-8
DGK
–40°C to +85°C
D26
ADS8326IB
ADS8326I
ADS8326IB
(1)
(2)
±1.5
16
±3
16
±1.5
16
MSOP-8
DGK
SON-8
DRB
SON-8
DRB
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
ORDERING
NUMBER
TRANSPORT
MEDIA,
QUANTITY
ADS8326IDGKT
Tape and Reel,
250
ADS8326IDGKR
Tape and Reel,
2500
ADS8326IBDGKT
Tape and Reel,
250
ADS8326IBDGKR
Tape and Reel,
2500
ADS8326IDRBT
Tape and Reel,
250
ADS8326IDRBR
Tape and Reel,
2500
ADS8326IBDRBT
Tape and Reel,
250
ADS8326IBDRBR
Tape and Reel,
2500
D26
D26
D26
For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet, or see
the TI website at www.ti.com.
Maximum Integral Linearity Error specifies a 5V power supply and reference voltage.
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range (unless otherwise noted).
ADS8326
UNIT
–0.3 to +7
V
–0.3 to VDD + 0.3
V
–0.3 to VDD + 0.3
V
Supply voltage, VDD to GND
Analog input voltage (2)
Reference input voltage
(2)
Digital input voltage (2)
–0.3 to VDD + 0.3
V
–20 to +20
mA
Input current to any pin except supply
Power dissipation
See Dissipation Ratings Table
Operating virtual junction temperature range, TJ
–40 to +150
°C
Operating free-air temperature range, TA
–40 to +85
°C
Storage temperature range, TSTG
–65 to +150
°C
+260
°C
Lead Temperature 1.6mm (1/16 inch) from case for 10sec
(1)
(2)
2
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum rated conditions for extended periods may affect device reliability.
All voltage values are with respect to ground terminal.
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ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
DISSIPATION RATINGS
PACKAGE
R θ JC
R θ JA
DERATING
FACTOR ABOVE
TA = +25°C
DGK
+39.1°C/W
+206.3°C/W
4.847mW/°C
606mW
388mW
315mW
DRB
+5°C/W
+45.8°C/W
3.7mW/°C
370mW
204mW
148mW
TA ≤ +25°C
POWER RATING
TA = +70°C
POWER RATING
TA = +85°C
POWER RATING
RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage, GND to VDD
Low-voltage levels
2.7
Supply voltage, GND to VDD
5V logic levels
4.5
Reference input voltage
TYP
5.0
0.1
Analog input voltage
UNIT
3.6
V
5.5
V
VDD
V
–IN to GND
–0.3
0.5
V
+IN to GND
–0.3
VDD + 0.2
V
+IN – (–IN)
0
VREF
V
–40
+125
°C
Operating junction temperature, TJ
0
MAX
ELECTRICAL CHARACTERISTICS: VDD = +5V
At –40°C to +85°C, VREF = +5V, –IN = GND, fSAMPLE = 250kHz, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8326I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8326IB
MAX
MIN
TYP
MAX
UNIT
V
ANALOG INPUT
Full-scale range
FSR +IN – (–IN)
Operating common-mode signal
Input resistance
RON
0
VREF
0
VREF
–0.3
0.5
–0.3
0.5
–IN = GND, off
5
V
5
50
–IN = GND, during sampling
48
48
pF
Input leakage current
–IN = GND
±50
±50
nA
Differential input capacitance
+IN to –IN, during sampling
20
20
pF
500
500
kHz
Full-power bandwidth
50
Ω
–IN = GND, on
Input capacitance
FS sinewave, SINAD =
FSBW
–60dB
100
GΩ
100
DC ACCURACY
Resolution
No missing codes
Integral linearity error
16
16
NMC
16
16
INL
–3
±2
Bits
Bits
+3
–1.5
±1
+1.5
LSB
LSB
Differential linearity error
DNL
–1
±0.5
+2
–1
±0.4
+1
Offset error
VOS
–1.5
±0.75
+1.5
–1
±0.5
+1
Offset error drift
Gain error
Gain error drift
TCVOS
±0.2
GERR
–24
TCGERR
+24
–12
±0.3
Noise
4.75V ≤ VDD ≤ 5.25V
Power-supply rejection
±0.2
mV
ppm/°C
+12
LSB
±0.3
ppm/°C
30
30
μVRMS
0.5
0.5
LSB
SAMPLING DYNAMICS
Conversion time
(16 DCLOCKs)
Acquisition time
(4.5 DCLOCKs)
tCONV 24kHz ≤ fDCLOCK ≤ 6MHz
tAQ fDCLOCK = 6MHz
2.667
0.75
Throughput rate
(22 DCLOCKs)
Clock frequency
666.7
2.667
0.024
6
0.024
250
kSPS
6
MHz
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μs
μs
0.75
250
fDCLOCK
666.7
3
ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
ELECTRICAL CHARACTERISTICS: VDD = +5V (continued)
At –40°C to +85°C, VREF = +5V, –IN = GND, fSAMPLE = 250kHz, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8326I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8326IB
MAX
MIN
TYP
MAX
UNIT
AC ACCURACY
Total harmonic distortion
Spurious-free dynamic
range
Signal-to-noise ratio
Signal-to-noise + distortion
Effective number of bits
THD
SFDR
SNR
SINAD
ENOB
5VPP sinewave at 2kHz
–98
–99
dB
5VPP sinewave at 10kHz
–90
–91
dB
5VPP sinewave at 2kHz
102
103
dB
5VPP sinewave at 10kHz
94
95
dB
5VPP sinewave at 2kHz
91
91.5
dB
5VPP sinewave at 10kHz
91
91.5
dB
5VPP sinewave at 2kHz
90
91
dB
5VPP sinewave at 10kHz
87.5
88
dB
5VPP sinewave at 2kHz
14.69
14.86
Bits
5VPP sinewave at 10kHz
14.28
14.35
Bits
VOLTAGE REFERENCE INPUT
Reference voltage
0.1
CS = GND, fSAMPLE = 0Hz
Reference input resistance
CS = VDD
Reference input capacitance
Reference input current
VDD
0.1
5
VDD
V
5
GΩ
5
5
GΩ
24
24
pF
fS = 250kHz
170
220
170
220
μA
fS = 200kHz
140
180
140
180
μA
fS = 100kHz
70
90
70
90
μA
fS = 10kHz
11
14
11
14
μA
CS = VDD
0.1
μA
0.1
DIGITAL INPUTS (1)
Logic family
CMOS
CMOS
High-level input voltage
VIH
0.7 × VDD
VDD + 0.3
0.7 × VDD
VDD + 0.3
Low-level input voltage
VIL
–0.3
0.3 × VDD
–0.3
0.3 × VDD
V
Input current
IIN VI = VDD or GND
–50
+50
–50
+50
nA
Input capacitance
CI
5
5
V
pF
DIGITAL OUTPUTS (1)
Logic family
CMOS
High-level output voltage
VOH VDD = 4.5V, IOH = –100μA
Low-level output voltage
VOL VDD = 4.5V, IOL = 100μA
High-impedance state
output current
IOZ CS = VDD, VI = VDD or GND
Output capacitance
CO
Load capacitance
CL
4
4.44
V
0.5
–50
+50
5
–50
Straight
binary
0.5
V
+50
nA
5
30
Data format
(1)
CMOS
4.44
pF
30
pF
Straight
binary
Applies for 5.0V nominal supply: VDD (min) = 4.5V and VDD (max) = 5.5V.
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Product Folder Link(s): ADS8326
ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
ELECTRICAL CHARACTERISTICS: VDD = +2.7V
At –40°C to +85°C, VREF = +2.5V, –IN = GND, fSAMPLE = 200kHz, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8326I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8326IB
MAX
MIN
TYP
MAX
UNIT
V
ANALOG INPUT
Full-scale range
FSR +IN – (–IN)
Operating common-mode signal
Input resistance
RON
0
VREF
0
VREF
–0.3
0.5
–0.3
0.5
–IN = GND, off
5
–IN = GND, on
100
Input capacitance
–IN = GND, during sampling
Input leakage current
–IN = GND
Differential input capacitance
+IN to –IN, during sampling
FS sinewave, SINAD =
FSBW
–60dB
Full-power bandwidth
V
5
150
100
GΩ
Ω
150
48
48
pF
±50
±50
nA
20
20
pF
60
60
kHz
DC ACCURACY
Resolution
No missing codes
Integral linearity error
16
16
NMC
16
16
INL
–3
±2
Bits
Bits
+3
–2.5
±1
+2.5
LSB
LSB
Differential linearity error
DNL
–1
±0.5
+2
–1
±0.4
+1
Offset error
VOS
–1.5
±0.75
+1.5
–1
±0.5
+1
Offset error drift
±0.2
±0.2
ppm/°C
GERR
±33
±16
LSB
TCGERR
±0.3
±0.3
ppm/°C
30
30
μVRMS
0.5
0.5
LSB
Gain error
Gain error drift
mV
TCVOS
Noise
2.7V ≤ VDD ≤ 3.6V
Power-supply rejection
SAMPLING DYNAMICS
Conversion time
(16 DCLOCKs)
Acquisition time
(4.5 DCLOCKs)
tCONV 24kHz ≤ fDCLOCK ≤ 4.8MHz
tAQ fDCLOCK = 4.8MHz
3.333
666.7
0.9375
666.7
200
fDCLOCK
0.024
4.8
μs
μs
0.9375
Throughput rate
(22 DCLOCKs)
Clock frequency
3.333
0.024
200
kSPS
4.8
MHz
AC ACCURACY
Total harmonic distortion
Spurious-free dynamic
range
Signal-to-noise ratio
Signal-to-noise + distortion
Effective number of bits
THD
SFDR
SNR
SINAD
ENOB
2.5VPP sinewave at 2kHz
–88
–88.5
dB
2.5VPP sinewave at 10kHz
–75
–75.5
dB
2.5VPP sinewave at 2kHz
91
91.5
dB
2.5VPP sinewave at 10kHz
77.5
78
dB
2.5VPP sinewave at 2kHz
86.5
87
dB
2.5VPP sinewave at 10kHz
86
86.5
dB
2.5VPP sinewave at 2kHz
85
85.5
dB
2.5VPP sinewave at 10kHz
74.5
75
dB
2.5VPP sinewave at 2kHz
13.86
13.94
Bits
2.5VPP sinewave at 10kHz
12.12
12.20
Bits
VOLTAGE REFERENCE INPUT
Reference voltage
Reference input resistance
0.1
CS = GND, fSAMPLE = 0Hz
CS = VDD
Reference input capacitance
Reference input current
VDD
5
0.1
VDD
V
5
GΩ
5
5
GΩ
24
24
pF
fS = 200kHz
70
90
70
90
μA
fS = 100kHz
25
33
25
33
μA
fS = 10kHz
5
7
5
7
μA
CS = VDD
0.1
0.1
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μA
5
ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
ELECTRICAL CHARACTERISTICS: VDD = +2.7V (continued)
At –40°C to +85°C, VREF = +2.5V, –IN = GND, fSAMPLE = 200kHz, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8326I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8326IB
MAX
MIN
TYP
MAX
UNIT
DIGITAL INPUTS (1)
Logic family
LVCMOS
LVCMOS
High-level input voltage
VIH VDD = 3.6V
2
VDD + 0.3
2
VDD + 0.3
Low-level input voltage
VIL VDD = 2.7V
–0.3
0.8
–0.3
0.8
V
Input current
IIN VI = VDD or GND
–50
+50
–50
+50
nA
Input capacitance
CI
5
5
V
pF
DIGITAL OUTPUTS (1)
Logic family
LVCMOS
High-level output voltage
VOH VDD = 2.7V, IOH = –100μA
Low-level output voltage
VOL VDD = 2.7V, IOL = 100μA
High-impedance state
output current
IOZ CS = VDD, VI = VDD or GND
Output capacitance
CO
Load capacitance
CL
VDD – 0.2
V
0.2
–50
+50
–50
5
0.2
V
+50
nA
5
30
pF
30
Straight
binary
Data format
(1)
LVCMOS
VDD – 0.2
pF
Straight
binary
Applies for 3.0V nominal supply: VDD (min) = 2.7V and VDD (max) = 3.6V.
ELECTRICAL CHARACTERISTICS
At –40°C to +85°C, –IN = GND, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8326I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8326IB
MAX
MIN
TYP
MAX
UNIT
ANALOG INPUT
Power supply
Operating supply current
Power-down supply current
Power dissipation
Power dissipation in power-down
6
VDD
IDD
IDD
Low-voltage levels
2.7
3.6
2.7
3.6
V
5V logic levels
4.5
5.5
4.5
5.5
V
VDD = 2.7V, fS = 10kHz,
fDCLOCK = 4.8MHz
0.065
0.085
0.065
0.085
mA
VDD = 2.7V, fS = 100kHz,
fDCLOCK = 4.8MHz
0.69
1.0
0.69
1.0
mA
VDD = 2.7V, fS = 200kHz,
fDCLOCK = 4.8MHz
1.38
2.0
1.38
2.0
mA
VDD = 5V, fS = 200kHz,
fDCLOCK = 6MHz
1.9
2.7
1.9
2.7
mA
VDD = 5V, fS = 250kHz,
fDCLOCK = 6MHz
2.0
3.0
2.0
3.0
mA
VDD = 2.7V
0.1
0.1
μA
VDD = 5V
0.2
0.2
μA
VDD = 2.7V, fS = 10kHz,
fDCLOCK = 4.8MHz
0.18
0.23
0.18
0.23
mW
VDD = 2.7V, fS = 100kHz,
fDCLOCK = 4.8MHz
1.86
2.7
1.86
2.7
mW
VDD = 2.7V, fS = 200kHz,
fDCLOCK = 4.8MHz
3.73
5.4
3.73
5.4
mW
VDD = 5V, fS = 200kHz,
fDCLOCK = 6MHz
9.5
13.5
9.5
13.5
mW
VDD = 5V, fS = 250kHz,
fDCLOCK = 6MHz
10
15
10
15
mW
VDD = 2.7V, CS = VDD
0.3
0.3
μW
VDD = 5V, CS = VDD
0.6
0.6
μW
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ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
PIN CONFIGURATION
DGK PACKAGE
MSOP-8
(TOP VIEW)
REF
1
+IN
2
8
VDD
7
DCLOCK
ADS8326
-IN
3
6
DOUT
GND
4
5
CS/SHDN
DRB PACKAGE
SON-8
(TOP VIEW)
REF
1
+IN
2
8
VDD
7
DCLOCK
6
DOUT
5
CS/SHDN
ADS8326
(1)
-IN
3
GND
4
(Thermal Pad)
The thermal pad is internally connected to the substrate. This pad can be connected to the analog ground or left
floating. Keep the thermal pad separate from the digital ground, if possible.
PIN ASSIGNMENTS
PIN
NAME
NO.
I/O
DESCRIPTION
REF
1
Analog input
Reference input
+IN
2
Analog input
Noninverting input
–IN
3
Analog input
Inverting analog input
GND
4
Power-supply connection
CS/SHDN
5
Digital input
DOUT
6
Digital output
DCLOCK
7
Digital input
VDD
8
Power-supply connection
Ground
Chip select when low; Shutdown mode when high.
Serial output data word
Data clock synchronizes the serial data transfer and determines conversion speed.
Power supply
Equivalent Input Circuit (VDD = 5.0V)
VDD
VDD
RON
50W
C(SAMPLE)
48pF
ANALOG IN
GND
Diode Turn-On Voltage: 0.35V
Equivalent Analog Input Circuit
VDD
RON
50W
REF
GND
Equivalent Reference Input Circuit
24pF
I/O
GND
Equivalent Digital Input/Output Circuit
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7
ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
TIMING INFORMATION
tCYC
CS/SHDN
Sample
Power Down
Conversion
tSUCS
DCLOCK
tCSD
Use positive clock edge for data transfer
Hi-Z
DOUT
0
tSMPL
B7
B15 B14 B13 B12 B11 B10 B9 B8
(MSB)
tCONV
B6
B5 B4
B3
B2
Hi-Z
(1)
B1 B0
(LSB)
NOTE: (1) A minimum of 22 clock cycles are required for 16-bit conversion; 24 clock cycles are shown.
If CS remains low at the end of conversion, a new data stream is shifted out with LSB-first data followed by zeroes indefinitely.
tCYC
CS/SHDN
tSUCS
Power Down
DCLOCK
tCSD
Hi-Z
DOUT
0
tSMPL
B15 B14 B6
(MSB)
B5
B4
B3
tCONV
B2
B1
B0
B1
B2
B3
B4
B5
(LSB)
B0
(2)
Hi-Z
B11 B12 B13 B14 B15
(MSB)
NOTE: (2) After completing the data transfer, if further clocks are applied with CS low, the A/D converter will output zeroes indefinitely.
1.4V
3kW
DOUT
90%
DOUT
10%
Test Point
tr
100pF
CLOAD
tf
Voltage Waveforms for DOUT Rise and Fall Times, tr, tf
Load Circuit for tdDO, tr, and tf
Test Point
DCLOCK
VDD
DOUT
tdDO
tdis Waveform 2, ten
3kW
tdis Waveform 1
100pF
CLOAD
DOUT
thDO
Load Circuit for tdis and ten
Voltage Waveforms for DOUT Delay Times, tdDO
90%
CS/SHDN
DOUT
Waveform 1(3)
CS/SHDN
90%
DCLOCK
1
4
5
tdis
DOUT
Waveform 2(4)
10%
DOUT
B15
ten
Voltage Waveforms for tdis
Voltage Waveforms for ten
NOTES: (3) Waveform 1 is for an output with internal conditions such that
the output is high unless disabled by the output control.
(4) Waveform 2 is for an output with internal conditions such that
the output is low unless disabled by the output control.
Figure 1. Timing Diagrams and Test Circuits for the Parameters in Table 1
8
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TIMING INFORMATION (continued)
Table 1. Timing Characteristics
SYMBOL
DESCRIPTION
tSMPL
Analog input sample time
tCONV
Conversion time
tCYC
Complete cycle time
MIN
TYP
4.5
MAX
5.0
16
UNIT
DCLOCKs
DCLOCKs
22
DCLOCKs
tCSD
CS falling to DCLOCK low
tSUCS
CS falling to DCLOCK rising
0
tHDO
DCLOCK falling to current DOUT not valid
tDIS
CS rising to DOUT tri-state
70
100
ns
tEN
DCLOCK falling to DOUT enabled
20
50
ns
tF
DOUT fall time
5
25
ns
tR
DOUT rise time
7
25
ns
20
5
ns
ns
15
ns
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TYPICAL CHARACTERISTICS: VDD = +5V
At TA = +25°C, VDD = +5V, VREF = +5V. fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR
vs
CODE
3
3
2
2
1
1
DLE (LSB)
ILE (LSB)
INTEGRAL LINEARITY ERROR
vs
CODE
0
0
-1
-1
-2
-2
-3
0000h
4000h
8000h
C000h
-3
0000h
FFFFh
4000h
8000h
Figure 2.
Figure 3.
CHANGE IN OFFSET
vs
TEMPERATURE
CHANGE IN GAIN
vs
TEMPERATURE
0.50
Delta from +25°C (LSB)
Delta from +25°C (LSB)
FFFFh
0.50
0.25
0
-0.25
-0.50
-0.75
-1.00
0.25
0
-0.25
-0.50
-0.75
-50
-25
0
25
50
75
100
-50
Temperature (°C)
-25
0
25
50
75
100
Temperature (°C)
Figure 4.
10
C000h
Output Code
Output Code
Figure 5.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = +25°C, VDD = +5V, VREF = +5V. fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
CHANGE IN OFFSET
vs
COMMON-MODE VOLTAGE
CHANGE IN GAIN
vs
COMMON-MODE VOLTAGE
30
Delta Relative to VCM = 0V (LSB)
Delta Relative to VCM = 0V (LSB)
30
25
20
15
10
5
0
-5
-10
-0.5 -0.4 -0.3 -0.2 -0.1
0
0.1 0.2 0.3
0.4
25
20
15
10
5
0
-5
-10
-0.5 -0.4 -0.3 -0.2 -0.1
0.5 0.6
0
0.1 0.2 0.3
0.4
0.5 0.6
VCM (V)
Figure 6.
Figure 7.
FREQUENCY SPECTRUM
(8192 point FFT, fIN = 1.9836kHz, –0.2dB)
FREQUENCY SPECTRUM
(8192 point FFT, fIN = 9.9792kHz, –0.2dB)
0
0
-20
-20
-40
-40
Amplitude (dB)
Amplitude (dB)
VCM (V)
-60
-80
-100
-60
-80
-100
-120
-120
-140
-140
-160
-160
0
25
50
75
Frequency (kHz)
100
125
0
Figure 8.
25
50
75
Frequency (kHz)
100
125
Figure 9.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = +25°C, VDD = +5V, VREF = +5V. fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SIGNAL-TO-NOISE AND
SIGNAL-TO-NOISE + DISTORTION
vs
INPUT FREQUENCY
SPURIOUS-FREE DYNAMIC RANGE AND
TOTAL HARMONIC DISTORTION
vs
INPUT FREQUENCY
105
SNR
90
SFDR (dB)
SNR and SINAD (dB)
95
85
SINAD
80
-105
100
75
-100
SFDR
95
-95
90
-90
85
-85
THD(1)
80
-80
75
70
THD (dB)
100
-75
70
-70
NOTE: (1) First nine harmonics of the input frequency.
65
65
1
10
100
-65
1
200
100
10
200
Frequency (kHz)
Frequency (kHz)
Figure 10.
Figure 11.
EFFECTIVE NUMBER OF BITS
vs
INPUT FREQUENCY
CHANGE IN SIGNAL-TO-NOISE + DISTORTION
vs
TEMPERATURE
16.0
0.25
fIN = 1.98364kHz, -0.2dB
0.20
Delta from +25°C (dB)
ENOB (Bits)
15.0
14.0
13.0
12.0
11.0
0.15
0.10
0.05
0
-0.05
-0.10
-0.15
10.0
-0.20
1
10
100
200
-50
Frequency (kHz)
0
25
50
75
100
Temperature (°C)
Figure 12.
12
-25
Figure 13.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = +25°C, VDD = +5V, VREF = +5V. fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SIGNAL-TO-NOISE + DISTORTION
vs
INPUT LEVEL
PEAK-TO-PEAK NOISE FOR A DC INPUT
vs
REFERENCE VOLTAGE
100
100
Peak-to-Peak Noise (LSB)
90
200
fIN = 1.98364kHz, -0.2dB
SINAD (dB)
80
70
60
50
40
30
10
20
10
1
-80
-70
-60
-50
-40
-30
-20
-10
0
0.1
Figure 14.
Figure 15.
SUPPLY CURRENT
vs
TEMPERATURE
SUPPLY CURRENT
vs
SAMPLING RATE
10
1.84
Supply Current (mA)
1.83
Supply Current (mA)
5
1
Reference Voltage (V)
Input Level (dB)
1.82
1.81
1.80
1
0.1
0.01
0.001
1.79
-50
-25
0
25
50
75
100
1
10
100
250
Sampling Rate (kHz)
Temperature (°C)
Figure 16.
Figure 17.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = +25°C, VDD = +5V, VREF = +5V. fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
REFERENCE CURRENT
vs
SAMPLING RATE
POWER-DOWN CURRENT
vs
TEMPERATURE
30
Power-Down Current (nA)
Reference Current (mA)
1000
100
10
1
0.1
28
26
24
22
20
18
1
10
100
250
-50
0
-25
Sampling Rate (kHz)
25
50
75
100
Temperature (°C)
Figure 18.
Figure 19.
OUTPUT CODE HISTOGRAM FOR A DC INPUT
(8192 Conversions)
Occurrence
6990
592
0
0
7FFC
7FFD
7FFE
610
7FFF
8000
0
0
8001
8002
Code
Figure 20.
14
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TYPICAL CHARACTERISTICS: VDD = +2.7V
At TA = +25°C, VDD = +2.7V, VREF = +2.5V. fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR
vs
CODE
3
3
2
2
1
1
DLE (LSB)
ILE (LSB)
INTEGRAL LINEARITY ERROR
vs
CODE
0
0
-1
-1
-2
-2
-3
0000h
4000h
8000h
C000h
-3
0000h
FFFFh
4000h
8000h
C000h
FFFFh
Output Code
Figure 21.
Figure 22.
CHANGE IN OFFSET
vs
TEMPERATURE
CHANGE IN GAIN
vs
TEMPERATURE
0.50
0.50
0.25
0.25
Delta from +25°C (LSB)
Delta from +25°C (LSB)
Output Code
0
-0.25
-0.50
-0.75
0
-0.25
-0.50
-0.75
-1.00
-1.00
-50
-25
0
25
50
75
100
-50
Temperature (°C)
-25
0
25
50
75
100
Temperature (°C)
Figure 23.
Figure 24.
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TYPICAL CHARACTERISTICS: VDD = +2.7V (continued)
At TA = +25°C, VDD = +2.7V, VREF = +2.5V. fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
CHANGE IN OFFSET
vs
COMMON-MODE VOLTAGE
CHANGE IN GAIN
vs
COMMON-MODE VOLTAGE
30
Delta Relative to VCM = 0V (LSB)
Delta Relative to VCM = 0V (LSB)
30
25
20
15
10
5
0
-5
-10
-0.5 -0.4 -0.3 -0.2 -0.1
0
25
20
15
10
5
0
-5
-10
-0.5 -0.4 -0.3 -0.2 -0.1
0.1 0.2 0.3 0.4 0.5 0.6
0.1 0.2 0.3 0.4 0.5 0.6
Figure 25.
Figure 26.
FREQUENCY SPECTRUM
(8192 point FFT, fIN = 1.9775kHz, –0.2dB)
FREQUENCY SPECTRUM
(8192 point FFT, fIN = 9.9854kHz, –0.2dB)
0
0
-20
-20
-40
-40
-60
-80
-100
-60
-80
-100
-120
-120
-140
-140
-160
-160
0
10
20
30
40
50
60
70
80
90
100
0
Frequency (kHz)
10
20
30
40
50
60
70
80
90
100
Frequency (kHz)
Figure 27.
16
0
VCM (V)
Amplitude (dB)
Amplitude (dB)
VCM (V)
Figure 28.
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TYPICAL CHARACTERISTICS: VDD = +2.7V (continued)
At TA = +25°C, VDD = +2.7V, VREF = +2.5V. fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SIGNAL-TO-NOISE AND
SIGNAL-TO-NOISE + DISTORTION
vs
INPUT FREQUENCY
SPURIOUS-FREE DYNAMIC RANGE AND
TOTAL HARMONIC DISTORTION
vs
INPUT FREQUENCY
SNR
85
100
-100
95
-95
90
-90
85
80
SFDR (dB)
SNR and SINAD (dB)
90
75
70
SINAD
65
-80
75
-75
70
-70
65
55
-55
50
-50
45
55
40
10
100
-60
55
50
1
200
-45
NOTE: (1) First nine harmonics of the input frequency.
-40
1
100
10
Frequency (kHz)
200
Frequency (kHz)
Figure 29.
Figure 30.
EFFECTIVE NUMBER OF BITS
vs
INPUT FREQUENCY
CHANGE IN SIGNAL-TO-NOISE + DISTORTION
vs
TEMPERATURE
15
0.4
fIN = 1.97754kHz, -0.2dB
14
0.2
Delta from +25°C (dB)
13
ENOB (Bits)
-65
THD(1)
60
60
-85
SFDR
80
THD (dB)
95
12
11
10
9
0
-0.2
-0.4
-0.6
8
7
-0.8
1
10
100
200
-50
Frequency (kHz)
-25
0
25
50
75
100
Temperature (°C)
Figure 31.
Figure 32.
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TYPICAL CHARACTERISTICS: VDD = +2.7V (continued)
At TA = +25°C, VDD = +2.7V, VREF = +2.5V. fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SIGNAL-TO-NOISE + DISTORTION
vs
INPUT LEVEL
100
SUPPLY CURRENT
vs
TEMPERATURE
1.38
fIN = 1.97754kHz, -0.2dB
90
1.37
Supply Current (mA)
SINAD (dB)
80
70
60
50
40
30
1.36
1.35
1.34
1.33
1.32
1.31
20
10
1.30
-80
-70
-60
-50
-40
-30
-20
-10
0
-50
-25
Input Level (dB)
25
50
75
100
Temperature (°C)
Figure 33.
Figure 34.
SUPPLY CURRENT
vs
SAMPLING RATE
REFERENCE CURRENT
vs
SAMPLING RATE
10
1000
Reference Current (mA)
1
Supply Current (mA)
0
0.1
0.01
0.001
0.0001
100
10
1
0.1
1
100
10
200
1
10
Sampling Rate (kHz)
Sampling Rate (kHz)
Figure 35.
Figure 36.
100
200
OUTPUT CODE HISTOGRAM FOR A DC INPUT
(8192 Conversions)
Occurrence
4791
1665
0
53
7FFC
7FFD
7FFE
1643
7FFF
8000
40
0
8001
8002
Code
Figure 37.
18
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THEORY OF OPERATION
The ADS8326 is a classic Successive Approximation
Register (SAR) Analog-to-Digital (A/D) converter. The
architecture is based on capacitive redistribution that
inherently includes a sample-and-hold function. The
converter is fabricated on a 0.6μ CMOS process. The
architecture and process allow the ADS8326 to
acquire and convert an analog signal at up to
250,000 conversions per second while consuming
less than 10mW from VDD.
Differential
linearity
for
the
ADS8326
is
factory-adjusted via a package-level trim procedure.
The state of the trim elements is stored in non-volatile
memory and is continuously updated after each
acquisition cycle, just prior to the start of the
successive approximation operation. This process
ensures that one complete conversion cycle always
returns the part to its factory-adjusted state in the
event of a power interruption.
The ADS8326 requires an external reference, an
external clock, and a single power source (VDD). The
external reference can be any voltage between 0.1V
and VDD. The value of the reference voltage directly
sets the range of the analog input. The reference
input current depends on the conversion rate of the
ADS8326.
The external clock can vary between 24kHz (1kHz
throughput) and 6.0MHz (250kHz throughput). The
duty cycle of the clock is essentially unimportant, as
long as the minimum high and low times are at least
200ns (VDD = 4.75V or greater). The minimum clock
frequency is set by the leakage on the internal
capacitors to the ADS8326.
The analog input is provided to two input pins: +IN
and –IN. When a conversion is initiated, the
differential input on these pins is sampled on the
internal capacitor array. While a conversion is in
progress, both inputs are disconnected from any
internal function.
The digital result of the conversion is clocked out by
the DCLOCK input and is provided serially (most
significant bit first) on the DOUT pin.
The digital data that is provided on the DOUT pin is for
the conversion currently in progress–there is no
pipeline delay. It is possible to continue to clock the
ADS8326 after the conversion is complete and to
obtain the serial data least significant bit first. See the
Timing Information section for more information.
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ANALOG INPUT
0V to +VREF
Peak-to-Peak
The analog input of ADS8326 is differential. The +IN
and –IN input pins allow for a differential input signal.
The amplitude of the input is the difference between
the +IN and –IN input, or (+IN) – (–IN). Unlike some
converters of this type, the –IN input is not resampled
later in the conversion cycle. When the converter
goes into Hold mode or conversion, the voltage
difference between +IN and –IN is captured on the
internal capacitor array.
ADS8326
Common-Mode
Voltage
Figure 38. Methods of Driving the ADS8326
Common Voltage Range (V)
The range of the –IN input is limited to –0.3V to
+0.5V. As a result of this limitation, the differential
input could be used to reject signals that are common
to both inputs in the specified range. Thus, the –IN
input is best used to sense a remote signal ground
that may move slightly with respect to the local
ground potential.
The general method for driving the analog input of the
ADS8326 is shown in Figure 38 and Figure 40. The
–IN input is held at the common-mode voltage. The
+IN input swings from –IN (or common-mode voltage)
to –IN + VREF (or common-mode voltage + VREF ),
and the peak-to-peak amplitude is +VREF . The value
of VREF determines the range over which the
common-mode voltage may vary, as shown in
Figure 39. Figure 6 and Figure 7 (+5V), and
Figure 25 and Figure 26 (+2.7V) illustrate the typical
change in gain and offset as a function of the
common-mode voltage applied to the –IN pin.
1
VDD = 5V
0.5
0
-0.3
-1
2
2.5
3
4
4.8 5
6
VREF (V)
Figure 39. +IN Analog Input: Common-Mode
Voltage Range vs VREF
+IN
Common-Mode Voltage + VREF
+VREF
t
Common-Mode Voltage
-IN = Common-Mode Voltage
NOTE: The maximum differential voltage between +IN and –IN of the ADS8326 is VREF. See Figure 39 for a further
explanation of the common-mode voltage range for differential inputs.
Figure 40. Differential Input Mode of the ADS8326
20
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The input current required by the analog inputs
depends on a number of factors: sample rate, input
voltage, source impedance, and power-down mode.
Essentially, the current into the ADS8326 charges the
internal capacitor array during the sample period.
After this capacitance has been fully charged, there is
no further input current. The source of the analog
input voltage must be able to charge the input
capacitance (48pF) to a 16-bit settling level within 4.5
clock cycles (0.750μs). When the converter goes into
Hold mode, or while it is in Power-Down mode, the
input impedance is greater than 1GΩ.
Care must be taken regarding the absolute analog
input voltage. To maintain the linearity of the
converter, the –IN input should not drop below GND –
0.3V or exceed GND + 0.5V. The +IN input should
always remain within the range of GND – 0.3V to VDD
+ 0.3V, or –IN to –IN + VREF , whichever limit is
reached first. Outside of these ranges, the converter
linearity may not meet specifications. To minimize
noise, low bandwidth input signals with low-pass
filters should be used. In each case, care should be
taken to ensure that the output impedance of the
sources driving the +IN and –IN inputs are matched.
Often, a small capacitor (20pF) between the positive
and negative inputs helps to match their impedance.
To obtain maximum performance from the ADS8326,
the input circuit from Figure 41 is recommended.
Single-Ended
10W
+IN
OPA365
50W
48pF
1000pF
ADS8326
-IN
50W
48pF
Differential
10W
+IN
OPA365
50W
48pF
1000pF
ADS8326
1nF
10W
-IN
OPA365
50W
48pF
1000pF
Figure 41. Single-Ended and Differential Methods
of Interfacing the ADS8326
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REFERENCE INPUT
ADS8326
The external reference sets the analog input range.
The ADS8326 operates with a reference in the range
of 0.1V to VDD. There are several important
implications to this.
As the reference voltage is reduced, the analog
voltage weight of each digital output code is reduced.
This is often referred to as the least significant bit
(LSB) size and is equal to the reference voltage
divided by 65,536. This means that any offset or gain
error inherent in the A/D converter will appear to
increase (in terms of LSB size) as the reference
voltage is reduced. For a reference voltage of 2.5V,
the value of the LSB is 38.15μV, and for a reference
voltage of 5V, the LSB is 76.3μV.
The noise inherent in the converter will also appear to
increase with a lower LSB size. With a 5V reference,
the internal noise of the converter typically contributes
only 1.5LSB peak-to-peak of potential error to the
output code. When the external reference is 2.5V, the
potential error contribution from the internal noise will
be two times larger (3LSB). The errors arising from
the internal noise are Gaussian in nature and can be
reduced by averaging consecutive conversion results.
For more information regarding noise, see Figure 15,
Peak-to-Peak Noise for a DC Input vs Reference
Voltage. Note that the Effective Number Of Bits
(ENOB) figure is calculated based on the converter
signal-to-(noise + distortion) ratio with a 1kHz, 0dB
input signal. SINAD is related to ENOB as follows:
SINAD = 6.02 × ENOB + 1.76
With lower reference voltages, extra care should be
taken to provide a clean layout including adequate
bypassing, a clean power supply, a low-noise
reference, and a low-noise input signal. Due to the
lower LSB size, the converter is also more sensitive
to external sources of error, such as nearby digital
signals and electromagnetic interference.
The equivalent input circuit for the reference voltage
is presented in Figure 42. During the conversion
process, an equivalent capacitor of 24pF is switched
on. To obtain optimum performance from the
ADS8326, special care must be taken in designing
the interface circuit to the reference input pin. To
ensure a stable reference voltage, a 47μF tantalum
capacitor with low ESR should be connected as close
as possible to the input pin. If a high output
impedance reference source is used, an additional
operational amplifier with a current-limiting resistor
must be placed in front of the capacitors.
VREF
50W
24pF
OPA350
47mF
Figure 42. Input Reference Circuit and Interface
When the ADS8326 is in Power-Down mode, the
input resistance of the reference pin will have a value
of 5GΩ. Since the input capacitors must be
recharged before the next conversion starts, an
operational
amplifier
with
good
dynamic
characteristics must be used to buffer the reference
input.
Noise
The transition noise of the ADS8326 itself is
extremely low, as shown in Figure 20 (+5V) and
Figure 37 (+2.7V); it is much lower than competing
A/D converters. These histograms were generated by
applying a low-noise DC input and initiating 8192
conversions. The digital output of the A/D converter
will vary in output code because of the internal noise
of the ADS8326. This is true for all 16-bit, SAR-type
A/D converters. Using a histogram to plot the output
codes, the distribution should appear bell-shaped with
the peak of the bell curve representing the nominal
code for the input value. The ±1σ, ±2σ, and ±3σ
distributions will represent 68.3%, 95.5%, and 99.7%,
respectively, of all codes. The transition noise can be
calculated by dividing the number of codes measured
by 6, which yields the ±3σ distribution, or 99.7%, of
all codes. Statistically, up to three codes could fall
outside the distribution when executing 1000
conversions. The ADS8326, with < 3 output codes for
the ±3σ distribution, yields < ±0.5LSB of transition
noise. Remember, to achieve this low-noise
performance, the peak-to-peak noise of the input
signal and reference must be < 50μV.
Averaging
The noise of the A/D converter can be compensated
by averaging the digital codes. By averaging
conversion results, transition noise is reduced by a
factor of 1/√n , where n is the number of averages.
For example, averaging four conversion results
reduces the transition noise from ±0.5LSB to
±0.25LSB. Averaging should only be used for input
signals with frequencies near DC.
For AC signals, a digital filter can be used to
low-pass filter and decimate the output codes. This
works in a similar manner to averaging; for every
decimation by 2, the signal-to-noise ratio improves by
3dB.
22
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ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
DIGITAL INTERFACE
Signal Levels
The ADS8326 has a wide range of power-supply
voltage. The A/D converter, as well as the digital
interface circuit, is designed to accept and operate
from 2.7V up to 5.5V. This voltage range will
accommodate different logic levels. When the
ADS8326 power-supply voltage is in the range of
4.5V to 5.5V (5V logic level), the ADS8326 can be
connected directly to another 5V, CMOS-integrated
circuit. When the ADS8326 power-supply voltage is in
the range of 2.7V to 3.6V (3V logic level), the
ADS8326 can be connected directly to another 3.3V
LVCMOS integrated circuit.
A falling CS signal initiates the conversion and data
transfer. The first 4.5 to 5.0 clock periods of the
conversion cycle are used to sample the input signal.
After the fifth falling DCLOCK edge, DOUT is enabled
and will output a low value for one clock period. For
the next 16 DCLOCK periods, DOUT will output the
conversion result, most significant bit first. After the
least significant bit (B0) has been output, subsequent
clocks will repeat the output data, but in a least
significant bit first format.
After the most significant bit (B15) has been
repeated, DOUT will tri-state. Subsequent clocks will
have no effect on the converter. A new conversion is
initiated only when CS has been taken high and
returned low.
Serial Interface
Data Format
The ADS8326 communicates with microprocessors
and other digital systems via a synchronous 3-wire
serial interface, as illustrated in the Timing
Information
section.
The
DCLOCK
signal
synchronizes the data transfer, with each bit being
transmitted on the falling edge of DCLOCK. Most
receiving systems will capture the bitstream on the
rising edge of DCLOCK. However, if the minimum
hold time for DOUT is acceptable, the system can use
the falling edge of DCLOCK to capture each bit.
The output data from the ADS8326 is in Straight
Binary format, as shown in Figure 43. This figure
represents the ideal output code for a given input
voltage and does not include the effects of offset,
gain error, or noise.
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ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
65535
1111 1111 1111 1111
65534
1111 1111 1111 1111
65533
1000 0000 0000 0001
32769
1000 0000 0000 0000
32768
0111 1111 1111 1111
32767
0000 0000 0000 0010
2
0000 0000 0000 0001
1
0000 0000 0000 0000
Step
Digital Output Code
Straight Binary
1111 1111 1111 1111
0
2.499962V
VZ = VCM = 0V
2.500038V
VFS = VCM + VREF = 5V
38.15mV
VFS - 1LSB = 4.999924V
VMS = VCM + VREF/2 = 2.5V
76.29mV
4.999847V
Unipolar Analog Input Voltage
1LSB = 76.29mV
152.58mV
VCM = 0V
VREF = 5V
16-BIT
Zero Code
Midscale Code
Full- Scale Code
Straight Binary Output
VZ = 0000h
VMS = 8000h
VFS = FFFFh
Unipolar Analog Input
VCODE = VCM
VCODE = VCM + VREF/2
VCODE = (VCM + VREF) - 1LSB
Figure 43. Ideal Conversion Characteristics (Conditions: VCM = 0V, VREF = 5V)
24
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ADS8326
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POWER DISSIPATION
Short Cycling
The architecture of the converter, the semiconductor
fabrication process, and a careful design allow the
ADS8326 to convert at up to a 250kHz rate while
requiring very little power. However, for the absolute
lowest power dissipation, there are several things to
keep in mind.
Another way to save power is to use the CS signal to
short-cycle the conversion. The ADS8326 places the
latest data bit on the DOUT line as it is generated;
therefore, the converter can easily be short-cycled.
This term means that the conversion can be
terminated at any time. For example, if only 14 bits of
the conversion result are needed, then the conversion
can be terminated (by pulling CS high) after the 14th
bit has been clocked out.
The power dissipation of the ADS8326 scales directly
with conversion rate. Therefore, the first step to
achieving the lowest power dissipation is to find the
lowest conversion rate that will satisfy the
requirements of the system.
In addition, the ADS8326 goes into Power-Down
mode under two conditions: when the conversion is
complete and whenever CS is high (see the Timing
Information section). Ideally, each conversion should
occur as quickly as possible, preferably at a 6.0MHz
clock rate. This way, the converter spends the
longest possible time in Power-Down mode. This is
very important because the converter not only uses
power on each DCLOCK transition (as is typical for
digital CMOS components), but also uses some
current for the analog circuitry, such as the
comparator. The analog section dissipates power
continuously until Power-Down mode is entered.
Figure 17 and Figure 18 (+5V), and Figure 35 and
Figure 36 illustrate the current consumption of the
ADS8326 versus sample rate. For these graphs, the
converter is clocked at maximum speed regardless of
the sample rate. CS is held high during the remaining
sample period.
This technique can also be used to lower the power
dissipation (or to increase the conversion rate) in
those applications where an analog signal is being
monitored until some condition becomes true. For
example, if the signal is outside a predetermined
range, the full 16-bit conversion result may not be
needed. If so, the conversion can be terminated after
the first n bits, where n might be as low as 3 or 4.
This results in lower power dissipation in both the
converter and the rest of the system because they
spend more time in Power-Down mode.
POWER-ON RESET
The ADS8326 bias circuit is self-starting. There may
be a static current (approximately 1.5mA with VDD =
5V) after power-on, unless the circuit is powered
down. It is recommended to run a single test
conversion (configured the same as any regular
conversion) after the power supply reaches at least
2.4V to ensure the device is put into power-down
mode.
There is an important distinction between the
power-down mode that is entered after a conversion
is complete and the full power-down mode that is
enabled when CS is high. CS low will only shut down
the analog section. The digital section is completely
shut down only when CS is high. Thus, if CS is left
low at the end of a conversion, and the converter is
continually clocked, the power consumption will not
be as low as when CS is high.
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ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
LAYOUT
For optimum performance, care should be taken with
the physical layout of the ADS8326 circuitry. This is
particularly true if the reference voltage is low and/or
the conversion rate is high. At a 250kHz conversion
rate, the ADS8326 makes a bit decision every 167ns.
That is, for each subsequent bit decision, the digital
output must be updated with the results of the last bit
decision, the capacitor array appropriately switched
and charged, and the input to the comparator settled
to a 16-bit level, all within one clock cycle.
The basic SAR architecture is sensitive to spikes on
the power supply, reference, and ground connections
that occur just prior to latching the comparator output.
Thus, during any single conversion for an n-bit SAR
converter, there are n windows in which large
external transient voltages can easily affect the
conversion result. Such spikes might originate from
switching power supplies, digital logic, and
high-power devices, to name a few potential sources.
This particular source of error can be very difficult to
track down if the glitch is almost synchronous to the
converter DCLOCK signal because the phase
difference between the two changes with time and
temperature, causing sporadic misoperation.
With this in mind, power to the ADS8326 should be
clean and well-bypassed. A 0.1μF ceramic bypass
capacitor should be placed as close as possible to
the ADS8326 package. In addition, a 1μF to 10μF
capacitor and a 5Ω or 10Ω series resistor may be
used to low-pass filter a noisy supply.
resistor can help in this case). Keep in mind that
while the ADS8326 draws very little current from the
reference on average, there are still instantaneous
current demands placed on the external input and
reference circuitry.
Texas Instruments' OPA365 op amp provides
optimum performance for buffering the signal inputs;
the OPA350 can be used to effectively buffer the
reference input.
Also, keep in mind that the ADS8326 offers no
inherent rejection of noise or voltage variation in
regards to the reference input. This is of particular
concern when the reference input is tied to the power
supply. Any noise and ripple from the supply will
appear directly in the digital results. While
high-frequency noise can be filtered out, as described
in the previous paragraph, voltage variation resulting
from the line frequency (50Hz or 60Hz) can be
difficult to remove.
The GND pin on the ADS8326 should be placed on a
clean ground point. In many cases, this will be the
analog ground. Avoid connecting the GND pin too
close to the grounding point for a microprocessor,
microcontroller, or digital signal processor. If needed,
run a ground trace directly from the converter to the
power-supply connection point. The ideal layout will
include an analog ground plane for the converter and
associated analog circuitry.
The reference should be similarly bypassed with a
47μF capacitor. Again, a series resistor and large
capacitor can be used to low-pass filter the reference
voltage. If the reference voltage originates from an op
amp, make sure that the op amp can drive the
bypass capacitor without oscillation (the series
26
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ADS8326
www.ti.com.................................................................................................................................................. SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009
APPLICATION CIRCUITS
high-frequency noise from the supply itself. The exact
values should be picked such that the filter provides
adequate rejection of noise. Operational amplifiers
and voltage reference are connected to analog power
supply, AVDD.
Figure 44 and Figure 45 show two examples of a
basic data acquisition system. The ADS8326 input
range is connected to 2.5V or 4.096V. The 5Ω
resistor and 1μF to 10μF capacitor filters the
microcontroller noise on the supply, as well as any
DVDD
2.7V to 3.6V
0.1mF
AVDD
2.7V to 5V
+
10mF
5W
REF3225
REF
OPA350
10W
IN
OUT
2.2mF
0.1mF
GND
0.47mF
VDD
47mF
+
10mF
ADS8326
DSP
10W
TMS320C6xx
or
TMS320C5xx
or
TMS320C2xx
+IN
OPA365
VCM + (0V to 2.5V)
1000pF
CS
1nF
DOUT
DCLOCK
10W
GND
-IN
OPA365
VCM
GND
1000pF
Figure 44. Basic Data Acquisition System: Example 1
DVDD
4.5V to 5.5V
0.1mF
10mF
5W
AVDD
4.3V to 5.5V
REF3240
REF
OPA350
10W
IN
OUT
VDD
0.1mF
47mF
2.2mF
0.47mF
+
GND
+
10mF
ADS8326
10W
Microcontroller
or
DSP
+IN
OPA365
0V to 4.096V
1000pF
CS
DOUT
DCLOCK
-IN
GND
GND
Figure 45. Basic Data Acquisition System: Example 2
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27
ADS8326
SBAS343C – MAY 2007 – REVISED SEPTEMBER 2009.................................................................................................................................................. www.ti.com
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (May, 2008) to Revision C ..................................................................................................... Page
•
Released SON-8 package; changed statements regarding SON-8 package availability ..................................................... 1
•
Deleted footnote about SON-8 package availability ............................................................................................................. 2
•
Deleted footnote about SON-8 package availability ............................................................................................................. 3
•
Deleted footnote about SON-8 package availability ............................................................................................................. 7
Changes from Revision A (August, 2007) to Revision B ............................................................................................... Page
•
Changed SON-8 package availability to Q3, 2008 ............................................................................................................... 1
•
Changed y-axis unit in Figure 35 from μA to mA ............................................................................................................... 18
•
Added Power-On Reset section ......................................................................................................................................... 25
28
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PACKAGE OPTION ADDENDUM
www.ti.com
7-Oct-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADS8326IBDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IBDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IBDGKTG4
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IBDRBR
ACTIVE
SON
DRB
8
3000
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IBDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IDRBR
ACTIVE
SON
DRB
8
3000
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
ADS8326IDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D26
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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