ADS8883IDRCR

ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 18-Bit, 680-kSPS, Serial Interface, microPower, Miniature, True-Differential Input, SAR Analog-to-Digital Converter Check for Samples: ADS8883 FEATURES APPLICATIONS • Sample Rate: 680 kHz • No Latency Output • Unipolar, True-Differential Input Range: –VREF to +VREF • Wide Common-Mode Voltage Range: 0 V to VREF with 90-dB CMRR (min) • SPI™-Compatible Serial Interface with Daisy-Chain Option • Excellent AC and DC Performance: – SNR: 100 dB, THD: –115 dB – INL: ±1.5 LSB (typ), ±3.0 LSB (max) – DNL: +1.5 and –1 LSB (max), 18-Bit NMC • Wide Operating Range: – AVDD: 2.7 V to 3.6 V – DVDD: 1.65 V to 3.6 V (Independent of AVDD) – REF: 2.5 V to 5 V (Independent of AVDD) – Operating Temperature: –40°C to +85°C • Low-Power Dissipation: – 4.2 mW at 680 kSPS – 0.6 mW at 100 kSPS – 60 µW at 10 kSPS • Power-Down Current (AVDD): 50 nA • Full-Scale Step Settling to 18 Bits: 540 ns • Packages: MSOP-10 and SON-10 • • • • 1 234 Traditional Input Range ADS8883 Input Range VDIFF Automatic Test Equipment (ATE) Instrumentation and Process Controls Precision Medical Equipment Low-Power, Battery-Operated Instruments DESCRIPTION The ADS8883 is an 18-bit, 680-kSPS, true-differential input, analog-to-digital converter (ADC). The device operates with a 2.5-V to 5-V external reference, offering a wide selection of signal ranges without additional input signal scaling. The reference voltage setting is independent of, and can exceed, the analog supply voltage (AVDD). The device that also cascading indicator bit easy. offers an SPI-compatible serial interface supports daisy-chain operation for multiple devices. An optional busymakes synchronizing with the digital host The device supports unipolar, true-differential analog input signals with a differential input swing of –VREF to +VREF. This true-differential analog input structure allows for a common-mode voltage of any value in the range of 0 V to +VREF (when both inputs are within the operating input range of –0.1 V to VREF + 0.1 V). Device operation is optimized for very low-power operation. Power consumption directly scales with speed. This feature makes the ADS8883 excellent for lower-speed applications. True-Differential Input Range +VREF 2.5 V to 5 V 2.7 V to 3.6 V REF AVDD 1.65 V to 3.6 V VREF VCM 0V 0 V - VREF DVDD AINP VCM DIN ADS8883 VREF SCLK DOUT 0V AINM CONVST 0 V - VREF VREF/2 GND -VREF 1 2 3 4 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. TINA is a trademark of Texas Instruments Inc.. SPI is a trademark of Motorola Inc. All other trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2013, Texas Instruments Incorporated ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. FAMILY INFORMATION (1) (1) (2) THROUGHPUT 18-BIT, TRUE-DIFFERENTIAL 16-BIT, SINGLE-ENDED 16-BIT, TRUE-DIFFERENTIAL 100 kSPS ADS8887 ADS8866 ADS8867 250 kSPS — — — 400 kSPS ADS8885 ADS8864 ADS8865 500 kSPS — ADS8319 (2) ADS8318 (2) 680 kSPS ADS8883 ADS8862 ADS8863 1 MSPS ADS8881 ADS8860 ADS8861 For the most current package and ordering information, see the Package Option Addendum at the end of this document, or visit the device product folder at www.ti.com. Pin-to-pin compatible device with AVDD = 5 V. ABSOLUTE MAXIMUM RATINGS (1) Over operating free-air temperature range, unless otherwise noted. VALUE MIN MAX AINP to GND or AINN to GND –0.3 REF + 0.3 V AVDD to GND or DVDD to GND –0.3 4 V REF to GND –0.3 5.7 V Digital input voltage to GND –0.3 DVDD + 0.3 V Digital output to GND –0.3 DVDD + 0.3 V Operating temperature range, TA –40 +85 °C Storage temperature range, Tstg –65 +150 °C (1) UNIT Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under electrical characteristics is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. THERMAL INFORMATION ADS8883 THERMAL METRIC (1) DGS DRC 10 PINS 10 PINS θJA Junction-to-ambient thermal resistance 151.9 111.1 θJCtop Junction-to-case (top) thermal resistance 45.4 46.4 θJB Junction-to-board thermal resistance 72.2 45.9 ψJT Junction-to-top characterization parameter 3.3 3.5 ψJB Junction-to-board characterization parameter 70.9 45.5 θJCbot Junction-to-case (bottom) thermal resistance N/A N/A (1) 2 UNITS °C/W For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 ELECTRICAL CHARACTERISTICS All minimum and maximum specifications are at TA = –40°C to +85°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, VCM = VREF / 2 V, and fSAMPLE = 680 kSPS, unless otherwise noted. Typical specifications are at TA = +25°C, AVDD = 3 V, and DVDD = 3 V. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT ANALOG INPUT Full-scale input span (1) (2) Operating input range (1) (2) AINP – AINN –VREF VREF V AINP –0.1 VREF + 0.1 V AINN –0.1 VREF + 0.1 V VCM Input common-mode range 0 CI Input capacitance AINP and AINN terminal to GND Input leakage current During acquisition for dc input VREF / 2 VREF V 59 pF 5 nA 18 Bits SYSTEM PERFORMANCE Resolution NMC No missing codes DNL Differential linearity INL Integral linearity (4) EO Offset error 18 (5) ±0.7 1.5 LSB (3) –3 ±1.5 3 LSB (3) ±1 4 –4 Offset error drift with temperature EG Bits –0.99 mV ±1.5 Gain error –0.01 Gain error drift with temperature ±0.005 µV/°C 0.01 ±0.15 CMRR Common-mode rejection ratio With common-mode input signal = 5 VPP at dc PSRR Power-supply rejection ratio At mid-code 90 Transition noise %FSR ppm/°C 100 dB 80 dB 0.7 LSB SAMPLING DYNAMICS tconv Conversion time 500 tACQ Acquisition time 540 930 ns Maximum throughput rate with or without latency (1) (2) (3) (4) (5) ns 680 kHz Aperture delay 4 ns Aperture jitter, RMS 5 ps Step response Settling to 18-bit accuracy 540 ns Overvoltage recovery Settling to 18-bit accuracy 540 ns Ideal input span, does not include gain or offset error. Specified for VCM = VREF / 2. Refer to the Analog Input section for the effect of VCM on the full-scale input range. LSB = least significant bit. This parameter is the endpoint INL, not best-fit. Measured relative to actual measured reference. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 3 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com ELECTRICAL CHARACTERISTICS (continued) All minimum and maximum specifications are at TA = –40°C to +85°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, VCM = VREF / 2 V, and fSAMPLE = 680 kSPS, unless otherwise noted. Typical specifications are at TA = +25°C, AVDD = 3 V, and DVDD = 3 V. PARAMETER TEST CONDITIONS MIN TYP 98 MAX UNIT DYNAMIC CHARACTERISTICS At 1 kHz, VREF = 5 V SINAD Signal-to-noise + distortion (6) 99.9 dB At 10 kHz, VREF = 5 V 98.7 dB At 100 kHz, VREF = 5 V 93.3 dB 100 dB At 10 kHz, VREF = 5 V 99.5 dB At 100 kHz, VREF = 5 V 93.5 dB At 1 kHz, VREF = 5 V –115 dB At 10 kHz, VREF = 5 V –112 dB At , VREF = 5 V –102 dB At 1 kHz, VREF = 5 V 115 dB At 10 kHz, VREF = 5 V 112 dB At 100 kHz, VREF = 5 V 102 dB 30 MHz At 1 kHz, VREF = 5 V Signal-to-noise ratio (6) SNR Total harmonic distortion (6) (7) THD SFDR BW–3dB Spurious-free dynamic range (6) 98.5 –3-dB small-signal bandwidth EXTERNAL REFERENCE INPUT VREF Input range 2.5 Reference input current During conversion, 680-kHz sample rate, mid-code Reference leakage current CREF Decoupling capacitor at the REF input 10 5 V 160 μA 250 nA 22 µF POWER-SUPPLY REQUIREMENTS PVA IAPD Power-supply voltage AVDD Analog supply DVDD Digital supply range Supply current AVDD Power dissipation 2.7 3 3.6 1.65 1.8 3.6 V 680-kHz sample rate, AVDD = 3 V 1.4 1.8 mA 680-kHz sample rate, AVDD = 3 V 4.2 5.4 mW 100-kHz sample rate, AVDD = 3 V 0.6 mW 10-kHz sample rate, AVDD = 3 V 60 μW 50 nA Device power-down current (8) V DIGITAL INPUTS: LOGIC FAMILY (CMOS) VIH High-level input voltage VIL Low-level input voltage ILK Digital input leakage current 1.65 V < DVDD < 2.3 V 0.8 × DVDD DVDD + 0.3 V 2.3 V < DVDD < 3.6 V 0.7 × DVDD DVDD + 0.3 V 1.65 V < DVDD < 2.3 V –0.3 0.2 × DVDD V 2.3 V < DVDD < 3.6 V –0.3 0.3 × DVDD V ±100 nA 0.8 × DVDD DVDD V 0 0.2 × DVDD V –40 +85 °C ±10 DIGITAL OUTPUTS: LOGIC FAMILY (CMOS) VOH High-level output voltage IO = 500-μA source, CLOAD = 20 pF VOL Low-level output voltage IO = 500-μA sink, CLOAD = 20 pF TEMPERATURE RANGE TA (6) (7) (8) 4 Operating free-air temperature All specifications expressed in decibels (dB) refer to the full-scale input (FSR) and are tested with an input signal 0.5 dB below full-scale, unless otherwise specified. Calculated on the first nine harmonics of the input frequency. The device automatically enters a power-down state at the end of every conversion, and remains in power-down during the acquisition phase. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 TIMING CHARACTERISTICS 3-WIRE OPERATION 1/fsample DIN = HIGH tconv-max tACQ tclkh th-CK-DO CONVST tclkl tquiet œœ SCLK 1 2 td-CNV-DO D17 DOUT 3 16 17 18 D1 D0 tSCLK œœ D16 D15 D2 œœ twh-CNV-min td-CK-DO td-CK-DOhz Figure 1. 3-Wire Operation: CONVST Functions as Chip Select NOTE Figure 1 shows the timing diagram for the 3-Wire CS Mode Without a Busy Indicator interface option. However, the timing parameters specified in Table 1 are also applicable for the 3-Wire CS Mode With a Busy Indicator interface option, unless otherwise specified. Refer to the Digital Interface section for specific details for each interface option. Table 1. TIMING REQUIREMENTS: 3-Wire Operation (1) PARAMETER MIN TYP MAX Acquisition time tconv Conversion time 1/fsample Time between conversions twh-CNV Pulse duration: CONVST high fSCLK SCLK frequency tSCLK SCLK period 27.8 tclkl SCLK low time tclkh SCLK high time th-CK-DO SCLK falling edge to current data invalid td-CK-DO SCLK falling edge to next data valid delay 13.4 ns td-CNV-DO Enable time: CONVST low to MSB valid 12.3 ns td-CNV-DOhz Disable time: CONVST high or last SCLK falling edge to DOUT 3-state (CS mode) 13.2 ns tquiet Quiet time (1) 540 UNIT tACQ 500 ns 930 ns 1470 ns 10 ns 36 MHz 0.45 0.55 tSCLK 0.45 0.55 tSCLK ns 3 ns 20 ns All specifications are at TA = –40°C to +85°C, AVDD = 3 V, and DVDD = 3 V, unless otherwise noted. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 5 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com 4-WIRE OPERATION 1/fsample tACQ tconv-max CONVST tsu-DI-CNV twl-CNV DIN œœ SCLK 1 DOUT D17 2 3 D16 D15 16 17 18 D1 D0 œœ twh-DI-min td-DI-DO D2 œœ td-DI-DOhz Figure 2. 4-Wire Operation: DIN Functions as Chip Select NOTE Figure 2 shows the timing diagram for the 4-Wire CS Mode Without a Busy Indicator interface option. However, the timing parameters specified in Table 2 are also applicable for the 4-Wire CS Mode With a Busy Indicator interface option, unless otherwise specified. Refer to the Digital Interface section for specific details for each interface option. Table 2. TIMING REQUIREMENTS: 4-Wire Operation (1) PARAMETER MIN TYP MAX UNIT tACQ Acquisition time 540 tconv Conversion time 500 1/fsample Time between conversions 1470 ns twh-DI Pulse duration: DIN high 10 ns twl-CNV Pulse width: CONVST low 20 td-DI-DO Delay time: DIN low to MSB valid td-DI-DOhz Delay time: DIN high or last SCLK falling edge to DOUT 3-state tsu-DI-CNV Setup time: DIN high to CONVST rising edge th-DI-CNV Hold time: DIN high from CONVST rising edge (see Figure 63) (1) 6 ns 930 ns ns 12.3 ns 13.2 ns 7.5 ns 0 ns All specifications are at TA = –40°C to +85°C, AVDD = 3 V, and DVDD = 3 V, unless otherwise noted. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 DAISY-CHAIN OPERATION 1/fsample tconv- tACQ max CONVST th-CK-CNV SCLK DIN 1 = LOW 1 2 17 18 19 20 35 36 D17 D16 D1 D0 tsu-DI-CK tsu-CK-CNV DOUT 1, DIN 2 D17 D16 D1 D0 DOUT 2 D17 D16 D1 D0 Device 2 Data Device 1 Data Figure 3. Daisy-Chain Operation: Two Devices NOTE Figure 3 shows the timing diagram for the Daisy-Chain Mode Without a Busy Indicator interface option. However, the timing parameters specified in Table 3 are also applicable for the Daisy-Chain Mode With a Busy Indicator interface option, unless otherwise specified. Refer to the Digital Interface section for specific details for each interface option. Table 3. TIMING REQUIREMENTS: Daisy-Chain (1) PARAMETER MIN TYP MAX UNIT tACQ Acquisition time 540 tconv Conversion time 500 1/fsample Time between conversions 1470 ns tsu-CK-CNV Setup time: SCLK valid to CONVST rising edge 5 ns th-CK-CNV Hold time: SCLK valid from CONVST rising edge 5 ns tsu-DI-CNV Setup time: DIN low to CONVST rising edge (see Figure 2) 7.5 ns th-DI-CNV Hold time: DIN low from CONVST rising edge (see Figure 63) 0 ns tsu-DI-CK Setup time: DIN valid to SCLK falling edge 1.5 ns (1) ns 930 ns All specifications are at TA = –40°C to +85°C, AVDD = 3 V, and DVDD = 3 V, unless otherwise noted. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 7 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com EQUIVALENT CIRCUITS 500 µA IOL 1.4 V DOUT 20 pF 500 µA IOH Figure 4. Load Circuit for Digital Interface Timing DIN CONVST SCLK VIH VIL VOH VOH VOL VOL SDO Figure 5. Voltage Levels for Timing 8 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 PIN CONFIGURATIONS DGS PACKAGE VSSOP-10 (TOP VIEW, Not to Scale) DRC PACKAGE SON-10 (TOP VIEW, Not to Scale) REF 1 10 DVDD AVDD 2 9 DIN AINP 3 8 SCLK AINN 4 7 DOUT GND 5 6 CONVST REF 1 10 DVDD AVDD 2 9 DIN AINP 3 8 SCLK AINN 4 7 DOUT GND 5 6 CONVST Thermal PAD PIN ASSIGNMENTS PIN NAME PIN NUMBER FUNCTION AINN 4 Analog input Inverting analog signal input AINP 3 Analog input Noninverting analog signal input AVDD 2 Analog CONVST 6 Digital input Convert input. This pin also functions as the CS input in 3-wire interface mode. Refer to the Description and Timing Characteristics sections for more details. Serial data input. The DIN level at the start of a conversion selects the mode of operation (such as CS or daisy-chain mode). This pin also serves as the CS input in 4-wire interface mode. Refer to the Description and Timing Characteristics sections for more details. DESCRIPTION Analog power supply. This pin must be decoupled to GND with a 1-μF capacitor. DIN 9 Digital input DOUT 7 Digital output Serial data output DVDD 10 Power supply Digital interface power supply. This pin must be decoupled to GND with a 1-μF capacitor. Device ground. Note that this pin is a common ground pin for both the analog power supply (AVDD) and digital I/O supply (DVDD). The reference return line is also internally connected to this pin. GND 5 Analog, digital REF 1 Analog SCLK 8 Digital input Clock input for serial interface. Data output (on DOUT) are synchronized with this clock. Thermal pad — Thermal pad Exposed thermal pad. Texas Instruments recommends connecting the thermal pad to the printed circuit board (PCB) ground. Positive reference input. This pin must be decoupled with a 10-μF or larger capacitor. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 9 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com TYPICAL CHARACTERISTICS At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 1 Typical Differential Nonlinearity (LSB) Typical Integral Nonlinearity (LSB) 3 AVDD = 3 V REF = 2.5 V TA = 25ƒC 2 1 0 ±1 ±2 ±3 ±131072 0.5 0.25 0 -0.25 -0.5 -0.75 -1 ±131072 131071 ADC Output Code 1 AVDD = 3 V REF = 5 V TA = 25ƒC 2 Typical Differential Nonlinearity (LSB) Typical Integral Nonlinearity (LSB) C002 Figure 7. TYPICAL DNL (VREF = 2.5 V) 3 1 0 ±1 ±2 ±3 ±131072 131071 ADC Output Code AVDD = 3 V REF = 5 V TA = 25ƒC 0.75 0.5 0.25 0 -0.25 -0.5 -0.75 -1 ±131072 131071 ADC Output Code C003 Figure 8. TYPICAL INL (VREF = 5 V) C004 Figure 9. TYPICAL DNL (VREF = 5 V) 3 2 AVDD = 3 V REF = 5 V 2 Differential Nonlinearity (LSB) Integral Nonlinearity (LSB) 131071 ADC Output Code C001 Figure 6. TYPICAL INL (VREF = 2.5 V) 1 0 -1 -2 -3 AVDD = 3 V REF = 5 V 1.5 1 0.5 0 -0.5 -1 -40 -15 10 35 Free-Air Temperature (oC) 60 85 -40 C00 Figure 10. INL vs TEMPERATURE 10 AVDD = 3 V REF = 2.5 V TA = 25ƒC 0.75 Submit Documentation Feedback -15 10 35 60 Free-Air Temperature (oC) 85 C00 Figure 11. DNL vs TEMPERATURE Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 2 AVDD = 3 V TA = 25oC 2 Differential Nonlinearity (LSB) Integral Nonlinearity (LSB) 3 1 0 -1 -2 AVDD = 3 V TA = 25oC 1.5 1 0.5 0 -0.5 -1 -3 2.5 3 3.5 4 Reference Voltage (V) 4.5 2.5 5 3 Figure 12. INL vs REFERENCE VOLTAGE 4.5 5 C00 Figure 13. DNL vs REFERENCE VOLTAGE 40 60 AVDD = 3 V REF = 2.5 V TA = 25oC AVDD = 3 V REF = 5 V TA = 25oC 50 Hits per Code (%) 30 Hits per Code (%) 3.5 4 Reference Voltage (V) C00 20 40 30 20 10 10 0 0 10 12 14 16 ADC Output Code 18 20 4 Figure 14. DC INPUT HISTOGRAM (VREF = 2.5 V) 8 ADC Output Code 10 C01 Figure 15. DC INPUT HISTOGRAM (VREF = 5 V) 0 0 AVDD = 3 V REF = 2.5 V TA = 25ƒC fIN = 1 kHz SNR = 95.4 dB THD = ±119 dB ±40 ±60 ±80 AVDD = 3 V REF = 5 V TA = 25ƒC fIN = 1 kHz SNR = 100 dB THD = ±119 dB ±20 ±40 ±60 Power (dB) ±20 Power (dB) 6 C00 ±100 ±120 ±80 ±100 ±120 ±140 ±140 ±160 ±160 ±180 ±180 ±200 ±200 0 68 136 204 Input Frequency (kHz) 272 340 0 C001 Figure 16. TYPICAL FFT (VREF = 2.5 V) 68 136 204 Input Frequency (kHz) 272 Product Folder Links: ADS8883 C002 Figure 17. TYPICAL FFT (VREF = 5 V) Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated 340 11 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com TYPICAL CHARACTERISTICS (continued) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 102 Signal-to-Noise and Distortion (dBFS) Signal-to-Noise Ratio (dBFS) 102 101 100 99 98 97 96 95 fIN = 1 kHz 94 101 100 99 98 97 96 95 fIN = 1 kHz 94 2.5 3 3.5 4 Reference Voltage (V) 4.5 5 2.5 Figure 18. SNR vs REFERENCE VOLTAGE Total Harmonic Distortion (dBFS) Effective Number of Bits 4.5 5 C01 -109 fIN = 1 kHz 17.5 17 16.5 16 15.5 15 14.5 fIN = 1 kHz -111 -113 -115 -117 -119 -121 -123 -125 14 2.5 3 3.5 4 Reference Voltage (V) 4.5 2.5 5 3 C01 Figure 20. ENOB vs REFERENCE VOLTAGE 3.5 4 Reference Voltage (V) 4.5 5 C01 Figure 21. THD vs REFERENCE VOLTAGE 102 126 fIN = 1 kHz 124 Signal-to-Noise Ratio (dBFS) Spurious-Free Dynamic Range (dBFS) 3.5 4 Reference Voltage (V) Figure 19. SINAD vs REFERENCE VOLTAGE 18 122 120 118 116 114 112 fIN = 1 kHz 101 100 99 98 97 96 95 94 110 2.5 3 3.5 4 Reference Voltage (V) 4.5 5 -40 C01 Figure 22. SFDR vs REFERENCE VOLTAGE 12 3 C01 Submit Documentation Feedback -15 10 35 Free-Air Temperature (oC) 60 85 C01 Figure 23. SNR vs TEMPERATURE Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 18 fIN = 1 kHz 101 fIN = 1 kHz 17.5 Effective Number of Bits Signal-to-Noise and Distortion (dBFS) 102 100 99 98 97 96 95 17 16.5 16 15.5 15 14.5 94 14 -40 -15 10 35 Free-Air Temperature (oC) 60 85 -40 Figure 24. SINAD vs TEMPERATURE Spurious-Free Dynamic Range (dBFS) Total Harmonic Distortion (dBFS) fIN = 1 kHz -113 -115 -117 -119 -121 -123 -125 60 85 C02 126 fIN = 1 kHz 124 122 120 118 116 114 112 110 -40 -15 10 35 Free-Air Temperature (oC) 60 85 -40 -15 C02 Figure 26. THD vs TEMPERATURE 10 35 Free-Air Temperature (oC) 60 85 C02 Figure 27. SFDR vs TEMPERATURE 101 Signal-to-Noise and Distortion (dBFS) 106 Signal-toNoise Ratio (dBFS) 10 35 Free-Air Temperature (oC) Figure 25. ENOB vs TEMPERATURE -109 -111 -15 C01 104 102 100 98 96 94 92 100 99 98 97 96 95 94 93 90 0 20 40 60 Input Frequency (kHz) 80 100 0 C02 Figure 28. SNR vs INPUT FREQUENCY 20 40 60 Input Frequency (kHz) 80 Product Folder Links: ADS8883 C02 Figure 29. SINAD vs INPUT FREQUENCY Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated 100 13 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com TYPICAL CHARACTERISTICS (continued) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 18 -100 Total Harmonic Distortion (dBFS) Effective Number of Bits 17.5 17 16.5 16 15.5 15 14.5 14 0 20 40 60 Input Frequency (kHz) 80 100 -103 -106 -109 -112 -115 -118 -121 -124 0 124 80 100 C02 2 121 118 115 112 109 106 1.6 1.2 0.8 0.4 103 100 0 0 20 40 60 Input Frequency (kHz) 80 100 -40 -15 10 35 60 Free-Air Temperature (oC) C02 Figure 32. SFDR vs INPUT FREQUENCY 85 C00 Figure 33. SUPPLY CURRENT vs TEMPERATURE 2 5.5 5 Analog Suppy Current (mA) Power Consumption (mW) 40 60 Input Frequency (kHz) Figure 31. THD vs INPUT FREQUENCY Analog Supply Current (mA) Spurious-Free Dynamic Range (dBFS) Figure 30. ENOB vs INPUT FREQUENCY 4.5 4 3.5 3 2.5 1.6 1.2 0.8 0.4 0 -40 -15 10 35 60 Free-Air Temperature (oC) 85 0 C00 Figure 34. POWER CONSUMPTION vs TEMPERATURE 14 20 C02 85 170 255 340 425 Throughput (kSPS) 510 595 680 C00 Figure 35. SUPPLY CURRENT vs THROUGHPUT Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 TYPICAL CHARACTERISTICS (continued) 6 200 5 175 Power-Down Current (mA) Power Consumption (mW) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 4 3 2 1 150 125 100 75 50 25 0 0 85 170 255 340 425 510 595 Throughput (kSPS) 0 680 -40 -15 C00 Figure 36. POWER CONSUMPTION vs THROUGHPUT 10 35 Free-Air Temperature (oC) 60 85 C03 Figure 37. POWER-DOWN CURRENT vs TEMPERATURE 4 0.01 3 0.006 Gain Error (%FS) Offset (mV) 2 1 0 -1 -2 0.002 -0.002 -0.006 -3 -4 -0.01 -40 -15 10 35 Free-Air Temperature (oC) 60 85 -40 10 35 Free-Air Temperature (oC) 60 85 C03 Figure 39. GAIN ERROR vs TEMPERATURE 105 18000 103 16000 101 14000 99 12000 Frequency Common-Mode Rejection Ratio (dB) Figure 38. OFFSET vs TEMPERATURE 97 95 93 AVDD = 3 V REF = 5 V TA = 25oC 15000 Devices 10000 8000 6000 91 4000 89 2000 87 85 0.001 -15 C03 0 0.01 0.1 1 Input Frequency (kHz) 10 100 -0.01 Figure 40. CMRR vs INPUT FREQUENCY -0.005 0 0.005 0.01 Gain Error (% FS) C03 C03 Figure 41. TYPICAL DISTRIBUTION OF GAIN ERROR Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 15 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com TYPICAL CHARACTERISTICS (continued) At TA = +25°C, AVDD = 3 V, DVDD = 3 V, VREF = 5 V, and fSAMPLE = 680 kSPS, unless otherwise noted. 16000 8000 AVDD = 3 V REF = 5 V TA = 25oC 15000 Devices 7000 12000 5000 Frequency Frequency 6000 AVDD = 3 V REF = 5 V TA = 25ƒC 15000 Devices 14000 4000 3000 10000 8000 6000 2000 4000 1000 2000 0 0 -4 -3 -2 -1 0 1 Offset (mV) 2 3 ±1.0 4 C03 Figure 42. TYPICAL DISTRIBUTION OF OFFSET ERROR 0.0 0.5 1.0 ±0.5 Differential Nonlinearity Min and Max (LSB) 1.5 C038 Figure 43. TYPICAL DISTRIBUTION OF DIFFERENTIAL NONLINEARITY (Minimum and Maximum) 14000 12000 Frequency 10000 AVDD = 3 V REF = 5 V TA = 25ƒC 15000 Devices 8000 6000 4000 2000 0 ±3.0 ±2.5 ±2.0 ±1.5 ±1.0 ±0.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0 Integral Nonlinearity Min and Max (LSB) C039 Figure 44. TYPICAL DISTRIBUTION OF INTEGRAL NONLINEARITY (Minimum and Maximum) 16 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 OVERVIEW The ADS8883 is a high-speed, successive approximation register (SAR), analog-to-digital converter (ADC) from a 16- and 18-bit product family. This compact device features high performance. Power consumption is inherently low and scales linearly with sampling speed. The architecture is based on charge redistribution, which inherently includes a sample-and-hold (S/H) function. The ADS8883 supports a true-differential analog input across two pins (INP and INN). When a conversion is initiated, the differential input on these pins is sampled on the internal capacitor array. While a conversion is in progress, both the INP and INN inputs are disconnected from the internal circuit. The ADS8883 uses an internal clock to perform conversions. The device reconnects the sampling capacitors to the INP and INN pins after conversion and then enters an acquisition phase. During the acquisition phase, the device is powered down and the conversion result can be read. The device digital output is available in SPI-compatible format, which makes interfacing with microprocessors, digital signal processors (DSPs), or field-programmable gate arrays (FPGAs) easy. ANALOG INPUT As shown in Figure 45, the device features a differential analog input. Both positive and negative inputs are individually sampled on 55-pF sampling capacitors and the device converts for the voltage difference between the two sampled values: VINP – VINN. AVDD REF DVDD REF CONVST AINP Sample and Hold AINN SCLK SAR ADC ADC SPI DOUT DIN AGND REFM DGND GND GND Figure 45. Detailed Block Diagram Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 17 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com Most differential input SAR ADCs prohibit the input common-mode voltage, VCM (that is, the average voltage between the inputs), at AINP or AINM from varying more than approximately 10% beyond the mid-scale input value. As shown in Figure 46, the device has a unique common-mode voltage detection and rejection block that does not have this restriction and thus allows VCM to be set to any value between 0 V and VREF without degrading device performance. REF AINP + ± 2 Binary Search Algorithm (SAR) + ± I N T E R F A C E AINM Common Mode Voltage Detection and Rejection Block Figure 46. Conceptual Diagram: True Differential Input Structure Table 4 shows the full-scale input range of the device as a function of input common-mode voltage. The device offers a maximum dynamic range for VCM = VREF / 2. The differential input with wide common-mode range allows connecting differential signals from sensors without any signal conditioning. Table 4. Full-Scale Input Range ABSOLUTE INPUT RANGE VCM VCM < VREF / 2 VAINP VAINN 0 to 2 × VCM 0 to 2 x VCM FULL SCALE INPUT RANGE (VFS) (–2 x VCM) to (2 x VCM) VCM = VREF / 2 0 to VREF 0 to VREF (–VREF) to (VREF) VCM > VREF / 2 (2 × VCM – VREF) to VREF (2 × VCM – VREF) to VREF (–2 × (VCM – VREF)) to (2 × (VCM – VREF)) Figure 47 shows an equivalent circuit of the input sampling stage. The sampling switch is represented by a 96-Ω resistance in series with the ideal switch. Refer to the ADC Input Driver section for more details on the recommended driving circuits. Device in Hold Mode 96 AINP 4 pF 55 pF REF 4 pF GND AINN 55 pF 96 GND Figure 47. Input Sampling Stage Equivalent Circuit Figure 45 and Figure 47 illustrate electrostatic discharge (ESD) protection diodes to REF and GND from both analog inputs. Make sure that these diodes do not turn on by keeping the analog inputs within the specified range. 18 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 REFERENCE The device operates with an external reference voltage and switches binary-weighted capacitors onto the reference terminal (REF pin) during the conversion process. The switching frequency is proportional to the internal conversion clock frequency but the dynamic charge requirements are a function of the absolute value of the input voltage and reference voltage. This dynamic load must be supported by a reference driver circuit without degrading the noise and linearity performance of the device. During the acquisition process, the device automatically powers down and does not take any dynamic current from the external reference source. The basic circuit diagram for such a reference driver circuit for precision ADCs is shown in Figure 48. Refer to the ADC Reference Driver section for more details on the application circuits. RREF_FLT Buffer CREF_FLT RBUF_FLT Voltage Reference REF CBUF_FLT ADC Figure 48. Reference Driver Schematic CLOCK The device uses an internal clock for conversion. Conversion duration may vary but is bounded by the minimum and maximum value of tconv, as specified in the Timing Characteristics section. An external SCLK is only used for a serial data read operation. Data are read after a conversion completes and when the device is in acquisition phase for the next sample. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 19 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com ADC TRANSFER FUNCTION The ADS8883 is a unipolar, differential input device. The device output is in twos compliment format. Figure 49 shows ideal characteristics for the device. The full-scale range for the ADC input (AINP – AINN) is equal to twice the reference input voltage to the ADC (2 × VREF). The LSB for the ADC is given by Equation 1. 1 LSB = [2 × (VREF / 218)] (1) ADC Code (Hex) 1FFFF 00000 3FFFF 20001 20000 ±VREF + 1 LSB ±1 LSB 0 VREF ± 1 LSB VIN Differential Analog Input (AINP  AINN) Figure 49. Differential Transfer Characteristics DIGITAL INTERFACE The ADS8883 is a low pin-count device. However, the device offers six different options for interfacing with the digital host. These options can be broadly classified as being either CS mode (in either a 3- or 4-wire interface) or daisychain mode. The device operates in CS mode if DIN is high at the CONVST rising edge. If DIN is low at the CONVST rising edge, or if DIN and CONVST are connected together, the device operates in daisy-chain mode. In both modes, the device can either operate with or without a busy indicator, where the busy indicator is a bit preceding the output data bits that can be used to interrupt the digital host and trigger the data transfer. The 3-wire interface in CS mode is useful for applications that need galvanic isolation on-board. The 4-wire interface in CS mode allows the user to sample the analog input independent of the serial interface timing and, therefore, allows easier control of an individual device while having multiple, similar devices on-board. The daisychain mode is provided to hook multiple devices in a chain similar to a shift register and is useful in reducing component count and the number of signal traces on the board. CS Mode CS mode is selected if DIN is high at the CONVST rising edge. There are four different interface options available in this mode: 3-wire CS mode without a busy indicator, 3-wire CS mode with a busy indicator, 4-wire CS mode without a busy indicator, and 4-wire CS mode with a busy indicator. The following sections discuss these interface options in detail. 20 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 3-Wire CS Mode Without a Busy Indicator This interface option is most useful when a single ADC is connected to an SPI-compatible digital host. In this interface option, DIN can be connected to DVDD and CONVST functions as CS (as shown in Figure 50). As shown in Figure 51, a CONVST rising edge forces DOUT to 3-state, samples the input signal, and causes the device to enter a conversion phase. Conversion is done with the internal clock and continues regardless of the state of CONVST. As a result, CONVST (functioning as CS) can be pulled low after the start of the conversion to select other devices on the board. However, CONVST must return high before the minimum conversion time (tconv-min) elapses and is held high until the maximum possible conversion time (tconv-max) elapses. A high level on CONVST at the end of the conversion ensures the device does not generate a busy indicator. DVDD DIN CONVST CNV SCLK CLK DOUT SDI ADC Digital Host Figure 50. Connection Diagram: 3-Wire CS Mode Without a Busy Indicator (DIN = 1) 1/fsample DIN = HIGH CONVST = 1 CONVST SCLK 1 DOUT D17 2 3 D16 D15 œœ 16 17 18 D1 D0 œœ tACQ tconv-max tconv-min ADC STATE D2 œœ Acquiring Sample N Conversion Result of Sample N Clocked-out while Acquiring Sample N+1 Converting Sample N End-of-Conversion Figure 51. Interface Timing Diagram: 3-Wire CS Mode Without a Busy Indicator (DIN = 1) When conversion is complete, the device enters an acquisition phase and powers down. CONVST (functioning as CS) can be brought low after the maximum conversion time (tconv-max) elapses. On the CONVST falling edge, DOUT comes out of 3-state and the device outputs the MSB of the data. The lower data bits are output on subsequent SCLK falling edges. Data are valid on both SCLK edges. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). DOUT goes to 3-state after the 18th SCLK falling edge or when CONVST goes high, whichever occurs first. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 21 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com 3-Wire CS Mode With a Busy Indicator This interface option is most useful when a single ADC is connected to an SPI-compatible digital host and an interrupt-driven data transfer is desired. In this interface option, DIN can be connected to DVDD and CONVST functions as CS (as shown in Figure 52). The pull-up resistor on the DOUT pin ensures that the IRQ pin of the digital host is held high when DOUT goes to 3-state. As shown in Figure 53, a CONVST rising edge forces DOUT to 3-state, samples the input signal, and causes the device to enter a conversion phase. Conversion is done with the internal clock and continues regardless of the state of CONVST. As a result, CONVST (functioning as CS) can be pulled low after the start of the conversion to select other devices on the board. However, CONVST must be pulled low before the minimum conversion time (tconv-min) elapses and must remain low until the maximum possible conversion time (tconv-max) elapses. A low level on the CONVST input at the end of a conversion ensures the device generates a busy indicator. DVDD CNV CONVST CLK SCLK DIN DVDD DOUT SDI ADC IRQ Digital Host Figure 52. Connection Diagram: 3-Wire CS Mode With a Busy Indicator 1/fsample DIN = DVDD CONVST CONVST = 0 SCLK 1 2 3 D17 D16 œœ 17 18 19 D1 D0 œœ SDO Pulled-up DOUT BUSY D2 œœ tACQ tconv-max tconv-min ADC STATE Acquiring Sample N Conversion Result of Sample N Clocked-out while Acquiring Sample N+1 Converting Sample N End-of-Conversion Figure 53. Interface Timing Diagram: 3-Wire CS Mode With a Busy Indicator (DIN = 1) When conversion is complete, the device enters an acquisition phase and powers down, DOUT comes out of 3state, and the device outputs a busy indicator bit (low level) on the DOUT pin. This configuration provides a highto-low transition on the IRQ pin of the digital host. The data bits are clocked out, MSB first, on the subsequent SCLK falling edges. Data are valid on both SCLK edges. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). DOUT goes to 3-state after the 19th SCLK falling edge or when CONVST goes high, whichever occurs first. 22 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 4-Wire CS Mode Without a Busy Indicator This interface option is useful when one or more ADCs are connected to an SPI-compatible digital host. Figure 54 shows the connection diagram for single ADC, Figure 56 shows the connection diagram for two ADCs. CS CNV DIN CONVST DOUT SDI SCLK CLK ADC Digital Host Figure 54. Connection Diagram: Single ADC with 4-Wire CS Mode Without a Busy Indicator In this interface option, DIN is controlled by the digital host and functions as CS. As shown in Figure 55, with DIN high, a CONVST rising edge selects CS mode, forces DOUT to 3-state, samples the input signal, and causes the device to enter a conversion phase. In this interface option, CONVST must be held at a high level from the start of the conversion until all data bits are read. Conversion is done with the internal clock and continues regardless of the state of DIN. As a result, DIN (functioning as CS) can be pulled low to select other devices on the board. However, DIN must be pulled high before the minimum conversion time (tconv-min) elapses and remains high until the maximum possible conversion time (tconv-max) elapses. A high level on DIN at the end of the conversion ensures the device does not generate a busy indicator. 1/fsample tconv-max tACQ tconv-min CONVST DIN = 1 DIN SCLK 1 2 DOUT D17 D16 œœ 17 18 œœ ADC STATE End-ofConversion Acquiring Sample N D1 D0 œœ Read Sample N Converting Sample N Acquiring Sample N+1 Figure 55. Interface Timing Diagram: Single ADC with 4-Wire CS Mode Without a Busy Indicator Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 23 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com When conversion is complete, the device enters acquisition phase and powers down. DIN (functioning as CS) can be brought low after the maximum conversion time (tconv-max) elapses. On the DIN falling edge, DOUT comes out of 3-state and the device outputs the MSB of the data. The lower data bits are output on subsequent SCLK falling edges. Data are valid on both SCLK edges. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). DOUT goes to 3-state after the 18th SCLK falling edge or when DIN goes high, whichever occurs first. As shown in Figure 56, multiple devices can be hooked together on the same data bus. In this case, as shown in Figure 57, the DIN of the second device (functioning as CS for the second device) can go low after the first device data are read and the DOUT of the first device is in 3-state. Care must be taken so that CONVST and DIN are not both low together at any time during the cycle. CS1 CS2 CNV CONVST DIN CONVST DIN DOUT DOUT SCLK SDI SCLK CLK ADC #1 ADC #2 Digital Host Figure 56. Connection Diagram: Two ADCs with 4-Wire CS Mode Without a Busy Indicator 1/fsample tconv-max tACQ tconv-min CONVST DIN = 1 DIN (ADC 1) DIN = 1 DIN (ADC 2) SCLK 1 2 DOUT D17 D16 œœ 17 18 19 20 D0 D17 D16 ADC STATE Acquiring Sample N Converting Sample N 36 œœ œœ End-ofConversion œœ 35 D1 D1 D0 œœ œœ Read Sample N ADC 1 Read Sample N ADC 2 Acquiring Sample N+1 Figure 57. Interface Timing Diagram: Two ADCs with 4-Wire CS Mode Without a Busy Indicator 24 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 4-Wire CS Mode With a Busy Indicator This interface option is most useful when a single ADC is connected to an SPI-compatible digital host and an interrupt-driven data transfer is desired. In this interface option, the analog sample is least affected by clock jitter because the CONVST signal (used to sample the input) is independent of the data read operation. In this interface option, DIN is controlled by the digital host and functions as CS (as shown in Figure 58). The pull-up resistor on the DOUT pin ensures that the IRQ pin of the digital host is held high when DOUT goes to 3-state. As shown in Figure 59, when DIN is high, a CONVST rising edge selects CS mode, forces DOUT to 3-state, samples the input signal, and causes the device to enter a conversion phase. In this interface option, CONVST must be held high from the start of the conversion until all data bits are read. Conversion is done with the internal clock and continues regardless of the state of DIN. As a result, DIN (acting as CS) can be pulled low to select other devices on the board. However, DIN must be pulled low before the minimum conversion time (tconv-min) elapses and remains low until the maximum possible conversion time (tconv-max) elapses. A low level on the DIN input at the end of a conversion ensures the device generates a busy indicator. CS DIN CNV CONVST CLK SCLK DVDD DOUT SDI IRQ ADC Digital Host Figure 58. Connection Diagram: 4-Wire CS Mode With a Busy Indicator 1/fsample tACQ tconv-max tconv-min CONVST DIN =0 DIN SCLK 1 2 3 D17 D16 œœ 17 18 19 D1 D0 œœ DOUT ADC STATE SDO Pulled-up Acquiring Sample N Converting Sample N BUSY D2 œœ Conversion Result of Sample N Clocked-out while Acquiring Sample N+1 Figure 59. Interface Timing Diagram: 4-Wire CS Mode With a Busy Indicator When conversion is complete, the device enters an acquisition phase and powers down, DOUT comes out of 3state, and the device outputs a busy indicator bit (low level) on the DOUT pin. This configuration provides a highto-low transition on the IRQ pin of the digital host. The data bits are clocked out, MSB first, on the subsequent SCLK falling edges. Data are valid on both SCLK edges. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). DOUT goes to 3-state after the 19th SCLK falling edge or when DIN goes high, whichever occurs first. Care must be taken so that CONVST and DIN are not both low together at any time during the cycle. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 25 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com DAISY-CHAIN MODE Daisy-chain mode is selected if DIN is low at the time of a CONVST rising edge or if DIN and CONVST are connected together. Similar to CS mode, this mode features operation with or without a busy indicator. The following sections discuss these interface modes in detail. Daisy-Chain Mode Without a Busy Indicator This interface option is most useful in applications where multiple ADC devices are used but the digital host has limited interfacing capability. Figure 60 shows a connection diagram with N ADCs connected in the daisy-chain. The CONVST pins of all ADCs in the chain are connected together and are controlled by a single pin of the digital host. Similarly, the SCLK pins of all ADCs in the chain are connected together and are controlled by a single pin of the digital host. The DIN pin for ADC 1 is connected to GND. The DOUT pin of ADC 1 is connected to the DIN pin of ADC 2, and so on. The DOUT pin of the last ADC in the chain (ADC N) is connected to the SDI pin of the digital host. CNV CONVST DIN DOUT SCLK CONVST DIN CONVST DOUT } DIN SCLK CONVST DOUT DIN SCLK SDI DOUT SCLK CLK ADC 1 ADC 2 } ADC N ADC N1 Digital Host Figure 60. Connection Diagram: Daisy-Chain Mode Without a Busy Indicator (DIN = 0) As shown in Figure 61, the device DOUT pin is driven low when DIN and CONVST are low together. With DIN low, a CONVST rising edge selects daisy-chain mode, samples the analog input, and causes the device to enter a conversion phase. In this interface option, CONVST must remain high from the start of the conversion until all data bits are read. When started, the conversion continues regardless of the state of SCLK, however SCLK must be low at the CONVST rising edge so that the device does not generate a busy indicator at the end of the conversion. tconv-min 1/fsample tconv-max tACQ CONVST SCLK 1 2 œœ 17 18 19 20 D17 D16 œœ 35 36 DIN 1 = LOW œœ DOUT 1 DIN 2 D17 D16 D1 D0 D1 D0 œœ œœ D17 DOUT 2 œœ œœ End-ofConversion ADC STATE D16 Acquiring Sample N Converting Sample N D1 D0 œœ Read Sample N ADC 2 Read Sample N ADC 1 Acquiring Sample N+1 Figure 61. Interface Timing Diagram: For Two devices in Daisy-Chain Mode Without a Busy Indicator 26 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 At the end of conversion, every ADC in the chain loads its own conversion result into the internal, 18-bit, shift register and also outputs the MSB bit of this conversion result on its own DOUT pin. All ADCs enter an acquisition phase and power-down. On every subsequent SCLK falling edge, the internal shift register of each ADC latches the data available on its DIN pin and shifts out the next bit of data on its DOUT pin. Therefore, the digital host receives the data of ADC N, followed by the data of ADC N–1, and so on (in MSB-first fashion). A total of 18 x N SCLK falling edges are required to capture the outputs of all N devices in the chain. Data are valid on both SCLK edges. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). Daisy-Chain Mode With a Busy Indicator This interface option is most useful in applications where multiple ADC devices are used but the digital host has limited interfacing capability and an interrupt-driven data transfer is desired. Figure 62 shows a connection diagram with N ADCs connected in the daisy-chain. The CONVST pins of all ADCs in the chain are connected together and are controlled by a single pin of the digital host. Similarly, the SCLK pins of all ADCs in the chain are connected together and are controlled by a single pin of the digital host. The DIN pin for ADC 1 is connected to its CONVST. The DOUT pin of ADC 1 is connected to the DIN pin of ADC 2, and so on. The DOUT pin of the last ADC in the chain (ADC N) is connected to the SDI and IRQ pins of the digital host. CNV CONVST DIN DOUT CONVST DIN DOUT SCLK SCLK ADC 1 ADC 2 CONVST } DIN DOUT CONVST DIN DOUT SCLK SCLK ADC N1 ADC N IRQ SDI CLK } Digital Host Figure 62. Connection Diagram: Daisy-Chain Mode With a Busy Indicator (DIN = 0) Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 27 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com As shown in Figure 63, the device DOUT pin is driven low when DIN and CONVST are low together. A CONVST rising edge selects daisy-chain mode, samples the analog input, and causes the device to enter a conversion phase. In this interface option, CONVST must remain high from the start of the conversion until all data bits are read. When started, the conversion continues regardless of the state of SCLK, however SCLK must be high at the CONVST rising edge so that the device generates a busy indicator at the end of the conversion. 1/fsample tconv-max tconv-min tACQ CONVST SCLK 2 1 th-DI-CNV œœ 18 19 20 21 D17 D16 œœ 36 37 DIN 1 = CONVST œœ DOUT 1 DIN 2 BUSY D17 D1 D0 D1 D0 œœ œœ BUSY DOUT 2 œœ œœ End-ofConversion ADC STATE D17 Acquiring Sample N Read Sample N ADC 2 Converting Sample N D1 D0 œœ Read Sample N ADC 1 Acquiring Sample N+1 Figure 63. Interface Timing Diagram: For Two Devices in Daisy-Chain Mode With a Busy Indicator At the end of conversion, every ADC in the chain loads its own conversion result into the internal, 18-bit, shift register and also forces its DOUT pin high, thereby providing a low-to-high transition on the IRQ pin of the digital host. All ADCs enter an acquisition phase and power-down. On every subsequent SCLK falling edge, the internal shift register of each ADC latches the data available on its DIN pin and shifts out the next bit of data on its DOUT pin. Therefore, the digital host receives the interrupt signal followed by the data of ADC N followed by the data of ADC N–1, and so on (in MSB-first fashion). A total of (18 x N) + 1 SCLK falling edges are required to capture the outputs of all N devices in the chain. Data are valid on both edges of SCLK and can be captured on either edge. However, a digital host capturing data on the SCLK falling edge can achieve a faster reading rate (provided th_CK_DO is acceptable). Note that the busy indicator bits of ADC 1 to ADC N–1 do not propagate to the next device in the chain. POWER SUPPLY The device has two separate power supplies: AVDD and DVDD. The internal circuits of the device operate on AVDD; DVDD is used for the digital interface. AVDD and DVDD can be independently set to any value within the permissible range. Decouple the AVDD and DVDD pins with GND, using individual 1-µF decoupling capacitors placed in close proximity to the pin, as shown in Figure 64. Digital Supply REF Analog Supply AVDD 1 µF DVDD DIN AINP SCLK AINN DOUT GND CONVST 1 µF Figure 64. Supply Decoupling 28 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 POWER SAVING The device has an auto power-down feature that powers down the internal circuitry at the end of every conversion. Referring to Figure 65, the input signal is acquired on the sampling capacitors when the device is in a power-down state (tacq); at the same time, the result for the previous conversion is available for reading. The device powers up on the start of the next conversion. During conversion phase (tconv), the device also consumes current from the reference source (connected to pin REF). tTHROUGHPUT Device Phase tCONV tACQ œœ IREF tACQ ~50000X ~50000X œœ IAVDD tCONV 2 * tTHROUGHPUT ~1200X ~1200X ~2X IAVG(AVDD+REF) Figure 65. Power Scaling with Throughput The conversion time, tconv, is independent of the SCLK frequency. When operating the device at speeds lower than the maximum rated throughput, the conversion time, tconv, does not change; the device spends more time in power-down state. Therefore, as shown in Figure 66, the device power consumption from the AVDD supply and the external reference source is directly proportional to the speed of operation. Extremely low AVDD power-down current (50 nA, typical) and extremely low external reference leakage current (250 nA, typical), make this device ideal for very low throughput applications (such as pulsed measurements). Analog Suppy Current (mA) 2 1.6 1.2 0.8 0.4 0 0 85 170 255 340 425 Throughput (kSPS) 510 595 680 C00 Figure 66. Power Scaling with Throughput Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 29 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com APPLICATION INFORMATION The two primary circuits required to maximize the performance of a high-precision, successive approximation register (SAR), analog-to-digital converter (ADC) are the input driver and the reference driver circuits. This section details some general principles for designing these circuits, followed by some application circuits designed using the ADS8883. ADC REFERENCE DRIVER The external reference source to the ADS8883 must provide low-drift and very accurate voltage for the ADC reference input and support the dynamic charge requirements without affecting the noise and linearity performance of the device. The output broadband noise of most references can be in the order of a few 100 μVRMS. Therefore, to prevent any degradation in the noise performance of the ADC, the output of the voltage reference must be appropriately filtered by using a low-pass filter with a cutoff frequency of a few hundred Hertz. After band-limiting the noise of the reference circuit, the next important step is to design a reference buffer that can drive the dynamic load posed by the reference input of the ADC. The reference buffer must regulate the voltage at the reference pin such that the value of VREF stays within the 1-LSB error at the start of each conversion. This condition necessitates the use of a large capacitor, CBUF_FLT (refer to Figure 48) for regulating the voltage at the reference input of the ADC. The amplifier selected to drive the reference pin should have an extremely low offset and temperature drift with a low output impedance to drive the capacitor at the ADC reference pin without any stability issues. Reference Driver Circuit for VREF = 4.5 V The application circuit in Figure 67 shows the schematic of a complete reference driver circuit that generates a voltage of 4.5 V dc using a single 5-V supply. This circuit is suitable to drive the reference of the ADS8883 at higher sampling rates up to 680 kSPS. The 4.5-V reference voltage in this design is generated by the highprecision, low-noise REF5045 circuit. The output broadband noise of the reference is heavily filtered by a lowpass filter with a 3-dB cutoff frequency of 160 Hz. 20 k 1 µF - AVDD REF5045 Vin + Vout 1 k + 1 k + OPA333 AVDD + THS4281 1 µF 0.2 AVDD Temp Gnd 1 µF AVDD REF AVDD V + AINP 10 µF Trim CONVST ADS8883 AINM GND 1 µF Figure 67. Schematic of Reference Driver Circuit with VREF = 4.5 V The reference buffer is designed with the THS4281 and OPA333 in a composite architecture to achieve superior dc and ac performance at a reduced power consumption, compared to using a single high-performance amplifier. The THS4281 is a high-bandwidth amplifier with a very low output impedance of 1 Ω at a frequency of 1 MHz. The low output impedance makes the THS4281 a good choice for driving a high capacitive load to regulate the voltage at the reference input of the ADC. The high offset and drift specifications of the THS4281 are corrected by using a dc-correcting amplifier (OPA333) inside the feedback loop. The composite scheme inherits the extremely low offset and temperature drift specifications of the OPA333. For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit Data Acquisition (DAQ) Block Optimized for 1-μs Full-Scale Step Response (SLAU512). 30 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 Reference Driver Circuit for VREF = 2.5 V in Ultralow Power, Lower Throughput Applications The application circuit in Figure 68 shows the schematic of a complete reference driver circuit that generates a voltage of 2.5 V dc using a single 3.3-V supply. This ultralow power reference block is suitable to drive the ADS8883 for power-sensitive applications at a relatively lower throughput. This design uses the high-precision REF3325 circuit that provides an accurate 2.5-V reference voltage at an extremely low quiescent current of 5 µA. The output broadband noise of the reference is heavily filtered by a low-pass filter with a 3-dB cutoff frequency of 16 Hz. 1 k  1 µF AVDD REF3325 IN AVDD 10 k  OUT + + 4.7  OPA333 REF AVDD AINP V+ GND 1 µF 22 µF CONVST ADS8883 AVDD AINM GND Figure 68. Schematic of Reference Driver Circuit with VREF = 2.5 V DC The reference buffer is designed using the low-power OPA333 that can operate from a 3.3-V supply at an extremely low quiescent current of 28 µA. The wideband noise contribution from the amplifier is limited by a lowpass filter of a cutoff frequency equal to 1.5 kHz, formed by a 4.7-Ω resistor in combination with a 22-µF capacitor. The 4.7-Ω series resistor creates an additional drop in the reference voltage, which is corrected by a dual-feedback configuration. For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit, 10kSPS Data Acquisition (DAQ) Block Optimized for Ultra Low Power < 1mW (SLAU514). Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 31 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com ADC INPUT DRIVER The input driver circuit for a high-precision ADC mainly consists of two parts: a driving amplifier and a fly-wheel RC filter. The amplifier is used for signal conditioning of the input voltage and its low output impedance provides a buffer between the signal source and the switched capacitor inputs of the ADC. The RC filter helps attenuate the sampling charge injection from the switched-capacitor input stage of the ADC and functions as an antialiasing filter to band-limit the wideband noise contributed by the front-end circuit. Careful design of the front-end circuit is critical to meet the linearity and noise performance of a high-precision, 18-bit ADC such as the ADS8883. Input Amplifier Selection Selection criteria for the input amplifiers is highly dependent on the input signal type as well as the performance goals of the data acquisition system. Some key amplifier specifications to consider while selecting an appropriate amplifier to drive the inputs of the ADC are: • Small-signal bandwidth. Select the small-signal bandwidth of the input amplifiers to be as high as possible after meeting the power budget of the system. Higher bandwidth reduces the closed-loop output impedance of the amplifier, thus allowing the amplifier to more easily drive the low cutoff frequency RC filter (refer to the Antialiasing Filter section) at the inputs of the ADC. Higher bandwidth also minimizes the harmonic distortion at higher input frequencies. In order to maintain the overall stability of the input driver circuit, the amplifier bandwidth should be selected as described in Equation 2: • § · 1 ¸¸ Unity  Gain Bandwidth t 4 u ¨¨ © 2S u RFLT u CFLT ¹ (2) Noise. Noise contribution of the front-end amplifiers should be as low as possible to prevent any degradation in SNR performance of the system. As a rule of thumb, to ensure that the noise performance of the data acquisition system is not limited by the front-end circuit, the total noise contribution from the front-end circuit should be kept below 20% of the input-referred noise of the ADC. Noise from the input driver circuit is bandlimited by designing a low cutoff frequency RC filter, as explained in Equation 3. 2 § V 1 _ AM P_ PP · S ¨ ¸ NG u 2 u ¨ f  en2 _ RM S u u f3dB ¸ 6 . 6 2 ¨ ¸ © ¹ d § SNRdB · ¸ 20 ¹ ¨ 1 VREF u u 10 © 5 2 where: • • • • • V1 / f_AMP_PP is the peak-to-peak flicker noise in µVRMS, en_RMS is the amplifier broadband noise density in nV/√Hz, f–3dB is the 3-dB bandwidth of the RC filter, and NG is the noise gain of the front-end circuit, which is equal to '1' in a buffer configuration. THD AMP d THD ADC  10 dB  • 32 (3) Distortion. Both the ADC and the input driver introduce nonlinearity in a data acquisition block. As a rule of thumb, to ensure that the distortion performance of the data acquisition system is not limited by the front-end circuit, the distortion of the input driver should be at least 10 dB lower than the distortion of the ADC, as shown in Equation 4. (4) Settling Time. For dc signals with fast transients that are common in a multiplexed application, the input signal must settle within an 18-bit accuracy at the device inputs during the acquisition time window. This condition is critical to maintain the overall linearity performance of the ADC. Typically, the amplifier data sheets specify the output settling performance only up to 0.1% to 0.001%, which may not be sufficient for the desired 18-bit accuracy. Therefore, the settling behavior of the input driver should always be verified by TINA™-SPICE simulations before selecting the amplifier. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 Antialiasing Filter Converting analog-to-digital signals requires sampling an input signal at a constant rate. Any higher frequency content in the input signal beyond half the sampling frequency is digitized and folded back into the low-frequency spectrum. This process is called aliasing. Therefore, an analog, antialiasing filter must be used to remove the harmonic content from the input signal before being sampled by the ADC. An antialiasing filter is designed as a low-pass, RC filter, for which the 3-dB bandwidth is optimized based on specific application requirements. For dc signals with fast transients (including multiplexed input signals), a high-bandwidth filter is designed to allow accurately settling the signal at the inputs of the ADC during the small acquisition time window. For ac signals, the filter bandwidth should be kept low to band-limit the noise fed into the input of the ADC, thereby increasing the signal-to-noise ratio (SNR) of the system. Besides filtering the noise from the front-end drive circuitry, the RC filter also helps attenuate the sampling charge injection from the switched-capacitor input stage of the ADC. A differential capacitor, CFLT, is connected across the inputs of the ADC (as shown in Figure 69). This capacitor helps reduce the sampling charge injection and provides a charge bucket to quickly charge the internal sample-and-hold capacitors during the acquisition process. As a rule of thumb, the value of this capacitor should be at least 10 times the specified value of the ADC sampling capacitance. For the ADS8883, the input sampling capacitance is equal to 59 pF, thus the value of CFLT should be greater than 590 pF. The capacitor should be a COG- or NPO-type because these capacitor types have a high-Q, low-temperature coefficient, and stable electrical characteristics under varying voltages, frequency, and time. RFLT ”22  f  3 dB 2S u R FLT 1  R FLT  u C FLT CFLT •590 pF V AINP + ADS8883 AINM GND RFLT ”22  Figure 69. Antialiasing Filter Note that driving capacitive loads can degrade the phase margin of the input amplifiers, thus making the amplifier marginally unstable. To avoid amplifier stability issues, series isolation resistors (RFLT) are used at the output of the amplifiers. A higher value of RFLT is helpful from the amplifier stability perspective, but adds distortion as a result of interactions with the nonlinear input impedance of the ADC. Distortion increases with source impedance, input signal frequency, and input signal amplitude. Therefore, the selection of RFLT requires balancing the stability and distortion of the design. For the ADS8883, TI recommends limiting the value of RFLT to a maximum of 22 Ω in order to avoid any significant degradation in linearity performance. The tolerance of the selected resistors can be chosen as 1% because the use of a differential capacitor at the input balances the effects resulting from any resistor mismatch. The input amplifier bandwidth should be much higher than the cutoff frequency of the antialiasing filter. TI strongly recommends performing a SPICE simulation to confirm that the amplifier has more than 40° phase margin with the selected filter. Simulation is critical because even with high-bandwidth amplifiers, some amplifiers might require more bandwidth than others to drive similar filters. If an amplifier has less than a 40° phase margin with 22-Ω resistors, using a different amplifier with higher bandwidth or reducing the filter cutoff frequency with a larger differential capacitor is advisable. Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 33 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com APPLICATION CIRCUIT EXAMPLES This section describes some common application circuits using the ADS8883. These data acquisition (DAQ) blocks are optimized for specific input types and performance requirements of the system. For simplicity, powersupply decoupling capacitors are not shown in these circuit diagrams; refer to the Power Supply section for suggested guidelines. DAQ Circuit for a 1.5-µs, Full-Scale Step Response The application circuit shown in Figure 70 is optimized for using the ADS8883 at the maximum-specified throughput of 680 kSPS for a full-scale step input voltage. Such step input signals are common in multiplexed applications when switching between different channels. In a worst-case scenario, one channel is at the negative full-scale (NFS) and the other channel is at the positive full-scale (PFS) voltage, in which case the step size is the full-scale range (FSR) of the ADC when the MUX channel is switched. In such applications, the primary design requirement is to ensure that the full-scale step input signal settles to 18bit accuracy at the ADC inputs. This condition is critical to achieve the excellent linearity specifications of the ADC. Therefore, the bandwidth of the antialiasing RC filter should be large enough to allow optimal settling of the input signal during the ADC acquisition time. The filter capacitor helps reduce the sampling charge injection at the ADC inputs, but degrades the phase margin of the driving amplifier, thereby leading to stability issues. Amplifier stability is maintained by the series isolation resistor. Therefore, the component values of the antialiasing filter should be carefully selected to meet the settling requirements of the system as well as to maintain the stability of the input driving amplifiers. For the input driving amplifiers, key specifications include rail-to-rail input and output swing, high bandwidth, high slew rate, and fast settling time. The OPA350 CMOS amplifier meets all these specification requirements for this circuit with a single-supply and low quiescent current. REFERENCE DRIVE CIRCUIT 20 k 1 µF THS4281 + - 1 k + AVDD OPA333 0.2 1 µF + 1 k + REF5045 Vout Vin AVDD AVDD 10 µF Temp 1 µF Trim Gnd 1 µF + AVDD VIN+ + AVDD + OPA350 10 REF AVDD VCM V + AINP 1 nF CONVST ADS8883 AINM GND + VIN- - CONVST 10 OPA350 + + AVDD INPUT DRIVER 18-Bit 680-kSPS SAR ADC Figure 70. DAQ Circuit for 1.5-µs, Full-Scale Step Response For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit Data Acquisition (DAQ) Block Optimized for 1-μs Full-Scale Step Response (SLAU512). 34 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 Low-Power DAQ Circuit for Excellent Dynamic Performance at 680 kSPS The application circuit shown in Figure 71 is optimized for using the ADS8883 at the maximum specified throughput of 680 kSPS for a full-scale sinusoidal signal of 10-kHz frequency. This circuit achieves excellent dynamic performance for the lowest power consumption. The differential ac input signal is processed through low-noise and low-power amplifiers configured as unity-gain buffers and a low-pass, RC filter before being fed into the ADC. In such applications, the input driver must be low in power and noise as well as able to support rail-to-rail input and output swing with a single supply. A high amplifier bandwidth is also preferred to help attenuate highfrequency distortion. However, oftentimes bandwidth and noise are traded off with the power consumption of the amplifier. This circuit uses the OPA320 as the front-end driving amplifier because this device has a relatively low noise density of 7 nV/√Hz for a maximum-specified quiescent current of 1.45 mA per channel. The noise contribution from the front-end amplifier is band-limited by the 3-dB bandwidth of the RC filter, which is designed to be 1.65 MHz in this application. Again, the component values of the antialiasing filter are carefully selected to maintain the stability of the input driving amplifiers. REFERENCE DRIVE CIRCUIT 20 k 1 µF THS4281 + - 1 k + AVDD OPA333 0.2 1 µF + 1 k + REF5045 Vout Vin AVDD AVDD 10 µF Temp 1 µF Trim Gnd 1 µF + AVDD VIN+ + AVDD + OPA320 22 REF AVDD VCM V + AINP 2.2 nF CONVST ADS8883 AINM GND + VIN- - CONVST 22 OPA320 + + AVDD INPUT DRIVER 18-Bit 680-kSPS SAR ADC Figure 71. DAQ Circuit for Lowest Power and Excellent Dynamic Performance at 680 kSPS For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit, 1-MSPS Data Acquisition (DAQ) Block Optimized for Lowest Power (SLAU513). Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 35 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com DAQ Circuit for Lowest Distortion and Noise Performance at 680 kSPS This section describes two application circuits (Figure 72 and Figure 73) that are optimized for using the ADS8883 with lowest distortion and noise performance at a throughput of 680 kSPS. In both applications, the input signal is processed through a high-bandwidth, low-distortion, fully-differential amplifier (FDA) designed in an inverting gain configuration and a low-pass RC filter before being fed into the ADC. As a rule of thumb, the distortion from the input driver should be at least 10 dB less than the ADC distortion. The distortion resulting from variation in the common-mode signal is eliminated by using the FDA in an inverting gain configuration that establishes a fixed common-mode level for the circuit. This configuration also eliminates the requirement of a rail-to-rail swing at the amplifier input. Therefore, these circuits use the low-power THS4521 as an input driver, which provides exceptional ac performance because of its extremely low-distortion and highbandwidth specifications. In addition, the components of the antialiasing filter are such that the noise from the front-end circuit is kept low without adding distortion to the input signal. Differential Input Configuration The circuit in Figure 72 shows a fully-differential DAQ block optimized for low distortion and noise using the THS4521 and ADS8883. This front-end circuit configuration requires a differential signal at the input of the FDA and provides a differential output to drive the ADC inputs. The common-mode voltage of the input signal provided to the ADC is set by the VOCM pin of the THS4521 (not shown in Figure 72). To use the complete dynamic range of the ADC, VOCM can be set to VREF / 2 by using a simple resistive divider. However, note that the ADS8883 allows the common-mode input voltage (VCM) to be set to any value in the range of 0 V to VREF. REFERENCE DRIVE CIRCUIT 20 k  1 µF THS4281 + - 1 k  + AVDD OPA333 0.2  1 µF + 1 k  + REF5045 Vout Vin AVDD AVDD 10 µF Temp 1 µF Trim Gnd 1 µF 1 K  1 K  AVDD AVDD VIN+ VCM + 10  + THS4521 + - 10 nF CONVST ADS8883 AINM 10  GND CONVST + VIN- REF AVDD V + AINP 1 K  1 K  INPUT DRIVER 18-Bit 680-kSPS SAR ADC Figure 72. Differential Input DAQ Circuit for Lowest Distortion and Noise at 680 kSPS For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit, 1-MSPS Data Acquisition (DAQ) Block Optimized for Lowest Distortion and Noise (SLAU515). 36 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 Single-Ended to Differential Configuration The circuit in Figure 73 shows a single-ended to differential DAQ block optimized for low distortion and noise using the THS4521 and the ADS8883. This front-end circuit configuration requires a single-ended ac signal at the input of the FDA and provides a fully-differential output to drive the ADC inputs. The common-mode voltage of the input signal provided to the ADC is set by the VOCM pin of the THS4521 (not shown in Figure 73). To use the complete dynamic range of the ADC, VOCM can be set to VREF / 2 by using a simple resistive divider. However, note that the ADS8883 allows the common-mode input voltage (VCM) to be set to any value in the range of 0 V to VREF. REFERENCE DRIVE CIRCUIT 20 k 1 µF THS4281 + - 1 k + AVDD OPA333 0.2 1 µF + 1 k + REF5045 Vout Vin AVDD AVDD 10 µF Temp 1 µF Trim Gnd 1 µF 1 K 1 K AVDD AVDD VIN+ + 10 + THS4521 + - 10 nF CONVST ADS8883 AINM 10 1 K REF AVDD V + AINP GND 1 K INPUT DRIVER CONVST 18-Bit 680-kSPS SAR ADC Figure 73. Single-Ended to Differential DAQ Circuit for Lowest Distortion and Noise at 680 kSPS Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 37 ADS8883 SBAS567A – MAY 2013 – REVISED DECEMBER 2013 www.ti.com Ultralow-Power DAQ Circuit at 10 kSPS The data acquisition circuit shown in Figure 74 is optimized for using the ADS8883 at a reduced throughput of 10 kSPS with ultralow-power consumption (< 1 mW) targeted at portable and battery-powered applications. In order to save power, this circuit is operated on a single 3.3-V supply. The circuit uses extremely low-power, dual amplifiers (such as the OPA2333) with a maximum quiescent current of 28 µA per channel to drive the ADC inputs. The input amplifiers are configured in a modified unity-gain buffer configuration. The filter capacitor at the ADC inputs attenuates the sampling charge-injection noise from the ADC but effects the stability of the input amplifiers by degrading the phase margin. This attenuation requires a series isolation resistor to maintain amplifier stability. The value of the series resistor is directly proportional to the open-loop output impedance of the driving amplifier to maintain stability, which is high (in the order of kΩ) in the case of low-power amplifiers such as the OPA333. Therefore, a high value of 1 kΩ is selected for the series resistor at the ADC inputs. However, this series resistor creates an additional voltage drop in the signal path, thereby leading to linearity and distortion issues. The dual-feedback configuration used in Figure 74 corrects for this additional voltage drop and maintains system performance at ultralow-power consumption. 1 k  REFERENCE DRIVE CIRCUIT 1 µF REF3325 IN AVDD 4.7  OPA333 10 k  + + OUT GND 22 µF 1 µF AVDD 10 k  AVDD 10 nF REF AVDD AINP CONVST ADS8883 AINM - 1 k  GND CONVST + OPA333 + + VIN+ VCM AVDD 4.7 nF VIN+ + + 1 k  OPA333 ADS8883 running at 10 kSPS - 10 nF 10 k  INPUT DRIVER Figure 74. Ultralow-Power DAQ Circuit at 10 kSPS For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, using a similar device, refer to 18-Bit, 10kSPS Data Acquisition (DAQ) Block Optimized for Ultra Low Power < 1mW (SLAU514). 38 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 ADS8883 www.ti.com SBAS567A – MAY 2013 – REVISED DECEMBER 2013 REVISION HISTORY NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Original (May 2013) to Revision A Page • Changed Wide Common-Mode Voltage Range Features bullet .......................................................................................... 1 • Changed sub-bullets of AC and DC Performance Features bullet ....................................................................................... 1 • Changed Full-Scale Step Settling Features bullet ................................................................................................................ 1 • Deleted last two Applications bullets .................................................................................................................................... 1 • Changed Description section ................................................................................................................................................ 1 • Changed front page graphic ................................................................................................................................................. 1 • Added Family Information, Absolute Maximum Ratings, and Thermal Information tables ................................................... 2 • Added Electrical Characteristics table .................................................................................................................................. 3 • Added Timing Characteristics section .................................................................................................................................. 5 • Added Pin Configurations section ......................................................................................................................................... 9 • Added Typical Characteristics section ................................................................................................................................ 10 • Added Overview section ..................................................................................................................................................... 17 • Added Application Information section ............................................................................................................................... 30 Submit Documentation Feedback Copyright © 2013, Texas Instruments Incorporated Product Folder Links: ADS8883 39 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) ADS8883IDGS ACTIVE VSSOP DGS 10 80 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 8883 ADS8883IDGSR ACTIVE VSSOP DGS 10 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 8883 ADS8883IDRCR ACTIVE VSON DRC 10 3000 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 8883 ADS8883IDRCT ACTIVE VSON DRC 10 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 8883 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of