Product
Folder
Sample &
Buy
Support &
Community
Tools &
Software
Technical
Documents
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
bq24753A Host-Controlled Li-Ion and Li-Polymer Battery Charger
With Low Iq and System Power Selector
1 Features
2 Applications
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
NMOS-NMOS Synchronous Buck Converter with
300 kHz Frequency and >95% Efficiency
30-ns Minimum Driver Dead-time and 99.5%
Maximum Effective Duty Cycle
High-Accuracy Voltage and Current Regulation
– ±0.5% Charge Voltage Accuracy
– ±3% Charge Current Accuracy
– ±3% Adapter Current Accuracy
– ±2% Input Current Sense Amp Accuracy
Integration
– Automatic System Power Selection From
AC/DC Adapter or Battery
– Internal Loop Compensation
– Internal Soft Start
Safety
– Programmable Input Overvoltage Protection
(OVP)
– Fast Dynamic Power Management (DPM)
– Programmable Inrush Adapter Power (ACOP)
and Overcurrent (ACOC) Limits
– Reverse-Conduction Protection Input FET
Supports Two, Three, or Four Li+ Cells
8 – 24 V AC/DC-Adapter Operating Range
Analog Inputs with Ratiometric Programming via
Resistors or DAC/GPIO Host Control
– Charge Voltage (4-4.512 V/cell)
– Charge Current (up to 10 A, with 10-mΩ)
– Adapter Current Limit (DPM)
Status and Monitoring Outputs
– Adapter Present with Programmable Voltage
Threshold
– Current Drawn from Input Source
Battery Learn Cycle Control
Charge Enable
28-pin, 5x5-mm VQFN package
Energy Star Low Iq
– < 10 μA Off-State Battery Discharge Current
– < 1.5 mA Off-State Input Quiescent Current
Notebook, Netbook and Ultra-Mobile Computers
Portable Data Capture Terminals
Portable Printers
Medical Diagnostics Equipment
Battery Bay Chargers
Battery Back-up Systems
3 Description
The bq24753A is a high-efficiency, synchronous
battery charger with integrated compensation and
system power selector logic, offering low component
count for space-constrained Li-Ion and Li-Polymer
battery charging applications. Ratiometric charge
current and voltage programming allows for high
regulation accuracies, and can be either hardwired
with resistors or programmed by the system powermanagement microcontroller using a DAC or GPIOs.
The bq24753A charges two, three, or four series Li+
cells, supporting up to 10 A of charge current, and is
available in a 28-pin, 5x5-mm thin VQFN package.
Device Information(1)
PART NUMBER
bq24753A
PACKAGE
VQFN (28)
BODY SIZE (NOM)
5.00 mm × 5.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
ADAPTER
SYSTEM
ACP
ACDRV
ACDET
OVPSET
ACN
BATDRV
HIDRV
VREF
PH
SRSET
ACSET
bq24753A
1
LODRV
SRP
SRN
GPIO
ACGOOD
CHGEN
CELL
LEARN
ADC
IADAPT Thermal Pad ACOP
Host
Battery
pack
BAT
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Description (continued).........................................
Pin Configuration and Functions .........................
Specifications.........................................................
7.1
7.2
7.3
7.4
7.5
7.6
7.7
8
1
1
1
2
3
4
6
Absolute Maximum Ratings ...................................... 6
ESD Ratings.............................................................. 6
Recommended Operating Conditions....................... 7
Thermal Information .................................................. 7
Electrical Characteristics........................................... 7
Timing Requirements .............................................. 11
Typical Characteristics ............................................ 11
Detailed Description ............................................ 18
8.1 Overview ................................................................. 18
8.2 Functional Block Diagram ....................................... 19
8.3 Feature Description................................................. 20
8.4 Device Functional Modes........................................ 25
9
Application and Implementation ........................ 29
9.1 Application Information............................................ 29
9.2 Typical Application .................................................. 29
10 Power Supply Recommendations ..................... 35
11 Layout................................................................... 35
11.1 Layout Guidelines ................................................. 35
11.2 Layout Example .................................................... 36
12 Device and Documentation Support ................. 37
12.1
12.2
12.3
12.4
12.5
12.6
Device Support......................................................
Documentation Support .......................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
37
37
37
37
37
37
13 Mechanical, Packaging, and Orderable
Information ........................................................... 37
4 Revision History
Changes from Original (July 2009) to Revision A
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section .................................................................................................. 1
•
Added text to the VREF pin description - Do not apply an external voltage source on this pin............................................. 6
•
Added 4 Cell to Figure 14 internal text................................................................................................................................. 13
•
Changed the Functional Block Diagram ............................................................................................................................... 19
•
Deleted bullet - Regulators are at 80% of final voltage ........................................................................................................ 21
2
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
5 Description (continued)
The bq24753A controls external switches to prevent battery discharge back to the input, connect the adapter to
the system, and to connect the battery to the system using 6-V gate drives for better system efficiency. For
maximum system safety, inrush-power limiting provides instantaneous response to high input voltage multiplied
by current. This AC Overpower protection (ACOP) feature limits the input-switch power to the programmed level
on the ACOP pin, and latches off if the high-power condition persists to prevent overheating.
The bq24753A features Dynamic Power Management (DPM) and input power limiting. These features reduce
battery charge current when the input power limit is reached to avoid overloading the AC adapter when supplying
the load and the battery charger simultaneously. A highly-accurate current-sense amplifier enables precise
measurement of input current from the AC adapter to monitor the overall system power.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
3
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
6 Pin Configuration and Functions
PGND
LODRV
REGN
HIDRV
PH
BTST
PVCC
RHD Package
28-Pin VQFN
Top View
28 27 26 25 24 23 22
LEARN
ACN
2
20
CELLS
ACP
3
19
SRP
ACDRV
4
18
SRN
ACDET
5
17
BAT
ACSET
6
16
SRSET
ACOP
7
15
IADAPT
BATDRV
ACGOOD
10 11 12 13 14
VADJ
9
VDAC
8
VREF
21
AGND
1
OVPSET
CHGEN
Pin Functions
PIN
NAME
NO.
ACDET
5
I/O
DESCRIPTION
I
Adapter detected voltage set input. Program the adapter detect threshold by connecting a resistor divider
from adapter input to ACDET pin to AGND pin. Adapter voltage is detected if ACDET-pin voltage is greater
than 2.4 V. The IADAPT current sense amplifier is active when the ACDET pin voltage is greater than 0.6 V.
ACDRV
4
O
AC adapter to system-switch driver output. Connect directly to the gate of the ACFET P-channel power
MOSFET and the reverse conduction blocking P-channel power MOSFET. Connect both FETs as
common-source. Connect the ACFET drain to the system-load side. The PVCC should be connected to the
common-source node to ensure that the driver logic is always active when needed. If needed, an optional
capacitor from gate to source of the ACFET is used to slow down the ON and OFF times. The internal gate
drive is asymmetrical, allowing a quick turn-off and slower turn-on in addition to the internal break-beforemake logic with respect to the BATDRV. The output goes into linear regulation mode when the input
sensed current exceeds the ACOC threshold. ACDRV is latched off after ACOP voltage exceeds 2 V, to
protect the charging system from an ACFET-overpower condition.
ACGOOD
13
O
Valid adapter active-low detect logic open-drain output. Pulled low when Input voltage is above
programmed ACDET. Connect a 10-kΩ pullup resistor from ACGOOD to VREF, or to a different pullupsupply rail.
ACN
2
I
Adapter current sense resistor, negative input. A 0.1-μF ceramic capacitor is placed from ACN to ACP to
provide differential-mode filtering. An optional 0.1-μF ceramic capacitor is placed from ACN pin to AGND
for common-mode filtering.
ACOP
7
I
Input power limit set input. Program the input overpower time constant by placing a ceramic capacitor from
ACOP to AGND. The capacitor sets the time that the input current limit, ACOC, can be sustained before
exceeding the power-MOSFET power limit. When the ACOP voltage exceeds 2 V, then the ACDRV latches
off to protect the charge system from an overpower condition, ACOP. Reset latch by toggling ACDET or
PVCC_UVLO.
ACP
3
I
Adapter current sense resistor, positive input. A 0.1-μF ceramic capacitor is placed from ACN to ACP to
provide differential-mode filtering. A 0.1-μF ceramic capacitor is placed from ACP pin to AGND for
common-mode filtering.
ACSET
6
I
Adapter current set input. The voltage ratio of ACSET voltage versus VDAC voltage programs the input
current regulation set-point during Dynamic Power Management (DPM). Program by connecting a resistor
divider from VDAC to ACSET to AGND; or by connecting the output of an external DAC to the ACSET pin
and connect the DAC supply to the VDAC pin.
4
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
Pin Functions (continued)
PIN
I/O
DESCRIPTION
NAME
NO.
AGND
9
–
Analog ground. Ground connection for low-current sensitive analog and digital signals. On PCB layout,
connect to the analog ground plane, and only connect to PGND through the thermal pad underneath the
IC.
BAT
17
–
Battery voltage remote sense. Directly connect a kelvin sense trace from the battery pack positive terminal
to the BAT pin to accurately sense the battery pack voltage. Place a 0.1-μF capacitor from BAT to AGND
close to the IC to filter high-frequency noise.
BATDRV
14
O
Battery to system switch driver output. Gate drive for the battery to system load BAT PMOS power FET to
isolate the system from the battery to prevent current flow from the system to the battery, while allowing a
low impedance path from battery to system and while discharging the battery pack to the system load.
Connect this pin directly to the gate of the input BAT P-channel power MOSFET. Connect the source of the
FET to the system load voltage node. Connect the drain of the FET to the battery pack positive node. An
optional capacitor is placed from the gate to the source to slow down the switching times. The internal gate
drive is asymmetrical to allow a quick turn-off and slower turn-on, in addition to the internal break-beforemake logic with respect to ACDRV.
BTST
27
–
PWM high side driver positive supply. Connect a 0.1-μF bootstrap ceramic capacitor from BTST to PH.
Connect a small bootstrap Schottky diode from REGN to BTST.
CELLS
20
I
2, 3 or 4 cells selection logic input. Logic low programs 3 cell. Logic high programs 4 cell. Floating
programs 2 cell.
CHGEN
1
I
Charge enable active-low logic input. LO enables charge. HI disables charge.
HIDRV
26
O
PWM high side driver output. Connect to the gate of the high-side power MOSFET with a short trace.
IADAPT
15
O
Adapter current sense amplifier output. IADAPT voltage is 20 times the differential voltage across ACPACN. Place a 100-pF or less ceramic decoupling capacitor from IADAPT to AGND.
LEARN
21
I
Learn mode logic input control pin — logic high to override system selector when adapter is present, the
battery is discharged to recalibrate the battery-pack gas gauge. When adapter is present and LEARN is
high, battery charging is disabled, the adapter is disconnected (ACDRV is off), and the battery is connected
to system (BATDRV is on). System selector automatically switches to adapter if battery is discharged
below LOWBAT (3 V). When adapter is present and LEARN is low, the adapter is connected to system in
normal selector logic (ACDRV is on and BATDRV is off), allowing battery charging. If adapter is not
present, the battery is always connected to the system (ACDRV is off and BATDRV is on).
LODRV
23
O
PWM low side driver output. Connect to the gate of the low-side power MOSFET with a short trace.
OVPSET
8
–
Set input over voltage protection threshold. Charge is disabled and ACDRV is turned off if adapter input
voltage is higher than the OVPSET programmed threshold. Input overvoltage, ACOV, disables charge and
ACDRV when OVPSET > 3.1 V. ACOV does not latch. Program the overvoltage protection threshold by
connecting a resistor divider from adapter input to OVPSET pin to AGND pin.
PGND
22
–
Power ground. Ground connection for high-current power converter node. On PCB layout, connect directly
to source of low-side power MOSFET, to ground connection of in put and output capacitors of the charger.
Only connect to AGND through the Thermal Pad underneath the IC.
PH
25
–
PWM high side driver negative supply. Connect to the phase switching node (junction of the low-side
power MOSFET drain, high-side power MOSFET source, and output inductor). Connect the 0.1-μF
bootstrap capacitor from PH to BTST.
PVCC
28
–
IC power positive supply. Connect to the common-source (diode-OR) point: source of high-side P-channel
MOSFET and source of reverse-blocking power P-channel MOSFET. Place a 0.1-μF ceramic capacitor
from PVCC to PGND pin close to the IC.
REGN
24
O
PWM low side driver positive 6-V supply output. Connect a 1-μF ceramic capacitor from REGN to PGND,
close to the IC. Use for high-side driver bootstrap voltage by connecting a small-signal Schottky diode from
REGN to BTST. REGN is disabled when CHGEN is high.
SRN
18
I
Charge current sense resistor, negative input. A 0.1-μF ceramic capacitor is placed from SRN to SRP to
provide differential-mode filtering. An optional 0.1-μF ceramic capacitor is placed from SRN pin to AGND
for common-mode filtering.
SRP
19
I
Charge current sense resistor, positive input. A 0.1-μF ceramic capacitor is placed from SRN to SRP to
provide differential-mode filtering. A 0.1-μF ceramic capacitor is placed from SRP pin to AGND for
common-mode filtering.
SRSET
16
I
Charge current set input. The voltage ratio of SRSET voltage versus VDAC voltage programs the charge
current regulation set-point. Program by connecting a resistor divider from VDAC to SRSET to AGND; or
by connecting the output of an external DAC to SRSET pin and connect the DAC supply to VDAC pin.
I
Charge voltage set input. The voltage ratio of VADJ voltage versus VDAC voltage programs the battery
voltage regulation set-point. Program by connecting a resistor divider from VDAC to VADJ, to AGND; or, by
connecting the output of an external DAC to VADJ, and connect the DAC supply to VDAC. VADJ
connected to REGN programs the default of 4.2 V per cell.
VADJ
12
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
5
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Pin Functions (continued)
PIN
NAME
NO.
I/O
DESCRIPTION
VDAC
11
I
Charge voltage set reference input. Connect the VREF or external DAC voltage source to the VDAC pin.
Battery voltage, charge current, and input current are programmed as a ratio of the VDAC pin voltage
versus the VADJ, SRSET, and ACSET pin voltages, respectively. Place resistor dividers from VDAC to
VADJ, SRSET, and ACSET pins to AGND for programming. A DAC could be used by connecting the DAC
supply to VDAC and connecting the output to VADJ, SRSET, or ACSET.
VREF
10
O
3.3-V regulated voltage output. Place a 1-μF ceramic capacitor from VREF to AGND pin close to the IC.
This voltage could be used for ratiometric programming of voltage and current regulation. Do not apply an
external voltage source on this pin.
Exposed pad beneath the IC. AGND and PGND star-connected only at the Thermal Pad plane. Always
solder Thermal Pad to the board, and have vias on the Thermal Pad plane connecting to AGND and PGND
planes. It also serves as a thermal pad to dissipate the heat.
Thermal Pad
7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
(2)
PVCC, ACP, ACN, SRP, SRN, BAT, BATDRV, ACDRV
PH
MIN
MAX
–0.3
30
–1
30
REGN, LODRV, VREF, VDAC, VADJ, ACSET, SRSET, ACDET, ACOP, CHGEN,
CELLS, STAT, ACGOOD, LEARN, OVPSET
–0.3
7
VREF, IADAPT
–0.3
3.6
BTST, HIDRV with respect to AGND and PGND
–0.3
36
ACP–ACN, SRP–SRN, AGND–PGND
–0.5
0.5
Junction temperature, TJ
–40
155
Storage temperature, Tstg
–55
155
Voltage range
Maximum difference
voltage
(1)
(2)
UNIT
V
V
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to GND if not specified. Currents are positive into, negative out of the specified terminal. Consult Packaging
Section of the data book for thermal limitations and considerations of packages.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
6
Electrostatic discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged device model (CDM), per JEDEC specification JESD22C101 (2)
±500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM MAX
PH
Voltage range
UNIT
–1
24
V
PVCC, ACP, ACN, SRP, SRN, BAT, BATDRV, ACDRV
0
24
V
REGN, LODRV, VADJ
0
6.5
V
VDAC, IADAPT
0
3.6
V
VREF
0
3.3
V
ACSET, SRSET, TS, ACDET, ACOP, CHGEN, CELLS, ACGOOD,
LEARN, OVPSET
0
5.5
V
BTST, HIDRV with respect to AGND and PGND
0
30
V
AGND, PGND
–0.3
0.3
V
ACP–ACN, SRP–SRN
–0.3
0.3
V
Junction temperature range, TJ
–40
125
Storage temperature range, Tstg
–55
150
Maximum difference voltage
°C
7.4 Thermal Information
bq24753A
THERMAL METRIC (1)
RHD (VQFN)
UNIT
28 PINS
RθJA
(1)
Junction-to-ambient thermal resistance
39
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
7.5 Electrical Characteristics
7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OPERATING CONDITIONS
VPVCC_OP
PVCC Input voltage operating range
8
24
V
CHARGE VOLTAGE REGULATION
VBAT_REG_RNG
BAT voltage regulation range
VVDAC_OP
VDAC reference voltage range
VADJ_OP
VADJ voltage range
4-4.512 V per cell, times 2,3,4 cells
Charge voltage regulation accuracy
Charge voltage regulation set to default
to 4.2 V per cell
8
18.048
V
2.6
3.6
V
0
REGN
V
8 V, 8.4 V, 9.024 V
–0.5%
0.5%
12 V, 12.6 V, 13.536 V
–0.5%
0.5%
16 V, 16.8 V, 18.048 V
–0.5%
0.5%
VADJ connected to REGN, 8.4 V, 12.6 V, 16.8 V
–0.5%
0.5%
VIREG_CHG = VSRP – VSRN
0
100
0
VDAC
VIREG_CHG = 40–100 mV
–3%
3%
CHARGE CURRENT REGULATION
VIREG_CHG
Charge current regulation differential
voltage range
VSRSET_OP
SRSET voltage range
Charge current regulation accuracy
VIREG_CHG = 20 mV
–5%
5%
VIREG_CHG = 5 mV
–25%
25%
VIREG_CHG = 1.5 mV (VBAT>4V)
–33%
33%
0
100
0
VDAC
mV
V
INPUT CURRENT REGULATION
VIREG_DPM
Adapter current regulation differential
voltage range
VACSET_OP
ACSET voltage range
VIREG_DPM = VACP – VACN
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
mV
V
7
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Electrical Characteristics (continued)
7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
VIREG_DPM = 40–100 mV
Input current regulation accuracy
TYP
–3%
MAX
UNIT
3%
VIREG_DPM = 20 mV
–5%
5%
VIREG_DPM = 5 mV
–25%
25%
VIREG_DPM = 1.5 mV
–33%
33%
3.267
VREF REGULATOR
VVREF_REG
VREF regulator voltage
VACDET > 0.6 V, 0-30 mA
IVREF_LIM
VREF current limit
VVREF = 0 V, VACDET > 0.6 V
35
3.3
3.333
VREGN_REG
REGN regulator voltage
VACDET > 0.6 V, 0-75 mA, PVCC > 10 V
5.6
6.2
V
IREGN_LIM
REGN current limit
VREGN = 0 V, VACDET > 0.6 V
90
135
mA
0
24
0
2
80
V
mA
REGN REGULATOR
5.9
ADAPTER CURRENT SENSE AMPLIFIER
VACP/N_OP
Input common mode range
VIADAPT
IADAPT output voltage range
IIADAPT
IADAPT output current
AIADAPT
Current sense amplifier voltage gain
Voltage on ACP/ACN
0
AIADAPT = VIADAPT / VIREG_DPM
Adapter current sense accuracy
1
20
–2%
VIREG_DPM = 20 mV
–3%
3%
VIREG_DPM = 5 mV
–25%
25%
VIREG_DPM = 1.5 mV
–33%
33%
Output current limit
VIADAPT = 0 V
CIADAPT_MAX
Maximum output load capacitance
For stability with 0 mA to 1 mA load
mA
V/V
VIREG_DPM = 40–100 mV
IIADAPT_LIM
V
2%
1
mA
100
pF
ACDET COMPARATOR
VACDET_CHG
ACDET adapter-detect rising threshold
Min voltage to enable charging, VACDET rising
VACDET_CHG_HYS
ACDET falling hysteresis
VACDET falling
2.376
2.40
2.424
deglitch time after VACDET rising
VACDET rising above 2.4V
518
700
908
ms
deglitch time after VACDET falling
VACDET falling below 2.4V
7
9
11
ms
VACDET_BIAS
ACDET enable-bias rising threshold
Min voltage to enable all bias, VACDET rising
0.56
0.62
0.68
VACDET_BIAS_HYS
Adapter present falling hysteresis
VACDET falling
20
mV
ACDET_BIAS rising deglitch
VACDET rising above 0.62V
10
μs
ACDET_BIAS falling deglitch
VACDET falling below 0.62V
10
μs
40
V
mV
V
PVCC / BAT COMPARATOR
VPVCC-BAT_OP
Differential Voltage from PVCC to BAT
VPVCC-BAT_FALL
PVCC to BAT falling threshold
VPVCC-BAT__HYS
PVCC to BAT hysteresis
–20
VPVCC – VBAT to turn off ACFET
140
185
24
V
240
mV
50
PVCC to BAT Rising Deglitch
VPVCC – VBAT > VPVCC-BAT_RISE
PVCC to BAT Falling Deglitch
VPVCC – VBAT < VPVCC-BAT_FALL
7
9
mV
11
ms
μs
6
OPEN-DRAIN LOGIC OUTPUT PIN CHARACTERISTICS (ACGOOD)
VOUT_LO
Output low saturation voltage
Sink Current = 5 mA
0.5
Delay between system power selector
(ACDRV and BATDRV) switching and
ACGOOD edge
V
μs
10
INPUT UNDERVOLTAGE LOCK-OUT COMPARATOR (UVLO)
UVLO
AC Undervoltage rising threshold to exit
Measured on PVCC
UVLO
UVLO(HYS)
AC Undervoltage hysteresis, falling
3.5
4
4.5
260
V
mV
AC LOWVOLTAGE COMPARATOR (ACLOWV)
AC low voltage rising threshold
VACLOWV
AC low voltage falling threshold
3.6
Measure on ACP pin
V
3
ACN / BAT COMPARATOR
VACN-BAT_FALL
ACN to BAT falling threshold
VACN-BAT_HYS
ACN to BAT hysteresis
8
VACN – VBAT to turn on BATDRV
175
285
50
Submit Documentation Feedback
340
mV
mV
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
Electrical Characteristics (continued)
7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ACN to BAT rising deglitch
VACN – VBAT > VACN-BAT_RISE
20
μs
ACN to BAT falling deglitch
VACN – VBAT < VACN-BAT_FALL
6
μs
BAT OVERVOLTAGE COMPARATOR
VOV_RISE
Overvoltage rising threshold
As percentage of VBAT_REG
104%
VOV_FALL
Overvoltage falling threshold
As percentage of VBAT_REG
102%
BAT SHORT (UNDERVOLTAGE) COMPARATOR
VBAT_SHORT_FALL
VBAT falling threshold to begin C/8
charge
VBAT falling
2.75
2.9
3.05
V/cell
VBAT_SHORT_HYS
BATSHORT hysteresis
VBAT rising
230
250
270
mV/cell
Deglitch time of VBAT rising to resume
full charge
VBAT > VBAT_SHORT+VBAT_SHORT_HYS
1.5
Deglitch time of VBAT falling to begin
C/8 charge
VBAT < VBAT_SHORT
1.5
s
BATSHORT EXIT delay to turn on
BATFET and turn off ACFET when
LEARN=HIGH
BATSHORT ENTRY delay to turn off
BATFET and turn on ACFET when
LEARN=HIGH
600
ms
10
μs
CHARGE OVERCURRENT COMPARATOR
VOC
Charge overcurrent rising threshold
As percentage of IREG_CHG
OCP Floor
Minimum overcurrent current limit
Measure at (SRP-SRN)
145%
50
mV
CHARGE UNDERCURRENT COMPARATOR (SYNCHRONOUS TO NON-SYNCHRONOUS TRANSITION)
VISYNSET_FALL
Charge undercurrent falling threshold
VISYNSET_HYS
Charge undercurrent rising hysteresis
Charge undercurrent, falling-current
deglitch
Changing from synchronous to non-synchronous
9.75
13
16.25
mV
8
mV
20
μs
VIREG_DPM < VISYNSET
Charge undercurrent, rising-current
deglitch
640
INPUT OVERPOWER COMPARATOR (ACOP)
VACOC
ACOC Gain for initial ACOC current
limit (Percentage of programmed
VIREG_DPM)
Begins 700 ms after ACDET, VACSET=1V
Input current limited to this threshold for fault
protection
VACOC_CEILING
Maximum ACOC input current limit
(VACP–VACN)max
Internally limited ceiling,
VACOC_MAX = (VACP–VACN)max
ACOP Latch Blankout Time with ACOC
active
(begins 700 ms after ACDET)
Begins 700 ms after ACDET
(does not allow ACOP latch-off, and no ACOP
source current)
250
265
283
%
VIREG_DPM
100
mV
2
ms
VACOP
ACOP pin latch-off threshold voltage
(See ACOP in Pin Configuration and
Functions)
KACOP
Gain for ACOP Source Current when in
ACOC
Current source on when in ACOC limit. Function of
voltage across power FET
IACOP_SOURCE = KACOP ×(VPVCC -VACP)
IACOP_SINK
ACOP Sink Current when not in ACOC
ACOP Latch is reset by going below
ACDET or UVLO
Current sink on when not in ACOC
VACN-SHORT
ACN Short protection threshold latching
ACN < 2.4 V, ACDET > 2.4 V
1.95
2
2.05
V
18
μA / V
5
μA
2.4
V
INPUT OVERVOLTAGE COMPARATOR (ACOV)
VACOV
VACOV_HYS
AC Overvoltage rising threshold on
OVPSET
(See OVPSET in Pin Configuration and
Functions)
Measured on OVPSET
3.007
3.1
AC Overvoltage rising deglitch
1.3
AC Overvoltage falling deglitch
1.3
3.193
V
ms
THERMAL SHUTDOWN COMPARATOR
TSHUT
Thermal shutdown rising temperature
TSHUT_HYS
Thermal shutdown hysteresis, falling
Temperature Increasing
155
°C
20
°C
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
9
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Electrical Characteristics (continued)
7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
BATTERY SWITCH (BATDRV) DRIVER
RDS(off)_BAT
BATFET Turn-off resistance
VACN > 5 V
160
Ω
RDS(on)_BAT
BATFET Turn-on resistance
VACN > 5 V
3
kΩ
VBATDRV_REG
BATFET drive voltage
VBATDRV_REG = VACN – VBATDRV when
VACN > 5 V and BATFET is on
BATFET Power-up delay
Delay to turn off BATFET after adapter is detected
(after VACDET > 2.4 V)
6.5
518
700
V
908
ms
AC SWITCH (ACDRV) DRIVER
RDS(off)_AC
ACFET turn-off resistance
VPVCC > 5 V
80
Ω
RDS(on)_AC
ACFET turn-on resistance
VPVCC > 5 V
2.5
kΩ
VACDRV_REG
ACFET drive voltage
VACDRV_REG = VPVCC – VACDRV when
VPVCC > 5 V and ACFET is on
ACFET Power-up Delay
Delay to turn on ACFET after adapter is detected
(after VACDET > 2.4 V)
6.5
518
700
V
908
ms
PWM HIGH SIDE DRIVER (HIDRV)
RDS(on)_HI
High side driver turn-on resistance
VBTST – VPH = 5.5 V, tested at 100 mA
3
6
Ω
RDS(off)_HI
High side driver turn-off resistance
VBTST – VPH = 5.5 V, tested at 100 mA
0.7
1.4
Ω
VBTST_REFRESH
Bootstrap refresh comparator threshold
voltage
VBTST – VPH when low side refresh pulse is
requested
4
V
PWM LOW SIDE DRIVER (LODRV)
RDS(on)_LO
Low side driver turn-on resistance
REGN = 6 V, tested at 100 mA
3
6
Ω
RDS(off)_LO
Low side driver turn-off resistance
REGN = 6 V, tested at 100 mA
0.6
1.2
Ω
300
360
PWM OSCILLATOR
FSW
PWM switching frequency
VRAMP_HEIGHT
PWM ramp height
240
As percentage of PVCC
6.6
kHz
%PVCC
QUIESCENT CURRENT
IOFF_STATE
Total off-state quiescent current into
pins SRP, SRN, BAT, BTST, PH,
PVCC, ACP, ACN
VBAT = 16.8 V, VACDET < 0.6 V,
VPVCC > 5 V, TJ = 0 to 85°C
IBATQ_CD
Total quiescent current into pins: SRP,
SRN, BAT, BTST, PH
Adapter present, VACDET > 2.4 V, charge disabled
IAC
Adapter quiescent current
VPVCC = 20 V, charge disabled
7
10
μA
100
200
μA
1
1.5
mA
INTERNAL SOFT START (8 steps to regulation current)
Soft start steps
Soft start step time
8
step
1.7
ms
CHARGER SECTION POWER-UP SEQUENCING
Charge-enable delay after power-up
Delay from when adapter is detected to when the
charger is allowed to turn on
518
700
908
ms
LOGIC INPUT PIN CHARACTERISTICS (CHGEN, LEARN)
VIN_LO
Input low threshold voltage
VIN_HI
Input high threshold voltage
0.8
IBIAS
Input bias current
VCHGEN = 0 to VREGN
tCHGEN_DEGLITCH
Charge enable deglitch time
ACDET > 2.4 V, CHGEN rising
2.1
1
2
V
μA
ms
LOGIC INPUT PIN CHARACTERISTICS (CELLS)
VIN_LO
Input low threshold voltage, 3 cells
CELLS voltage falling edge
VIN_MID
Input mid threshold voltage, 2 cells
CELLS voltage rising for MIN,
CELLS voltage falling for MAX
0.8
VIN_HI
Input high threshold voltage, 4 cells
CELLS voltage rising
2.5
IBIAS_FLOAT
Input bias float current for 2-cell
selection
VCHGEN = 0 to VREGN
–1
10
Submit Documentation Feedback
0.5
1.8
V
1
μA
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
7.6 Timing Requirements
7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted)
MIN NOM MAX
UNIT
AC / BAT MOSFET DRIVERS TIMING
Driver dead time – Dead time when switching between ACDRV and BATDRV
μs
10
PWM DRIVERS TIMING
Driver Dead Time – Dead time when switching between LODRV and HIDRV. No load at LODRV
and HIDRV
30
ns
7.7 Typical Characteristics
Table 1. Table of Graphs (1)
FIGURE
VREF Load and Line Regulation
vs Load Current
Figure 1
REGN Load and Line Regulation
vs Load Current
Figure 2
BAT Voltage
vs VADJ/VDAC Ratio
Figure 3
Charge Current
vs SRSET/VDAC Ratio
Figure 4
Input Current
vs ACSET/VDAC Ratio
Figure 5
BAT Voltage Regulation Accuracy
vs Charge Current
Figure 6
BAT Voltage Regulation Accuracy
Figure 7
Charge Current Regulation Accuracy
Figure 8
Input Current Regulation (DPM) Accuracy
Figure 9
VIADAPT Input Current Sense Amplifier Accuracy
Figure 10
Input Regulation Current (DPM), and Charge Current
vs System Current
Figure 11
Transient System Load (DPM) Response
Figure 12
Fast (DPM) Response
Figure 13
Charge Current Regulation
vs BAT Voltage
Figure 14
Efficiency
vs Battery Charge Current
Figure 15
Battery Removal (from Constant Current Mode)
Figure 16
ACDRV and BATDRV Startup
Figure 17
REF and REGN Startup
Figure 18
System Selector on Adapter Insertion with 390-μF SYS-to-PGND System Capacitor
Figure 19
System Selector on Adapter Removal with 390-μF SYS-to-PGND System Capacitor
Figure 20
System Selector LEARN Turn-On with 390-μF SYS-to-PGND System Capacitor
Figure 21
System Selector LEARN Turn-Off with 390-μF SYS-to-PGND System Capacitor
Figure 22
System Selector on Adapter Insertion
Figure 23
Selector Gate Drive Voltages, 700 ms delay after ACDET
Figure 24
System Selector when Adapter Removed
Figure 25
Charge Enable / Disable and Current Soft-Start
Figure 26
Nonsynchronous to Synchronous Transition
Figure 27
Synchronous to Nonsynchronous Transition
Figure 28
Near 100% Duty Cycle Bootstrap Recharge Pulse
Figure 29
Battery Shorted Charger Response, Over Current Protection (OCP) and Charge Current Regulation
Figure 30
Continuous Conduction Mode (CCM) Switching Waveforms
Figure 32
Discontinuous Conduction Mode (DCM) Switching Waveforms
Figure 31
BATSHORT Entry
Figure 33
BATSHORT Exit
Figure 34
(1)
Test results based on Figure 37 application schematic. VIN = 20 V, VBAT = 3-cell Li-Ion, ICHG = 3 A, IADAPTER_LIMIT = 4 A, TA = 25°C,
unless otherwise specified.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
11
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
0
0.50
-0.50
Regulation Error - %
Regulation Error - %
0.40
0.30
PVCC = 10 V
0.20
0.10
0
-1
-1.50
PVCC = 10 V
-2
PVCC = 20 V
-0.10
-2.50
-0.20
-3
PVCC = 20 V
0
10
20
30
VREF - Load Current - mA
40
50
Figure 1. VREF Load and Line Regulation vs Load Current
0
20
70
80
10
VADJ = 0 -VDAC,
4-Cell,
No Load
17.8
SRSET Varied,
4-Cell,
Vbat = 16 V
9
Charge Current Regulation - A
18
17.6
17.4
17.2
17
16.8
16.6
16.4
8
7
6
5
4
3
2
1
16.2
0
16
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0
1
0.1
0.2
VADJ/VDAC Ratio
0.3
0.4
0.5
0.6
0.7
SRSET/VDAC Ratio
0.8
0.9
1
Figure 4. Charge Current vs SRSET/VDAC Ratio
Figure 3. BAT Voltage vs VADJ/VDAC Ratio
0.2
10
ACSET Varied,
4-Cell,
Vbat = 16 V
8
Vreg = 16.8 V
Regulation Error - %
9
Input Current Regulation - A
30
40
50
60
REGN - Load Current - mA
Figure 2. REGN Load and Line Regulation vs Load Current
18.2
Voltage Regulation - V
10
7
6
5
4
3
0.1
0
-0.1
2
1
-0.2
0
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
ACSET/VDAC Ratio
0.8
0.9
Figure 5. Input Current vs ACSET/VDAC Ratio
12
1
0
2000
4000
Charge Current - mA
6000
8000
Figure 6. BAT Voltage Regulation Accuracy vs Charge
Current
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
0.10
2
SRSET Varied
0
0.06
-1
0.04
Regulation Error - %
Regulation Error - %
4-Cell, VBAT = 16 V
1
VADJ = 0 -VDAC
0.08
4-Cell, no load
0.02
0
-0.02
-0.04
-0.06
-2
-3
-4
-5
-6
-7
-8
-0.08
-9
-0.10
16.5
-10
17
17.5
18
18.5
0
19
2
4
I(CHRG) - Setpoint - A
V(BAT) - Setpoint - V
8
Figure 8. Charge Current Regulation Accuracy
Figure 7. BAT Voltage Regulation Accuracy
5
10
ACSET Varied
9
0
8
7
4-Cell, VBAT = 16 V
6
Percent Error
Regulation Error - %
6
5
4
3
2
VI = 20 V, CHG = EN
-5
VI = 20 V, CHG = DIS
-10
-15
1
0
-20
-1
-2
-25
Iadapt Amplifier Gain
0
1
2
3
4
Input Current Regulation Setpoint - A
5
6
Figure 9. Input Current Regulation (DPM) Accuracy
0
1
2
3
4
5
6
I(ACPWR) - A
7
8
9
10
Figure 10. VIADAPT Input Current Sense Amplifier Accuracy
5
Input Current
ISYS
3
Ch4
2 A/div
Ichrg and Iin - A
4
Ch2
2 A/div
VI = 20 V,
4-Cell,
Vbat = 16 V
2
System Current
IIN
Charge Current
Ch3
2 A/div
1
0
0
1
2
System Current - A
3
IBAT
4
t − Time = 4 ms/div
Figure 11. Input Regulation Current (DPM), and Charge
Current vs System Current
Figure 12. Transient System Load (DPM) Response
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
13
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Ch2
2 A/div
5
Vin = 20 V,
Ichrg_set = 4 A,
TA = 20°C,
4 Cell
Charge Current - A
4
ISYS
Ch4
1 A/div
IBAT
3
2
1
0
0
Efficiency - %
90
4
6
8
10
12
14
16
18
Figure 14. Charge Current Regulation vs BAT Voltage
Figure 13. Fast DPM Response
100
2
Battery Voltage - V
t − Time = 40 ms/div
V(BAT) = 16.8 V
Vreg = 12.6 V
Vreg = 8.4 V
80
70
0
2000
8000
6000
4000
Battery Charge Current - mA
Ch4
5 V/div
Ch3
5 V/div
Ch2
20 V/div
Ch1
2 V/div
Figure 15. Efficiency vs Battery Charge Current
Figure 16. Battery Removal
VACDET
VBATDRV
VACDRV
VACGOOD
t − Time = 100 ms/div
Figure 17. ACDRV and BATDRV Startup
14
Submit Documentation Feedback
Figure 18. REF and REGN Startup
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
Ch3
Ch2
Ch1
10 V/div 20 V/div 20 V/div
SLUS920A – JULY 2009 – REVISED JULY 2015
Ch3
Ch2
Ch1
10 V/div 20 V/div 20 V/div
www.ti.com
VBAT
VSYS
Ch4
10 V/div
Ch4
10 V/div
VACDRV
VBATDRV
VBAT
VSYS
VACDRV
VBATDRV
t − Time = 400 ms/div
t − Time = 2 ms/div
Figure 22. System Selector Learn Turnoff With 390 μF SYSto-PGND System Capacitor
VACDRV
VACGOOD
IL
VACDRV
Ch1
10.8 V
Ch3
Ch4
Ch2
5 A/div 500 mV/div 5 V/div
Ch4
5 A/div
VACPWR
Ch1
5 V/div
Figure 21. System Selector Learn Turnon With 390 μF SYSto-PGND System Capacitor
Ch1
7.2 V
Figure 20. System Selector On Adapter Removal With 390
μF SYS-to-PGND System Capacitor
Ch3
Ch2
Ch1
5 V/div 20 V/div 20 V/div
Figure 19. System Selector On Adapter Insertion With 390
μF SYS-to-PGND System Capacitor
VSYS
VACOP
IIN
t − Time = 400 ms/div
t − Time = 1 ms/div
Figure 23. System Selector On Adapter Insertion
Figure 24. Selector Gate Drive Voltages, 700 ms Delay After
ACDET
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
15
bq24753A
Ch1
10 V/div
www.ti.com
Ch4
1 V/div
ACDET
Ch2
20 V/div
ACDRV
VSYS
Ch3
2 A/div
Ch4
5 V/div
Ch3
Ch1
Ch2
10 V/div 10 V/div 1 V/div
SLUS920A – JULY 2009 – REVISED JULY 2015
ACGOOD
VCHGEN
VBAT
VPH
IBAT
t − Time = 4 ms/div
t − Time = 2 ms/div
Figure 26. Charge Enable / Disable and Current Soft-Start
VHIDRV
Ch2
10 V/div
VPH
VLODRV
Ch4
5 V/div
VPH
VLDDRV
Ch3
2 A/div
Ch2
Ch3
Ch4
2 A/div 5 V/div 10 V/div
Ch1
10 V/div
Figure 25. System Selector On Adapter Removal
IL
IL
t − Time = 2 ms/div
t − Time = 4 ms/div
Ch4
10 V/div
Figure 28. Synchronous to Nonsynchronous Transition
VPH
VHIDRV
VLODRV
IL
IL
t − Time = 400 ms/div
t − Time = 4 ms/div
Figure 29. Near 100% Duty Cycle Bootstrap Recharge Pulse
16
VBAT
Ch3
2 A/div
Ch4
Ch3
2 A/div 5 V/div
Ch2
Ch1
20 V/div 20 V/div
Figure 27. Nonsynchronous to Synchronous Transition
Figure 30. Battery Shorted Charger Response, Overcurrent
Protection (OCP) and Charge Current Regulation
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
Ch1
20 V/div
SLUS920A – JULY 2009 – REVISED JULY 2015
Ch2
20 V/div
VPH
Ch3
5 V/div
VHIDRV
VLODRV
Ch4
2 A/div
Ch4
5 A/div
Ch3
5 V/div
Ch2
20 V/div
Ch1
20 V/div
www.ti.com
IL
VPH
VHIDRV
VLODRV
IL
t − Time = 1 ms/div
t − Time = 1 ms/div
Ch3
5 V/div
Figure 32. Discontinuous Conduction Mode (DCM)
Switching Waveforms
Ch1
10 V/div
VBAT
BATDRV
Ch2
10 V/div
Ch2
10 V/div
Ch1
10 V/div
Ch3
5 V/div
Figure 31. Continuous Conduction Mode (CCM) Switching
Waveforms
ACDRV
VBAT
BATDRV
ACDRV
t − Time = 100 ms/div
t − Time = 200 ms/div
Figure 33. BATSHORT Entry
Figure 34. BATSHORT Exit
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
17
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
8 Detailed Description
8.1 Overview
The bq24753A device is a host-controlled, Li-ion or Li-polymer battery charger with integrated compensation and
system power selector logic. The device employs a switched-mode synchronous buck PWM controller with
constant switching frequency. The device controls external switches to prevent battery discharge back to the
input, connect the adapter to the system, and connect the battery to the system using 6-V gate drives for better
system efficiency. The bq24753A features Dynamic Power Management (DPM) which reduces battery charge
current when the input power limit is reached to avoid overloading the AC adapter when supplying current to the
system and the battery charger simultaneously. A highly accurate current-sense amplifier enables precise
measurement of input current from the AC adapter to monitor the overall system power. The input current limit
can be configured through the ACSET pin of the device. Ratiometric charge current and voltage programming
allows for high regulation accuracies, and can be either hardwired with resistors or programmed by the system
power-management microcontroller using a DAC or GPIOs.
18
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
8.2 Functional Block Diagram
ENA_BIAS
-
0.6 V
+
+
ACDET
acvgood_dg
ADAPTER
DETECTED
-
2.4 V
700ms Rising
8ms Falling
Delay
ACVGOOD
/ACGOOD
DRV_/ACGOOD
VREF
PVCC
ACOP
BAT
ACOPDET
+
+
_
+
Isrc=K*V(PVCC-ACP)
K=18 µA/V
ENA_SRC
-
185 mV
-
5 µA
2V
ENA_SNK
8ms Rising
6µs Falling
Delay
PVCC- BAT
pvcc_bat_dg
PVCC-6V
PVCC-6V
LDO
PVCC
ENA_BIAS
+
-
S
Q
R
Q
SYSTEM
POWER
SELECTOR
LOGIC
ACOP_LATCH
ACDET
PVCC_UVLO
/CHGEN
PVCC
/ACDRV
PVCC-6V
ACN
ENA_BIAS
3.3 V
LDO
VREF
EAI
PVCC
EAO
/BATDRV
ACN-6V
ACP
ACFET_ON
FBO
+
V(ACP-ACN)
-
IIN_REG
COMP
ERROR
AMPLIFIER
+
ACN
_
+
+
V(ACN-BAT)
-
+
1V
BAT_OVP
BAT_ER
VBAT_REG
BTST
/CHGEN
-
285 mV
BAT
LEARN
IIN_ER
-
LEVEL
SHIFTER
CHG_OCP
+
20uA
HIDRV
ACOV
ACOP
SRP
V(ACN-BAT)
3.5 mA
+
20X
-
V(SRP-SRN)
ICH_ER
IBAT_ REG +
SRN
BAT_SHORT
UVLO
20 µA
PVCC
ACLOWV
3.5 mA
SYNCH
CHRG_ON
BTST
+
V(SRP - SRN)
-
REGN
6V LDO
ENA_BIAS
/CHGEN
REFRESH
CBTST
LODRV
+
SYNCH
+
4V _
13 mV +-
ACSET
PH
DC-DC
CONVERTER
PWM LOGIC
PH
IC Tj
+
155 degC
-
PGND
TSHUT
SRSET
VBATSET
IBATSET
IINSET
VADJ
BAT
+
104% X VBAT_REG
-
VBAT_REG
BAT_OVP
ACP
ACN
IBAT_REG
RATIO
IIN_REG
PROGRAM
V(SRP-SRN)
+
145% X IBAT_REG
+
VDAC
ACOV
PVCC
-
ACP
3.0 V +
3.1 V +-
CELLS
IADAPT
ACLOWV
+
-
-
UVLO
BAT
+
4 V +-
AGND
V(IADAPT)
CHG_OCP
OVPSET
+
20x
-
2.9 V/cell +-
-
BAT_SHORT
+
bq24753A
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
19
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
8.3 Feature Description
8.3.1 Battery Voltage Regulation
The bq24753A uses a high-accuracy voltage regulator for charging voltage. Internal default battery voltage
setting VBATT = 4.2 V × cell count. The regulation voltage is ratiometric with respect to VADC. The ratio of VADJ
and VDAC provides extra 12.5% adjust range on VBATT regulation voltage. By limiting the adjust range to 12.5%
of the regulation voltage, the external resistor mismatch error is reduced from ±1% to ±0.1%. Therefore, an
overall voltage accuracy as good as 0.5% is maintained, while using 1% mismatched resistors. Ratiometric
conversion also allows compatibility with D/As or microcontrollers (μC). The battery voltage is programmed
through VADJ and VDAC using Equation 1:
é
æ
öù
V
VBATT = cell count ´ ê 4 V + ç 0.512 ´ VADJ ÷ ú
VVDAC ø úû
è
ëê
(1)
The input voltage range of VDAC is between 2.6 V and 3.6 V. VADJ is set between 0 and VDAC. VBATT defaults
to 4.2 V × cell count when VADJ is connected to REGN.
The CELLS pin is the logic input for selecting the cell count. Connect CELLS to the appropriate voltage level to
charge 2,3, or 4 Li+ cells, as shown in Table 2. When charging other cell chemistries, use CELLS to select an
output voltage range for the charger.
Table 2. Cell-Count Selection
CELLS
CELL COUNT
Float
2
AGND
3
VREF
4
The per-cell charge-termination voltage is a function of the battery chemistry. Consult the battery manufacturer to
determine this voltage.
The BAT pin is used to sense the battery voltage for voltage regulation and should be connected as close to the
battery as possible, or directly on the output capacitor. A 0.1-μF ceramic capacitor from BAT to AGND is
recommended to be as close to the BAT pin as possible to decouple high-frequency noise.
8.3.2 Battery Current Regulation
The SRSET input sets the maximum charge current. Battery current is sensed by resistor RSR connected
between SRP and SRN. The full-scale differential voltage between SRP and SRN is 100 mV. Thus, for a 0.010Ω sense resistor, the maximum charging current is 10 A. SRSET is ratiometric with respect to VDAC using
Equation 2:
V
0.10
ICHARGE = SRSET ´
VVDAC RSR
(2)
The input voltage range of SRSET is between 0 and VDAC, up to 3.6 V.
The SRP and SRN pins are used to sense across RSR, with a default value of 10 mΩ. However, resistors of other
values can also be used. A larger sense-resistor value yields a larger sense voltage, and a higher regulation
accuracy. However, this is at the expense of a higher conduction loss.
8.3.3 Input Adapter Current Regulation
The total input current from an AC adapter or other DC sources is a function of the system supply current and
the battery charging current. System current normally fluctuates as portions of the systems are powered up or
down. Without Dynamic Power Management (DPM), the source must be able to supply the maximum system
current and the maximum charger input current simultaneously. By using DPM, the input current regulator
reduces the charging current when the input current exceeds the input current limit set by ACSET. The current
capacity of the AC adapter can be lowered, reducing system cost.
Similar to setting battery-regulation current, adapter current is sensed by resistor RAC connected between ACP
and ACN. Its maximum value is set by ACSET, which is ratiometric with respect to VDAC, using Equation 3:
20
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
IADAPTER =
SLUS920A – JULY 2009 – REVISED JULY 2015
VACSET 0.10
´
VVDAC R AC
(3)
The input voltage range of ACSET is between 0 and VDAC, up to 3.6 V.
The ACP and ACN pins are used to sense RAC with a default value of 10 mΩ. However, resistors of other values
can also be used. A larger sense-resistor value yields a larger sense voltage, and a higher regulation accuracy.
However, this is at the expense of a higher conduction loss.
8.3.4 Adapter Detect and Power Up
An external resistor voltage divider attenuates the adapter voltage to the ACDET pin. The adapter detect
threshold should typically be programmed to a value greater than the maximum battery voltage and lower than
the minimum-allowed adapter voltage. The ACDET divider should be placed before the ACFET in order to sense
the true adapter input voltage whether the ACFET is on or off. Before the adapter is detected, BATFET stays on
and ACFET turns off.
If PVCC is below 4 V, the device is disabled. If ACDET is below 0.6 V but PVCC is above 4 V, part of the bias is
enabled, including a crude bandgap reference, ACFET drive and BATFET drive. IADAPT is disabled and pulled
down to GND. The total quiescent current is less than 10 μA.
When ACDET rises above 0.6 V and PVCC is above 4 V, all the bias circuits are enabled and VREF rises to 3.3
V, and the REGN output rises to 6 V when CHGEN is LOW. IADAPT becomes valid to proportionally reflect the
adapter current.
When ACDET keeps rising and passes 2.4 V, a valid AC adapter is present. 700 ms later, the following occurs:
• ACGOOD is pulled high through the external pull-up resistor to the host digital voltage rail;
• ACFET is allowed to turn on and BATFET turns off consequently (refer to System Power Selector);
• Charging begins if all the conditions are satisfied (refer to Enable and Disable Charging).
8.3.5 Enable and Disable Charging
The following conditions must be valid before the charge function is enabled:
• CHGEN is LOW
• PVCC > UVLO
• Adapter is detected
• Adapter voltage is higher than BAT + 185 mV
• Adapter is not over voltage (ACOV)
• 700 ms delay is complete after the adapter is detected plus 10 ms ACOC time
• Thermal Shut (TSHUT) is not valid
• TS is within the temperature qualification window
• VDAC > 2.4 V
• LEARN is low
8.3.6 System Power Selector
The bq24753A automatically switches between connecting the adapter or battery power to the system load. By
default, the battery is connected to the system during power up or when a valid adapter is not present. When the
adapter is detected, the battery is first disconnected from the system, then the adapter is connected. An
automatic break-before-make algorithm prevents shoot-through currents when the selector transistors switch.
The ACDRV signal drives a pair of back-to-back p-channel power MOSFETs (with sources connected together
and to PVCC) connected between the adapter and ACP. The FET connected to the adapter prevents reverse
discharge from the battery to the adapter when it is turned off. The p-channel FET with the drain connected to
the adapter input provides reverse battery discharge protection when off; and also minimizes system power
dissipation, with its low RDS(on), compared to a Schottky diode. The other p-channel FET connected to ACP
separates the battery from the adapter, and provides both ACOC current limit and ACOP power limit to the
system. The BATDRV signal controls a p-channel power MOSFET placed between BAT and the system.
When the adapter is not detected, the ACDRV output is pulled to PVCC to turn off the ACFET, disconnecting the
adapter from system. BATDRV stays at ACN – 6 V to connect the battery to the system.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
21
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
At 700 ms after adapter is detected, the system begins to switch from the battery to the adapter. The PVCC
voltage must be 185 mV above BAT to enable the switching. The break-before-make logic turns off both ACFET
and BATFET for 10μs before ACFET turns on. This isolates the battery from shoot-through current or any large
discharging current. The BATDRV output is pulled up to ACN and the ACDRV pin is set to PVCC – 6 V by an
internal regulator to turn on the p-channel ACFET, connecting the adapter to the system.
When the adapter is removed, the system waits till ACN drops back to within 285 mV above BAT to switch from
the adapter back to the battery. The break-before-make logic ensures a 10-μs dead time. The ACDRV output is
pulled up to PVCC and the BATDRV pin is set to ACN – 6 V by an internal regulator to turn on the p-channel
BATFET, connecting the battery to the system.
Asymmetrical gate drive for the ACDRV and BATDRV drivers provides fast turn-off and slow turn-on of the
ACFET and BATFET to help the break-before-make logic and to allow a soft-start at turn-on of either FET. The
soft-start time can be further increased, by putting a capacitor from gate to source of the p-channel power
MOSFETs.
To keep BATDRV on when adaptor is removed, do not apply extermal voltage source on VREF pin.
8.3.7 Battery Learn Cycles
A battery Learn cycle can be implemented using the LEARN pin. A logic low on LEARN keeps the system power
selector logic in its default states dependant on the adapter. If adapter is not detected, then; the ACFET is kept
off, and the BATFET is kept on. If the adapter is detected, the BATFET is kept off, and the ACFET is kept on.
When the LEARN pin is at logic high, the system power selector logic is overridden, keeping the ACFET off and
the BATFET on when the adapter is present. This is used to allow the battery to discharge in order to calibrate
the battery gas gauge over a complete discharge/charge cycle. Charge turns off when LEARN is high. The
controller automatically exits the learn cycle when BAT < 2.9 V per cell. BATDRV turns off and ACDRV turns on.
8.3.8 Automatic Internal Soft-Start Charger Current
The charger automatically soft-starts the charger regulation current every time the charger is enabled to ensure
there is no overshoot or stress on the output capacitors or the power converter. The soft-start consists of
stepping-up the charge regulation current into 8 evenly-divided steps up to the programmed charge current. Each
step lasts approximately 1 ms, for a typical rise time of 8 ms. No external components are needed for this
function.
8.3.9 Converter Operation
The synchronous-buck PWM converter uses a fixed-frequency (300 kHz) voltage mode with a feed-forward
control scheme. A Type-III compensation network allows the use of ceramic capacitors at the output of the
converter. The compensation input stage is internally connected between the feedback output (FBO) and the
error-amplifier input (EAI). The feedback compensation stage is connected between the error amplifier input
(EAI) and error amplifier output (EAO). The LC output filter is selected for a nominal resonant frequency of 8
kHz–12.5 kHz.
The resonant frequency, fo, is given by:
1
fO =
2p LOCO
where (from Figure 37):
CO = C11 + C12
LO = L1
An internal sawtooth ramp is compared to the internal EAO error-control signal to vary the duty cycle of the
converter. The ramp height is one-fifteenth of the input adapter voltage, making it always directly proportional to
the input adapter voltage. This cancels out any loop-gain variation due to a change in input voltage, and
simplifies the loop compensation. The ramp is offset by 200 mV in order to allow a 0% duty cycle when the EAO
signal is below the ramp. The EAO signal is also allowed to exceed the sawtooth ramp signal in order to operate
with a 100% duty-cycle PWM request. Internal gate-drive logic allows a 99.98% duty-cycle while ensuring that
22
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
the N-channel upper device always has enough voltage to stay fully on. If the BTST-to-PH voltage falls below 4 V
for more than 3 cycles, the high-side N-channel power MOSFET is turned off and the low-side N-channel power
MOSFET is turned on to pull the PH node down and recharge the BTST capacitor. Then the high-side driver
returns to 100% duty-cycle operation until the (BTST-PH) voltage is detected falling low again due to leakage
current discharging the BTST capacitor below 4 V, and the reset pulse is reissued.
The 300-kHz fixed-frequency oscillator tightly controls the switching frequency under all conditions of input
voltage, battery voltage, charge current, and temperature. This simplifies output-filter design, and keeps it out of
the audible noise region. The charge-current sense resistor RSR should be designed with at least half or more of
the total output capacitance placed before the sense resistor, contacting both sense resistor and the output
inductor; and the other half, or remaining capacitance placed after the sense resistor. The output capacitance
should be divided and placed on both sides of the charge-current sense resistor. A ratio of 50:50 percent gives
the best performance; but the node in which the output inductor and sense resistor connect should have a
minimum of 50% of the total capacitance. This capacitance provides sufficient filtering to remove the switching
noise and give better current-sense accuracy. The Type-III compensation provides phase boost near the crossover frequency, giving sufficient phase margin.
8.3.10 Synchronous and Non-Synchronous Operation
The charger operates in non-synchronous mode when the sensed charge current is below the ISYNSET internal
setting value. Otherwise, the charger operates in synchronous mode.
During synchronous mode, the low-side N-channel power MOSFET is on when the high-side N-channel power
MOSFET is off. The internal gate-drive logic uses break-before-make switching to prevent shoot-through
currents. During the 30-ns dead time where both FETs are off, the back-diode of the low-side power MOSFET
conducts the inductor current. Having the low-side FET turn-on keeps the power dissipation low, and allows safe
charging at high currents. During synchronous mode, the inductor current always flows, and the device operates
in Continuous Conduction Mode (CCM), creating a fixed two-pole system.
During non-synchronous operation, after the high-side n-channel power MOSFET turns off, and after the breakbefore-make dead-time, the low-side n-channel power MOSFET turns on for approximately 80 ns, then the lowside power MOSFET turns off and stays off until the beginning of the next cycle, when the high-side power
MOSFET is turned on again. The 80-ns low-side MOSFET on-time is required to ensure that the bootstrap
capacitor is always recharged and able to keep the high-side power MOSFET on during the next cycle. This is
important for battery chargers, where unlike regular dc-dc converters, there is a battery load that maintains a
voltage and can both source and sink current. The 80-ns low-side pulse pulls the PH node (connection between
high and low-side MOSFET) down, allowing the bootstrap capacitor to recharge up to the REGN LDO value.
After the 80 ns, the low-side MOSFET is kept off to prevent negative inductor current from flowing. The inductor
current is blocked by the turned-off low-side MOSFET, and the inductor current becomes discontinuous. This
mode is called Discontinuous Conduction Mode (DCM).
During the DCM mode, the loop response automatically changes and has a single-pole system at which the pole
is proportional to the load current, because the converter does not sink current, and only the load provides a
current sink. This means that at low currents, the loop response is slower, because there is less sinking current
available to discharge the output voltage. At low currents during non-synchronous operation, there may be a
small amount of negative inductor current during the 80-ns recharge pulse. The charge should be low enough to
be absorbed by the input capacitance. When BTST – PH < 4 V, the 80-ns recharge pulse occurs on LODRV, the
high-side MOSFET does not turn on, and the low-side MOSFET does not turn on (only 80-ns recharge pulse).
In the bq24753A, VISYNSET = ISYN × RSR is internally set to 13 mV as the charge-current threshold at which the
charger changes from non-synchronous operation to synchronous operation. The low-side driver turns on for only
80 ns to charge the boost capacitor. This is important to prevent negative inductor current, which may cause a
boost effect in which the input voltage increases as power is transferred from the battery to the input capacitors.
This boost effect can lead to an overvoltage on the PVCC node and potentially damage the system. The inductor
ripple current is given by:
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
23
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
IRIPPLE_MAX
2
www.ti.com
£ ISYN £ IRIPPLE_ MAX
and
(VIN - VBAT )´
IRIPPLE =
VBAT 1
´
VIN fs
L
VIN ´ (1- D )´ D ´
=
1
fs
L
(4)
where:
VIN = adapter voltage
VBAT = BAT voltage
fS
=
switching frequency
L = output inductor
D = VBAT/VIN, duty-cycle
IRIPPLE_MAX happens when the duty cycle (D) value is close to 0.5 at given VIN, fS, and L.
The ISYNSET comparator, or charge undercurrent comparator, compares the voltage between SRP-SRN and
the internal threshold. The threshold is set to 13 mV on the falling edge with an 8-mV hysteresis on the rising
edge with a 10% variation.
8.3.11 High Accuracy IADAPT Using Current Sense Amplifier (CSA)
An industry-standard, high-accuracy current sense amplifier (CSA) is used by the host or some discrete logic to
monitor the input current through the analog voltage output of the IADAPT pin. The CSA amplifies the sensed
input voltage of ACP – ACN by 20x through the IADAPT pin. The IADAPT output is a voltage source 20 times
the input differential voltage. When PVCC is above 5 V and ACDET is above 0.6 V, IADAPT no longer stays at
ground, but becomes active. If the user wants to lower the voltage, they can use a resistor divider from IOUT to
AGND, and still achieve accuracy over temperature as the resistors can be matched according to their thermal
coefficients.
A 100-pF capacitor connected on the output is recommended for decoupling high-frequency noise. An additional
RC filter is optional, after the 100-pF capacitor, if additional filtering is desired. Note that adding filtering also
adds additional response delay.
8.3.12 Input Overvoltage Protection (ACOV)
ACOV provides protection to prevent system damage due to high input voltage. Once the adapter voltage is
above the programmable OVPSET voltage (3.1 V), charge is disabled, the adapter is disconnected from the
system by turning off ACDRV, and the battery is connected to the system by turning on BATDRV. ACOV is not
latched—normal operation resumes when the OVPSET voltage returns below 3.1 V.
8.3.13 Input Undervoltage Lockout (UVLO)
The system must have 4 V minimum of PVCC voltage for proper operation. This PVCC voltage can come from
either the input adapter or the battery, using a diode-OR input. When the PVCC voltage is below 4 V, the bias
circuits REGN, VREF, and the gate drive bias to ACFET and BATFET stay inactive, even with ACDET above
0.6 V.
8.3.14 AC Lowvoltage (ACLOWV)
ACLOWV clears the break-before-make protection latch when ACP < 3 V in addition to UVLO clearing this latch
when PVCC < UVLO. It ensures the BATDRV is off when ACP < 3 V, and thus this function allows the ACDRV to
turn on the ACFET again when ACP < 3 V or PVCC 2.4 V), there is a 700-ms delay before ACGOOD is asserted low, and Q3 (BATFET) is
turned-off. Then Q1/Q2 (ACFET) are turned on by the ACDRV pin. When Q1/Q2 (ACFET) are turned on, the
ACFET allows operation in linear-regulation mode to limit the maximum input current, ACOC, to a safe level. The
ACOC current limit is 2.65 times the programmed DPM input current limit set by the ratio of ACSET/VDAC. The
maximum allowable current limit is 100 mV across ACP – ACN (10 A for a 10-mΩ sense resistor).
The first 2 ms after the ACDRV signal begins to turn on, ACOC may limit the current; but the controller is not
allowed to latch off in order to allow a reasonable time for the system voltage to rise.
After 2 ms, ACOP is enabled. ACOP allows the ACFET to latch off before the ACFET can be damaged by
excessive thermal dissipation. The controller only latches if the ACOP pin voltage exceeds 2 V with respect to
AGND. In ACOP, a current source begins to charge the ACOP capacitor when the input current is being limited
by ACOC. This current source is proportional to the voltage across the source-drain of the ACFET (VPVCC-ACP) by
an 18-μA/V ratio. This dependency allows faster capacitor charging if the voltage is larger (more power
dissipation). It allows the time to be programmed by the ACOP capacitor selected. If the controller is not limiting
current, a fixed 5-μA sink current into the ACOP pin to discharge the ACOP capacitor. This charge and discharge
effect depends on whether there is a current-limit condition, and has a memory effect that averages the power
over time, protecting the system from potentially hazardous repetitive faults. Whenever the ACOP threshold is
exceeded, the charge is disabled and the adapter is disconnected from the system to protect the ACFET and the
whole system. If the ACFET is latched off, the BATFET is turned on to connect the battery to the system.
The capacitor provides a predictable time to limit the power dissipation of the ACFET. Since the input current is
constant at the ACOC current limit, the designer can calculate the power dissipation on the ACFET.
The ACOC current Limit threshold is equal to:
Power = Id × Vsd = IACOC _ LIM × V(PVCC - ACP)
The time it takes to charge to 2 V can be calculated from:
C
C ACOP × 2V
× DVACOP
Dt = ACOP
=
i ACOP
18mA/V × V(PVCC - ACP)
(5)
An ACOP fault latch off can only be cleared by bringing the ACDET pin voltage below 2.4 V, then above 2.4 V
(for example, remove adapter and reinsert), or by reducing the PVCC voltage below the UVLO threshold and
raising it.
8.4.1.1
Conditions For ACOP Latch Off
702 ms after ACDET (adapter detected), and:
space
a. ACOP voltage > 2V. The ACOP pin charges the ceramic capacitor when in an ACOC current-limit condition.
The ACOP pin discharges the capacitor when not in ACOC current-limit.
b. ACOP protects from a single-pulse ACOC condition depending on duration and source-drain voltage of
ACFET. Larger voltage across ACFET creates more power dissipation so latch-off protection occurs faster,
by increasing the current source out of ACOP pin.
c. Memory effect (capacitor charging and discharging) allows protection from repetitive ACOC conditions,
depending on duration and frequency (see Figure 36).
d. In short conditions when the system is shorted to ground (ACN < 2.4 V) after the initial 2-ms ACDET.
26
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
Device Functional Modes (continued)
In all cases, after 700ms
delay, have input overcurrent protection,
ACOC, by linearly limiting
input current.
Threshold is equal to the
lower of Idpm x 2.65, or
10A.
ACOC, No Latch-off
ACOC, with ACOP Latch-off,
700ms delay after
ACDET, before allow
ACDRV to turn-on
After Latch-Off, Latch
can only clear by:
Latch-off time accumulates
only when in current limit
regulation, ACOC. The time
before latch-off is
programmable with Cacop,
and is inversely proportional to
source-drain voltage of
ACFET (power). Cacop
charge/discharge per time
also provides memory for
power averaging over time.
1) bringing ACDET below
2.4V, then above 2. 4V; or
2) bringing PVCC below
UVLO, then above
UVLO.
700ms
2ms
8ms
Allow Charge to Turn-on
Vin
Vadapter
ACDET
0V
ACGOOD
BATDRV
ACDRV
Vadapter
Vsystem
Vbattery
Ilim = 2.65 x Idpm
(100 mV max
Across ACP_ACN)
Input Current
Allow Charge
Charge Current
V(ACOP)
ACFET overpower protection; initial current limit allows safe soft-start without system voltage droop.
Figure 35. ACOC Protection During Adapter Insertion
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
27
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Device Functional Modes (continued)
Ilim = 2.65xIdpm
Iin
ACOC_REG
V(PVCC-ACP)
LATCH-OFF
Iacop_pin
LATCH-OFF
2V
Memory Effect
Averages Power
V(ACOP)
ACDRV_ON
ON
OFF
LATCH-OFF
Figure 36. ACOC Protection and ACOP Latch Off with Memory Effect Example
28
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The bq24753A battery charger is ideal for a host control high current charging (up to 10 A) and can charge
battery packs consisting of single cells or multiple cells in series.
9.2 Typical Application
RPVCC
10Ω
ADAPTER +
SYSTEM
R10
2Ω
ADAPTER -
C1
2.2 µF
P
Q1 (ACFET)
SI4435
C6
10 µF
RAC
0.010 Ω
P
C7
10 µF
Q2 (ACFET)
SI4435
C3
C2
0.1 µF
0.1 µF
ACN
PVCC
C8
0.1 µF
ACP
ACDRV
ACDET
R2
66.5 kΩ
1%
R5
10 kΩ
/ACGOOD
bq24753A
OVPSET
422 kΩ
1%
L1
8.2 µH
PH
RSR
0.010 Ω
BTST
ACGOOD
R3
P
Q4
FDS6680A
HIDRV
AGND
VREF
N
432 kΩ
1%
Q3(BATFET)
SI4435
BATDRV
R1
REGN
R4
71 kΩ
1%
PACK+
D1
BAT54
C9
0.1 µF
C10
1 µF
C13
0.1 µF
Q5
FDS6680A
SRSET
DAC
LODRV
ACSET
N
C4
1 µF
SRN
CELLS
BAT
C15
0.1 µF
CHGEN
ACOP
VDAC
DAC
ADC
C14
0.1 µF
SRP
LEARN
GPIO
PACK-
PGND
VREF
HOST
C12
10 µF
C11
10 µF
C16
0.47 µF
VADJ
IADAPT
Thermal Pad
C5
100 pF
(1) Pull-up rail could be either VREF or other system rail.
(2) SRSET/ACSET could come from either DAC or resistor dividers.
VIN = 20 V, VBAT = 3-cell Li-Ion, Icharge = 3 A, Iadapter_limit = 3.5 A
Figure 37. Typical System Schematic, Voltage and Current Programmed by DAC
Table 3. Component List For Typical System Circuit of Figure 37
PART DESIGNATOR
QTY
DESCRIPTION
Q1, Q2, Q3
3
P-channel MOSFET, –30 V, –6 A, SO-8, Vishay-Siliconix, Si4435
Q4, Q5
2
N-channel MOSFET, 30 V, 12.5 A, SO-8, Fairchild, FDS6680A
D1
1
Diode, Dual Schottky, 30 V, 200 mA, SOT23, Fairchild, BAT54C
RAC, RSR
2
Sense Resistor, 10 mΩ, 1%, 1 W, 2010, Vishay-Dale, WSL2010R0100F
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
29
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Typical Application (continued)
Table 3. Component List For Typical System Circuit of Figure 37 (continued)
PART DESIGNATOR
QTY
DESCRIPTION
L1
1
Inductor, 8.2 μH, 8.5 A, 24.8 mΩ, Vishay-Dale, IHLP5050CE-01
C1
1
Capacitor, Ceramic, 2.2 μF, 25 V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E225M
C6, C7, C11, C12
4
Capacitor, Ceramic, 10 μF, 35 V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E106M
C4, C10
2
Capacitor, Ceramic, 1 μF, 25 V, 10%, X7R, 2012, TDK, C2012X7R1E105K
C2, C3, C8, C9, C13, C14, C15
7
Capacitor, Ceramic, 0.1 μF, 50 V, 10%, X7R, 0805, Kemet, C0805C104K5RACTU
C5
1
Capacitor, Ceramic, 100 pF, 25 V, 10%, X7R, 0805, Kemet
C16
1
Capacitor, Ceramic, 0.47 μF, 25 V, 10%, X7R, 0805, Kemet
R1
1
Resistor, Chip, 432 kΩ, 1/16 W, 1%, 0402
R2
1
Resistor, Chip, 66.5 kΩ, 1/16 W, 1%, 0402
R3
1
Resistor, Chip, 422 kΩ, 1/16 W, 1%, 0402
R4
1
Resistor, Chip, 71 kΩ, 1/16 W, 5%, 0402
R10
1
Resistor, Chip, 2 Ω, 1 W, 5%, 2010
RPVCC
1
Resistor, Chip, 10 Ω, 1/4 W, 5%, 2016
9.2.1 Design Requirements
For this design example, use the parameters listed in Table 4 as the input parameters.
Table 4. Design Parameters
PARAMETER
EXAMPLE VALUE
AC adapter voltage (VIN)
19 V
AC adapter current limit
4A
Battery charge voltage (number of cells in series)
12.6 V (3 cells)
Battery charge current (during constant current phase)
3A
9.2.2 Detailed Design Procedure
9.2.2.1 Input Capacitance Calculation
During the adapter hot plug-in, the ACDRV has not been enabled. The AC switch is off and the simplified
equivalent circuit of the input is shown in Figure 38.
IIN
Ri
VIN
Li
Vi
Rc
Ci Vc
Figure 38. Simplified Equivalent Circuit During Adapter Insertion
The voltage on the charger input side VIN is given by:
Ri
t
ù
2Li é Ri - RC
VIN (t) = IIN (t) ´ RC + VCi (t) = Ve
sin wt + cos wt ú
ê
i
w
L
i
ë
û
(6)
in which,
R t = Ri + R C w =
30
1
L i Ci
-
æ Rt ö
ç ÷
è 2Li ø
2
IIN (t) =
Vi
Ri
t
2Li
sin w t
e
wL i
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
Rt
t
2Li
VCi (t) = Vi - Vie
æ Rt
ö
s in w t + cosw t ÷
ç
è 2w L i
ø
(7)
The damping conditions is:
Ri ) Rc u 2
Ǹ
Li
Ci
(8)
Figure 39 (a) demonstrates a higher Ci helps dampen the voltage spike. Figure 39 (b) demonstrates the effect of
the input stray inductance Li upon the input voltage spike. Figure 39 (c) shows how increased resistance helps to
suppress the input voltage spike.
35
35
Ci = 20 mF
Ci = 40 mF
30
Input Capacitor Voltage - V
30
Input Capacitor Voltage - V
Li = 5 m H
Ri = 0.21 W
Li = 9.3 mH
25
20
15
10
5
0
Li = 12 mH
Ri = 0.15 W
Ci = 40 mF
25
20
15
10
5
0
0.5
1
1.5
2
2.5
3
3.5
Time - ms
(a) Vc with various Ci values
4
4.5
0
5
0
0.5
1
1.5
2
2.5
3
3.5
Time - ms
(b) Vc with various Li values
4
4.5
5
35
Ri = 0.15 W
Li = 9.3 mH
Ci = 40 mF
Input Capacitor Voltage - V
30
Ri = 0.5 W
25
20
15
10
5
0
0
0.5
1
1.5
2
2.5
3
Time - ms
3.5
4
4.5
5
(c) Vc with various Ri values
Figure 39. Parametric Study of the Input Voltage
As shown in Figure 39, minimizing the input stray inductance, increasing the input capacitance, and adding
resistance (including using higher ESR capacitors) helps suppress the input voltage spike. However, a user often
cannot control input stray inductance and increasing capacitance can increase costs. Therefore, the most
efficient and cost-effective approach is to add an external resistor.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
31
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
Figure 40 depicts the recommended input filter design. The measured input voltage and current waveforms are
shown in Figure 41. The input voltage spike has been well damped by adding a 2-Ω resistor, while keeping the
capacitance low.
VPVCC
VIN
2W
(0.5 W, 1210 anti-surge)
2.2 mF
(25 V, 1210)
Rext
C1
C2
0.1 mF
(50 V, 0805, very close to PVCC)
Figure 40. Recommended Input Filter Design
9.2.2.2 Inductor Selection
The bq24753A has three selectable fixed switching frequency. Higher switching frequency allows the use of
smaller inductor and capacitor values. Inductor saturation current should be higher than the charging current
(ICHG) plus half the ripple current (IRIPPLE):
ISAT ³ ICHG + (1/2) IRIPPLE
(9)
The inductor ripple current depends on input voltage (VIN), duty cycle (D = VOUT/VIN), switching frequency (fS),
and inductance (L):
V ´ D ´ (1 - D)
IRIPPLE = IN
fS ´ L
(10)
The maximum inductor ripple current happens with D = 0.5 or close to 0.5. For example, the battery charging
voltage range is from 9 V to 12.6 V for 3-cell battery pack. For 20-V adapter voltage, 10-V battery voltage gives
the maximum inductor ripple current. Another example is 4-cell battery, the battery voltage range is from 12 V to
16.8 V, and 12-V battery voltage gives the maximum inductor ripple current.
Usually inductor ripple is designed in the range of (20 – 40%) maximum charging current as a trade-off between
inductor size and efficiency for a practical design.
The bq24753A has charge under current protection (UCP) by monitoring charging current sensing resistor cycleby-cycle. The typical cycle-by-cycle UCP threshold is 5-mV falling edge corresponding to 0.5-A falling edge for a
10-mΩ charging current sensing resistor. When the average charging current is less than 125 mA for a 10-mΩ
charging current sensing resistor, the low side MOSFET is off until BTST capacitor voltage needs to refresh the
charge. As a result, the converter relies on low side MOSFET body diode for the inductor freewheeling current.
9.2.2.3 Input Capacitor
Input capacitor should have enough ripple current rating to absorb input switching ripple current. The worst case
RMS ripple current is half of the charging current when duty cycle is 0.5. If the converter does not operate at
50% duty cycle, then the worst case capacitor RMS current occurs where the duty cycle is closest to 50% and
can be estimated by Equation 11:
ICIN = ICHG ´
D × (1 - D)
(11)
Low ESR ceramic capacitor such as X7R or X5R is preferred for input decoupling capacitor and should be
placed to the drain of the high side MOSFET and source of the low side MOSFET as close as possible. Voltage
rating of the capacitor must be higher than normal input voltage level. 25-V rating or higher capacitor is preferred
for 19 – 20-V input voltage. 10 – 20-μF capacitance is suggested for typical of 3 – 4-A charging current.
32
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
Ceramic capacitors show a dc-bias effect. This effect reduces the effective capacitance when a dc-bias voltage is
applied across a ceramic capacitor, as on the input capacitor of a charger. The effect may lead to a significant
capacitance drop, especially for high input voltages and small capacitor packages. See the manufacturer's data
sheet about the performance with a dc bias voltage applied. It may be necessary to choose a higher voltage
rating or nominal capacitance value in order to get the required value at the operating point.
9.2.2.4 Output Capacitor
Output capacitor also should have enough ripple current rating to absorb output switching ripple current. The
output capacitor RMS current is given by:
I
ICOUT = RIPPLE » 0.29 ´ IRIPPLE
2 ´ 3
(12)
The bq24753A has internal loop compensator. To get good loop stability, the resonant frequency of the output
inductor and output capacitor should be designed between 10 kHz and 20 kHz. The preferred ceramic capacitor
is 25-V X7R or X5R for output capacitor. 10 – 20-μF capacitance is suggested for a typical of 3 – 4-A charging
current. Place the capacitors after charging current sensing resistor to get the best charge current regulation
accuracy.
Ceramic capacitors show a dc-bias effect. This effect reduces the effective capacitance when a dc-bias voltage is
applied across a ceramic capacitor, as on the output capacitor of a charger. The effect may lead to a significant
capacitance drop, especially for high output voltages and small capacitor packages. See the manufacturer's data
sheet about the performance with a dc bias voltage applied. It may be necessary to choose a higher voltage
rating or nominal capacitance value in order to get the required value at the operating point.
9.2.2.5 Power MOSFET Selection
Two external N-channel MOSFETs are used for a synchronous switching battery charger. The gate drivers are
internally integrated into the IC with 6 V of gate drive voltage. 30 V or higher voltage rating MOSFETs are
preferred for 19 – 20-V input voltage.
Figure-of-merit (FOM) is usually used for selecting proper MOSFET based on a tradeoff between the conduction
loss and switching loss. For the top side MOSFET, FOM is defined as the product of a MOSFET's on-resistance,
RDS(ON), and the gate-to-drain charge, QGD. For the bottom side MOSFET, FOM is defined as the product of the
MOSFET's on-resistance, RDS(ON), and the total gate charge, QG.
FOMtop = RDS(on) x QGD; FOMbottom = RDS(on) x QG
(13)
The lower the FOM value, the lower the total power loss. Usually lower RDS(ON) has higher cost with the same
package size.
The top-side MOSFET loss includes conduction loss and switching loss. It is a function of duty cycle (D =
VOUT/VIN), charging current (ICHG), MOSFET's on-resistance (RDS(ON)), input voltage (VIN), switching frequency
(fS), turn on time (ton), and turn off time (toff):
1
Ptop = D ´ ICHG2 ´ RDS(on) +
´ VIN ´ ICHG ´ (t on + t off ) ´ f s
2
(14)
The first item represents the conduction loss. Usually MOSFET RDS(ON) increases by 50% with 100°C junction
temperature rise. The second term represents the switching loss. The MOSFET turn-on and turn-off times are
given by:
Q
Q
t on = SW , t off = SW
Ion
Ioff
(15)
Where Qsw is the switching charge, Ion is the turn-on gate driving current, and Ioff is the turn-off gate driving
current. If the switching charge is not given in the MOSFET data sheet, it can be estimated by gate-to-drain
charge (QGD) and gate-to-source charge (QGS):
1
QSW = QGD +
´ QGS
2
(16)
Gate driving current can be estimated by the REGN voltage (VREGN), MOSFET plateau voltage (Vplt), total turn-on
gate resistance (Ron), and turn-off gate resistance (Roff) of the gate driver:
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
33
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
Ion =
VREGN - Vplt
Ron
, Ioff =
www.ti.com
Vplt
Roff
(17)
The conduction loss of the bottom-side MOSFET is calculated with the following equation when it operates in
synchronous continuous conduction mode:
Pbottom = (1 - D) x ICHG 2 x RDS(on)
(18)
When the charger operates in non-synchronous mode, the bottom-side MOSFET is off. As a result, all of the
freewheeling current goes through the body-diode of the bottom-side MOSFET. The body diode power loss
depends on its forward voltage drop (VF), non-synchronous mode charging current (INONSYNC), and duty cycle (D):
PD = VF x INONSYNC x (1 - D)
(19)
The maximum charging current in non-synchronous mode can be up to 0.25 A for a 10-mΩ charging current
sensing resistor or 0.5 A if battery voltage is below 2.5 V. The minimum duty cycle happens at lowest battery
voltage. Choose the bottom-side MOSFET with either an internal Schottky or body diode capable of carrying the
maximum non-synchronous mode charging current.
9.2.3 Application Curve
Figure 41. Adapter DC Side Hot Plug-in Test Waveforms
34
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
10 Power Supply Recommendations
For proper operation of the bq24753A, VCC must be from 8 V to 24 V. When an adapter
ACGOOD goes LOW, the system is connected to an adapter through ACFET/RBFET. An
voltage divider attenuates the adapter voltage before it goes to ACDET. The adapter detect
typically be programmed to a value greater than the maximum battery voltage, but lower than
allowed input voltage and system maximum allowed voltage.
is attached, and
external resistor
threshold should
the IC maximum
11 Layout
11.1 Layout Guidelines
1. It is critical that the exposed thermal pad on the backside of the IC package be soldered to the PCB ground.
Ensure that there are sufficient thermal vias directly under the IC, connecting to the ground plane on the
other layers.
2. The control stage and the power stage should be routed separately. At each layer, the signal ground and the
power ground are connected only at the thermal pad.
3. The AC current-sense resistor must be connected to ACP (pin 3) and ACN (pin 2) with a Kelvin contact. The
area of this loop must be minimized. An additional 0.1-μF decoupling capacitor for ACN is required to further
reduce noise. The decoupling capacitors for these pins should be placed as close to the IC as possible.
4. The charge-current sense resistor must be connected to SRP (pin 19), SRN (pin 18) with a Kelvin contact.
The area of this loop must be minimized. An additional 0.1-μF decoupling capacitor for SRN is required to
further reduce noise. The decoupling capacitors for these pins should be placed as close to the IC as
possible.
5. Decoupling capacitors for PVCC (pin 28), VREF (pin 10), REGN (pin 24) should be placed underneath the IC
(on the bottom layer) with the interconnections to the IC as short as possible.
6. Decoupling capacitors for BAT (pin 17), IADAPT (pin 15) must be placed close to the corresponding IC pins
with the interconnections to the IC as short as possible.
7. Decoupling capacitor CX for the charger input must be placed close to the Q4 drain and Q5 source.
Figure 42 shows the recommended component placement with trace and via locations.
For VQFN information, refer to the following links: SCBA017 and SLUA271.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
35
bq24753A
SLUS920A – JULY 2009 – REVISED JULY 2015
www.ti.com
11.2 Layout Example
(a) Top Layer
(b) Bottom Layer
Figure 42. PCB Layout Layers
36
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
bq24753A
www.ti.com
SLUS920A – JULY 2009 – REVISED JULY 2015
12 Device and Documentation Support
12.1 Device Support
12.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
12.2 Documentation Support
12.2.1 Related Documentation
Quad Flatpack No-Lead Logic Packages Application Report (SCBA017)
QFN/SON PCB Attachment Application Report (SLUA271)
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Submit Documentation Feedback
Copyright © 2009–2015, Texas Instruments Incorporated
Product Folder Links: bq24753A
37
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
BQ24753ARHDR
ACTIVE
VQFN
RHD
28
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BQ
24753A
BQ24753ARHDT
ACTIVE
VQFN
RHD
28
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BQ
24753A
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
