DAC0832LCWM

DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 DAC0830/DAC0832 8-Bit μP Compatible, Double-Buffered D to A Converters Check for Samples: DAC0830, DAC0832 FEATURES KEY SPECIFICATIONS • Double-Buffered, Single-Buffered or FlowThrough Digital Data Inputs • Easy Interchange and Pin-Compatible with 12bit DAC1230 Series • Direct Interface to All Popular Microprocessors • Linearity Specified with Zero and Full Scale Adjust Only—NOT BEST STRAIGHT LINE FIT. • Works with ±10V Reference-Full 4-Quadrant Multiplication • Can Be Used in the Voltage Switching Mode • Logic Inputs Which Meet TTL Voltage Level Specs (1.4V Logic Threshold) • Operates “STAND ALONE” (without μP) if Desired • Available in 20-Pin SOIC or PLCC Package • • • 1 234 • • • Current Settling Time: 1 μs Resolution: 8 bits Linearity: 8, 9, or 10 bits (Ensured Over Temp.) Gain Tempco: 0.0002% FS/°C Low Power Dissipation: 20 mW Single Power Supply: 5 to 15 VDC DESCRIPTION The DAC0830 is an advanced CMOS/Si-Cr 8-bit multiplying DAC designed to interface directly with the 8080, 8048, 8085, Z80®, and other popular microprocessors. A deposited silicon-chromium R-2R resistor ladder network divides the reference current and provides the circuit with excellent temperature tracking characteristics (0.05% of Full Scale Range maximum linearity error over temperature). The circuit uses CMOS current switches and control logic to achieve low power consumption and low output leakage current errors. Special circuitry provides TTL logic input voltage level compatibility. Double buffering allows these DACs to output a voltage corresponding to one digital word while holding the next digital word. This permits the simultaneous updating of any number of DACs. The DAC0830 series are the 8-bit members of a family of microprocessor-compatible DACs (MICRODAC). Typical Application 1 2 3 4 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. BI-FET is a trademark of Texas Instruments. Z80 is a registered trademark of Zilog Corporation. All other trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1999–2013, Texas Instruments Incorporated DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Connection Diagrams (Top Views) Figure 1. PDIP, CDIP, and SOIC Packages Figure 2. PLCC Package These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1) (2) (3) Supply Voltage (VCC) 17 VDC Voltage at Any Digital Input VCC to GND Voltage at VREF Input Storage Temperature Range ±25V −65°C to +150°C Package Dissipation at TA=25°C (4) 500 mW DC Voltage Applied to IOUT1 or IOUT2 (5) −100 mV to VCC ESD Susceptability (5) (6) Lead Temperature (Soldering, 10 sec.) (1) (2) (3) (4) (5) (6) 2 800V PDIP Package (plastic) 260°C CDIP Package (ceramic) 300°C SOIC Package Vapor Phase (60 sec.) 215°C Infrared (15 sec.) 220°C All voltages are measured with respect to GND, unless otherwise specified. Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. DC and AC electrical specifications do not apply when operating the device beyond its specified operating conditions. If Military/Aerospace specified devices are required, please contact the TI Sales Office/ Distributors for availability and specifications. The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any temperature is PD = (TJMAX − TA)/θJA or the number given in the Absolute Maximum Ratings, whichever is lower. For this device, TJMAX = 125°C (plastic) or 150°C (ceramic), and the typical junction-to-ambient thermal resistance of the J package when board mounted is 80°C/W. For the NFH package, this number increases to 100°C/W and for the FN package this number is 120°C/W. For current switching applications, both IOUT1 and IOUT2 must go to ground or the “Virtual Ground” of an operational amplifier. The linearity error is degraded by approximately VOS ÷ VREF. For example, if VREF = 10V then a 1 mV offset, VOS, on IOUT1 or IOUT2 will introduce an additional 0.01% linearity error. Human body model, 100 pF discharged through a 1.5 kΩ resistor. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Operating Conditions TMIN≤TA≤TMAX Temperature Range Part numbers with “LCN” suffix 0°C to +70°C Part numbers with “LCWM” suffix 0°C to +70°C Part numbers with “LCV” suffix 0°C to +70°C Part numbers with “LCJ” suffix −40°C to +85°C −55°C to +125°C Part numbers with “LJ” suffix Voltage at Any Digital Input VCC to GND Electrical Characteristics VREF=10.000 VDC unless otherwise noted. Boldface limits apply over temperature, TMIN≤TA≤TMAX. (1) For all other limits TA=25°C. Parameter Conditions See Note VCC = 4.75 VDC VCC = 15.75 VDC VCC = 5 VDC ±5% VCC = 12 VDC ±5% to 15 VDC ±5% Limit Units (2) Typ (3) Tested Limit (4) Design Limit (5) 8 8 8 bits DAC0830LJ & LCJ 0.05 0.05 % FSR DAC0832LJ & LCJ 0.2 0.2 % FSR DAC0830LCN, LCWM & LCV 0.05 0.05 % FSR DAC0831LCN 0.1 0.1 % FSR DAC0832LCN, LCWM & LCV 0.2 0.2 % FSR DAC0830LJ & LCJ 0.1 0.1 % FSR DAC0832LJ & LCJ 0.4 0.4 % FSR DAC0830LCN, LCWM & LCV 0.1 0.1 % FSR DAC0831LCN 0.2 0.2 % FSR DAC0832LCN, LCWM & LCV 0.4 0.4 % FSR 8 8 bits CONVERTER CHARACTERISTICS Resolution Linearity Error Max Differential Nonlinearity Max Zero and full scale adjusted −10V≤VREF≤+10V Zero and full scale adjusted −10V≤VREF≤+10V Monotonicity −10V≤VREF Gain Error Max Using Internal Rfb ≤+10V LJ & LCJ See (6) and (2) See (6) and (2) See (6) LCN, LCWM & LCV See (7) ±0.2 8 8 bits ±1 ±1 % FS 0.0006 % FS/°C −10V≤VREF≤+10V Gain Error Tempco Max (1) (2) (3) (4) (5) (6) (7) Using internal Rfb 0.0002 Boldface tested limits apply to the LJ and LCJ suffix parts only. The unit “FSR” stands for “Full Scale Range.” “Linearity Error” and “Power Supply Rejection” specs are based on this unit to eliminate dependence on a particular VREF value and to indicate the true performance of the part. The “Linearity Error” specification of the DAC0830 is “0.05% of FSR (MAX)”. This ensures that after performing a zero and full scale adjustment (see sections Zero Adjustment and Full-Scale Adjustment), the plot of the 256 analog voltage outputs will each be within 0.05%×VREF of a straight line which passes through zero and full scale. Typicals are at 25°C and represent most likely parametric norm. Tested limits are ensured to TI's AOQL (Average Outgoing Quality Level). Ensured, but not 100% production tested. These limits are not used to calculate outgoing quality levels. For current switching applications, both IOUT1 and IOUT2 must go to ground or the “Virtual Ground” of an operational amplifier. The linearity error is degraded by approximately VOS ÷ VREF. For example, if VREF = 10V then a 1 mV offset, VOS, on IOUT1 or IOUT2 will introduce an additional 0.01% linearity error. Specified at VREF=±10 VDC and VREF=±1 VDC. Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 3 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Electrical Characteristics (continued) VREF=10.000 VDC unless otherwise noted. Boldface limits apply over temperature, TMIN≤TA≤TMAX.(1) For all other limits TA=25°C. Parameter Conditions VCC=14.5V to 15.5V 0.0002 0.0025 11.5V to 12.5V 0.0006 4.5V to 5.5V 0.013 0.015 Max 15 20 20 kΩ Min 15 10 10 kΩ All digital inputs latched high VREF=20 Vp-p, f=100 kHz All data inputs latched low Output Leakage Current Max (8) All data inputs IOUT1 IOUT1 IOUT2 LJ & LCJ % FSR/V 3 mVp-p See (8) 100 100 LCN, LCWM & LCV 50 100 All data inputs LJ & LCJ 100 100 latched high LCN, LCWM & LCV 50 100 latched low IOUT2 Design Limit (5) Limit Units (2) Tested Limit (4) Output Feedthrough Error Output Capacitance VCC = 5 VDC ±5% VCC = 12 VDC ±5% to 15 VDC ±5% Typ (3) Power Supply Rejection Reference Input See Note VCC = 4.75 VDC VCC = 15.75 VDC All data inputs 45 latched low All data inputs 130 IOUT2 latched high 30 nA pF 115 IOUT1 nA pF DIGITAL AND DC CHARACTERISTICS Digital Input Voltages Max Min Digital Input Currents Supply Current Drain (8) 4 Max Logic Low Logic High LJ: 4.75V 0.6 LJ: 15.75V 0.8 LCJ: 4.75V 0.7 LCJ: 15.75V 0.8 LCN, LCWM, LCV 0.95 0.8 LJ & LCJ 2.0 2.0 LCN, LCWM, LCV 1.9 2.0 −200 −200 μA −160 −200 μA +10 +10 μA +8 +10 3.5 3.5 1.7 2.0 Digital inputs 2.0V LJ & LCJ Max VDC −50 LCN, LCWM, LCV 0.1 LCN, LCWM, LCV LJ & LCJ 1.2 LCN, LCWM, LCV VDC mA A 100nA leakage current with Rfb=20k and VREF=10V corresponds to a zero error of (100×10−9×20×103)×100/10 which is 0.02% of FS. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Electrical Characteristics VREF=10.000 VDC unless otherwise noted. Boldface limits apply over temperature, TMIN≤TA≤TMAX. (1) For all other limits TA=25°C. Symbol Parameter AC CHARACTERISTICS See Note Conditions Typ (2) Current Setting Time VIL=0V, VIH=5V tW Write and XFER VIL=0V, VIH=5V Data Setup Time See ( 5) 100 See ( Data Hold Time VIL=0V, VIH=5V See ( VIL=0V, VIH=5V See ( VIL=0V, VIH=5V See ( See ( 1) 100 Control Setup Time 1) 1) 110 Min Control Hold Time Tested Limit (3) VIL=0V, VIH=5V 1) 0 375 320 Design Limit (4) Limit Units μs 600 900 250 375 320 900 50 30 50 250 0 600 320 0 0 900 ns 900 1100 10 900 600 30 320 Min (1) (2) (3) (4) (5) 250 320 Min tCH Typ (2) VCC=5 VDC±5% 1.0 320 1) Min tCS Design Limit (4) VCC=4.75 VDC 1.0 Pulse Width Min tDH Tested Limit (3) VCC=12 VDC±5% to 15 VDC ±5% (5) ts tDS VCC=15.75 VDC 1100 0 0 Boldface tested limits apply to the LJ and LCJ suffix parts only. Typicals are at 25°C and represent most likely parametric norm. Tested limits are ensured to TI's AOQL (Average Outgoing Quality Level). Ensured, but not 100% production tested. These limits are not used to calculate outgoing quality levels. The entire write pulse must occur within the valid data interval for the specified tW, tDS, tDH, and tS to apply. Switching Waveform Definition of Package Pinouts Control Signals (All control signals level actuated) CS: Chip Select (active low). The CS in combination with ILE will enable WR1. ILE: Input Latch Enable (active high). The ILE in combination with CS enables WR1. WR1: Write 1. The active low WR1 is used to load the digital input data bits (DI) into the input latch. The data in the input latch is latched when WR1 is high. To update the input latch–CS and WR1 must be low while ILE Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 5 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com is high. WR2: Write 2 (active low). This signal, in combination with XFER, causes the 8-bit data which is available in the input latch to transfer to the DAC register. XFER: Transfer control signal (active low). The XFER will enable WR2. Other Pin Functions DI0-DI7: Digital Inputs. DI0 is the least significant bit (LSB) and DI7 is the most significant bit (MSB). IOUT1: DAC Current Output 1. IOUT1 is a maximum for a digital code of all 1's in the DAC register, and is zero for all 0's in DAC register. IOUT2: DAC Current Output 2. IOUT2 is a constant minus IOUT1 , or IOUT1 + IOUT2 = constant (I full scale for a fixed reference voltage). Rfb: Feedback Resistor. The feedback resistor is provided on the IC chip for use as the shunt feedback resistor for the external op amp which is used to provide an output voltage for the DAC. This on-chip resistor should always be used (not an external resistor) since it matches the resistors which are used in the on-chip R-2R ladder and tracks these resistors over temperature. VREF: Reference Voltage Input. This input connects an external precision voltage source to the internal R-2R ladder. VREF can be selected over the range of +10 to −10V. This is also the analog voltage input for a 4quadrant multiplying DAC application. VCC: Digital Supply Voltage. This is the power supply pin for the part. VCC can be from +5 to +15VDC. Operation is optimum for +15VDC GND: The pin 10 voltage must be at the same ground potential as IOUT1 and IOUT2 for current switching applications. Any difference of potential (VOS pin 10) will result in a linearity change of : (1) For example, if VREF = 10V and pin 10 is 9mV offset from IOUT1 and IOUT2 the linearity change will be 0.03%. Pin 3 can be offset ±100mV with no linearity change, but the logic input threshold will shift. Linearity Error Figure 3. a) End Point Test After Zero and fs adj. Figure 4. b) Best Straight Line Figure 5. c) Shifting fs adj. to Pass Best Straight Line Test Definition of Terms Resolution: Resolution is directly related to the number of switches or bits within the DAC. For example, the DAC0830 has 28 or 256 steps and therefore has 8-bit resolution. Linearity Error: Linearity Error is the maximum deviation from a straight line passing through the endpoints of the DAC transfer characteristic. It is measured after adjusting for zero and full-scale. Linearity error is a parameter intrinsic to the device and cannot be externally adjusted. 6 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 TI's linearity “end point test” (a) and the “best straight line” test (b,c) used by other suppliers are illustrated above. The “end point test'' greatly simplifies the adjustment procedure by eliminating the need for multiple iterations of checking the linearity and then adjusting full scale until the linearity is met. The “end point test'' ensures that linearity is met after a single full scale adjust. (One adjustment vs. multiple iterations of the adjustment.) The “end point test'' uses a standard zero and F.S. adjustment procedure and is a much more stringent test for DAC linearity. Power Supply Sensitivity: Power supply sensitivity is a measure of the effect of power supply changes on the DAC full-scale output. Settling Time: Settling time is the time required from a code transition until the DAC output reaches within ±½LSB of the final output value. Full-scale settling time requires a zero to full-scale or full-scale to zero output change. Full Scale Error: Full scale error is a measure of the output error between an ideal DAC and the actual device output. Ideally, for the DAC0830 series, full scale is VREF −1LSB. For VREF = 10V and unipolar operation, VFULLSCALE = 10,0000V–39mV 9.961V. Full-scale error is adjustable to zero. Differential Nonlinearity: The difference between any two consecutive codes in the transfer curve from the theoretical 1 LSB to differential nonlinearity. Monotonic: If the output of a DAC increases for increasing digital input code, then the DAC is monotonic. An 8bit DAC which is monotonic to 8 bits simply means that increasing digital input codes will produce an increasing analog output. Figure 6. DAC0830 Functional Diagram Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 7 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Typical Performance Characteristics 8 Digital Input Threshold vs. Temperature Digital Input Threshold vs. VCC Figure 7. Figure 8. Gain and Linearity Error Variation vs. Temperature Gain and Linearity Error Variation vs. Supply Voltage Figure 9. Figure 10. Write Pulse Width Data Hold Time Figure 11. Figure 12. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 DAC0830 SERIES APPLICATION HINTS These DAC's are the industry's first microprocessor compatible, double-buffered 8-bit multiplying D to A converters. Double-buffering allows the utmost application flexibility from a digital control point of view. This 20pin device is also pin for pin compatible (with one exception) with the DAC1230, a 12-bit MICRO-DAC. In the event that a system's analog output resolution and accuracy must be upgraded, substituting the DAC1230 can be easily accomplished. By tying address bit A0 to the ILE pin, a two-byte μP write instruction (double precision) which automatically increments the address for the second byte write (starting with A0=“1”) can be used. This allows either an 8-bit or the 12-bit part to be used with no hardware or software changes. For the simplest 8-bit application, this pin should be tied to VCC (also see other uses in Double-Buffered Operation). Analog signal control versatility is provided by a precision R-2R ladder network which allows full 4-quadrant multiplication of a wide range bipolar reference voltage by an applied digital word. DIGITAL CONSIDERATIONS A most unique characteristic of these DAC's is that the 8-bit digital input byte is double-buffered. This means that the data must transfer through two independently controlled 8-bit latching registers before being applied to the R2R ladder network to change the analog output. The addition of a second register allows two useful control features. First, any DAC in a system can simultaneously hold the current DAC data in one register (DAC register) and the next data word in the second register (input register) to allow fast updating of the DAC output on demand. Second, and probably more important, double-buffering allows any number of DAC's in a system to be updated to their new analog output levels simultaneously via a common strobe signal. The timing requirements and logic level convention of the register control signals have been designed to minimize or eliminate external interfacing logic when applied to most popular microprocessors and development systems. It is easy to think of these converters as 8-bit “write-only” memory locations that provide an analog output quantity. All inputs to these DAC's meet TTL voltage level specs and can also be driven directly with high voltage CMOS logic in non-microprocessor based systems. To prevent damage to the chip from static discharge, all unused digital inputs should be tied to VCC or ground. If any of the digital inputs are inadvertantly left floating, the DAC interprets the pin as a logic “1”. Double-Buffered Operation Updating the analog output of these DAC's in a double-buffered manner is basically a two step or double write operation. In a microprocessor system two unique system addresses must be decoded, one for the input latch controlled by the CS pin and a second for the DAC latch which is controlled by the XFER line. If more than one DAC is being driven, Figure 13, the CS line of each DAC would typically be decoded individually, but all of the converters could share a common XFER address to allow simultaneous updating of any number of DAC's. The timing for this operation is shown, Figure 14. It is important to note that the analog outputs that will change after a simultaneous transfer are those from the DAC's whose input register had been modified prior to the XFER command. Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 9 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com *TIE TO LOGIC 1 IF NOT NEEDED (SEE Double-Buffered Operation). Figure 13. Controlling Mutiple DACs Figure 14. The ILE pin is an active high chip select which can be decoded from the address bus as a qualifier for the normal CS signal generated during a write operation. This can be used to provide a higher degree of decoding unique control signals for a particular DAC, and thereby create a more efficient addressing scheme. Another useful application of the ILE pin of each DAC in a multiple DAC system is to tie these inputs together and use this as a control line that can effectively “freeze” the outputs of all the DAC's at their present value. Pulling this line low latches the input register and prevents new data from being written to the DAC. This can be particularly useful in multiprocessing systems to allow a processor other than the one controlling the DAC's to take over control of the data bus and control lines. If this second system were to use the same addresses as those decoded for DAC control (but for a different purpose) the ILE function would prevent the DAC's from being erroneously altered. In a “Stand-Alone” system the control signals are generated by discrete logic. In this case double-buffering can be controlled by simply taking CS and XFER to a logic “0”, ILE to a logic “1” and pulling WR1 low to load data to the input latch. Pulling WR2 low will then update the analog output. A logic “1” on either of these lines will prevent the changing of the analog output. 10 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 ILE=LOGIC “1”; WR2 and XFER GROUNDED Figure 15. Single-Buffered Operation In a microprocessor controlled system where maximum data throughput to the DAC is of primary concern, or when only one DAC of several needs to be updated at a time, a single-buffered configuration can be used. One of the two internal registers allows the data to flow through and the other register will serve as the data latch. Digital signal feedthrough (see Section Digital Signal Feedthrough) is minimized if the input register is used as the data latch. Timing for this mode is shown in Figure 15. Single-buffering in a “stand-alone” system is achieved by strobing WR1 low to update the DAC with CS, WR2 and XFER grounded and ILE tied high. Flow-Through Operation Though primarily designed to provide microprocessor interface compatibility, the MICRO-DAC's can easily be configured to allow the analog output to continuously reflect the state of an applied digital input. This is most useful in applications where the DAC is used in a continuous feedback control loop and is driven by a binary updown counter, or in function generation circuits where a ROM is continuously providing DAC data. Simply grounding CS, WR1, WR2, and XFER and tying ILE high allows both internal registers to follow the applied digital inputs (flow-through) and directly affect the DAC analog output. Control Signal Timing When interfacing these MICRO-DAC to any microprocessor, there are two important time relationships that must be considered to insure proper operation. The first is the minimum WR strobe pulse width which is specified as 900 ns for all valid operating conditions of supply voltage and ambient temperature, but typically a pulse width of only 180ns is adequate if VCC=15VDC. A second consideration is that the specified minimum data hold time of 50ns should be met or erroneous data can be latched. This hold time is defined as the length of time data must be held valid on the digital inputs after a qualified (via CS) WR strobe makes a low to high transition to latch the applied data. If the controlling device or system does not inherently meet these timing specs the DAC can be treated as a slow memory or peripheral and utilize a technique to extend the write strobe. A simple extension of the write time, by adding a wait state, can simultaneously hold the write strobe active and data valid on the bus to satisfy the minimum WR pulsewidth. If this does not provide a sufficient data hold time at the end of the write cycle, a negative edge triggered one-shot can be included between the system write strobe and the WR pin of the DAC. This is illustrated in Figure 16 for an exemplary system which provides a 250ns WR strobe time with a data hold time of less than 10ns. The proper data set-up time prior to the latching edge (LO to HI transition) of the WR strobe, is insured if the WR pulsewidth is within spec and the data is valid on the bus for the duration of the DAC WR strobe. Digital Signal Feedthrough When data is latched in the internal registers, but the digital inputs are changing state, a narrow spike of current may flow out of the current output terminals. This spike is caused by the rapid switching of internal logic gates that are responding to the input changes. Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 11 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com There are several recommendations to minimize this effect. When latching data in the DAC, always use the input register as the latch. Second, reducing the VCC supply for the DAC from +15V to +5V offers a factor of 5 improvement in the magnitude of the feedthrough, but at the expense of internal logic switching speed. Finally, increasing CC (Figure 19) to a value consistent with the actual circuit bandwidth requirements can provide a substantial damping effect on any output spikes. Figure 16. Accommodating a High Speed System ANALOG CONSIDERATIONS The fundamental purpose of any D to A converter is to provide an accurate analog output quantity which is representative of the applied digital word. In the case of the DAC0830, the output, IOUT1, is a current directly proportional to the product of the applied reference voltage and the digital input word. For application versatility, a second output, IOUT2, is provided as a current directly proportional to the complement of the digital input. Basically: where the digital input is the decimal (base 10) equivalent of the applied 8-bit binary word (0 to 255), VREF is the voltage at pin 8 and 15 kΩ is the nominal value of the internal resistance, R, of the R-2R ladder network (discussed in Section The Current Switching R-2R Ladder). Several factors external to the DAC itself must be considered to maintain analog accuracy and are covered in subsequent sections. The Current Switching R-2R Ladder The analog circuitry, Figure 17, consists of a silicon-chromium (SiCr or Si-chrome) thin film R-2R ladder which is deposited on the surface oxide of the monolithic chip. As a result, there are no parasitic diode problems with the ladder (as there may be with diffused resistors) so the reference voltage, VREF, can range −10V to +10V even if VCC for the device is 5VDC. The digital input code to the DAC simply controls the position of the SPDT current switches and steers the available ladder current to either IOUT1 or IOUT2 as determined by the logic input level (“1” or “0”) respectively, as shown in Figure 17. The MOS switches operate in the current mode with a small voltage drop across them and can therefore switch currents of either polarity. This is the basis for the 4-quadrant multiplying feature of this DAC. 12 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Basic Unipolar Output Voltage To maintain linearity of output current with changes in the applied digital code, it is important that the voltages at both of the current output pins be as near ground potential (0VDC) as possible. With VREF=+10V every millivolt appearing at either IOUT1 or IOUT2 will cause a 0.01% linearity error. In most applications this output current is converted to a voltage by using an op amp as shown in Figure 18. The inverting input of the op amp is a “virtual ground” created by the feedback from its output through the internal 15 kΩ resistor, Rfb. All of the output current (determined by the digital input and the reference voltage) will flow through Rfb to the output of the amplifier. Two-quadrant operation can be obtained by reversing the polarity of VREF thus causing IOUT1 to flow into the DAC and be sourced from the output of the amplifier. The output voltage, in either case, is always equal to IOUT1×Rfb and is the opposite polarity of the reference voltage. The reference can be either a stable DC voltage source or an AC signal anywhere in the range from −10V to +10V. The DAC can be thought of as a digitally controlled attenuator: the output voltage is always less than or equal to the applied reference voltage. The VREF terminal of the device presents a nominal impedance of 15 kΩ to ground to external circuitry. Always use the internal Rfb resistor to create an output voltage since this resistor matches (and tracks with temperature) the value of the resistors used to generate the output current (IOUT1). Figure 17. Figure 18. Op Amp Considerations The op amp used in Figure 18 should have offset voltage nulling capability (See Section Zero Adjustment). The selected op amp should have as low a value of input bias current as possible. The product of the bias current times the feedback resistance creates an output voltage error which can be significant in low reference voltage applications. BI-FET™ op amps are highly recommended for use with these DACs because of their very low input current. Transient response and settling time of the op amp are important in fast data throughput applications. The largest stability problem is the feedback pole created by the feedback resistance, Rfb, and the output capacitance of the DAC. This appears from the op amp output to the (−) input and includes the stray capacitance at this node. Addition of a lead capacitance, CC in Figure 19, greatly reduces overshoot and ringing at the output for a step change in DAC output current. Finally, the output voltage swing of the amplifier must be greater than VREF to allow reaching the full scale output voltage. Depending on the loading on the output of the amplifier and the available op amp supply voltages (only ±12 volts in many development systems), a reference voltage less than 10 volts may be necessary to obtain the full analog output voltage range. Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 13 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Bipolar Output Voltage with a Fixed Reference The addition of a second op amp to the previous circuitry can be used to generate a bipolar output voltage from a fixed reference voltage. This, in effect, gives sign significance to the MSB of the digital input word and allows two-quadrant multiplication of the reference voltage. The polarity of the reference can also be reversed to realize full 4-quadrant multiplication: ±VREF×±Digital Code=±VOUT. This circuit is shown in Figure 20. This configuration features several improvements over existing circuits for bipolar outputs with other multiplying DACs. Only the offset voltage of amplifier 1 has to be nulled to preserve linearity of the DAC. The offset voltage error of the second op amp (although a constant output voltage error) has no effect on linearity. It should be nulled only if absolute output accuracy is required. Finally, the values of the resistors around the second amplifier do not have to match the internal DAC resistors, they need only to match and temperature track each other. A thin film 4-resistor network available from Beckman Instruments, Inc. (part no. 694-3-R10K-D) is ideally suited for this application. These resistors are matched to 0.1% and exhibit only 5 ppm/°C resistance tracking temperature coefficient. Two of the four available 10 kΩ resistors can be paralleled to form R in Figure 20 and the other two can be used independently as the resistances labeled 2R. Zero Adjustment For accurate conversions, the input offset voltage of the output amplifier must always be nulled. Amplifier offset errors create an overall degradation of DAC linearity. The fundamental purpose of zeroing is to make the voltage appearing at the DAC outputs as near 0VDC as possible. This is accomplished for the typical DAC — op amp connection (Figure 18) by shorting out Rfb, the amplifier feedback resistor, and adjusting the VOS nulling potentiometer of the op amp until the output reads zero volts. This is done, of course, with an applied digital code of all zeros if IOUT1 is driving the op amp (all one's for IOUT2). The short around Rfb is then removed and the converter is zero adjusted. Figure 19. ts OP Amp CC (O to Full Scale) LF356 22 pF 4 μs LF351 22 pF 5 μs LF357 (1) 10 pF 2 μs (1) 2.4 kΩ RESISTOR ADDED FROM−INPUT TO GROUND TO INSURE STABILITY *THESE RESISTORS ARE AVAILABLE FROM BECKMAN INSTRUMENTS, INC. AS THEIR PART NO. 694-3-R10KD Figure 20. 14 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Input Code MSB IDEAL VOUT LSB 11 1 1 1 1 1 1 11 0 0 0 0 0 0 10 0 0 0 0 0 0 01 1 1 1 1 1 1 00 1 1 1 1 1 1 00 0 0 0 0 0 0 +VREF −VREF Full-Scale Adjustment In the case where the matching of Rfb to the R value of the R-2R ladder (typically ±0.2%) is insufficient for fullscale accuracy in a particular application, the VREF voltage can be adjusted or an external resistor and potentiometer can be added as shown in Figure 21 to provide a full-scale adjustment. The temperature coefficients of the resistors used for this adjustment are of an important concern. To prevent degradation of the gain error temperature coefficient by the external resistors, their temperature coefficients ideally would have to match that of the internal DAC resistors, which is a highly impractical constraint. For the values shown in Figure 21, if the resistor and the potentiometer each had a temperature coefficient of ±100 ppm/°C maximum, the overall gain error temperature coefficent would be degraded a maximum of 0.0025%/°C for an adjustment pot setting of less than 3% of Rfb. Using the DAC0830 in a Voltage Switching Configuration The R-2R ladder can also be operated as a voltage switching network. In this mode the ladder is used in an inverted manner from the standard current switching configuration. The reference voltage is connected to one of the current output terminals (IOUT1 for true binary digital control, IOUT2 is for complementary binary) and the output voltage is taken from the normal VREF pin. The converter output is now a voltage in the range from 0V to 255/256 VREF as a function of the applied digital code as shown in Figure 22. Figure 21. Adding Full-Scale Adjustment Figure 22. Voltage Mode Switching Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 15 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com This configuration offers several useful application advantages. Since the output is a voltage, an external op amp is not necessarily required but the output impedance of the DAC is fairly high (equal to the specified reference input resistance of 10 kΩ to 20 kΩ) so an op amp may be used for buffering purposes. Some of the advantages of this mode are illustrated in Figure 23, Figure 24, Figure 25, and Figure 26. There are two important things to keep in mind when using this DAC in the voltage switching mode. The applied reference voltage must be positive since there are internal parasitic diodes from ground to the IOUT1 and IOUT2 terminals which would turn on if the applied reference went negative. There is also a dependence of conversion linearity and gain error on the voltage difference between VCC and the voltage applied to the normal current output terminals. This is a result of the voltage drive requirements of the ladder switches. To ensure that all 8 switches turn on sufficiently (so as not to add significant resistance to any leg of the ladder and thereby introduce additional linearity and gain errors) it is recommended that the applied reference voltage be kept less than +5VDC and VCC be at least 9V more positive than VREF. These restrictions ensure less than 0.1% linearity and gain error change. Figure 27, Figure 28, and Figure 29 characterize the effects of bringing VREF and VCC closer together as well as typical temperature performance of this voltage switching configuration. • Voltage switching mode eliminates output signal inversion and therefore a need for a negative power supply. • Zero code output voltage is limited by the low level output saturation voltage of the op amp. The 2 kΩ pull-down resistor helps to reduce this voltage. • VOS of the op amp has no effect on DAC linearity. Figure 23. Single Supply DAC Figure 24. Obtaining a Bipolar Output from a Fixed Reference with a Single Op Amp 16 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Figure 25. Bipolar Output with Increased Output Voltage Swing Figure 26. Single Supply DAC with Level Shift and Span- Adjustable Output Gain and Linearity Error Variation vs. Supply Voltage Gain and Linearity Error Variation vs. Reference Voltage Note: For these curves, VREF is the voltage applied to pin 11 (IOUT1) with pin 12 (IOUT2) grounded. Figure 27. Figure 28. Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 17 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Gain and Linearity Error Variation vs. Temperature Figure 29. Miscellaneous Application Hints These converters are CMOS products and reasonable care should be exercised in handling them to prevent catastrophic failures due to static discharge. Conversion accuracy is only as good as the applied reference voltage so providing a stable source over time and temperature changes is an important factor to consider. A “good” ground is most desirable. A single point ground distribution technique for analog signals and supply returns keeps other devices in a system from affecting the output of the DACs. During power-up supply voltage sequencing, the −15V (or −12V) supply of the op amp may appear first. This will cause the output of the op amp to bias near the negative supply potential. No harm is done to the DAC, however, as the on-chip 15 kΩ feedback resistor sufficiently limits the current flow from IOUT1 when this lead is internally clamped to one diode drop below ground. Careful circuit construction with minimization of lead lengths around the analog circuitry, is a primary concern. Good high frequency supply decoupling will aid in preventing inadvertant noise from appearing on the analog output. Overall noise reduction and reference stability is of particular concern when using the higher accuracy versions, the DAC0830 and DAC0831, or their advantages are wasted. GENERAL APPLICATION IDEAS The connections for the control pins of the digital input registers are purposely omitted. Any of the control formats discussed in Digital Considerations of the accompanying text will work with any of the circuits shown. The method used depends on the overall system provisions and requirements. The digital input code is referred to as D and represents the decimal equivalent value of the 8-bit binary input, for example: Binary Input 18 D Pin 13 Pin 7 Decimal MSB LSB Equivalent 1 1 1 1 1 1 1 1 255 1 0 0 0 0 0 0 0 128 0 0 0 1 0 0 0 0 16 0 0 0 0 0 0 1 0 2 0 0 0 0 0 0 0 0 0 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Applications Figure 30. DAC Controlled Amplifier (Volume Control) Figure 31. Capacitance Multiplier Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 19 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com Figure 32. Variable fO, Variable QO, Constant BW Bandpass Filter Figure 33. DAC Controlled Function Generator 20 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 DAC0830, DAC0832 www.ti.com SNAS534B – MAY 1999 – REVISED MARCH 2013 Figure 34. Two Terminal Floating 4 to 20 mA Current Loop Controller • • • • DAC0830 linearly controls the current flow from the input terminal to the output terminal to be 4 mA (for D=0) to 19.94 mA (for D=255). Circuit operates with a terminal voltage differential of 16V to 55V. P2 adjusts the magnitude of the output current and P1 adjusts the zero to full scale range of output current. Digital inputs can be supplied from a processor using opto isolators on each input or the DAC latches can flow-through (connect control lines to pins 3 and 10 of the DAC) and the input data can be set by SPST toggle switches to ground (pins 3 and 10). Figure 35. DAC Controlled Exponential Time Response • • • Output responds exponentially to input changes and automatically stops when VOUT=VIN Output time constant is directly proportional to the DAC input code and capacitor C Input voltage must be positive (See Using the DAC0830 in a Voltage Switching Configuration Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 Submit Documentation Feedback 21 DAC0830, DAC0832 SNAS534B – MAY 1999 – REVISED MARCH 2013 www.ti.com REVISION HISTORY Changes from Revision A (March 2013) to Revision B • 22 Page Changed layout of National Data Sheet to TI format .......................................................................................................... 21 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: DAC0830 DAC0832 PACKAGE OPTION ADDENDUM www.ti.com 22-Dec-2017 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (°C) Device Marking (4/5) DAC0830LCN LIFEBUY PDIP NFH 20 18 TBD Call TI Call TI 0 to 70 DAC0830LCN DAC0830LCN/NOPB LIFEBUY PDIP NFH 20 18 Pb-Free (RoHS) CU SN Level-1-NA-UNLIM 0 to 70 DAC0830LCN DAC0832LCN LIFEBUY PDIP NFH 20 18 TBD Call TI Call TI 0 to 70 DAC0832LCN DAC0832LCN/NOPB LIFEBUY PDIP NFH 20 18 Green (RoHS & no Sb/Br) CU SN Level-1-NA-UNLIM 0 to 70 DAC0832LCN DAC0832LCWM LIFEBUY SOIC DW 20 36 TBD Call TI Call TI 0 to 70 DAC0832 LCWM DAC0832LCWM/NOPB ACTIVE SOIC DW 20 36 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR 0 to 70 DAC0832 LCWM DAC0832LCWMX LIFEBUY SOIC DW 20 1000 TBD Call TI Call TI 0 to 70 DAC0832 LCWM DAC0832LCWMX/NOPB ACTIVE SOIC DW 20 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR 0 to 70 DAC0832 LCWM (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of