INA225-Q1
SBOS728A – FEBRUARY 2015 – REVISED MARCH 2021
INA225-Q1 AEC-Q100, 36-V, Bidirectional Current Sense Amplifier With Four PinSelectable Gain Settings
1 Features
3 Description
•
The INA225-Q1 is a voltage-output, current-sense
amplifier that senses drops across current-sensing
resistors at common-mode voltages that vary from 0
V to 36 V, independent of the supply voltage. The
device is a bidirectional, current-shunt monitor that
allows an external reference to be used to measure
current flowing in both directions across a currentsensing resistor.
•
•
•
•
•
•
•
•
•
AEC-Q100 qualified:
– Temperature grade 1: –40 °C to +125 °C
– HBM ESD classification 2
– CDM ESD classification C4B
Functional Safety-Capable
– Documentation available to aid functional safety
system design
Wide common-mode range: 0 V to 36 V
Offset voltage: ±150 μV (maximum, all gains)
Offset voltage drift: 0.5 μV/°C (maximum)
Gain accuracy, over temperature (maximum):
– 25 V/V, 50 V/V: ±0.15%
– 100 V/V: ±0.2%
– 200 V/V: ±0.3%
– 10-ppm/°C gain drift
Bandwidth: 250 kHz (gain = 25 V/V)
Programmable gains:
– G1 = 25 V/V
– G2 = 50 V/V
– G3 = 100 V/V
– G4 = 200 V/V
Quiescent current: 350 μA (maximum)
Package: 8-pin VSSOP
2 Applications
•
•
•
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•
Automotive lighting
Body control module
Motor control
Valve control
Cluster
Central control module
Four discrete gain levels are selectable using the
two gain-select terminals (GS0 and GS1) to program
gains of 25 V/V, 50 V/V, 100 V/V, and 200 V/V. The
low-offset, zero-drift architecture and precision gain
values enable current-sensing with maximum drops
across the shunt as low as 10 mV of full-scale, while
maintaining very high accuracy measurements over
the entire operating temperature range.
The device operates from a single +2.7-V to +36V power supply, drawing a maximum of 350 μA
of supply current. The device is specified over the
extended operating temperature range of –40 °C to
+125 °C, and is offered in a VSSOP-8 package.
Device Information(1)
PART NUMBER
PACKAGE
INA225-Q1
(1)
VSSOP (8)
BODY SIZE (NOM)
3.00 mm x 3.00 mm
For all available packages, see the package option
addendum at the end of the data sheet.
RSHUNT
5-V Supply
Load
CBYPASS
0.1µF
VS
INA225
IN-
OUT
ADC
Microcontroller
+
IN+
GPIO
REF
GAIN SELECT
GS0
GS0
GS1
GAIN
GND
GND
VS
VS
GND
VS
GND
VS
25
50
100
200
GS1
GND
Typical Application
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
INA225-Q1
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SBOS728A – FEBRUARY 2015 – REVISED MARCH 2021
Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
6 Specifications.................................................................. 4
6.1 Absolute Maximum Ratings(1) .................................... 4
6.2 ESD Ratings............................................................... 4
6.3 Recommended Operating Conditions.........................4
6.4 Thermal Information....................................................4
6.5 Electrical Characteristics.............................................5
6.6 Typical Characteristics................................................ 7
7 Detailed Description......................................................13
7.1 Overview................................................................... 13
7.2 Functional Block Diagram......................................... 13
7.3 Feature Description...................................................13
7.4 Device Functional Modes..........................................16
8 Applications and Implementation................................ 19
8.1 Application Information............................................. 19
8.2 Typical Applications.................................................. 19
9 Power Supply Recommendations................................25
10 Layout...........................................................................25
10.1 Layout Guidelines................................................... 25
10.2 Layout Example...................................................... 25
11 Device and Documentation Support..........................26
11.1 Documentation Support.......................................... 26
11.2 Receiving Notification of Documentation Updates.. 26
11.3 Support Resources................................................. 26
11.4 Trademarks............................................................. 26
11.5 Electrostatic Discharge Caution.............................. 26
11.6 Glossary.................................................................. 26
12 Mechanical, Packaging, and Orderable
Information.................................................................... 26
4 Revision History
Changes from Revision * (February 2015) to Revision A (March 2021)
Page
• Updated the numbering format for tables, figures, and cross-references throughout the document..................1
• Added Functional Safety bullets to the Features ............................................................................................... 1
• Added title to key graphic................................................................................................................................... 1
• Added 25 kΩ value to RINT in Input Filtering ....................................................................................................16
2
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5 Pin Configuration and Functions
IN+
1
8
IN-
GND
2
7
REF
VS
3
6
GS1
OUT
4
5
GS0
Figure 5-1. DGK Package VSSOP-8 (Top View)
Table 5-1. Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
1
IN+
Analog input
2
GND
Analog
Connect to supply side of shunt resistor.
Ground
Power supply, 2.7 V to 36 V
3
VS
Analog
4
OUT
Analog output
5
GS0
Digital input
Gain select. Connect to VS or GND.
Table 7-3 lists terminal settings and the corresponding gain value.
6
GS1
Digital input
Gain select. Connect to VS or GND.
Table 7-3 lists terminal settings and the corresponding gain value.
7
REF
Analog input
Reference voltage, 0 V to VS
8
IN–
Analog input
Connect to load side of shunt resistor.
Output voltage
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6 Specifications
6.1 Absolute Maximum Ratings(1)
Over operating free-air temperature range, unless otherwise noted.
MIN
Supply voltage
UNIT
+40
V
–40
+40
V
GND – 0.3
+40
V
REF, GS0, and GS1 inputs
GND – 0.3
(VS) + 0.3
V
Output
GND – 0.3
(VS) + 0.3
V
–55
+150
°C
+150
°C
+150
°C
Analog inputs, VIN+, VIN– (2)
Differential (VIN+) – (VIN–)
MAX
Common-mode(3)
Operating, TA
Junction, TJ
Temperature
Storage, Tstg
(1)
(2)
(3)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress
ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
VIN+ and VIN– are the voltages at the IN+ and IN– terminals, respectively.
Input voltage at any terminal may exceed the voltage shown if the current at that terminal is limited to 5 mA.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per AEC Q100-002(1)
±2500
Charged-device model (CDM), per AEC Q100-011
±1000
UNIT
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
Over operating free-air temperature range, unless otherwise noted.
MIN
VCM
Common-mode input voltage
VS
Operating supply voltage
TA
Operating free-air temperature
NOM
MAX
UNIT
12
V
5
V
–40
+125
°C
6.4 Thermal Information
INA225-Q1
THERMAL METRIC(1)
DGK (VSSOP)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
163.6
RθJC(top)
Junction-to-case (top) thermal resistance
57.7
RθJB
Junction-to-board thermal resistance
84.7
ψJT
Junction-to-top characterization parameter
6.5
ψJB
Junction-to-board characterization parameter
83.2
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
(1)
4
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
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6.5 Electrical Characteristics
At TA = +25 °C, VSENSE = VIN+ – VIN–, VS = +5 V, VIN+ = 12 V, and VREF = VS / 2, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
INPUT
VCM
CMR
Common-mode input range
TA = –40 °C to +125 °C
Common-mode rejection
VIN+ = 0 V to +36 V, VSENSE = 0 mV,
TA = –40 °C to +125 °C
RTI(1)
0
95
36
105
V
dB
VOS
Offset voltage,
VSENSE = 0 mV
±75
±150
dVOS/dT
RTI vs. temperature
TA = –40 °C to +125 °C
0.2
0.5
μV/°C
PSRR
Power-supply rejection ratio
VSENSE = 0 mV, VREF = 2.5 V,
VS = 2.7 V to 36 V
±0.1
±1
μV/V
IB
Input bias current
VSENSE = 0 mV
72
85
μA
IOS
Input offset current
VSENSE = 0 mV
VREF
Reference input range
TA = –40 °C to +125 °C
55
±0.5
0
μV
μA
VS
V
OUTPUT
G
EG
Gain
Gain error
Gain error vs. temperature
25, 50, 100, 200
V/V
Gain = 25 V/V and 50 V/V, VOUT = 0.5 V to
VS – 0.5 V, TA = –40 °C to +125 °C
±0.05%
±0.15%
Gain = 100 V/V, VOUT = 0.5 V to VS – 0.5 V,
TA = –40 °C to +125 °C
±0.1%
±0.2%
Gain = 200 V/V, VOUT = 0.5 V to VS – 0.5 V,
TA = –40 °C to +125 °C
±0.1%
±0.3%
G = 25 V/V, 50 V/V, 100 V/V,
TA = –40 °C to +125 °C
3
10
G = 200 V/V, TA = –40 °C to +125 °C
5
15
Nonlinearity error
VOUT = 0.5 V to VS – 0.5 V
Maximum capacitive load
No sustained oscillation
ppm/°C
±0.01%
1
nF
VOLTAGE OUTPUT(2)
Swing to VS power-supply rail RL = 10 kΩ to GND, TA = –40 °C to +125 °C
VS – 0.05
VS – 0.2
VREF = VS / 2, all gains, RL = 10 kΩ to GND,
TA = –40 °C to +125 °C
VGND + 5
VGND + 10
VREF = GND, gain = 25 V/V, RL = 10 kΩ to GND,
TA = –40 °C to +125 °C
VGND + 7
mV
VREF = GND, gain = 50 V/V, RL = 10 kΩ to GND,
TA = –40 °C to +125 °C
VGND + 15
mV
VREF = GND, gain = 100 V/V, RL = 10 kΩ to GND,
TA = –40 °C to +125 °C
VGND + 30
mV
VREF = GND, gain = 200 V/V, RL = 10 kΩ to GND,
TA = –40 °C to +125 °C
VGND + 60
mV
Gain = 25 V/V, CLOAD = 10 pF
250
kHz
Gain = 50 V/V, CLOAD = 10 pF
200
kHz
Gain = 100 V/V, CLOAD = 10 pF
125
kHz
Swing to GND(3)
V
mV
FREQUENCY RESPONSE
BW
Bandwidth
SR
Slew rate
Gain = 200 V/V, CLOAD = 10 pF
70
kHz
0.4
V/μs
50
nV/√ Hz
NOISE, RTI(1)
Voltage noise density
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At TA = +25 °C, VSENSE = VIN+ – VIN–, VS = +5 V, VIN+ = 12 V, and VREF = VS / 2, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
DIGITAL INPUT
Ci
Input capacitance
Leakage input current
3
0 ≤ VIN ≤ VS
1
pF
2
μA
VIL
Low-level input logic level
0
0.6
V
VIH
High-level input logic level
2
VS
V
POWER SUPPLY
VS
Operating voltage range
TA = –40 °C to +125 °C
IQ
Quiescent current
VSENSE = 0 mV
IQ over temperature
TA = –40 °C to +125 °C
+2.7
300
+36
V
350
μA
375
μA
TEMPERATURE RANGE
(1)
(2)
(3)
6
Specified range
–40
+125
°C
Operating range
–55
+150
°C
RTI = referred-to-input.
See Typical Characteristic curve, Output Voltage Swing vs. Output Current (Figure 6-10).
See Typical Characteristic curve, Unidirectional Output Voltage Swing vs. Temperature (Figure 6-14).
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6.6 Typical Characteristics
At TA = +25 °C, VS = +5 V, VIN+ = 12 V, and VREF = VS / 2, unless otherwise noted.
175
Poplulation
Offset Voltage (µV)
150
125
100
75
50
200
175
150
125
75
100
50
0
25
-25
-50
-75
-100
-125
-150
-175
-200
25
0
±50
Offset Voltage (µV)
0
±25
25
Figure 6-1. Input Offset Voltage Production
Distribution
50
75
100
125
150
Temperature (ƒC)
C001
C002
Figure 6-2. Input Offset Voltage vs. Temperature
8
Population
CMRR (µV/V)
7
6
5
4
20
18
16
14
12
8
10
6
4
2
0
-2
-4
-6
-8
-10
3
2
±50
Common-Mode Rejection Ratio (µV/V)
0
±25
25
75
100
125
150
C004
Gain Error (%)
0.1
0.08
0.06
0.04
0
0.02
-0.02
-0.04
-0.06
-0.1
-0.08
-0.12
-0.14
-0.16
-0.18
Population
Figure 6-4. Common-Mode Rejection Ratio vs.
Temperature
-0.2
0.1
0.08
0.06
0.04
0.02
0
-0.02
-0.04
-0.06
-0.08
-0.1
-0.12
-0.14
-0.16
-0.18
-0.2
Population
Figure 6-3. Common-Mode Rejection Production
Distribution
50
Temperature (ƒC)
C003
Gain Error (%)
C005
Figure 6-5. Gain Error Production Distribution
(Gain = 25 V/V)
C006
Figure 6-6. Gain Error Production Distribution
(Gain = 50 V/V)
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Figure 6-7. Gain Error Production Distribution
(Gain = 100 V/V)
0.5
0.3
Figure 6-8. Gain Error Production Distribution
(Gain = 200 V/V)
50
45
40
0.2
0.1
Gain (dB)
Gain Error (%)
C008
25 V/V
50 V/V
100 V/V
200 V/V
0.4
0.0
-0.1
-0.2
35
30
25
200 V/V
100 V/V
50 V/V
25 V/V
-0.3
20
-0.4
-0.5
15
±50
0
±25
25
50
75
100
125
Temperature (ƒC)
150
1
10
100
C009
VCM = 0 V
1k
10k
Frequency (Hz)
100k
1M
C010
VSENSE = 15 mVPP
Figure 6-10. Gain vs. Frequency
Figure 6-9. Gain Error vs. Temperature
140
120
120
100
100
80
CMR (dB)
PSR (dB)
0.1
Gain Error (%)
C007
80
60
60
40
40
20
20
0
0
10
100
1,000
10,000
Frequency (Hz)
100,000
1,000,000
10
Figure 6-11. Power-Supply Rejection Ratio vs.
Frequency
100
1,000
10,000
Frequency (Hz)
C011
VCM = 0 V
VREF = 2.5 V
VSENSE = 0 mV, Shorted
VS = 5 V + 250-mV Sine Disturbance
8
0.08
0.06
0.04
0
0.02
-0.02
-0.04
-0.06
-0.1
Gain Error (%)
-0.08
-0.12
-0.14
-0.16
-0.18
Population
-0.2
0.1
0.08
0.06
0.04
0
0.02
-0.02
-0.04
-0.06
-0.1
-0.08
-0.12
-0.14
-0.16
-0.18
-0.2
Population
SBOS728A – FEBRUARY 2015 – REVISED MARCH 2021
VS = 5 V
VREF = 2.5 V
VCM = 1-V Sine Wave
100,000
1,000,000
C012
VSENSE = 0 mV, Shorted
Figure 6-12. Common-Mode Rejection Ratio vs.
Frequency
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100
Vs
90
Unidirectional, G = 200
80
Output Voltage (mV)
Output Voltage Swing (V)
(Vs) -1
(Vs) -2
(Vs) -3
GND +3
GND +2
- 40ƒC
25ƒC
125ƒC
GND +1
GND
0
2
4
6
8
10 12
Current (mA)
14
16
18
70
Unidirectional, G = 100
60
50
Unidirectional, G = 50
40
Unidirectional, G = 25
30
Bidirectional, All Gains
20
10
20
0
C013
±50
±25
0
25
50
75
100
125
Temperature (ƒC)
Unidirectional, REF = GND
Figure 6-13. Output Voltage Swing vs Output
Current
Bidirectional, REF > GND
80
70
120
IB+, IB-, VREF = 0V
Input Bias Current (µA)
Input Bias Current (µA)
C038
Figure 6-14. Unidirectional Output Voltage Swing
vs. Temperature
140
100
80
60
IB+, IB-, VREF = 2.5V
40
20
60
50
40
30
IB+, IB-, VREF=0V
20
10
0
±20
0
150
0
5
10
15
20
25
30
35
Common-Mode Voltage (V)
±10
0
40
5
10
Figure 6-15. Input Bias Current vs. Common-Mode
Voltage (Supply Voltage = +5 V)
15
20
25
30
35
Common-Mode Voltage (V)
C014
40
C015
Figure 6-16. Input Bias Current vs. Common-Mode
Voltage (Supply Voltage = 0 V, Shutdown)
85
550
80
500
VS = 5V
450
VS = 2.7V
Quiescent Current (µA)
Input Bias Current (µA)
VS = 36V
75
70
65
60
400
350
300
250
200
55
±50
±25
0
25
50
75
Temperature (ƒC)
VS = 5 V
100
125
150
±50
±25
0
25
50
75
Temperature (ƒC)
C016
100
125
150
C017
VCM = 12 V
Figure 6-17. Input Bias Current vs. Temperature
Figure 6-18. Quiescent Current vs. Temperature
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100
400
Gain = 100 V/V
375
Input-Referred
Voltage Noise (nV/¥+]
Quiescent Current (µA)
Gain = 200 V/V
350
325
300
275
200 V/V
250
100 V/V
Gain = 50 V/V
Gain = 25 V/V
50 V/V
225
25 V/V
200
0
10
5
10
15
20
25
30
35
Supply Voltage (V)
40
1
10
100
1k
10k
Frequency (Hz)
C018
VREF = 0 V
VSENSE = 0 mV, Shorted
Output
(0.5 V/div)
Referred-to-Input
Voltage Noise (200 nV/div)
Input
(25 mV/div)
Figure 6-20. Input-Referred Voltage Noise vs.
Frequency
Time (1 s/div)
Time (25 µs/div)
C020
VS = ± 2.5 V
VCM = 0 V
C021
VSENSE = 0 mV, Shorted
Figure 6-21. 0.1-Hz to 10-Hz Voltage Noise
(Referred-to-Input)
Input
(10 mV/div)
Output
(1 V/div)
Output
(1 V/div)
Input
(20 mV/div)
Figure 6-22. Step Response (Gain = 25 V/V, 2-VPP
Output Step)
Time (25 µs/div)
Time (25 µs/div)
C022
Figure 6-23. Step Response (Gain = 50 V/V, 2-VPP
Output Step)
10
1M
C019
VS = ± 2.5 V
Figure 6-19. Quiescent Current vs. Supply Voltage
100k
C023
Figure 6-24. Step Response (Gain = 100 V/V, 2-VPP
Output Step)
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Output
(1 V/div)
Output
(250 mV/div
Input
(5 mV/div)
Gain Select (GS1)
(2 V/div)
SBOS728A – FEBRUARY 2015 – REVISED MARCH 2021
Time (25 µs/div)
Time (5 µs/div)
C024
C025
VDIFF = 20 mV
VOUT at 50-V/V Gain = 1 V
Figure 6-26. Gain Change Output Response (Gain
= 25 V/V to 50 V/V)
Output
(1 V/div)
Output
(500 mV/div)
Gain Select (GS0)
(2 V/div)
Gain Select (GS0)
(2 V/div)
Figure 6-25. Step Response (Gain = 200 V/V, 2-VPP
Output Step)
VOUT at 25-V/V Gain = 500 mV
Time (5 µs/div)
Time (5 µs/div)
C026
VDIFF = 20 mV
VOUT at 100-V/V Gain = 2 V
VOUT at 25-V/V Gain = 500 mV
VOUT at 50-V/V Gain = 1 V
Figure 6-28. Gain Change Output Response (Gain
= 50 V/V to 200 V/V)
Output
(1 V/div)
Output
(0.5 V/div)
Gain Select (GS1)
(2 V/div)
Gain Select (GS1)
(2 V/div)
Figure 6-27. Gain Change Output Response (Gain
= 25 V/V to 100 V/V)
C027
VDIFF = 20 mV
VOUT at 200-V/V Gain = 4 V
Time (25 µs/div)
Time (5 µs/div)
C029
C028
VDIFF = 20 mV
VOUT at 200-V/V Gain = 4 V
VOUT at 100-V/V Gain = 2 V
Figure 6-29. Gain Change Output Response (Gain
= 100 V/V to 200 V/V)
Figure 6-30. Gain Change Output Response From
Saturation (Gain = 50 V/V to 25 V/V)
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Gain Select (GS0)
(2 V/div)
Output
(0.5 V/div)
Output
(0.5 V/div)
Gain Select (GS0)
(2 V/div)
SBOS728A – FEBRUARY 2015 – REVISED MARCH 2021
Time (25 µs/div)
Time (25 µs/div)
C030
C031
Figure 6-32. Gain Change Output Response From
Saturation (Gain = 200 V/V to 50 V/V)
Common-Mode
Voltage (10 V/div)
Gain Select (GS1)
(2 V/div)
Figure 6-31. Gain Change Output Response From
Saturation (Gain = 100 V/V to 25 V/V)
Output
(0.5 V/div )
Output
(1 V/div)
Gain = 25 V/V
Gain = 100 V/V
Gain = 200 V/V
Gain = 50 V/V
Time (25 µs/div)
Time (5 µs/div)
C032
C033
Figure 6-34. Common-Mode Voltage Transient
Response
Supply Voltage
(2.5 V/div)
Output
(0.5 V/div)
Figure 6-33. Gain Change Output Response From
Saturation (Gain = 200 V/V to 100 V/V)
Time (25 µs/div)
C034
Figure 6-35. Start-Up Response
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7 Detailed Description
7.1 Overview
The INA225-Q1 is a 36-V, common-mode, zero-drift topology, current-sensing amplifier. This device features
a significantly higher signal bandwidth than most comparable precision, current-sensing amplifiers, reaching
up to 125 kHz at a gain of 100 V/V. A very useful feature present in the device is the built-in programmable
gain selection. To increase design flexibility with the device, a programmable gain feature is added that allows
changing device gain during operation in order to accurately monitor wider dynamic input signal ranges. Four
discrete gain levels (25 V/V, 50 V/V, 100 V/V, and 200 V/V) are available in the device and are selected using the
two gain-select terminals, GS0 and GS1.
7.2 Functional Block Diagram
VS
INA225
-
IN-
OUT
+
IN+
REF
Gain Select
GS0
GS1
GND
7.3 Feature Description
7.3.1 Selecting A Shunt Resistor
The device measures the differential voltage developed across a resistor when current flows through it. This
resistor is commonly referred to as a current-sensing resistor or a current-shunt resistor, with each term
commonly used interchangeably. The flexible design of the device allows a wide range of input signals to be
measured across this current-sensing resistor.
Selecting the value of this current-sensing resistor is based primarily on two factors: the required accuracy of the
current measurement and the allowable power dissipation across the resistor. The larger the voltage developed
across this resistor the more accurate of a measurement that can be made because of the fixed internal amplifier
errors. These fixed internal amplifier errors, which are dominated by the internal offset voltage of the device,
result in a larger measurement uncertainty when the input signal gets smaller. When the input signal gets larger,
the measurement uncertainty is reduced because the fixed errors become a smaller percentage of the signal
being measured.
A system design trade-off for improving the measurement accuracy through the use of the larger input signals
is the increase in the power dissipated across the current-sensing resistor. Increasing the value of the currentshunt resistor increases the differential voltage developed across the resistor when current passes through it.
However, the power that is then dissipated across this component also increases. Decreasing the value of
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the current-shunt resistor value reduces the power dissipation requirements of the resistor, but increases the
measurement errors resulting from the decreasing input signal. Finding the optimal value for the shunt resistor
requires factoring both the accuracy requirement of the application and allowable power dissipation into the
selection of the component. An increasing amount of very low ohmic value resistors are becoming available with
values reaching down to 200 μΩ with power dissipations of up to 5 W, thus enabling very large currents to be
accurately monitored using sensing resistors.
The maximum value for the current-sensing resistor that can be chosen is based on the full-scale current to
be measured, the full-scale input range of the circuitry following the device, and the device gain selected.
The minimum value for the current-sensing resistor is typically a design-based decision because maximizing
the input range of the circuitry following the device is commonly preferred. Full-scale output signals that are
significantly less than the full input range of the circuitry following the device output can limit the ability of the
system to exercise the full dynamic range of system control based on the current measurement.
7.3.1.1 Selecting A Current-Sense Resistor Example
The example in Table 7-1 is based on a set of application characteristics, including a 10-A full-scale current
range and a 4-V full-scale output requirement. The calculations for selecting a current-sensing resistor of an
appropriate value are shown in Table 7-1.
Table 7-1. Calculating the Current-Sense Resistor, RSENSE
PARAMETER
IMAX
Full-scale current
VOUT
Full-scale output voltage
EQUATION
RESULT
10 A
4V
Gain
Gain selected
VDIFF
Ideal maximum differential input voltage
RSHUNT
Shunt resistor value
PRSENSE
Current-sense resistor power dissipation
VOS Error
Offset voltage error
Initial selection based on default
gain setting.
25 V/V
VDiff = VOUT / Gain
160 mV
RSHUNT = VDiff / IMAX
16 mΩ
RSENSE x IMAX 2
1.6 W
(VOS / VDIFF ) x 100
0.094%
7.3.1.2 Optimizing Power Dissipation versus Measurement Accuracy
The example shown in Table 7-1 results in a maximum current-sensing resistor value of 16 mΩ to develop the
160 mV required to achieve the 4-V full-scale output with the gain set to 25 V/V. The power dissipated across
this 16-mΩ resistor at the 10-A current level is 1.6 W, which is a fairly high power dissipation for this component.
Adjusting the device gain allows alternate current-sense resistor values to be selected to ease the power
dissipation requirement of this component.
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Changing the gain setting from 25 V/V to 100 V/V, as shown in Table 7-2, decreases the maximum differential
input voltage from 160 mV down to 40 mV, thus requiring only a 4-mΩ current-sensing resistor to achieve the
4-V output at the 10-A current level. The power dissipated across this resistor at the 10-A current level is
400 mW, significantly increasing the availability of component options to select from.
The increase in gain by a factor of four reduces the power dissipation requirement of the current-sensing resistor
by this same factor of four. However, with this smaller full-scale signal, the measurement uncertainty resulting
from the device fixed input offset voltage increases by the same factor of four. The measurement error resulting
from the device input offset voltage is approximately 0.1% at the 160-mV full-scale input signal for the 25-V/V
gain setting. Increasing the gain to 100 V/V and decreasing the full-scale input signal to 40 mV increases the
offset induced measurement error to 0.38%.
Table 7-2. Accuracy and RSENSE Power Dissipation vs. Gain Setting
PARAMETER
EQUATION
IMAX
Full-scale current
VOUT
Full-scale output voltage
Gain
Gain selected
VDIFF
Ideal maximum differential input voltage
RSENSE
Current-sense resistor value
RESULT
10 A
4V
100 V/V
PRSENSE
Current-sense resistor power dissipation
VOS Error
Offset voltage error
VDiff = VOUT / Gain
40 mV
RSENSE = VDiff / IMAX
4 mΩ
RSENSE x IMAX
2
0.4 W
(VOS / VDIFF ) x 100
0.375%
7.3.2 Programmable Gain Select
The device features a terminal-controlled gain selection in determining the device gain setting. Four discrete
gain options are available (25 V/V, 50 V/V, 100 V/V, and 200 V/V) on the device and are selected based on the
voltage levels applied to the gain-select terminals (GS0 and GS1). These terminals are typically fixed settings
for most applications but the programmable gain feature can be used to adjust the gain setting to enable wider
dynamic input range monitoring as well as to create an automatic gain control (AGC) network.
Table 7-3 shows the corresponding gain values and gain-select terminal values for the device.
Table 7-3. Gain Select Settings
GAIN
GS0
GS1
25 V/V
GND
GND
50 V/V
GND
VS
100 V/V
VS
GND
200 V/V
VS
VS
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7.4 Device Functional Modes
7.4.1 Input Filtering
An obvious and straightforward location for filtering is at the device output; however, this location negates the
advantage of the low output impedance of the internal buffer. The input then represents the best location for
implementing external filtering. Figure 7-1 shows the typical implementation of the input filter for the device.
RSHUNT
5-V Supply
Power
Supply
Load
RS
” 10
CBYPASS
0.1 µF
RS
” 10
VS
Device
¦-3dB =
1
CF
2ŒRSCF
RINT
¦-3dB
OUT
BIAS
Output
+
RINT
REF
GS0
GS1
GND
Figure 7-1. Input Filter
Care must be taken in the selection of the external filter component values because these components can
affect device measurement accuracy. Placing external resistance in series with the input terminals creates an
additional error so these resistors should be kept as low of a value as possible with a recommended maximum
value of
10 Ω or less. Increasing the value of the input filter resistance beyond 10 Ω results in a smaller voltage signal
present at the device input terminals than what is developed across the current-sense shunt resistor.
The internal bias network shown in Figure 7-1 creates a mismatch in the two input bias current paths when a
differential voltage is applied between the input terminals. Under normal conditions, where no external resistance
is added to the input paths, this mismatch of input bias currents has little effect on device operation or accuracy.
However, when additional external resistance is added (such as for input filtering), the mismatch of input bias
currents creates unequal voltage drops across these external components. The mismatched voltages result
in a signal reaching the input terminals that is lower in value than the signal developed directly across the
current-sensing resistor.
The amount of variance in the differential voltage present at the device input relative to the voltage developed
at the shunt resistor is based both on the external series resistance value (RS) and the internal input resistors
(RINT = 25 kΩ). The reduction of the shunt voltage reaching the device input terminals appears as a gain error
when comparing the output voltage relative to the voltage across the shunt resistor. A factor can be calculated to
determine the amount of gain error that is introduced by the addition of external series resistance.
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The amount of error these external filter resistors introduce into the measurement can be calculated using the
simplified gain error factor in Equation 1, where the gain error factor is calculated with Equation 2.
Gain Error Factor =
50,000
(41 x RS) + 50,000
(1)
(1250 ´ RINT)
Gain Error Factor =
(1250 ´ RS) + (1250 ´ RINT) + (RS ´ RINT)
(2)
where:
•
•
RINT is the internal input impedance, and
RS is the external series resistance.
For example, using the gain error factor (Equation 1), a 10-Ω series resistance results in a gain error factor
of 0.992. The corresponding gain error is then calculated using Equation 3, resulting in a gain error of
approximately 0.81% solely because of the external 10-Ω series resistors. Using 100-Ω filter resistors increases
this gain error to approximately 7.58% from these resistors alone.
Gain Error (%) = 1 ± Gain Error Factor
(3)
7.4.2 Shutting Down the Device
Although the device does not have a shutdown terminal, the low-power consumption allows for the device to be
powered from the output of a logic gate or transistor switch that can turn on and turn off the voltage connected to
the device power-supply terminal.
However, in current-shunt monitoring applications, there is also a concern for how much current is drained from
the shunt circuit in shutdown conditions. Evaluating this current drain involves considering the device simplified
schematic in shutdown mode, as shown in Figure 7-2.
CBYPASS
0.1 µF
Supply
Shutdown
Control
Load
VS
Device
IN-
OUT
Output
Reference
Voltage
+
IN+
GS0
GS1
+
REF
-
GND
Figure 7-2. Shutting Down the Device
Note that there is typically a 525-kΩ impedance (from the combination of the 500-kΩ feedback and 25-kΩ input
resistors) from each device input to the REF terminal. The amount of current flowing through these terminals
depends on the respective configuration. For example, if the REF terminal is grounded, calculating the effect
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of the 525-kΩ impedance from the shunt to ground is straightforward. However, if the reference or op amp
is powered while the device is shut down, the calculation is direct. Instead of assuming 525 kΩ to ground,
assume 525 kΩ to the reference voltage. If the reference or op amp is also shut down, some knowledge of the
reference or op amp output impedance under shutdown conditions is required. For instance, if the reference
source behaves similar to an open circuit when un-powered, little or no current flows through the 525-kΩ path.
7.4.3 Using the Device with Common-Mode Transients Above 36 V
With a small amount of additional circuitry, the device can be used in circuits subject to transients higher than
36 V (such as automotive applications). Use only zener diodes or zener-type transient absorbers (sometimes
referred to as transzorbs); any other type of transient absorber has an unacceptable time delay. Start by adding
a pair of resistors, as shown in Figure 7-3, as a working impedance for the zener. Keeping these resistors
as small as possible is preferable, most often around 10 Ω. This value limits the impact on accuracy with
the addition of these external components, as described in the Input Filtering section. Larger values can be
used if necessary with the result having an impact on gain error. Because this circuit limits only short-term
transients, many applications are satisfied with a 10-Ω resistor along with conventional zener diodes of the
lowest power rating available. This combination uses the least amount of board space. These diodes can be
found in packages as small as SOT-523 or SOD-523.
RSHUNT
Power
Supply
5-V Supply
Load
CBYPASS
0.1µF
RPROTECT
” 10
VS
Device
IN-
OUT
Output
+
IN+
REF
GS0
GS1
GND
Figure 7-3. Device Transient Protection
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8 Applications and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
8.1 Application Information
The INA225-Q1 measures the voltage developed across a current-sensing resistor when current passes through
it. The ability to drive the reference terminal to adjust the functionality of the output signal offers multiple
configurations discussed throughout this section.
8.2 Typical Applications
8.2.1 Microcontroller-Configured Gain Selection
RSHUNT
Power
Supply
5-V Supply
Load
CBYPASS
0.1 µF
VS
Device
IN-
OUT
+
ADC
Microcontroller
IN+
GPIO
REF
GS0
GS1
GND
Figure 8-1. Microcontroller-Configured Gain Selection Schematic
8.2.1.1 Design Requirements
Figure 8-1 shows the typical implementation of the device interfacing with an analog-to-digital converter (ADC)
and microcontroller.
8.2.1.2 Detailed Design Procedure
In this application, the device gain setting is selected and controlled by the microcontroller to ensure the device
output is within the linear input range of the ADC. Because the output range of the device under a specific gain
setting approaches the linear output range of the INA225-Q1 itself or the linear input range of the ADC, the
microcontroller can adjust the device gain setting to ensure the signal remains within both the device and the
ADC linear signal range.
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8.2.1.3 Application Curve
Figure 8-2 illustrates how the microcontroller can monitor the ADC measurements to determine if the device gain
setting should be adjusted to ensure the output of the device remains within the linear output range as well as
the linear input range of the ADC. When the output of the device rises to a level near the desired maximum
voltage level, the microcontroller can change the GPIO settings connected to the G0 and G1 gain-select
terminals to adjust the device gain setting, thus resulting in the output voltage dropping to a lower output range.
When the input current increases, the output voltage increases again to the desired maximum voltage level. The
microcontroller can again change the device gain setting to drop the output voltage back to a lower range.
Output Voltage
200
Gain (V/V)
5
Gain
4
150
3
100
2
50
1
Output Voltage (V)
250
0
0
0
1
2
3
4
5
6
7
8
Load Current (A)
9
10
C035
Figure 8-2. Microcontroller-Configured Gain Selection Response
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8.2.2 Unidirectional Operation
Supply
2.7-V to 36-V
Supply
Load
CBYPASS
0.1 µF
VS
Device
IN-
OUT
Output
+
IN+
REF
VS
GS0
GS1
GND
Figure 8-3. Unidirectional Application Schematic
8.2.2.1 Design Requirements
The device can be configured to monitor current flowing in one direction or in both directions, depending on
how the REF terminal is configured. For measuring current in one direction, only the REF terminal is typically
connected to ground as shown in Figure 8-3. With the REF terminal connected to ground, the output is low
with no differential input signal applied. When the input signal increases, the output voltage at the OUT terminal
increases above ground based on the device gain setting.
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8.2.2.2 Detailed Design Procedure
The linear range of the output stage is limited in how close the output voltage can approach ground under zero
input conditions. Resulting from an internal node limitation when the REF terminal is grounded (unidirectional
configuration) the device gain setting determines how close to ground the device output voltage can achieve
when no signal is applied; see Figure 6-14. To overcome this internal node limitation, a small reference voltage
(approximately 10 mV) can be applied to the REF terminal to bias the output voltage above this voltage level.
The device output swing capability returns to the 10-mV saturation level with this small reference voltage
present.
At the lowest gain setting, 25 V/V, the device is capable of accurately measuring input signals that result in
output voltages below this 10-mV saturation level of the output stage. For these gain settings, a reference
voltage can be applied to bias the output voltage above this lower saturation level to allow the device to monitor
these smaller input signals. To avoid common-mode rejection errors, buffer the reference voltage connected to
the REF terminal.
A less frequently-used output biasing method is to connect the REF terminal to the supply voltage, VS. This
method results in the output voltage saturating at 200 mV below the supply voltage when no differential input
signal is present. This method is similar to the output saturated low condition with no input signal when the
REF terminal is connected to ground. The output voltage in this configuration only responds to negative currents
that develop negative differential input voltage relative to the device IN– terminal. Under these conditions, when
the differential input signal increases negatively, the output voltage moves downward from the saturated supply
voltage. The voltage applied to the REF terminal must not exceed the device supply voltage.
8.2.2.3 Application Curve
Output Voltage
(0.5 V/div)
An example output response of a unidirectional configuration is shown in Figure 8-4. With the REF terminal
connected directly to ground, the output voltage is biased to this zero output level. The output rises above the
reference voltage for positive differential input signals but cannot fall below the reference voltage for negative
differential input signals because of the grounded reference voltage.
0V
Output
Vref
Time (500 µs/div)
C036
Figure 8-4. Unidirectional Application Output Response
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8.2.3 Bidirectional Operation
Supply
2.7-V to 36-V
Supply
Load
CBYPASS
0.1µF
VS
Device
IN-
OUT
Output
Reference
Voltage
+
IN+
+
VS
GS0
GS1
-
-
REF
GND
Figure 8-5. Bidirectional Application Schematic
8.2.3.1 Design Requirements
The device is a bidirectional, current-sense amplifier capable of measuring currents through a resistive shunt
in two directions. This bidirectional monitoring is common in applications that include charging and discharging
operations where the current flow-through resistor can change directions.
8.2.3.2 Detailed Design Procedure
The ability to measure this current flowing in both directions is enabled by applying a voltage to the REF
terminal, as shown in Figure 8-5. The voltage applied to REF (VREF) sets the output state that corresponds to
the zero-input level state. The output then responds by increasing above VREF for positive differential signals
(relative to the IN– terminal) and responds by decreasing below VREF for negative differential signals. This
reference voltage applied to the REF terminal can be set anywhere between 0 V to VS. For bidirectional
applications, VREF is typically set at mid-scale for equal range in both directions. In some cases, however, VREF
is set at a voltage other than half-scale when the bidirectional current is non-symmetrical.
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8.2.3.3 Application Curve
Output Voltage
(0.5 V/div)
An example output response of a bidirectional configuration is shown in Figure 8-6. With the REF terminal
connected to a reference voltage, 2.5 V in this case, the output voltage is biased upwards by this reference level.
The output rises above the reference voltage for positive differential input signals and falls below the reference
voltage for negative differential input signals.
Output
0V
Vref
Time (500 µs/div)
C037
Figure 8-6. Bidirectional Application Output Response
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9 Power Supply Recommendations
The input circuitry of the device can accurately measure signals on common-mode voltages beyond its power
supply voltage, VS. For example, the voltage applied to the VS power supply terminal can be 5 V, whereas the
load power-supply voltage being monitored (the common-mode voltage) can be as high as +36 V. Note also that
the device can withstand the full –0.3-V to +36-V range at the input terminals, regardless of whether the device
has power applied or not.
Power-supply bypass capacitors are required for stability and should be placed as closely as possible to
the supply and ground terminals of the device. A typical value for this supply bypass capacitor is 0.1 μF.
Applications with noisy or high-impedance power supplies may require additional decoupling capacitors to reject
power-supply noise.
10 Layout
10.1 Layout Guidelines
•
•
Connect the input terminals to the sensing resistor using a Kelvin or 4-wire connection. This connection
technique ensures that only the current-sensing resistor impedance is detected between the input terminals.
Poor routing of the current-sensing resistor commonly results in additional resistance present between the
input terminals. Given the very low ohmic value of the current resistor, any additional high-current carrying
impedance can cause significant measurement errors.
The power-supply bypass capacitor should be placed as closely as possible to the supply and ground
terminals. The recommended value of this bypass capacitor is 0.1 μF. Additional decoupling capacitance can
be added to compensate for noisy or high-impedance power supplies.
10.2 Layout Example
VIA to Power or Ground Plane
VIA to Ground Plane
Supply Bypass
Capacitor
Supply
Voltage
Output Signal Trace
IN+
IN-
GND
REF
VS
GS1
OUT
GS0
Figure 10-1. Recommended Layout
Note
The layout shown has REF connected to ground for unidirectional operation. Gain-select terminals
(GS0 and GS1) are also connected to ground, indicating a 25-V/V gain setting.
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11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
For related documentation see the following:
•
INA225EVM User's Guide, SBOU140
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
11.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.4 Trademarks
TI E2E™ is a trademark of Texas Instruments.
All trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
INA225AQDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
IAAQ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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